AD7226 [ADI]
LC2MOS Quad 8-Bit D/A Converter; LC2MOS四通道8位D / A转换器
| 型号： | AD7226 |
| 厂家： | 
ADI |
| 描述： | LC2MOS Quad 8-Bit D/A Converter |
| 文件： | 总12页 (文件大小：249K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
2
LC MOS
Quad 8-Bit D/A Converter
a
AD7226
FEATURES
FUNCTIO NAL BLO CK D IAGRAM
Four 8-Bit DACs w ith Output Am plifiers
Skinny 20-Pin DIP, SOIC and 20-Term inal
Surface Mount Packages
Microprocessor Com patible
TTL/ CMOS Com patible
No User Trim s
Extended Tem perature Range Operation
Single Supply Operation Possible
APPLICATIONS
Process Control
Autom atic Test Equipm ent
Autom atic Calibration of Large System Param eters,
e.g., Gain/ Offset
GENERAL D ESCRIP TIO N
P RO D UCT H IGH LIGH TS
T he AD7226 contains four 8-bit voltage-output digital-to-
analog converters, with output buffer amplifiers and interface
logic on a single monolithic chip. No external trims are required
to achieve full specified performance for the part.
1. DAC-to-DAC Matching
Since all four DACs are fabricated on the same chip at the
same time, precise matching and tracking between the DACs
is inherent.
Separate on-chip latches are provided for each of the four D/A
converters. Data is transferred into one of these data latches
through a common 8-bit T T L/CMOS (5 V) compatible input
port. Control inputs A0 and A1 determine which DAC is loaded
when WR goes low. T he control logic is speed-compatible with
most 8-bit microprocessors.
2. Single Supply Operation
T he voltage mode configuration of the DACs allows the
AD7226 to be operated from a single power supply rail.
3. Microprocessor Compatibility
T he AD7226 has a common 8-bit data bus with individual
DAC latches, providing a versatile control architecture for
simple interface to microprocessors. All latch enable signals
are level triggered.
Each D/A converter includes an output buffer amplifier capable
of driving up to 5 mA of output current. T he amplifiers’ offsets
are laser-trimmed during manufacture, thereby eliminating any
requirement for offset nulling.
4. Small Size
Combining four DACs and four op amps plus interface logic
into a 20-pin DIP or SOIC or a 20-terminal surface mount
package allows a dramatic reduction in board space require-
ments and offers increased reliability in systems using mul-
tiple converters. Its pinout is aimed at optimizing board
layout with all the analog inputs and outputs at one end of the
package and all the digital inputs at the other.
Specified performance is guaranteed for input reference voltages
from +2 V to +12.5 V with dual supplies. T he part is also speci-
fied for single supply operation at a reference of +10 V.
T he AD7226 is fabricated in an all ion-implanted high speed
Linear Compatible CMOS (LC2MOS) process which has been
specifically developed to allow high speed digital logic circuits
and precision analog circuits to be integrated on the same chip.
REV. A
Inform ation furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assum ed by Analog Devices for its
use, nor for any infringem ents of patents or other rights of third parties
which m ay result from its use. No license is granted by im plication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norw ood, MA 02062-9106, U.S.A.
Tel: 617/ 329-4700
Fax: 617/ 326-8703
AD7226–SPECIFICATIONS
(V = 11.4 V to 16.5 V, V = –5 V ؎ 10%; AGND = DGND = O V; V = +2 V to (V – 4 V)1 unless otherwise noted.
DD
SS
REF
DD
DUAL SUPPLY
P aram eter
All specifications TMIN to TMAX unless otherwise noted.)
K, B, T Versions2
Units
Conditions/Com m ents
ST AT IC PERFORMANCE
Resolution
8
Bits
T otal Unadjusted Error
Relative Accuracy
Differential Nonlinearity
Full Scale Error
Full Scale T emperature Coefficient
Zero Code Error
Zero Code Error T emperature Coefficient
±2
±1
±1
±1 1/2
±20
±30
±50
LSB max
LSB max
LSB max
LSB max
ppm/°C typ
mV max
µV/°C typ
VDD = +15 V ± 5%, VREF = +10 V
Guaranteed Monotonic
VDD = 14 V to 16.5 V, VREF = +10 V
REFERENCE INPUT
Voltage Range
2 to (VDD – 4)
2
65
300
V min to V max
kΩ min
pF min
Input Resistance
Input Capacitance3
Occurs when each DAC is loaded with all 0s.
Occurs when each DAC is loaded with all 1s.
pF max
DIGIT AL INPUT S
Input High Voltage, VINH
Input Low Voltage, VINL
Input Leakage Current
Input Capacitance
2.4
0.8
±1
V min
V max
µA max
pF max
VIN = 0 V or VDD
8
Input Coding
Binary
DYNAMIC PERFORMANCE
Voltage Output Slew Rate4
Voltage Output Settling T ime4
Positive Full Scale Change
Negative Full Scale Change
Digital Crosstalk
2.5
V/µs min
5
7
50
2
µs max
µs max
nV secs typ
kΩ min
VREF = +10 V; Settling T ime to ±1/2 LSB
VREF = +10 V; Settling T ime to ±1/2 LSB
Minimum Load Resistance
VOUT = +10 V
POWER SUPPLIES
VDD Range
IDD
ISS
11.4/16.5
13
11
V min/V max
mA max
mA max
For Specified Performance
Outputs Unloaded; VIN = VINL or VINH
Outputs Unloaded; VIN = VINL or VINH
SWITCH ING CH ARACTERISTICS4, 5
Address to Write Setup T ime, tAS
@ 25°C
0
0
ns min
ns min
T MIN to T MAX
Address to Write Hold T ime, tAH
@ 25°C
T MIN to T MAX
10
10
ns min
ns min
Data Valid to Write Setup T ime, tDS
@ 25°C
T MIN to T MAX
90
100
ns min
ns min
Data Valid to Write Hold T ime, tDH
@ 25°C
T MIN to T MAX
10
10
ns min
ns min
Write Pulse Width, tWR
@ 25°C
T MIN to T MAX
150
200
ns min
ns min
NOT ES
1Maximum possible reference voltage.
2T emperature ranges are as follows:
K Version: –40°C to +85°C
B Version: –40°C to +85°C
T Version: –55°C to +125°C
3Guanteed by design. Not production tested.
4Sample T ested at 25°C to ensure compliance.
5Switching Characteristics apply for single and dual supply operation.
Specifications subject to change without notice.
–2–
REV. A
AD7226
1
(V = +15 V ؎ 5%; V = AGND = DGND = O V; V = +10 V unless otherwise noted.
DD
SS
REF
SINGLE SUPPLY
P aram eter
All specifications TMIN to TMAX unless otherwise noted.)
K, B, T Versions2
Units
Conditions/Com m ents
ST AT IC PERFORMANCE
Resolution
8
Bits
T otal Unadjusted Error
Differential Nonlinearity
±2
±1
LSB max
LSB max
Guaranteed Monotonic
REFERENCE INPUT
Input Resistance
2
65
300
kΩ min
pF min
pF max
Input Capacitance3
Occurs when each DAC is loaded with all 0s.
Occurs when each DAC is loaded with all 1s.
DIGIT AL INPUT S
Input High Voltage, VINH
Input Low Voltage, VINL
Input Leakage Current
Input Capacitance
2.4
0.8
±1
V min
V max
µA max
pF max
VIN = 0 V or VDD
8
Input Coding
Binary
DYNAMIC PERFORMANCE
Voltage Output Slew Rate4
Voltage Output Settling T ime4
Positive Full Scale Change
Negative Full Scale Change
Digital Crosstalk
2
V/µs min
5
µs max
µs max
nV secs typ
kΩ min
Settling T ime to ±1/2 LSB
Settling T ime to ±1/2 LSB
20
50
2
Minimum Load Resistance
VOUT = +10 V
POWER SUPPLIES
VDD Range
IDD
14.25/15.75
13
V min/V max
mA max
For Specified Performance
Outputs Unloaded; VIN = VINL or VINH
NOT ES
1Maximum possible reference voltage.
2T emperature ranges are as follows:
K Version: –40°C to +85°C
B Version: –40°C to +85°C
T Version: –55°C to +125°C
3Guanteed by design. Not production tested.
4Sample T ested at 25°C to ensure compliance.
5Switching Characteristics apply for single and dual supply operation.
Specifications subject to change without notice.
O RD ERING GUID E
Total
Tem perature
Range
Unadjusted
Error
P ackage
O ption2
Model1
AD7226KN
AD7226KP
AD7226KR
AD7226BQ
AD7226T Q
AD7226T E
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–55°C to +125°C
–55°C to +125°C
±2 LSB
±2 LSB
±2 LSB
±2 LSB
±2 LSB
±2 LSB
N-20
P-20A
R-20
Q-20
Q-20
E-20A
NOT ES
1T o order MIL-ST D-883, Class B processed parts, add /883B to part number.
Contact your local sales office for Military data sheet, for U.S. Standard Military
Drawing (SMD), see DESC drawing # 5962–87802.
2E = Leadless Ceramic Chip Carrier; N = Plastic DIP;
P = Plastic Leaded Chip Carrier; Q = Cerdip; R = SOIC.
–3–
REV. A
AD7226
ABSO LUTE MAXIMUM RATINGS*
Operating T emperature
VDD to AGND . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V, +17 V
VDD to DGND . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V, +17 V
VSS to AGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –7 V, VDD
VSS to DGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –7 V, VDD
VDD to VSS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V, +24 V
AGND to DGND . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V, VDD
Digital Input Voltage to DGND . . . . . . . –0.3 V, VDD + 0.3 V
VREF to AGND . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V, VDD
VOUT to AGND1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . VSS, VDD
Power Dissipation (Any Package) to +75°C . . . . . . . . 500 mW
Derates above 75°C by . . . . . . . . . . . . . . . . . . . . . 2.0 mW/°C
Commercial (K Version) . . . . . . . . . . . . . . –40°C to +85°C
Industrial (B Version) . . . . . . . . . . . . . . . . –40°C to +85°C
Extended (T Version) . . . . . . . . . . . . . . . –55°C to +125°C
Storage T emperature . . . . . . . . . . . . . . . . . . –65°C to +150°C
Lead T emperature (Soldering, 10 secs) . . . . . . . . . . . +300°C
NOT ES
*Stresses above those listed under “Absolute Maximum Ratings” may cause
permanent damage to the device. T his is a stress rating only and functional
operation of the device at these or any other conditions above those indicated in
the operational sections of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
1Outputs may be shorted to AGND provided that the power dissipation of the
package is not exceeded. T ypically short circuit current to AGND is 60 mA.
CAUTIO N
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD7226 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. T herefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
P IN CO NFIGURATIO NS
D IP and SO IC
LCCC
P LCC
TERMINO LO GY
TO TAL UNAD JUSTED ERRO R
T his is a comprehensive specification which includes full-scale
error, relative accuracy and zero code error. Maximum output
voltage is VREF – 1 LSB (ideal), where 1 LSB (ideal) is VREF
D IFFERENTIAL NO NLINEARITY
Differential Nonlinearity is the difference between the measured
change and the ideal 1 LSB change between any two adjacent
codes. A specified differential nonlinearity of ±1 LSB max over
the operating temperature range ensures monotonicity.
/
256. T he LSB size will vary over the VREF range. Hence the zero
code error will, relative to the LSB size, increase as VREF de-
creases. Accordingly, the total unadjusted error, which includes
the zero code error, will also vary in terms of LSB’s over the
VREF range. As a result, total unadjusted error is specified for a
fixed reference voltage of +10 V.
D IGITAL CRO SSTALK
T he glitch impulse transferred to the output of one converter
due to a change in the digital input code to another of the con-
verters. It is specified in nV secs and is measured at VREF = 0 V.
FULL SCALE ERRO R
Full-Scale Error is defined as:
Measured Value – Zero Code Error – Ideal Value
RELATIVE ACCURACY
Relative Accuracy or endpoint nonlinearity, is a measure of the
maximum deviation from a straight line passing through the
endpoints of the DAC transfer function. It is measured after al-
lowing for zero and full-scale error and is normally expressed in
LSB’s or as a percentage of full-scale reading.
–4–
REV. A
AD7226
CIRCUIT INFO RMATIO N
D /A SECTIO N
T he AD7226 contains four, identical, 8-bit, voltage mode
digital-to-analog converters. T he output voltages from the con-
verters have the same polarity as the reference voltage allowing
single supply operation. A novel DAC switch pair arrangement
on the AD7226 allows a reference voltage range from +2 V to
+12.5 V.
Each DAC consists of a highly stable, thin-film, R-2R ladder
and eight high speed NMOS, single-pole, double-throw
switches. T he simplified circuit diagram for one channel is
shown in Figure 1. Note that VREF (Pin 4) and AGND (Pin 5)
are common to all four DACs.
Figure 2. Am plifier Output Stage
the current load ceases to act as a current sink and begins to act
as a resistive load of approximately 2 kΩ to AGND. T his occurs
as the NMOS transistors come out of saturation. T his means
that, in single supply operation, the sink capability of the ampli-
fiers is reduced when the output voltage is at or near AGND. A
typical plot of the variation of current sink capability with out-
put voltage is shown in Figure 3.
Figure 1. D/A Sim plified Circuit Diagram
T he input impedance at the VREF pin of the AD7226 is the par-
allel combination of the four individual DAC reference input
impedances. It is code dependent and can vary from 2 kΩ to in-
finity. T he lowest input impedance (i.e., 2 kΩ) occurs when all
four DACs are loaded with the digital code 01010101. T here-
fore, it is important that the reference presents a low output im-
pedance under changing load conditions. T he nodal capacitance
at the reference terminals is also code dependent and typically
varies from 100 pF to 250 pF.
Each VOUT pin can be considered as a digitally programmable
voltage source with an output voltage of:
VOUT X = DX VREF
where DX is fractional representation of the digital input code
and can vary from 0 to 255/256.
T he source impedance is the output resistance of the buffer
amplifier.
Figure 3. Variation of ISINK with VOUT
If the full sink capability is required with output voltages at or
near AGND (=0 V), then VSS can be brought below 0 V by 5 V
and thereby maintain the 400 µA current sink as indicated in
Figure 3. Biasing VSS below 0 V also gives additional headroom
in the output amplifier which allows for better zero code error
performance on each output. Also improved is the slew-rate
and negative-going settling-time of the amplifiers (discussed
later).
O P AMP SECTIO N
Each voltage-mode D/A converter output is buffered by a unity
gain, noninverting CMOS amplifier. T his buffer amplifier is
capable of developing +10 V across a 2 kΩ load and can drive
capacitive loads of 3300 pF. T he output stage of this amplifier
consists of a bipolar transistor from the VDD line and a current
load to the VSS, the negative supply for the output amplifiers.
T his output stage is shown in Figure 2.
Each amplifier offset is laser trimmed during manufacture to
eliminate any requirement for offset nulling.
T he NPN transistor supplies the required output current drive
(up to 5 mA). T he current load consists of NMOS transistors
which normally act as a constant current sink of 400 µA to VSS
giving each output a current sink capability of approximately
400 µA if required.
,
D IGITAL SECTIO N
T he digital inputs of the AD7226 are both T T L and CMOS
(5 V) compatible from VDD = +11.4 V to +16.5 V. All logic in-
puts are static protected MOS gates with typical input currents
of less than 1 nA. Internal input protection is achieved by an
on-chip distributed diode from DGND to each MOS gate. T o
minimize power supply currents, it is recommended that the
digital input voltages be driven as close to the supply rails (VDD
and DGND) as practically possible.
T he AD7226 can be operated single or dual supply resulting
in different performance in some parameters from the output
amplifiers.
In single supply operation (VSS = 0 V = AGND), with the out-
put approaching AGND (i.e., digital code approaching all 0s)
–5–
REV. A
AD7226
INTERFACE LO GIC INFO RMATIO N
Typical Performance Characteristics
Address lines A0 and A1 select which DAC will accept data
from the input port. T able I shows the selection table for the
four DACs with Figure 4 showing the input control logic. When
the WR signal is LOW, the input latches of the selected DAC
are transparent and its output responds to activity on the data
bus. T he data is latched into the addressed DAC latch on the
rising edge of WR. While WR is high the analog outputs remain
at the value corresponding to the data held in their respective
latches.
(T = 25؇C, V = +15 V, V = –5 V)
A
DD
SS
Table I. AD 7226 Truth Table
AD 7226 Control Inputs
AD 7226
WR
A1
A0
O peration
H
L
g
L
g
L
g
L
g
X
L
X
L
No Operation Device Not Selected
DAC A T ransparent
DAC A Latched
L
L
L
H
H
L
DAC B T ransparent
DAC B Latched
L
Figure 6. Channel-to-Channel Matching
H
H
H
H
DAC C T ransparent
DAC C Latched
L
H
H
DAC D T ransparent
DAC D Latched
L = Low State, H = High State, X = Don’t Care
Figure 4. Input Control Logic
Figure 7. Relative Accuracy vs. VREF
Figure 5. Write Cycle Tim ing Diagram
Figure 8. Differential Nonlinearity vs. VREF
–6–
REV. A
AD7226
Figure 10. Dynam ic Response (VSS = –5 V)
Figure 9. Zero Code Error vs. Tem perature
SP ECIFICATIO N RANGES
In order for the DACs to operate to their specifications, the ref-
erence voltage must be at least 4 V below the VDD power supply
voltage. T his voltage differential is required for correct genera-
tion of bias voltages for the DAC switches.
T he AD7226 is specified to operate over a VDD range from
+12 V ± 5% to +15 V ± 10% (i.e., from +11.4 V to +16.5 V)
with a VSS of –5 V ± 10%. Operation is also specified for a
single +15 V ± 5% VDD supply. Applying a VSS of –5 V results
in improved zero code error, improved output sink capability
with outputs near AGND and improved negative-going settling-
time.
Figure 11a. Positive-Step Settling-Tim e (VSS = –5 V)
Performance is specified over a wide range of reference voltages
from 2 V to (VDD – 4 V) with dual supplies. T his allows a range
of standard reference generators to be used such as the AD580,
a +2.5 V bandgap reference and the AD584, a precision +10 V
reference. Note that in order to achieve an output voltage range
of 0 V to +10 V a nominal +15 V ± 5% power supply voltage is
required by the AD7226.
SETTLING TIME
T he output stage of the buffer amplifiers consists of a bipolar
NPN transistor from the VDD line and a constant current load to
VSS. VSS is the negative power supply for the output buffer am-
plifiers. As mentioned in the op amp section, in single supply
operation the NMOS transistor will come out of saturation as
the output voltage approaches AGND and will act as a resistive
load of approximately 2 kΩ to AGND. As a result, the settling-
time for negative-going signals approaching AGND in single
supply operation will be longer than for dual supply operation
where the current load of 400 µA is maintained all the way down
to AGND. Positive-going settling-time is not affected by VSS.
Figure 11b. Negative-Step Settling-Tim e (VSS = –5 V)
GRO UND MANAGEMENT
AC or transient voltages between AGND and DGND can cause
noise at the analog output. T his is especially true in micropro-
cessor systems where digital noise is prevalent. T he simplest
method of ensuring that voltages at AGND and DGND are
equal is to tie AGND and DGND together at the AD7226. In
more complex systems where the AGND and DGND intertie is
on the backplane, it is recommended that two diodes be con-
nected in inverse parallel between the AD7226 AGND and
DGND pins (IN914 or equivalent).
T he settling-time for the AD7226 is limited by the slew-rate of
the output buffer amplifiers. T his can be seen from Figure 10
which shows the dynamic response for the AD7226 for a full
scale change. Figures 11a and 11b show expanded settling-time
photographs with the output waveforms derived from a differen-
tial input to an oscilloscope. Figure 11a shows the settling-time
for a positive-going step and Figure 11b shows the settling-time
for a negative-going output step.
–7–
REV. A
AD7226
Unipolar O utput O per ation
operation) with DAC A of the AD7226. In this case
T his is the basic mode of operation for each channel of the
AD7226, with the output voltage having the same positive
polarity as +VREF. T he AD7226 can be operated single supply
(VSS = AGND) or with positive/negative supplies (see op-amp
section which outlines the advantages of having negative VSS).
T he code table for unipolar output operation is shown in T able
II. Note that the voltage at VREF must never be negative with re-
spect to DGND in order to prevent parasitic transistor turn-on.
Connections for the unipolar output operation are shown in Fig-
ure 12.
R2
R1
R2
R1
VOUT = 1 +
D V
–
V
(
REF
(
)
)
A
REF
With R1 = R2
VOUT = (2 DA – 1) • VREF
where DA is a fractional representation of the digital word in
latch A.
Mismatch between R1 and R2 causes gain and offset errors and
therefore these resistors must match and track over temperature.
Once again the AD7226 can be operated in single supply or
from positive/negative supplies. T able III shows the digital code
versus output voltage relationship for the circuit of Figure 13
with R1 = R2.
Figure 13. AD7226 Bipolar Output Circuit
Table III. Bipolar (O ffset Binary) Code Table
D AC Latch Contents
Figure 12. AD7226 Unipolar Output Circuit
MSB
LSB
Analog O utput
Table II. Unipolar Code Table
D AC Latch Contents
127
+VREF
1 1 1 1
1 0 0 0
1 0 0 0
0 1 1 1
0 0 0 0
0 0 0 0
1 1 1 1
0 0 0 1
0 0 0 0
1 1 1 1
0 0 0 1
0 0 0 0
128
MSB
LSB
Analog O utput
1
+VREF
128
255
+VREF
1 1 1 1 1 1 1 1
1 0 0 0 0 0 0 1
1 0 0 0 0 0 0 0
256
0 V
129
+VREF
256
1
–VREF
128
VREF
2
128
256
+VREF
= +
127
–VREF
128
127
256
+VREF
+VREF
0 1 1 1 1 1 1 1
0 0 0 0 0 0 0 1
0 0 0 0 0 0 0 0
128
128
–VREF
= –VREF
1
256
AGND BIAS
0 V
T he AD7226 AGND pin can be biased above system GND
(AD7226 DGND) to provide an offset “zero” analog output
voltage level. Figure 14 shows a circuit configuration to achieve
this for channel A of the AD7226. T he output voltage, VOUT A
can be expressed as:
1
256
Note: 1 LSB = V
2−8 = V
(
)
(
)
REF
REF
,
Bipolar O utput O per ation
V
OUTA = VBIAS + DA (VIN)
Each of the DACs of the AD7226 can be individually config-
ured to provide bipolar output operation. T his is possible using
one external amplifier and two resistors per channel. Figure 13
shows a circuit used to implement offset binary coding (bipolar
where DA is a fractional representation of the digital input
word (0 ≤ D ≤ 255/256).
–8–
REV. A
AD7226
where G = RF/R
and DD is a fractional representation of the digital word
in latch D.
Alternatively, for a given VIN and resistance ratio, the required
value of DD for a given value of VREF can be determined from
the expression
V IN
VREF RF
R
DD = (1 + R / RF )
–
Figure 16 shows typical plots of VREF versus digital code for
three different values of RF. With VIN = +2.5 V and RF = 3 R
the peak-to-peak sine wave voltage from the converter outputs
will vary between +2.5 V and +10 V over the digital input code
range of 0 to 255.
Figure 14. AGND Bias Circuit
For a given VIN, increasing AGND above system GND will re-
duce the effective VDD–VREF which must be at least 4 V to en-
sure specified operation. Note that because the AGND pin is
common to all four DACs, this method biases up the output
voltages of all the DACs in the AD7226. Note that VDD and VSS
of the AD7226 should be referenced to DGND.
3-P H ASE SINE WAVE
T he circuit of Figure 15 shows an application of the AD7226 in
the generation of 3-phase sine waves which can be used to con-
trol small 3-phase motors. T he proper codes for synthesizing a
full sine wave are stored in EPROM, with the required phase-
shift of 120° between the three D/A converter outputs being
generated in software.
Data is loaded into the three D/A converters from the sine
EPROM via the microprocessor or control logic. T hree loops
are generated in software with each D/A converter being loaded
from a separate loop. T he loops run through the look-up table
producing successive triads of sinusoidal values with 120° sepa-
ration which are loaded to the D/A converters producing 3 sine
wave voltages 120° apart. A complete sine wave cycle is gener-
ated by stepping through the full look-up table. If a 256-element
sine wave table is used then the resolution of the circuit will be
1.4° (360°/256). Figure 17 shows typical resulting waveforms.
T he sine waves can be smoothed by filtering the D/A converter
outputs.
Figure 16. Variation of VREF with Feedback Configuration
T he fourth D/A converter of the AD7226, DAC D, may be used
in a feedback configuration to provide a programmable refer-
ence voltage for itself and the other three converters. T his con-
figuration is shown in Figure 15. T he relationship of VREF to VIN
is dependent upon digital code and upon the ratio of RF to R
and is given by the formula
(1 + G)
(1 + G. DD )
VREF
=
V IN
Figure 17. 3-Phase Sine Wave Output
Figure 15. 3-Phase Sine Wave Generation Circuit
–9–
REV. A
AD7226
STAIRCASE WIND O W CO MP ARATO R
In many test systems, it is important to be able to determine
whether some parameter lies within defined limits. T he staircase
window comparator of Figure 18a is a circuit which can be
used, for example, to measure the VOH and VOL thresholds of a
T T L device under test. Upper and lower limits on both VOH
and VOL can be programmably set using the AD7226. Each ad-
jacent pair of comparators forms a window of programmable
size. If VT EST lies within a window then the output for that win-
dow will be high. With a reference of +2.56 V applied to the
VREF input, the minimum window size is 10 mV.
Figure 19a. Overlapping Windows
Figure 19b. Window Structure
Figure 18a. Logic Level Measurem ent
Figure 20. Varying Reference Signal
VARYING REFERENCE SIGNAL
In some applications, it may be desirable to have a varying sig-
nal applied to the reference input of the AD7226. T he AD7226
has multiplying capability within upper and lower limits of refer-
ence voltage when operated with dual supplies. T he upper and
lower limits are those required by the AD7226 to achieve its lin-
earity specification. Figure 20 shows a sine wave signal applied
to the reference input of the AD7226. For input signal frequen-
cies up to 50 kHz the output distortion typically remains less
than 0.1%. T ypical 3 dB bandwidth figure is 700 kHz.
Figure 18b. Window Structure
T he circuit can easily be adapted to allow for overlapping of
windows as shown in Figure 19a. If the three outputs from this
circuit are decoded then five different nonoverlapping program-
mable windows can again be defined.
–10–
REV. A
AD7226
O FFSET AD JUST
Figure 21 shows how the AD7226 can be used to provide pro-
grammable input offset voltage adjustment for the AD544 op
amp. Each output of the AD7226 can be used to trim the input
offset voltage on one AD544. T he 620 kΩ resistor tied to +10 V
provides a fixed bias current to one offset node. For symmetrical
adjustment, this bias current should equal the current in the
other offset node with the half-full scale code (i.e., 10000000)
on the DAC. Changing the code on the DAC varies the bias
current and hence provides offset adjust for the AD544. For ex-
ample, the input offset voltage on the AD544J, which has a
maximum of ±2 mV, can be programmably trimmed to ±10 µV.
Figure 21. Offset Adjust for AD544
Microprocessor Interface
Figure 22. AD7226 to 8085A Interface
Figure 24. AD7226 to 6502 Interface
Figure 25. AD7226 to Z-80 Interface
Figure 23. AD7226 to 6809 Interface
–11–
REV. A
AD7226
O UTLINE D IMENSIO NS
D imensions shown in inches and (mm).
20-P in P lastic (N-20)
20-Ter m inal P lastic Leaded
Chip Car r ier (P -20A)
20-P in Cer dip (Q -20)
20-Ter m inal Leadless
Cer am ic Chip Car r ier (E-20A)
20-P in SO IC (R-20)
–12–
REV. A
相关型号：
AD7226BRS
IC PARALLEL, 8 BITS INPUT LOADING, 8-BIT DAC, PDSO20, MO-150AE, SSOP-20, Digital to Analog Converter
ADI
©2020 ICPDF网 联系我们和版权申明