AD7712ANZ_技术文档

LC²MOS

Signal Conditioning ADC

AD7712

FEATURES

FUNCTIONAL BLOCK DIAGRAM

Charge Balancing ADC

24 Bits No Missing Codes

REF REF

DD IN (–) IN (+)

DV

V

BIAS

AV

REF OUT

DD

؎0.0015% Nonlinearity

High Level and Low Level Analog Input Channels

Programmable Gain for Both Inputs

Gains from 1 to 128

AV

DD

2.5V REFERENCE

4.5␮A

CHARGE-BALANCING A/D

CONVERTER

Differential Input for Low Level Channel

Low-Pass Filter with Programmable Filter Cutoffs

Ability to Read/Write Calibration Coefficients

Bidirectional Microcontroller Serial Interface

Internal/External Reference Option

Single- or Dual-Supply Operation

Low Power (25 mW typ) with Power-Down Mode

(100 ␮W typ)

AIN1(+)

AIN1(–)

AUTO-ZEROED

DIGITAL

SYNC

M

U

X

PGA

A = 1 – 128

⌺–⌬

FILTER

MODULATOR

STANDBY

MCLK

IN

CLOCK

GENERATION

VOLTAGE

ATTENUATION

AIN2

TP

MCLK

OUT

SERIAL INTERFACE

CONTROL

REGISTER

OUTPUT

REGISTER

APPLICATIONS

AD7712

Process Control

Smart Transmitters

Portable Industrial Instruments

AGND DGND V

DRDY A0

MODE SDATA SCLK

RFS TFS

SS

GENERAL DESCRIPTION

The AD7712 is a complete analog front end for low frequency

measurement applications. The device has two analog input

channels and accepts either low level signals directly from a trans-

ducer or high level ( 4 ϫ V_REF) signals, and outputs a serial

digital word. It employs a sigma-delta conversion technique to

realize up to 24 bits of no missing codes performance. The low

level input signal is applied to a proprietary programmable gain

front end based around an analog modulator. The high level

analog input is attenuated before being applied to the same

modulator. The modulator output is processed by an on-chip

digital filter. The first notch of this digital filter can be programmed

via the on-chip control register, allowing adjustment of the filter

cutoff and settling time.

port. The AD7712 also contains self-calibration, system calibra-

tion, and background calibration options, and allows the user to

read and to write the on-chip calibration registers.

CMOS construction ensures low power dissipation, and a hard-

ware programmable power-down mode reduces the standby power

consumption to only 100 µW typical. The part is available in a

24-lead, 0.3 inch wide, plastic and hermetic dual-in-line pack-

age (DIP), as well as a 24-lead small outline (SOIC) package.

PRODUCT HIGHLIGHTS

1. The low level analog input channel allows the AD7712 to

accept input signals directly from a strain gage or transducer,

removing a considerable amount of signal conditioning. To

maximize the flexibility of the part, the high level analog

input accepts signals of 4 ϫ V_REF/GAIN.

Normally, one of the channels will be used as the main channel

with the second channel used as an auxiliary input to periodi-

cally measure a second voltage. The part can be operated from a

single supply (by tying the V_SSpin to AGND), provided that the

input signals on the low level analog input are more positive

than –30 mV. By taking the V_SSpin negative, the part can con-

vert signals down to –V_REFon this low level input. This low level

input, as well as the reference input, features differential input

capability.

2. The AD7712 is ideal for microcontroller or DSP processor

applications with an on-chip control register that allows

control over filter cutoff, input gain, channel selection, signal

polarity, and calibration modes.

3. The AD7712 allows the user to read and to write the on-chip

calibration registers. This means that the microcontroller has

much greater control over the calibration procedure.

The AD7712 is ideal for use in smart, microcontroller based

systems. Input channel selection, gain settings, and signal polar-

ity can be configured in software using the bidirectional serial

4. No missing codes ensures true, usable, 23-bit dynamic range

coupled with excellent 0.0015% accuracy. The effects of

temperature drift are eliminated by on-chip self-calibration,

which removes zero-scale and full-scale errors.

REV. F

Information furnished by Analog Devices is believed to be accurate and

reliable. However, no responsibility is assumed by Analog Devices for its

use, norforanyinfringementsofpatentsorotherrightsofthirdpartiesthat

may result from its use. No license is granted by implication or otherwise

under any patent or patent rights of Analog Devices. Trademarks and

registered trademarks are the property of their respective owners.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.

Tel: 781/329-4700

Fax: 781/326-8703

www.analog.com

(AV = +5 V ؎ 5%; DV = +5 V ؎ 5%; V = 0 V or –5 V ؎ 5%; REF IN(+) = +2.5 V;

AD7712–SPECIFICATIONS

DD

SS

REF IN(–) = AGND; MCLK IN = 10 MHz unless otherwise stated. All specifications T_MINto T_MAX, unless otherwise noted.)

Parameter

A, S Versions¹

Unit

Conditions/Comments

STATIC PERFORMANCE

No Missing Codes

24

22

18

15

Bits min

Guaranteed by Design. For Filter Notches ≤ 60 Hz

For Filter Notch = 100 Hz

For Filter Notch = 250 Hz

For Filter Notch = 500 Hz

For Filter Notch = 1 kHz

12

Output Noise

Integral Nonlinearity @ 25°C

See Tables I and II

0.0015

0.003

Depends on Filter Cutoffs and Selected Gain

Filter Notches ≤ 60 Hz

% FSR max

T

_MINto T_MAX

Typically 0.0003%

Positive Full-Scale Error^{2, 3, 4}

Full-Scale Drift⁵

Excluding Reference

Excluding Reference. For Gains of 1, 2

Excluding Reference. For Gains of 4, 8, 16, 32, 64, 128

1

0.3

µV/°C typ

Unipolar Offset Error^{2, 4}

Unipolar Offset Drift⁵

0.5

0.25

µV/°C typ

For Gains of 1, 2

For Gains of 4, 8, 16, 32, 64, 128

Bipolar Zero Error^{2, 4}

Bipolar Zero Drift⁵

0.5

0.25

2

0.003

0.006

1

µV/°C typ

ppm/°C typ

% FSR max

µV/°C typ

For Gains of 1, 2

For Gains of 4, 8, 16, 32, 64, 128

Gain Drift

Bipolar Negative Full-Scale Error²@ 25°C

T_MINto T_MAX

Excluding Reference

Typically 0.0006%

Excluding Reference. For Gains of 1, 2

Excluding Reference. For Gains of 4, 8, 16, 32, 64, 128

Bipolar Negative Full-Scale Drift⁵

0.3

ANALOG INPUTS/REFERENCE INPUTS

Normal-Mode 50 Hz Rejection⁶

Normal-Mode 60 Hz Rejection⁶

AIN1/REF IN

100

dB min

For Filter Notches of 10 Hz, 25 Hz, 50 Hz, 0.02 ϫ f_NOTCH

For Filter Notches of 10 Hz, 30 Hz, 60 Hz, 0.02 ϫ f_NOTCH

DC Input Leakage Current @ 25°C⁶

T_MINto T_MAX

10

1

20

pA max

nA max

pF max

Sampling Capacitance⁶

Common-Mode Rejection (CMR)

100

dB min

At dc and AV_DD= 5 V

90

150

V_SSto AV_DD

dB min

V min to V max

At dc and AV_DD= 10 V

For Filter Notches of 10 Hz, 25 Hz, 50 Hz, 0.02 ϫ f_NOTCH

For Filter Notches of 10 Hz, 30 Hz, 60 Hz, 0.02 ϫ f_NOTCH

Common-Mode 50 Hz Rejection⁶

Common-Mode 60 Hz Rejection⁶

Common-Mode Voltage Range⁷

Analog Inputs⁸

Input Sampling Rate, f_S

See Table III

AIN1 Input Voltage Range⁹

For Normal Operation. Depends on Gain Selected

Unipolar Input Range (B/U Bit of Control Register = 1)

Bipolar Input Range (B/U Bit of Control Register = 0)

For Normal Operation. Depends on Gain Selected

Unipolar Input Range (B/U Bit of Control Register = 1)

Bipolar Input Range (B/U Bit of Control Register = 0)

10

0 V to V_REF

V max

V_REF

AIN2 Input Voltage Range⁹

10

0 V to 4 ϫ V_REF

4 ϫ V_REF

V max

AIN2 DC Input Impedance

AIN2 Gain Error¹¹

30

0.05

1

10

20

kΩ

% typ

Additional Error Contributed by Resistor Attenuator

Additional Drift Contributed by Resistor Attenuator

Additional Error Contributed by Resistor Attenuator

AIN2 Gain Drift

ppm/°C typ

mV max

µV/°C typ

AIN2 Offset Error¹¹

AIN2 Offset Drift

Reference Inputs

REF IN(+) – REF IN(–) Voltage¹²

2.5 to 5

V min to V max

For Specified Performance. Part Is Functional with

Lower V_REFVoltages

Input Sampling Rate, f_S

NOTES

f_{CLK IN}/256

¹Temperature range is as follows: A Version, –40°C to +85°C; S Version –55°C to +125°C. See also Note 18.

²Applies after calibration at the temperature of interest.

³Positive full-scale error applies to both unipolar and bipolar input ranges.

⁴These errors will be of the order of the output noise of the part as shown in Table I after system calibration. These errors will be 20 µV typical after self-calibration

or background calibration.

⁵Recalibration at any temperature or use of the background calibration mode will remove these drift errors.

⁶These numbers are guaranteed by design and/or characterization.

⁷This common-mode voltage range is allowed, provided that the input voltage on AIN1(+) and AIN1(–) does not exceed AV _DD+ 30 mV and V_SS– 30 mV.

⁸The AIN1 analog input presents a very high impedance dynamic load that varies with clock frequency and input sample rate. The maximum recommended

source resistance depends on the selected gain (see Tables IV and V).

⁹The analog input voltage range on the AIN1(+) input is given here with respect to the voltage on the AIN1(–) input. The input voltage range on the AIN2

input is with respect to AGND. The absolute voltage on the AIN1 input should not go more positive than AV_DD+ 30 mV or more negative than V_SS– 30 mV.

10

V

= REF IN(+) – REF IN(–).

REF

¹¹This error can be removed using the system calibration capabilities of the AD7712. This error is not removed by the AD7712’s self-calibration features. The offset

drift on the AIN2 input is 4 times the value given in the Static Performance section.

¹²The reference input voltage range may be restricted by the input voltage range requirement on the V_BIASinput.

–2–

REV. F

AD7712

SPECIFICATIONS (continued)

Parameter

A, S Versions¹

Unit

Conditions/Comments

REFERENCE OUTPUT

Output Voltage

Initial Tolerance

Drift

Output Noise

Line Regulation (AV_DD

Load Regulation

2.5

1

20

30

1

V nom

% max

ppm/°C typ

µV typ

mV/V max

mV/mA max

mA max

pk-pk Noise; 0.1 Hz to 10 Hz Bandwidth

Maximum Load Current 1 mA

)

1.5

1

External Current

V_BIASINPUT¹³

Input Voltage Range

AV_DD– 0.85 ϫ V_REF

See V_BIASInput Section

or AV_DD– 3.5

V max

V min

Whichever Is Smaller: +5 V/–5 V or +10 V/0 V

Nominal AV_DD/V_SS

Whichever Is Smaller: +5 V/0 V Nominal AV_DD/V_SS

See V_BIASInput Section

Whichever Is Greater: +5 V/–5 V or +10 V/0 V

Nominal AV_DD/V_SS

or AV_DD– 2.1

V_SS+ 0.85 ϫ V_REF

or V_SS+ 3

or V_SS+ 2.1

65 to 85

V min

dB typ

Whichever Is Greater: +5 V/0 V Nominal AV_DD/V_SS

Increasing with Gain

V_BIASRejection

LOGIC INPUTS

Input Current

10

µA max

All Inputs except MCLK IN

_INL, Input Low Voltage

V

0.8

2.0

V max

V min

V_INH, Input High Voltage

MCLK IN Only

V

_INL, Input Low Voltage

0.8

3.5

V max

V min

V_INH, Input High Voltage

LOGIC OUTPUTS

V_OL, Output Low Voltage

_OH, Output High Voltage

0.4

4.0

10

9

V max

V min

µA max

pF typ

I_SINK= 1.6 mA

I_SOURCE= 100 µA

V

Floating State Leakage Current

Floating State Output Capacitance¹⁴

TRANSDUCER BURNOUT

Current

Initial Tolerance

Drift

4.5

10

0.1

µA nom

% typ

%/°C typ

SYSTEM CALIBRATION

AIN1

Positive Full-Scale Calibration Limit¹⁵

Negative Full-Scale Calibration Limit¹⁵

Offset Calibration Limit^{16, 17}

Input Span¹⁵

(1.05 ϫ V_REF)/GAIN

–(1.05 ϫ V_REF)/GAIN

0.8 ϫ V_REF/GAIN

V max

V min

V max

GAIN Is the Selected PGA Gain (Between 1 and 128)

(2.1 ϫ V_REF)/GAIN

AIN2

Positive Full-Scale Calibration Limit¹⁵

Negative Full-Scale Calibration Limit¹⁵

Offset Calibration Limit¹⁷

Input Span¹⁵

(4.2 ϫ V_REF)/GAIN

–(4.2 ϫ V_REF)/GAIN

3.2 ϫ V_REF/GAIN

V max

V min

V max

GAIN Is the Selected PGA Gain (Between 1 and 128)

(8.4 ϫ V_REF)/GAIN

NOTES

¹³The AD7712 is tested with the following V_BIASvoltages. With AV_DD= 5 V and V_SS= 0 V, V_BIAS= 2.5 V; with AV_DD= 10 V and V_SS= 0 V, V_BIAS= 5 V and

with AV_DD= 5 V and V_SS= –5 V, V_BIAS= 0 V.

¹⁴Guaranteed by design, not production tested.

¹⁵After calibration, if the analog input exceeds positive full scale, the converter will output all 1s. If the analog input is less than negative full scale, then the device will

output all 0s.

¹⁶These calibration and span limits apply provided the absolute voltage on the AIN1 analog inputs does not exceed AV _DD+ 30 mV or does not go more negative

than V_SS– 30 mV.

¹⁷The offset calibration limit applies to both the unipolar zero point and the bipolar zero point.

REV. F

–3–

AD7712–SPECIFICATIONS

Parameter

A, S Versions¹

Unit

Conditions/Comments

POWER REQUIREMENTS

Power Supply Voltages

AV_DDVoltage¹⁸

+5 to +10

+5

V nom

V max

5% for Specified Performance

For Specified Performance

DV_DDVoltage¹⁹

AV_DD– V_SSVoltage

Power Supply Currents

AV_DDCurrent

+10.5

4

4.5

1.5

mA max

DV_DDCurrent

V

_SSCurrent

V_SS= –5 V

Power Supply Rejection²⁰

Rejection w.r.t. AGND; Assumes V_BIASIs Fixed

21

Positive Supply (AV_DDand DV_DD

)

dB typ

Negative Supply (V_SS

Power Dissipation

Normal Mode

)

90

45

52.5

200

mW max AV_DD= DV_DD= +5 V, V_SS= 0 V; Typically 25 mW

mW max AV_DD= DV_DD= +5 V, V_SS= –5 V; Typically 30 mW

µW max

Normal Mode

Standby (Power-Down) Mode²²

AV_DD= DV_DD= +5 V, V_SS= 0 V or –5 V; Typically 100 µW

NOTES

¹⁸The AD7712 is specified with a 10 MHz clock for AV_DDvoltages of +5 V 5%. It is specified with an 8 MHz clock for AV_DDvoltages greater than 5.25 V and less

than 10.5 V. Operating with AV_DDvoltages in the range 5.25 V to 10.5 V is guaranteed only over the 0ЊC to 70ЊC temperature range.

¹⁹The 5% tolerance on the DV_DDinput is allowed provided that DV_DDdoes not exceed AV_DDby more than 0.3 V.

²⁰Measured at dc and applies in the selected passband. PSRR at 50 Hz will exceed 120 dB with filter notches of 10 Hz, 25 Hz, or 50 Hz. PSRR at 60 Hz will

exceed 120 dB with filter notches of 10 Hz, 30 Hz, or 60 Hz.

²¹PSRR depends on gain: gain of 1 = 70 dB typ; gain of 2 = 75 dB typ; gain of 4 = 80 dB typ; gains of 8 to 128 = 85 dB typ. These numbers can be improved

(to 95 dB typ) by deriving the V_BIASvoltage (via Zener diode or reference) from the AV_DDsupply.

²²Using the hardware STANDBY pin. Standby power dissipation using the software standby bit (PD) of the Control Register is 8 mW typ.

Specifications subject to change without notice.

ABSOLUTE MAXIMUM RATINGS*

(T_A= 25°C, unless otherwise noted.)

Digital Input Voltage to DGND . . . . . –0.3 V to AV_DD+ 0.3 V

Digital Output Voltage to DGND . . . . –0.3 V to DV_DD+ 0.3 V

Operating Temperature Range

Commercial (A Version) . . . . . . . . . . . . . . . –40°C to +85°C

Extended (S Version) . . . . . . . . . . . . . . . . . –55°C to +125°C

Storage Temperature Range . . . . . . . . . . . . . –65°C to +150°C

Lead Temperature (Soldering, 10 secs) . . . . . . . . . . . . . 300°C

Power Dissipation (Any Package) to 75°C . . . . . . . . . . 450 mW

*Stresses above those listed under Absolute Maximum Ratings may cause perma-

nent damage to the device. This is a stress rating only; functional operation of the

device at these or any other conditions above those listed in the operational

sections of the specification is not implied. Exposure to absolute maximum rating

conditions for extended periods may affect device reliability.

AV_DDto DV_DD. . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +12 V

AV_DDto V_SS. . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +12 V

AV_DDto AGND . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +12 V

AV_DDto DGND . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +12 V

DV_DDto AGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +6 V

DV_DDto DGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +6 V

V_SSto AGND . . . . . . . . . . . . . . . . . . . . . . . . . +0.3 V to –6 V

V_SSto DGND . . . . . . . . . . . . . . . . . . . . . . . . . +0.3 V to –6 V

AIN1 Input Voltage to AGND . . .

V_SS– 0.3 V to AV_DD+ 0.3 V

Reference Input Voltage to AGND . . V_SS– 0.3 V to AV_DD+ 0.3 V

REF OUT to AGND . . . . . . . . . . . . . . . . . . . . –0.3 V to AV_DD

ORDERING GUIDE

Temperature Range Package Options*

Model

AD7712AN

AD7712AR

–40°C to +85°C

N-24

RW-24

Q-24

AD7712AR-REEL –40°C to +85°C

AD7712AR-REEL7 –40°C to +85°C

AD7712AQ

AD7712SQ

–40°C to +85°C

–55°C to +125°C

Q-24

EVAL-AD7712EB Evaluation Board

*N = PDIP, Q = CERDIP; RW = SOIC.

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily

accumulate on the human body and test equipment and can discharge without detection. Although the

AD7712 features proprietary ESD protection circuitry, permanent damage may occur on devices

subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended

to avoid performance degradation or loss of functionality.

–4–

REV. F

AD7712

(DV_DD= +5 V ؎ 5%; AV_DD= +5 V or +10 V³؎ 5%; V_SS= 0 V or –5 V ؎ 5%; AGND = DGND =

TIMING CHARACTERISTICS^{1, 2}

0 V; f_CLKIN=10 MHz; Input Logic 0 = 0 V, Logic 1 = DV_DD, unless otherwise noted.)

Limit at T_MIN, T_MAX

Parameter

(A, S Versions)

Unit

Conditions/Comments

4, 5

f_{CLK IN}

Master Clock Frequency: Crystal Oscillator or

Externally Supplied

400

10

8

kHz min

MHz max

MHz

AV_DD= 5 V 5%

For Specified Performance

AV_DD= 5.25 V to 10.5 V

t_{CLK IN LO}

0.4 ϫ t_{CLK IN}

50

ns min

ns max

ns min

Master Clock Input Low Time; t_{CLK IN}= 1/f_{CLK IN}

Master Clock Input High Time

Digital Output Rise Time; Typically 20 ns

Digital Output Fall Time; Typically 20 ns

SYNC Pulse Width

t_{CLK IN HI}

t_r⁶

t_f⁶

t₁

1000

Self-Clocking Mode

t₂

t₃

t₄

0

ns min

ns max

ns min

ns max

ns nom

ns min

ns max

ns min

DRDY to RFS Setup Time; t_{CLK IN}= 1/f_{CLK IN}

DRDY to RFS Hold Time

A0 to RFS Setup Time

2 ϫ t_{CLK IN}

0

4 ϫ t_{CLK IN}+ 20

t_{CLK IN}/2

t₅

A0 to RFS Hold Time

t₆₇

t₇₇

t₈

RFS Low to SCLK Falling Edge

Data Access Time (RFS Low to Data Valid)

SCLK Falling Edge to Data Valid Delay

t

_{CLK IN}/2 + 30

t₉

t_{CLK IN}/2

3 ϫ t_{CLK IN}/2

50

0

4 ϫ t_{CLK IN}+ 20

4 ϫ t_{CLK IN}

0

10

SCLK High Pulse Width

SCLK Low Pulse Width

A0 to TFS Setup Time

A0 to TFS Hold Time

TFS to SCLK Falling Edge Delay Time

TFS to SCLK Falling Edge Hold Time

Data Valid to SCLK Setup Time

Data Valid to SCLK Hold Time

t₁₀

t₁₄

t₁₅

t₁₆

t₁₇

t₁₈

t₁₉

NOTES

¹Guaranteed by design, not production tested. Sample tested during initial release and after any redesign or process change that may affect this parameter. All input

signals are specified with tr = tf = 5 ns (10% to 90% of 5 V) and timed from a voltage level of 1.6 V.

²See Figures 11 to 14.

³The AD7712 is specified with a 10 MHz clock for AV_DDvoltages of 5 V 5%. It is specified with an 8 MHz clock for AV_DDvoltages greater than 5.25 V and less

than 10.5 V.

⁴CLK IN duty cycle range is 45% to 55%. CLK IN must be supplied whenever the AD7712 is not in STANDBY mode. If no clock is present in this case, the

device can draw higher current than specified and possibly become uncalibrated.

⁵The AD7712 is production tested with f_{CLK IN}at 10 MHz (8 MHz for AV_DD< 5.25 V). It is guaranteed by characterization to operate at 400 kHz.

⁶Specified using 10% and 90% points on waveform of interest.

⁷These numbers are measured with the load circuit of Figure 1 and defined as the time required for the output to cross 0.8 V or 2.4 V.

REV. F

–5–

AD7712

TIMING CHARACTERISTICS (continued)

Limit at T_MIN, T_MAX

(A, S Versions)

Parameter

Unit

Conditions/Comments

External Clocking Mode

f_SCLK

t₂₀

t₂₁

t₂₂

f_{CLK IN}/5

0

2 ϫ t_{CLK IN}

0

4 ϫ t_{CLK IN}

10

2 ϫ t_{CLK IN}+ 20

2 ϫ t_{CLK IN}

t_{CLK IN}+ 10

MHz max

ns min

ns max

ns min

ns max

ns min

ns max

ns min

ns max

ns min

ns max

ns min

Serial Clock Input Frequency

DRDY to RFS Setup Time

DRDY to RFS Hold Time

A0 to RFS Setup Time

A0 to RFS Hold Time

t₂₃

t₂₄

t₂₅

7

Data Access Time (RFS Low to Data Valid)

SCLK Falling Edge to Data Valid Delay

7

t₂₆

t₂₇

t₂₈

SCLK High Pulse Width

SCLK Low Pulse Width

SCLK Falling Edge to DRDY High

SCLK to Data Valid Hold Time

8

t₂₉

10

t_{CLK IN}+ 10

10

5 ϫ t_{CLK IN}/2 + 50

0

t₃₀

t₃₁

RFS/TFS to SCLK Falling Edge Hold Time

RFS to Data Valid Hold Time

A0 to TFS Setup Time

8

t₃₂

t₃₃

t₃₄

t₃₅

t₃₆

0

A0 to TFS Hold Time

4 ϫ t_{CLK IN}

2 ϫ t_{CLK IN}– SCLK High

30

SCLK Falling Edge to TFS Hold Time

Data Valid to SCLK Setup Time

Data Valid to SCLK Hold Time

NOTES

⁸These numbers are derived from the measured time taken by the data output to change 0.5 V when loaded with the circuit of Figure 1. The measured number

is then extrapolated back to remove effects of charging or discharging the 100 pF capacitor. This means that the times quoted in the timing characteristics are

the true bus relinquish times of the part and, as such, are independent of external bus loading capacitances.

Specifications subject to change without notice.

PIN CONFIGURATION

DIP and SOIC

SCLK

MCLK IN

MCLK OUT

A0

DGND

24

1

2

3

4

5

6

7

8

9

1.6mA

23 DV

DD

22 SDATA

21

20

19

DRDY

TO OUTPUT

2.1V

SYNC

RFS

TFS

AD7712

100pF

MODE

TOP VIEW

(Not to Scale)

AIN1(+)

AIN1(–)

18 AGND

17 AIN2

200␮A

REF OUT

16

15

STANDBY

TP 10

REF IN(+)

11

12

V

14 REF IN(–)

Figure 1. Load Circuit for Access Time and

Bus Relinquish Time

SS

AV

V

BIAS

13

DD

–6–

REV. F

AD7712

PIN FUNCTION DESCRIPTION

Pin Mnemonic

Function

1

SCLK

Serial Clock. Logic input/output, depending on the status of the MODE pin. When MODE is high, the

device is in its self-clocking mode, and the SCLK pin provides a serial clock output. This SCLK becomes

active when RFS or TFS goes low, and it goes high impedance when either RFS or TFS returns high or when

the device has completed transmission of an output word. When MODE is low, the device is in its external

clocking mode, and the SCLK pin acts as an input. This input serial clock can be a continuous clock with all

data transmitted in a continuous train of pulses. Alternatively, it can be a noncontinuous clock with the

information being transmitted to the AD7712 in smaller batches of data.

2

MCLK IN

Master Clock Signal for the Device. This can be provided in the form of a crystal or external clock. A crystal can

be tied across the MCLK IN and MCLK OUT pins. Alternatively, the MCLK IN pin can be driven with a

CMOS-compatible clock and MCLK OUT left unconnected. The clock input frequency is nominally 10 MHz.

3

4

MCLK OUT When the master clock for the device is a crystal, the crystal is connected between MCLK IN and MCLK OUT.

A0

Address Input. With this input low, reading and writing to the device is to the control register. With this input

high, access is to either the data register or the calibration registers.

5

6

7

SYNC

MODE

AIN1(+)

Logic Input. Allows for synchronization of the digital filters when using a number of AD7712s. It resets

the nodes of the digital filter.

Logic Input. When this pin is high, the device is in its self-clocking mode. With this pin low, the device is in its

external clocking mode.

Analog Input Channel 1. Positive input of the programmable gain differential analog input. The AIN1(+) input

is connected to an output current source that can be used to check that an external transducer has burned out

or gone open circuit. This output current source can be turned on/off via the control register.

8

9

AIN1(–)

Analog Input Channel 1. Negative input of the programmable gain differential analog input.

STANDBY

Logic Input. Taking this pin low shuts down the internal analog and digital circuitry, reducing power

consumption to less than 50 µW.

10 TP

11 V_SS

Test Pin. Used when testing the device. Do not connect anything to this pin.

Analog Negative Supply, 0 V to –5 V. Tied to AGND for single-supply operation. The input voltage on AIN1

should not go > 30 mV negative w.r.t. V_SSfor correct operation of the device.

12 AV_DD

13 V_BIAS

Analog Positive Supply Voltage, 5 V to 10 V.

Input Bias Voltage. This input voltage should be set such that V_BIAS+ 0.85 ϫ V_REF< AV_DDand V_BIAS– 0.85

ϫ V_REF> V_SSwhere V_REFis REF IN(+) – REF IN(–). Ideally, this should be tied halfway between AV_DD

and V_SS. Thus, with AV_DD= +5 V and V_SS= 0 V, it can be tied to REF OUT; with AV_DD= +5 V and V_SS

–5 V, it can be tied to AGND, while with AV_DD= +10 V, it can be tied to +5 V.

=

14 REF IN(–)

15 REF IN(+)

16 REF OUT

17 AIN2

Reference Input. The REF IN(–) can lie anywhere between AV_DDand V_SSprovided REF IN(+) is greater

than REF IN(–).

Reference Input. The reference input is differential providing that REF IN(+) is greater than REF IN(–).

REF IN(+) can lie anywhere between AV_DDand V_SS.

Reference Output. The internal 2.5 V reference is provided at this pin. This is a single-ended output

that is referred to AGND.

Analog Input Channel 2. High level analog input that accepts an analog input voltage range of 4 ϫ

V

_REF/GAIN. At the nominal V_REFof +2.5 V and a gain of 1, the AIN2 input voltage range is 10 V.

18 AGND

Ground Reference Point for Analog Circuitry.

19 TFS

Transmit Frame Synchronization. Active low logic input used to write serial data to the device with serial

data expected after the falling edge of this pulse. In the self-clocking mode, the serial clock becomes active

after TFS goes low. In the external clocking mode, TFS must go low before the first bit of the data-word

is written to the part.

20 RFS

Receive Frame Synchronization. Active low logic input used to access serial data from the device. In the

self-clocking mode, both the SCLK and SDATA lines become active after RFS goes low. In the external

clocking mode, the SDATA line becomes active after RFS goes low.

REV. F

–7–

AD7712

Pin Mnemonic

Function

21

DRDY

Logic Output. A falling edge indicates that a new output word is available for transmission. The DRDY pin

will return high upon completion of transmission of a full output word. DRDY is also used to indicate

when the AD7712 has completed its on-chip calibration sequence.

22 SDATA

Serial Data. Input/output with serial data being written to either the control register or the calibration

registers and serial data being accessed from the control register, calibration registers, or the data register.

During an output data read operation, serial data becomes active after RFS goes low (provided DRDY is

low). During a write operation, valid serial data is expected on the rising edges of SCLK when TFS is low.

The output data coding is natural binary for unipolar inputs and offset binary for bipolar inputs.

23 DV_DD

Digital Supply Voltage, 5 V. DV_DDshould not exceed AV_DDby more than 0.3 V in normal operation.

Ground Reference Point for Digital Circuitry.

24 DGND

Positive Full-Scale Overrange

TERMINOLOGY

Positive full-scale overrange is the amount of overhead available

Integral Nonlinearity

to handle input voltages on AIN1(+) input greater than

(AIN1(–) + V_REF/GAIN) or on the AIN2 of greater than +4 ϫ

V_REF/GAIN (for example, noise peaks or excess voltages due to

system gain errors in system calibration routines) without intro-

ducing errors due to overloading the analog modulator or to

overflowing the digital filter.

This is the maximum deviation of any code from a straight line

passing through the endpoints of the transfer function. The end-

points of the transfer function are zero-scale (not to be confused

with bipolar zero), a point 0.5 LSB below the first code transi-

tion (000 . . . 000 to 000 . . . 001) and full scale, a point 0.5 LSB

above the last code transition (111 . . . 110 to 111 . . . 111). The

error is expressed as a percentage of full scale.

Negative Full-Scale Overrange

This is the amount of overhead available to handle voltages on

AIN1(+) below (AIN1(–) – V_REF/GAIN) or on AIN2 below

–4 ϫ V_REF/GAIN without overloading the analog modulator or

overflowing the digital filter. Note that the analog input will

accept negative voltage peaks on AIN1(+) even in the unipolar

mode provided that AIN1(+) is greater than AIN1(–) and greater

than V_SS– 30 mV.

Positive Full-Scale Error

Positive full-scale error is the deviation of the last code transi-

tion (111 . . . 110 to 111 . . . 111) from the ideal input full-scale

voltage. For AIN1(+), the ideal full-scale input voltage is

(AIN1(–) + V_REF/GAIN – 3/2 LSBs); for AIN2, the ideal full-

scale voltage is +4 ϫ V_REF/GAIN – 3/2 LSBs. Positive full-scale

error applies to both unipolar and bipolar analog input ranges.

Offset Calibration Range

Unipolar Offset Error

In the system calibration modes, the AD7712 calibrates its

offset with respect to the analog input. The offset calibration

range specification defines the range of voltages that the AD7712

can accept and still accurately calibrate offset.

Unipolar offset error is the deviation of the first code transition

from the ideal voltage. For AIN1(+), the ideal input voltage is

(AIN1(–) + 0.5 LSB); for AIN2, the ideal input is 0.5 LSB

when operating in the unipolar mode.

Full-Scale Calibration Range

Bipolar Zero Error

This is the range of voltages that the AD7712 can accept in the

system calibration mode and still correctly calibrate full scale.

This is the deviation of the midscale transition (0111 . . . 111

to 1000 . . . 000) from the ideal input voltage. For AIN1(+), the

ideal input voltage is (AIN1(–) – 0.5 LSB); for AIN2, the ideal

input is –0.5 LSB when operating in the bipolar mode.

Input Span

In system calibration schemes, two voltages applied in sequence

to the AD7712’s analog input define the analog input range.

The input span specification defines the minimum and maxi-

mum input voltages from zero to full scale that the AD7712 can

accept and still accurately calibrate gain.

Bipolar Negative Full-Scale Error

This is the deviation of the first code transition from the ideal

input voltage. For AIN1(+), the ideal input voltage is (AIN1(–)

– V_REF/GAIN + 0.5 LSB); for AIN2, the ideal input voltage is

(–4 ϫ V_REF/GAIN + 0.5 LSB) when operating in the bipolar

mode.

–8–

REV. F

AD7712

Control Register (24 Bits)

A write to the device with the A0 input low writes data to the control register. A read to the device with the A0 input low accesses the

contents of the control register. The control register is 24 bits wide and when writing to the register 24 bits of data must be written

otherwise the data will not be loaded to the control register. In other words, it is not possible to write just the first 12 bits of data into

the control register. If more than 24 clock pulses are provided before TFS returns high, then all clock pulses after the 24th clock

pulse are ignored. Similarly, a read operation from the control register should access 24 bits of data.

MSB

MD2 MD1 MD0 G2

G1

G0

CH

PD

WL

FS3

X

BO

B/U

FS0

LSB

FS11 FS10 FS9

FS8

FS7

FS6

FS5

FS4

FS2

FS1

X = Don’t Care.

Operating Mode

MD2

MD1

MD0 Operating Mode

0

Normal Mode. This is the normal mode of operation of the device whereby a read to the device accesses

data from the data register. This is the default condition of these bits after the internal power-on reset.

0

1

Activate Self-Calibration. This activates self-calibration on the channel selected by CH. This is a one-step

calibration sequence, and when complete, the part returns to normal mode (with MD2, MD1, MD0 of

the control registers returning to 0, 0, 0). The DRDY output indicates when this self-calibration is complete.

For this calibration type, the zero-scale calibration is done internally on shorted (zeroed) inputs, and the

full-scale calibration is done on V_REF

.

0

Activate System Calibration. This activates system calibration on the channel selected by CH. This is a

two-step calibration sequence, with the zero-scale calibration done first on the selected input channel and

DRDY indicating when this zero-scale calibration is complete. The part returns to normal mode at the

end of this first step in the two-step sequence.

0

1

0

1

0

Activate System Calibration. This is the second step of the system calibration sequence with full-scale

calibration being performed on the selected input channel. Once again, DRDY indicates when the full-

scale calibration is complete. When this calibration is complete, the part returns to normal mode.

Activate System Offset Calibration. This activates system offset calibration on the channel selected by

CH. This is a one-step calibration sequence and, when complete, the part returns to normal mode with

DRDY indicating when this system offset calibration is complete. For this calibration type, the zero-scale

calibration is done on the selected input channel, and the full-scale calibration is done internally on V_REF

.

1

0

1

Activate Background Calibration. This activates background calibration on the channel selected by CH. If

the background calibration mode is on, then the AD7712 provides continuous self-calibration of the

reference and shorted (zeroed) inputs. This calibration takes place as part of the conversion sequence,

extending the conversion time and reducing the word rate by a factor of 6. Its major advantage is that

the user does not have to worry about recalibrating the device when there is a change in the ambient

temperature. In this mode, the shorted (zeroed) inputs and V_REF, as well as the analog input voltage, are

continuously monitored, and the calibration registers of the device are automatically updated.

1

0

1

Read/Write Zero-Scale Calibration Coefficients. A read to the device with A0 high accesses the contents

of the zero-scale calibration coefficients of the channel selected by CH. A write to the device with A0 high

writes data to the zero-scale calibration coefficients of the channel selected by CH. The word length for

reading and writing these coefficients is 24 bits, regardless of the status of the WL bit of the control

register. Therefore, when writing to the calibration register, 24 bits of data must be written; otherwise the

new data will not be transferred to the calibration register.

Read/Write Full-Scale Calibration Coefficients. A read to the device with A0 high accesses the contents of

the full-scale calibration coefficients of the channel selected by CH. A write to the device with A0 high

writes data to the full-scale calibration coefficients of the channel selected by CH. The word length for

reading and writing these coefficients is 24 bits, regardless of the status of the WL bit of the control

register. Therefore, when writing to the calibration register, 24 bits of data must be written; otherwise the

new data will not be transferred to the calibration register.

REV. F

–9–

AD7712

PGA Gain

G2

0

Gl

0

G0

0

Gain

1

(Default Condition after the Internal Power-On Reset)

0

1

0

2

4

0

1

8

1

0

1

0

1

0

1

16

32

64

128

Channel Selection

CH Channel

0

1

AIN1

AIN2

Low Level Input

High Level Input

(Default Condition after the Internal Power-On Reset)

(Default Condition after Internal Power-On Reset)

Power-Down

PD

0

1

Normal Operation

Power-Down

Word Length

WL Output Word Length

0

1

16-Bit

24-Bit

Burnout Current

BO

0

1

Off

On

(Default Condition after Internal Power-On Reset)

Bipolar/Unipolar Selection (Both Inputs)

B/U

0

1

Bipolar

Unipolar

(Default Condition after Internal Power-On Reset)

Filter Selection (FS11–FS0)

first notch of the filter. For example, if the first notch of the filter

is selected at 50 Hz, then a new word is available at a 50 Hz rate

or every 20 ms. If the first notch is at 1 kHz, a new word is avail-

able every 1 ms.

The on-chip digital filter provides a sinc³(or (sinx/x)³) filter

response. The 12 bits of data programmed into these bits deter-

mine the filter cutoff frequency, the position of the first notch of

the filter, and the data rate for the part. In association with the

gain selection, it also determines the output noise (and therefore

the effective resolution) of the device.

The settling time of the filter to a full-scale step input change is

worst case 4 ϫ 1/(output data rate). This settling time is to

100% of the final value. For example, with the first filter notch

at 50 Hz, the settling time of the filter to a full-scale step input

change is 80 ms max. If the first notch is at 1 kHz, the settling

time of the filter to a full-scale input step is 4 ms max. This

settling time can be reduced to 3 ϫ l/(output data rate) by syn-

chronizing the step input change to a reset of the digital filter. In

other words, if the step input takes place with SYNC low, the

settling time will be 3 ϫ l/(output data rate). If a change of

channels takes place, the settling time is 3 ϫ l/(output data rate)

regardless of the SYNC input.

The first notch of the filter occurs at a frequency determined by

the relationship filter first notch frequency = (f_{CLK IN}/512)/code

where code is the decimal equivalent of the code in bits FS0 to

FS11 and is in the range 19 to 2,000. With the nominal f_{CLK IN}of

10 MHz, this results in a first notch frequency range from 9.76 Hz

to 1.028 kHz. To ensure correct operation of the AD7712, the

value of the code loaded to these bits must be within this range.

Failure to do this will result in unspecified operation of the device.

Changing the filter notch frequency, as well as the selected gain,

impacts resolution. Tables I and II and Figure 2 show the effect

of the filter notch frequency and gain on the effective resolution

of the AD7712. The output data rate (or effective conversion

time) for the device is equal to the frequency selected for the

The –3 dB frequency is determined by the programmed first

notch frequency according to the relationship filter –3 dB

frequency = 0.262 ϫ first notch frequency.

–10–

REV. F

AD7712

Tables I and II show the output rms noise for some typical

notch and –3 dB frequencies. The numbers given are for the

bipolar input ranges with a V_REFof 2.5 V. These numbers are

typical and are generated with an analog input voltage of 0 V.

The output noise from the part comes from two sources. First,

there is the electrical noise in the semiconductor devices used in

the implementation of the modulator (device noise). Second,

when the analog input signal is converted into the digital do-

main, quantization noise is added. The device noise is at a low

level and is largely independent of frequency. The quantization

noise starts at an even lower level but rises rapidly with increasing

frequency to become the dominant noise source. Consequently,

lower filter notch settings (below 60 Hz approximately) tend to

be device noise dominated while higher notch settings are domi-

nated by quantization noise. Changing the filter notch and cutoff

frequency in the quantization noise dominated region results in a

more dramatic improvement in noise performance than it does

in the device noise dominated region as shown in Table I.

Furthermore, quantization noise is added after the PGA, so

effective resolution is independent of gain for the higher filter

notch frequencies. Meanwhile, device noise is added in the PGA

and, therefore, effective resolution suffers a little at high gains

for lower notch frequencies.

At the lower filter notch settings (below 60 Hz), the no missing

codes performance of the device is at the 24-bit level. At the

higher settings, more codes will be missed until at the 1 kHz

notch setting; no missing codes performance is guaranteed only

to the 12-bit level. However, since the effective resolution of the

part is 10.5 bits for this filter notch setting, this no missing codes

performance should be more than adequate for all applications.

The effective resolution of the device is defined as the ratio of

the output rms noise to the input full scale. This does not

remain constant with increasing gain or with increasing band-

width. Table II is the same as Table I except that the output is

expressed in terms of effective resolution (the magnitude of the

rms noise with respect to 2

؋

V_REF/GAIN, i.e., the input full

scale). It is possible to do post filtering on the device to improve

the output data rate for a given –3 dB frequency and to further

reduce the output noise (see the Digital Filtering section).

Table I. Output Noise vs. Gain and First Notch Frequency

Typical Output RMS Noise (␮V)

First Notch of

Filter and O/P –3 dB

Gain of

2

Gain of

4

Gain of

8

Gain of

16

Gain of

32

Gain of

64

Gain of

128

Data Rate¹

Frequency 1

10 Hz²

25 Hz²

30 Hz²

50 Hz²

60 Hz²

100 Hz³

250 Hz³

500 Hz³

1 kHz³

2.62 Hz

6.55 Hz

7.86 Hz

13.1 Hz

15.72 Hz

26.2 Hz

65.5 Hz

131 Hz

262 Hz

1.0

1.8

2.5

4.33

5.28

13

0.78

1.1

1.31

2.06

2.36

6.4

0.48

0.63

0.84

1.2

1.33

3.7

0.33

0.5

0.57

0.64

0.87

1.8

0.25

0.44

0.46

0.54

0.63

1.1

7.5

35

180

0.25

0.41

0.43

0.46

0.62

0.9

4

25

120

0.25

0.38

0.4

0.46

0.6

0.65

2.7

15

0.25

0.38

0.4

0.46

0.56

0.65

1.7

130

75

25

12

0.6 ϫ 10³

3.1 ϫ 10³

0.26 ϫ 10³

1.6 ϫ 10³

140

70

8

40

0.7 ϫ 10³

0.29 ϫ 10³

70

NOTES

¹The default condition (after the internal power-on reset) for the first notch of filter is 60 Hz.

²For these filter notch frequencies, the output rms noise is primarily dominated by device noise, and, as a result, is independent of the value of the reference voltage.

Therefore, increasing the reference voltage will give an increase in the effective resolution of the device (i.e., the ratio of the rms noise to the input full scale is

increased since the output rms noise remains constant as the input full scale increases).

³For these filter notch frequencies, the output rms noise is dominated by quantization noise, and, as a result, is proportional to the value of the reference voltage.

Table II. Effective Resolution vs. Gain and First Notch Frequency

Effective Resolution* (Bits)

First Notch of

Filter and O/P –3 dB

Gain of

1

Gain of

2

Gain of

4

Gain of

8

Gain of

16

Gain of

32

Gain of

64

Gain of

128

Data Rate

Frequency

10 Hz

25 Hz

30 Hz

50 Hz

60 Hz

100 Hz

250 Hz

500 Hz

1 kHz

2.62 Hz

6.55 Hz

7.86 Hz

13.1 Hz

15.72 Hz

26.2 Hz

65.5 Hz

131 Hz

262 Hz

22.5

21.5

21

20

18.5

15

13

21.5

21

20

18.5

15.5

13

21.5

21

20.5

20

21

20

19.5

18.5

15.5

13

20.5

19.5

19

18

15.5

13

11

19.5

18.5

18

17.5

15.5

12.5

10.5

18.5

17.5

17

15

12.5

10

17.5

16.5

16

20

18.5

15.5

13

16

14.5

12.5

10

10.5

11

*Effective resolution is defined as the magnitude of the output rms noise with respect to the input full scale (i.e., 2 ϫ V_REF/GAIN). The above table applies for

a V_REFof 2.5 V and resolution numbers are rounded to the nearest 0.5 LSB.

REV. F

–11–

AD7712

Figures 2a and 2b give information similar to that outlined in Table I. In these plots, the output rms noise is shown for the full range

of available cutoffs frequencies rather than for some typical cutoff frequencies as in Tables I and II. The numbers given in these plots

are typical values at 25°C.

10000

1000

100

10

GAIN OF 1

GAIN OF 2

GAIN OF 4

GAIN OF 8

GAIN OF 16

GAIN OF 32

1000

GAIN OF 64

100

10

1

GAIN OF 128

1

0.1

10

0.1

10

100

1000

10000

100

1000

10000

NOTCH FREQUENCY – Hz

Figure 2b. Plot of Output Noise vs. Gain and

Notch Frequency (Gains of 16 to 128)

Figure 2a. Plot of Output Noise vs. Gain and

Notch Frequency (Gains of 1 to 8)

The basic connection diagram for the part is shown in Figure 3.

This shows the AD7712 in the external clocking mode with both

the AV_DDand DV_DDpins of the AD7712 being driven from the

analog 5 V supply. Some applications will have separate supplies

for both AV_DDand DV_DD, and in some of these cases, the analog

supply will exceed the 5 V digital supply (see the Power Supplies

and Grounding section).

CIRCUIT DESCRIPTION

The AD7712 is a sigma-delta A/D converter with on-chip digital

filtering, intended for the measurement of wide dynamic range,

low frequency signals such as those in industrial control or process

control applications. It contains a sigma-delta (or charge-

balancing) ADC, a calibration microcontroller with on-chip

static RAM, a clock oscillator, a digital filter, and a bidirectional

serial communications port.

ANALOG

5V SUPPLY

The part contains two analog input channels, one programmable

gain differential input, and one programmable gain high level

single-ended input. The gain range on both inputs is from 1 to

128. For the AIN1 input, this means that the input can accept

unipolar signals of between 0 mV and 20 mV and 0 mV and

+2.5 V or bipolar signals in the range from 20 mV to 2.5 V

when the reference input voltage equals 2.5 V. The input volt-

age range for the AIN2 input is 4 ϫ V_REF/GAIN and is 10 V

with the nominal reference of 2.5 V and a gain of 1. The input

signal to the selected analog input channel is continuously

sampled at a rate determined by the frequency of the master

clock, MCLK IN, and the selected gain (see Table III). A

chargebalancing A/D converter (sigma-delta modulator) converts

the sampled signal into a digital pulse train whose duty cycle

contains the digital information. The programmable gain

function on the analog input is also incorporated in this sigma-

delta modulator with the input sampling frequency being

modified to give the higher gains. A sinc³digital low-pass filter

processes the output of the sigma-delta modulator and updates

the output register at a rate determined by the first notch

frequency of this filter. The output data can be read from the

serial port randomly or periodically at any rate up to the output

register update rate. The first notch of this digital filter (and

therefore its –3 dB frequency) can be programmed via an on-chip

control register. The programmable range for this first notch

frequency is from 9.76 Hz to 1.028 kHz, giving a programmable

range for the –3 dB frequency of 2.58 Hz to 269 Hz.

0.1␮F

10␮F

0.1␮F

AV

DV

DD

DATA

DRDY

TFS

READY

AIN1(+)

AIN1(–)

DIFFERENTIAL

TRANSMIT

(WRITE)

ANALOG INPUT

RECEIVE

(READ)

RFS

SINGLE-ENDED

ANALOG INPUT

AIN2

AD7712

SERIAL

DATA

SDATA

SCLK

SERIAL

CLOCK

DV

DD

STANDBY

ADDRESS

INPUT

ANALOG

GROUND

A0

AGND

V

SS

MODE

DIGITAL

GROUND

DGND

5V

SYNC

REF OUT

REF IN(+)

MCLK OUT

V

BIAS

MCLK IN

REF IN(–)

Figure 3. Basic Connection Diagram

–12–

REV. F

AD7712

The AD7712 provides a number of calibration options that can

be programmed via the on-chip control register. A calibration

cycle can be initiated at any time by writing to this control regis-

ter. The part can perform self-calibration using the on-chip

calibration microcontroller and SRAM to store calibration

parameters. Other system components can also be included in

the calibration loop to remove offset and gain errors in the input

channel using the system calibration mode. Another option is a

background calibration mode where the part continuously

performs self-calibration and updates the calibration coeffi-

cients. Once the part is in this mode, the user does not have to

worry about issuing periodic calibration commands to the

device or asking the device to recalibrate when there is a change

in the ambient temperature or power supply voltage.

Oversampling is fundamental to the operation of sigma-delta

ADCs. Using the quantization noise formula for an ADC:

SNR = (6.02 3 number of bits + 1.76) dB

A 1-bit ADC or comparator yields an SNR of 7.78 dB.

The AD7712 samples the input signal at a frequency of 39 kHz

or greater (see Table III). As a result, the quantization noise is

spread over a much wider frequency than that of the band of

interest. The noise in the band of interest is reduced still further

by analog filtering in the modulator loop, which shapes the

quantization noise spectrum to move most of the noise energy

to frequencies outside the bandwidth of interest. The noise

performance is thus improved from this 1-bit level to the perfor-

mance outlined in Tables I and II and in Figure 2.

The AD7712 gives the user access to the on-chip calibration

registers, allowing the microprocessor to read the device’s

calibration coefficients and also to write its own calibration

coefficients to the part from prestored values in E²PROM. This

gives the microprocessor much greater control over the AD7712’s

calibration procedure. It also means that the user can verify that

the device has performed its calibration correctly by comparing

the coefficients after calibration with prestored values in

E²PROM.

The output of the comparator provides the digital input for the

1-bit DAC, so that the system functions as a negative feedback

loop that tries to minimize the difference signal. The digital data

that represents the analog input voltage is contained in the duty

cycle of the pulse train appearing at the output of the compara-

tor. It can be retrieved as a parallel binary data-word using a

digital filter.

Sigma-delta ADCs are generally described by the order of the

analog low-pass filter. A simple example of a first-order sigma-

delta ADC is shown in Figure 5. This contains only a first order

low-pass filter or integrator. It also illustrates the derivation of

the alternative name for these devices: charge-balancing ADCs.

The AD7712 can be operated in single-supply systems, provided

that the analog input voltage on the AIN1 input does not go

more negative than –30 mV. For larger bipolar signals on the

AIN1 input, a V_SSof –5 V is required by the part. For battery

operation or low power systems, the AD7712 offers a standby

mode (controlled by the STANDBY pin) that reduces idle

power consumption to typically 100 µW.

DIFFERENTIAL

AMPLIFIER

COMPARATOR

V

IN

THEORY OF OPERATION

The general block diagram of a sigma-delta ADC is shown in

Figure 4. It contains the following elements:

+FS

–FS

DAC

•

A sample-hold amplifier

A differential amplifier or subtracter

An analog low-pass filter

A 1-bit A/D converter (comparator)

A 1-bit DAC

Figure 5. Basic Charge-Balancing ADC

It consists of a differential amplifier (whose output is the differ-

ence between the analog input and the output of a 1-bit DAC),

an integrator, and a comparator. The term charge balancing,

comes from the fact that this system is a negative feedback loop

that tries to keep the net charge on the integrator capacitor at

zero by balancing charge injected by the input voltage with

charge injected by the 1-bit DAC. When the analog input is

zero, the only contribution to the integrator output comes from

the 1-bit DAC. For the net charge on the integrator capacitor to

be zero, the DAC output must spend half its time at +FS and

half its time at –FS. Assuming ideal components, the duty cycle

of the comparator will be 50%.

A digital low-pass filter

S/H AMP

COMPARATOR

ANALOG

LOW-PASS

FILTER

DIGITAL

FILTER

DAC

DIGITAL DATA

Figure 4. General Sigma-Delta ADC

When a positive analog input is applied, the output of the 1-bit

DAC must spend a larger proportion of the time at +FS, so the

duty cycle of the comparator increases. When a negative input

voltage is applied, the duty cycle decreases.

In operation, the analog signal sample is fed to the subtracter,

along with the output of the 1-bit DAC. The filtered difference

signal is fed to the comparator, whose output samples the differ-

ence signal at a frequency many times that of the analog signal

sampling frequency (oversampling).

The AD7712 uses a second-order sigma-delta modulator and a

digital filter that provides a rolling average of the sampled out-

put. After power-up, or if there is a step change in the input

voltage, there is a settling time that must elapse before valid

data is obtained.

REV. F

–13–

AD7712

0

–20

Input Sample Rate

The modulator sample frequency for the device remains at

f_{CLK IN}/512 (19.5 kHz @ f_{CLK IN}= 10 MHz) regardless of the

selected gain. However, gains greater than ϫ1 are achieved by

a combination of multiple input samples per modulator cycle

and scaling the ratio of reference capacitor to input capacitor.

As a result of the multiple sampling, the input sample rate of

the device varies with the selected gain (see Table III). The

effective input impedance is 1/C ϫ f_Swhere C is the input

sampling capacitance and f_Sis the input sample rate.

–40

–60

–80

–100

–120

–140

–160

–180

–200

–220

–240

Table III. Input Sampling Frequency vs. Gain

Gain

Input Sampling Frequency (f_S)

0

10

20

30

40

50

60

1

f_{CLK IN}/256 (39 kHz @ f_{CLK IN}= 10 MHz)

FREQUENCY – Hz

2

4

8

16

32

64

128

2 ϫ f_{CLK IN}/256 (78 kHz @ f_{CLK IN}= 10 MHz)

4 ϫ f_{CLK IN}/256 (156 kHz @ f_{CLK IN}= 10 MHz)

8 ϫ f_{CLK IN}/256 (312 kHz @ f_{CLK IN}= 10 MHz)

Figure 6. Frequency Response of AD7712 Filter

Since the AD7712 contains this on-chip, low-pass filtering,

there is a settling time associated with step function inputs, and

data on the output will be invalid after a step change until the

settling time has elapsed. The settling time depends upon the

notch frequency chosen for the filter. The output data rate

equates to this filter notch frequency, and the settling time of

the filter to a full-scale step input is four times the output data

period. In applications using both input channels, the settling

time of the filter must be allowed to elapse before data from the

second channel is accessed.

DIGITAL FILTERING

The AD7712’s digital filter behaves like a similar analog filter,

with a few minor differences.

First, since digital filtering occurs after the A-to-D conversion

process, it can remove noise injected during the conversion

process. Analog filtering cannot do this.

Post Filtering

The on-chip modulator provides samples at a 19.5 kHz output

rate. The on-chip digital filter decimates these samples to pro-

vide data at an output rate that corresponds to the programmed

first notch frequency of the filter. Since the output data rate

exceeds the Nyquist criterion, the output rate for a given band-

width will satisfy most application requirements. However, there

may be some applications that require a higher data rate for a

given bandwidth and noise performance. Applications that need

this higher data rate will require some post filtering following

the digital filter of the AD7712.

On the other hand, analog filtering can remove noise superim-

posed on the analog signal before it reaches the ADC. Digital

filtering cannot do this, and noise peaks riding on signals near

full scale have the potential to saturate the analog modulator

and digital filter, even though the average value of the signal

is within limits. To alleviate this problem, the AD7712 has

overrange headroom built into the sigma-delta modulator and

digital filter, which allows overrange excursions of 5% above

the analog input range. If noise signals are larger than this,

consideration should be given to analog input filtering, or to

reducing the input channel voltage so that its full scale is half

that of the analog input channel full scale. This will provide

an overrange capability greater than 100% at the expense of

reducing the dynamic range by 1 bit (50%).

For example, if the required bandwidth is 7.86 Hz but the

required update rate is 100 Hz, the data can be taken from the

AD7712 at the 100 Hz rate giving a –3 dB bandwidth of

26.2 Hz. Post filtering can be applied to this to reduce the

bandwidth and output noise, to the 7.86 Hz bandwidth level,

while maintaining an output rate of 100 Hz.

Filter Characteristics

Post filtering can also be used to reduce the output noise from

the device for bandwidths below 2.62 Hz. At a gain of 128, the

output rms noise is 250 nV. This is essentially device noise or

white noise, and since the input is chopped, the noise has a flat

frequency response. By reducing the bandwidth below 2.62 Hz,

the noise in the resultant passband can be reduced. A reduction

in bandwidth by a factor of 2 results in a √2 reduction in the

output rms noise. This additional filtering will result in a longer

settling time.

The cutoff frequency of the digital filter is determined by the

value loaded to bits FS0 to FS11 in the control register. At

the maximum clock frequency of 10 MHz, the minimum

cutoff frequency of the filter is 2.58 Hz while the maximum

programmable cutoff frequency is 269 Hz.

Figure 6 shows the filter frequency response for a cutoff

frequency of 2.62 Hz, which corresponds to a first filter notch

frequency of 10 Hz. This is a (sinx/x)³response (also called

sinc³), that provides >100 dB of 50 Hz and 60 Hz rejection.

Programming a different cutoff frequency via FS0–FS11 does

not alter the profile of the filter response; it changes the fre-

quency of the notches as outlined in the Control Register

section.

–14–

REV. F

AD7712

Antialias Considerations

Table IV. Typical External Series Resistance That Will Not

Introduce 16-Bit Gain Error

The digital filter does not provide any rejection at integer mul-

tiples of the modulator sample frequency (n ϫ 19.5 kHz, where

n = 1, 2, 3 . . . ). This means that there are frequency bands,

f_{3 dB}wide (f_{3 dB}is cutoff frequency selected by FS0 to FS11),

where noise passes unattenuated to the output. However, due to

the AD7712’s high oversampling ratio, these bands occupy only

a small fraction of the spectrum, and most broadband noise is

filtered. In any case, because of the high oversampling ratio, a

simple, RC, single-pole filter is generally sufficient to attenuate

the signals in these bands on the analog input and thus provide

adequate antialiasing filtering.

External Capacitance (pF)

Gain

0

50

100

500

1000

5000

1

2

4

184 kΩ 45.3 kΩ 27.1 kΩ 7.3 kΩ 4.1 kΩ 1.1 kΩ

88.6 kΩ 22.1 kΩ 13.2 kΩ 3.6 kΩ 2.0 kΩ 560 Ω

41.4 kΩ 10.6 kΩ 6.3 kΩ 1.7 kΩ 970 Ω

8–128 17.6 kΩ 4.8 kΩ 2.9 kΩ 790 Ω 440 Ω

270 Ω

120 Ω

Table V. Typical External Series Resistance That Will Not

Introduce 20-Bit Gain Error

If passive components are placed in front of the AIN1 input of the

AD7712, care must be taken to ensure that the source impedance

is low enough so as not to introduce gain errors in the system. The

dc input impedance for the AIN1 input is over 1 GΩ. The input

appears as a dynamic load that varies with the clock frequency

and with the selected gain (see Figure 7). The input sample

rate, as shown in Table III, determines the time allowed for the

analog input capacitor, C_IN, to be charged. External impedances

result in a longer charge time for this capacitor, which may result

in gain errors being introduced on the analog inputs. Table IV

shows the allowable external resistance/capacitance values

such that no gain error to the 16-bit level is introduced, while

Table V shows the allowable external resistance/capacitance

values such that no gain error to the 20-bit level is introduced.

Both inputs of the differential input channels (AIN1) look into

similar input circuitry.

External Capacitance (pF)

Gain

0

50

100

500

1000

5000

1

2

4

145 kΩ 34.5 kΩ 20.4 kΩ 5.2 kΩ 2.8 kΩ 700 Ω

70.5 kΩ 16.9 kΩ 10 kΩ

2.5 kΩ 1.4 kΩ 350 Ω

31.8 kΩ 8.0 kΩ 4.8 kΩ 1.2 kΩ 670 Ω

170 Ω

80 Ω

8–128 13.4 kΩ 3.6 kΩ 2.2 kΩ 550 Ω 300 Ω

The numbers in Tables IV and V assume a full-scale change on

the analog input. In any case, the error introduced due to longer

charging times is a gain error that can be removed using the

system calibration capabilities of the AD7712 provided that the

resultant span is within the span limits of the system calibration

techniques for the AD7712.

The AIN2 input contains a resistive attenuation network as

outlined in Figure 8. The typical input impedance on this input

is 44 kΩ. As a result, the AIN2 input should be driven from a

low impedance source.

R

INT

HIGH

IMPEDANCE

>1G⍀

(7k⍀ TYP)

AIN

C

INT

(11.5pF TYP)

33k⍀

V

BIAS

AIN2

11k⍀

SWITCHING FREQUENCY DEPENDS ON

AND SELECTED GAIN

MODULATOR

CIRCUIT

f

CLKIN

V

BIAS

Figure 7. AIN1 Input Impedance

Figure 8. AIN2 Input Impedance

REV. F

–15–

AD7712

ANALOG INPUT FUNCTIONS

Analog Input Ranges

to 1 mA to an external load. In applications where REF OUT

is connected directly to REF IN(+), REF IN(–) should be tied

to AGND to provide the nominal 2.5 V reference for the

AD7712.

The analog inputs on the AD7712 provide the user with consid-

erable flexibility in terms of analog input voltage ranges. One of

the inputs is a differential, programmable gain, input channel

that can handle either unipolar or bipolar input signals. The

common-mode range of this input is from V_SSto AV_DDprovided

that the absolute value of the analog input voltage lies between

V_SS– 30 mV and AV_DD+ 30 mV. The second analog input is a

single-ended, programmable gain, high level input that accepts

analog input ranges of 0 to +4 ϫ V_REF/GAIN or 4 ϫ V_REF/GAIN.

The reference inputs of the AD7712, REF IN(+) and

REF IN(–), provide a differential reference input capability.

The common-mode range for these differential inputs is from

V_SSto AV_DD. The nominal differential voltage, V_REF(REF

IN(+) – REF IN(–)), is 2.5 V for specified operation, but the

reference voltage can go to 5 V with no degradation in perfor-

mance provided that the absolute value of REF IN(+) and REF

IN(–) does not exceed its AV_DDand V_SSlimits and the V_BIAS

input voltage range limits are obeyed. The part is also functional

with V_REFvoltages down to 1 V but with degraded performance

as the output noise will, in terms of LSB size, be larger. REF

IN(+) must always be greater than REF IN(–) for correct opera-

tion of the AD7712.

The dc input leakage current on the AIN1 input is 10 pA maxi-

mum at 25°C ( 1 nA over temperature). This results in a dc

offset voltage developed across the source impedance. However,

this dc offset effect can be compensated for by a combination of

the differential input capability of the part and its system cali-

bration mode. The dc input current on the AIN2 input depends

on the input voltage. For the nominal input voltage range of

10 V, the input current is 225 µA typ.

Both reference inputs provide a high impedance, dynamic load

similar to the AIN1 analog inputs. The maximum dc input

leakage current is 10 pA ( 1 nA over temperature), and source

resistance may result in gain errors on the part. The reference

inputs look like the AIN1 analog input (see Figure 7). In this

case, R_INTis 5 kΩ typ and C_INTvaries with gain. The input

sample rate is f_{CLK IN}/256 and does not vary with gain. For gains

of 1 to 8, C_INTis 20 pF; for a gain of 16, it is 10 pF; for a gain

of 32, it is 5 pF; for a gain of 64, it is 2.5 pF; and for a gain of

128, it is 1.25 pF.

Burnout Current

The AIN1(+) input of the AD7712 contains a 4.5 µA current

source that can be turned on/off via the control register. This

current source can be used in checking that a transducer has not

burned out or gone open circuit before attempting to take mea-

surements on that channel. If the current is turned on and is

allowed to flow into the transducer and a measurement of the

input voltage on the AIN1 input is taken, it can indicate that the

transducer is not functioning correctly. For normal operation,

this burnout current is turned off by writing a 0 to the BO bit in

the control register.

The digital filter of the AD7712 removes noise from the refer-

ence input just as it does with the analog input, and the same

limitations apply regarding lack of noise rejection at integer

multiples of the sampling frequency. The output noise perfor-

mance outlined in Tables I and II assumes a clean reference. If

the reference noise in the bandwidth of interest is excessive, it

can degrade the performance of the AD7712. Using the on-chip

reference as the reference source for the part (i.e., connecting

REF OUT to REF IN) results in somewhat degraded output

noise performance from the AD7712 for portions of the noise

table that are dominated by the device noise. The on-chip refer-

ence noise effect is eliminated in ratiometric applications where

the reference is used to provide its excitation voltage for the analog

front end. The connection scheme shown in Figure 9 between

the REF OUT and REF IN pins of the AD7712 is recommended

when using the on-chip reference. Recommended reference

voltage sources for the AD7712 include the AD780 and AD680

2.5 V references.

Bipolar/Unipolar Inputs

The two analog inputs on the AD7712 can accept either unipo-

lar or bipolar input voltage ranges. Bipolar or unipolar options

are chosen by programming the B/U bit of the control register.

This programs both channels for either unipolar or bipolar

operation. Programming the part for either unipolar or bipolar

operation does not change any of the input signal conditioning;

it simply changes the data output coding. The data coding is

binary for unipolar inputs and offset binary for bipolar inputs.

The AIN1 input channel is differential and, as a result, the

voltage to which the unipolar and bipolar signals are referenced

is the voltage on the AIN1(–) input. For example, if AIN1(–) is

1.25 V and the AD7712 is configured for unipolar operation

with a gain of 1 and a V_REFof 2.5 V, the input voltage range on

the AIN1(+) input is 1.25 V to 3.75 V. If AIN1(–) is 1.25 V and

the AD7712 is configured for bipolar mode with a gain of 1 and

a V_REFof 2.5 V, the analog input range on the AIN1(+) input is

–1.25 V to +3.75 V. For the AIN2 input, the input signals are

referenced to AGND.

REF OUT

REF IN(+)

REF IN(–)

REFERENCE INPUT/OUTPUT

AD7712

The AD7712 contains a temperature compensated 2.5 V refer-

ence, which has an initial tolerance of 1%. This reference

voltage is provided at the REF OUT, pin and can be used as the

reference voltage for the part by connecting the REF OUT pin

to the REF IN(+) pin. This REF OUT pin is a single-ended

output, referenced to AGND, which is capable of providing up

Figure 9. REF OUT/REF IN Connection

–16–

REV. F

AD7712

V_BIASInput

The current drawn from the DV_DDpower supply is also directly

related to f_{CLK IN}. Reducing f_{CLK IN}by a factor of 2 will halve the

DV_DDcurrent but will not affect the current drawn from the

AV_DDpower supply.

The V_BIASinput determines at what voltage the internal analog

circuitry is biased. It essentially provides the return path for

analog currents flowing in the modulator, and as such it should

be driven from a low impedance point to minimize errors.

System Synchronization

For maximum internal headroom, the V_BIASvoltage should be

set halfway between AV_DDand V_SS. The difference between

AV_DDand (V_BIAS+ 0.85 ϫ V_REF) determines the amount of

headroom the circuit has at the upper end, while the difference

between V_SSand (V_BIAS– 0.85 ϫ V_REF) determines the amount

of headroom the circuit has at the lower end. Care should be

taken in choosing a V_BIASvoltage to ensure that it stays within

prescribed limits. For single +5 V operation, the selected V_BIAS

voltage must ensure that V_BIAS0.85 ϫ V_REFdoes not exceed

AV_DDor V_SSor that the V_BIASvoltage itself is greater than

V_SS+ 2.1 V and less than AV_DD– 2.1 V. For single +10 V

operation or dual 5 V operation, the selected V_BIASvoltage

must ensure that V_BIAS0.85 ϫ V_REFdoes not exceed AV_DD

or V_SSor that the V_BIASvoltage itself is greater than V_SS+ 3 V

or less than AV_DD– 3 V. For example, with AV_DD= +4.75 V,

V_SS= 0 V and V_REF= +2.5 V, the allowable range for the V_BIAS

voltage is +2.125 V to +2.625 V. With AV_DD= +9.5 V, V_SS= 0 V

and V_REF= +5 V, the range for V_BIASis +4.25 V to +5.25 V.

With AV_DD= +4.75 V, V_SS= –4.75 V, and V_REF= +2.5 V, the

V_BIASrange is –2.625 V to +2.625 V.

If multiple AD7712s are operated from a common master clock,

they can be synchronized to update their output registers simul-

taneously. A falling edge on the SYNC input resets the filter

and places the AD7712 into a consistent, known state. A com-

mon signal to the AD7712’s SYNC inputs will synchronize their

operation. This would normally be done after each AD7712 has

performed its own calibration or has had calibration coefficients

loaded to it.

The SYNC input can also be used to reset the digital filter in

systems where the turn-on time of the digital power supply

(DV_DD) is very long. In such cases, the AD7712 will start oper-

ating internally before the DV_DDline has reached its minimum

operating level, 4.75 V. With a low DV_DDvoltage, the AD7712’s

internal digital filter logic does not operate correctly. Thus, the

AD7712 may have clocked itself into an incorrect operating

condition by the time that DV_DDhas reached its correct level.

The digital filter will be reset upon issue of a calibration

command (whether it is self-calibration, system calibration, or

background calibration) to the AD7712. This ensures correct

operation of the AD7712. In systems where the power-on

default conditions of the AD7712 are acceptable, and no calibra-

tion is performed after power-on, issuing a SYNC pulse to the

AD7712 will reset the AD7712’s digital filter logic. An R, C on

the SYNC line, with R, C time constant longer than the DV_DD

power-on time, will perform the SYNC function.

The V_BIASvoltage does have an effect on the AV_DDpower supply

rejection performance of the AD7712. If the V_BIASvoltage tracks

the AV_DDsupply, it improves the power supply rejection from

the AV_DDsupply line from 80 dB to 95 dB. Using an external

Zener diode connected between the AV_DDline and V_BIASas the

source for the V_BIASvoltage gives the improvement in AV_DD

power supply rejection performance.

Accuracy

Sigma-delta ADCs, like VFCs and other integrating ADCs, do

not contain any source of nonmonotonicity and inherently offer

no missing codes performance. The AD7712 achieves excellent

linearity by the use of high quality, on-chip silicon dioxide

capacitors, which have a very low capacitance/voltage coefficient.

The device also achieves low input drift through the use of chop-

per stabilized techniques in its input stage. To ensure excellent

performance over time and temperature, the AD7712 uses digital

calibration techniques that minimize offset and gain error.

USING THE AD7712

SYSTEM DESIGN CONSIDERATIONS

The AD7712 operates differently from successive approximation

ADCs or integrating ADCs. Since it samples the signal continu-

ously, like a tracking ADC, there is no need for a start convert

command. The output register is updated at a rate determined

by the first notch of the filter, and the output can be read at any

time, either synchronously or asynchronously.

Autocalibration

Clocking

Autocalibration on the AD7712 removes offset and gain errors

from the device. A calibration routine should be initiated on the

device whenever there is a change in the ambient operating

temperature or supply voltage. It should also be initiated if there

is a change in the selected gain, filter notch, or bipolar/unipolar

input range. However, if the AD7712 is in its background cali-

bration mode, the above changes are all automatically taken

care of (after the settling time of the filter has been allowed for).

The AD7712 requires a master clock input, which may be an

external TTL/CMOS compatible clock signal applied to the

MCLK IN pin with the MCLK OUT pin left unconnected.

Alternatively, a crystal of the correct frequency can be connected

between MCLK IN and MCLK OUT, in which case the clock

circuit will function as a crystal controlled oscillator. For lower

clock frequencies, a ceramic resonator may be used instead of

the crystal. For these lower frequency oscillators, external

capacitors may be required on either the ceramic resonator or

on the crystal.

The AD7712 offers self-calibration, system calibration, and

background calibration facilities. For calibration to occur on the

selected channel, the on-chip microcontroller must record the

modulator output for two different input conditions. These are

zero-scale and full-scale points. With these readings, the micro-

controller can calculate the gain slope for the input to output

transfer function of the converter. Internally, the part works

with a resolution of 33 bits to determine its conversion result of

either 16 bits or 24 bits.

The input sampling frequency, the modulator sampling frequency,

the –3 dB frequency, the output update rate, and the calibration

time are all directly related to the master clock frequency, f_CLK

_IN.Reducing the master clock frequency by a factor of 2 will

halve the above frequencies and update rate and will double

the calibration time.

REV. F

–17–

AD7712

The AD7712 also provides the facility to write to the on-chip

calibration registers, and, in this manner, the span and offset for

the part can be adjusted by the user. The offset calibration regis-

ter contains a value that is subtracted from all conversion

results, while the full-scale calibration register contains a value

that is multiplied by all conversion results. The offset calibration

coefficient is subtracted from the result prior to the multiplication

by the full-scale coefficient. In the first three modes outlined

here, the DRDY line indicates that calibration is complete by

going low. If DRDY is low before (or goes low during) the

calibration command, it may take up to one modulator cycle

before DRDY goes high to indicate that calibration is in progress.

Therefore, the DRDY line should be ignored for up to one

modulator cycle after the last bit of the calibration command is

written to the control register.

unipolar mode, the system calibration is performed between the

two endpoints of the transfer function; in the bipolar mode, it is

performed between midscale and positive full scale.

This two-step system calibration mode offers another feature.

After the sequence has been completed, additional offset or gain

calibrations can be performed by themselves to adjust the zero

reference point or the system gain. This is achieved by perform-

ing the first step of the system calibration sequence (by writing

0, 1, 0 to MD2, MD1, MD0). This will adjust the zero-scale or

offset point but will not change the slope factor from what was

set during a full system calibration sequence.

System calibration can also be used to remove any errors from

an antialiasing filter on the analog input. A simple R, C anti-

aliasing filter on the front end may introduce a gain error on the

analog input voltage but the system calibration can be used to

remove this error.

Self-Calibration

In the self-calibration mode with a unipolar input range, the

zero-scale point used in determining the calibration coefficients

is with both inputs shorted (i.e., AIN1(+) = AIN1(–) = V_BIAS

for AIN1 and AIN2 = V_BIASfor AIN2 ) and the full-scale point

is V_REF. The zero-scale coefficient is determined by converting

an internal shorted inputs node. The full-scale coefficient is

determined from the span between this shorted inputs conversion

and a conversion on an internal V_REFnode. The self-calibra-

tion mode is invoked by writing the appropriate values (0, 0, 1)

to the MD2, MD1, and MD0 bits of the control register. In this

calibration mode, the shorted inputs node is switched in to the

modulator first and a conversion is performed; the V_REFnode is

then switched in, and another conversion is performed. When

the calibration sequence is complete, the calibration coefficients

updated and the filter resettled to the analog input voltage, the

DRDY output goes low. The self-calibration procedure takes

into account the selected gain on the PGA.

System Offset Calibration

System offset calibration is a variation of both the system cali-

bration and self-calibration. In this case, the zero-scale point

for the system is presented to the AIN input of the converter.

System offset calibration is initiated by writing 1, 0, 0 to MD2,

MD1, MD0. The system zero-scale coefficient is determined by

converting the voltage applied to the AIN input, while the full-

scale coefficient is determined from the span between this AIN

conversion and a conversion on V_REF. The zero-scale point

should be applied to the AIN input for the duration of the cali-

bration sequence. This is a one-step calibration sequence with

DRDY going low when the sequence is completed. In the uni-

polar mode, the system offset calibration is performed between

the two endpoints of the transfer function; in the bipolar mode,

it is performed between midscale and positive full scale.

Background Calibration

For bipolar input ranges in the self-calibrating mode, the

sequence is very similar to that just outlined. In this case, the

two points that the AD7712 calibrates are midscale (bipolar

zero) and positive full scale.

The AD7712 also offers a background calibration mode where

the part interleaves its calibration procedure with its normal

conversion sequence. In the background calibration mode, the

same voltages used as the calibration points in the self-calibration

mode are used, i.e., shorted inputs and V_REF. The background

calibration mode is invoked by writing 1, 0, 1 to MD2, MD1,

MD0 of the control register. When invoked, the background

calibration mode reduces the output data rate of the AD7712 by

a factor of 6, while the –3 dB bandwidth remains unchanged. Its

advantage is that the part is continually performing calibration

and automatically updating its calibration coefficients. As a

result, the effects of temperature drift, supply sensitivity, and

time drift on zero-scale and full-scale errors are automatically

removed. When the background calibration mode is turned on,

the part will remain in this mode until bits MD2, MD1, and

MD0 of the control register are changed. With background

calibration mode on, the first result from the AD7712 will be

incorrect as the full-scale calibration will not have been per-

formed. For a step change on the input, the second output

update will have settled to 100% of the final value.

System Calibration

System calibration allows the AD7712 to compensate for system

gain and offset errors as well as its own internal errors. System

calibration performs the same slope factor calculations as self-

calibration but uses voltage values presented by the system to

the AIN inputs for the zero-scale and full-scale points. System

calibration is a two-step process. The zero-scale point must be

presented to the converter first. It must be applied to the con-

verter before the calibration step is initiated and remain stable

until the step is complete. System calibration is initiated by

writing the appropriate values (0, 1, 0) to the MD2, MD1, and

MD0 bits of the control register. The DRDY output from the

device will signal when the step is complete by going low. After

the zero-scale point is calibrated, the full-scale point is applied

and the second step of the calibration process is initiated by

again writing the appropriate values (0, 1, 1) to MD2, MD1,

and MD0. Again the full-scale voltage must be set up before the

calibration is initiated, and it must remain stable throughout the

calibration step. DRDY goes low at the end of this second step

to indicate that the system calibration is complete. In the

Table VI summarizes the calibration modes and the calibration

points associated with them. It also gives the duration from

when the calibration is invoked to when valid data is available to

the user.

–18–

REV. F

AD7712

Table VI. Calibration Truth Table

Cal Type

MD2, MD1, MD0

Zero-Scale Cal

Full-Scale Cal

Sequence

Duration

Self-Cal

System Cal

System Offset Cal

Background Cal

0, 0, 1

0, 1, 0

0, 1, 1

1, 0, 0

1, 0, 1

Shorted Inputs

AIN

–

AIN

V_REF

–

AIN

V_REF

One-Step

Two-Step

One-Step

9 ϫ 1/Output Rate

4 ϫ 1/Output Rate

9 ϫ 1/Output Rate

6 ϫ 1/Output Rate

Shorted Inputs

Measurement errors due to offset drift or gain drift can be elimi-

nated at any time by recalibrating the converter or by operating

the part in the background calibration mode. Using the system

calibration mode can also minimize offset and gain errors in the

signal conditioning circuitry. Integral and differential linearity

errors are not significantly affected by temperature changes.

Span and Offset Limits

Whenever a system calibration mode is used, there are limits on

the amount of offset and span that can be accommodated. The

range of input span in both the unipolar and bipolar modes for

AIN1 has a minimum value of 0.8 ϫ V_REF/GAIN and a maxi-

mum value of 2.1 ϫ V_REF/GAIN. For AIN2, both numbers are

a factor of 4 higher.

POWER SUPPLIES AND GROUNDING

The amount of offset that can be accommodated depends on

whether the unipolar or bipolar mode is being used. This offset

range is limited by the requirement that the positive full-scale

calibration limit is ≤ 1.05 ϫ V_REF/GAIN for AIN1. Therefore,

the offset range plus the span range cannot exceed 1.05 ϫ V_REF

GAIN for AIN1. If the span is at its minimum (0.8 ϫ V_REF

GAIN), the maximum the offset can be is (0.25 ϫ V_REF/GAIN)

Since the analog inputs and reference input are differential,

most of the voltages in the analog modulator are common-mode

voltages. V_BIASprovides the return path for most of the analog

currents flowing in the analog modulator. As a result, the V_BIAS

input should be driven from a low impedance to minimize

errors due to charging/discharging impedances on this line.

When the internal reference is used as the reference source for

the part, AGND is the ground return for this reference voltage.

/

for AIN1. For AIN2, both ranges are multiplied by a factor of 4.

In the bipolar mode, the system offset calibration range is again

restricted by the span range. The span range of the converter in

bipolar mode is equidistant around the voltage used for the

zero-scale point, thus the offset range plus half the span range

cannot exceed (1.05 × V_REF/GAIN) for AIN1. If the span is

set to 2 ×V_REF/GAIN, the offset span cannot move more than

(0.05 × V_REF/GAIN) before the endpoints of the transfer func-

tion exceed the input overrange limits (1.05 × V_REF/GAIN) for

The analog and digital supplies to the AD7712 are independent

and separately pinned out to minimize coupling between the

analog and digital sections of the device. The digital filter will

provide rejection of broadband noise on the power supplies,

except at integer multiples of the modulator sampling frequency.

The digital supply (DV_DD) must not exceed the analog positive

supply (AV_DD) by more than 0.3 V in normal operation. If sepa-

rate analog and digital supplies are used, the decoupling scheme

AIN1. If the span range is set to the minimum (0.4 × V_REF

/

shown in Figure 10 is recommended. In systems where AV_DD

=

GAIN), the maximum allowable offset range is (0.65 × V_REF

GAIN) for AIN1. Once again, for AIN2, both ranges are

multiplied by a factor of 4.

/

5 V and DV_DD= 5 V, it is recommended that AV_DDand DV_DD

are driven from the same 5 V supply, although each supply

should be decoupled separately as shown in Figure 10. It is

preferable that the common supply is the system’s analog

5 V supply.

POWER-UP AND CALIBRATION

On power-up, the AD7712 performs an internal reset, which

sets the contents of the control register to a known state. How-

ever, to ensure correct calibration for the device, a calibration

routine should be performed after power-up.

It is also important that power is applied to the AD7712 before

signals at REF IN, AIN, or the logic input pins in order to avoid

excessive current. If separate supplies are used for the AD7712

and the system digital circuitry, then the AD7712 should be

powered up first. If it is not possible to guarantee this, then

current limiting resistors should be placed in series with the

logic inputs.

The power dissipation and temperature drift of the AD7712 are

low and no warm-up time is required before the initial calibra-

tion is performed. However, if an external reference is being

used, this reference must have stabilized before calibration is

initiated.

DIGITAL 5V

SUPPLY

ANALOG

SUPPLY

Drift Considerations

The AD7712 uses chopper stabilization techniques to minimize

input offset drift. Charge injection in the analog switches and dc

leakage currents at the sampling node are the primary sources of

offset voltage drift in the converter. The dc input leakage cur-

rent is essentially independent of the selected gain. Gain drift

within the converter depends primarily upon the temperature

tracking of the internal capacitors. It is not affected by leakage

currents.

10␮F

0.1␮F

AV

DV

DD

AD7712

Figure 10. Recommended Decoupling Scheme

REV. F

–19–

AD7712

DIGITAL INTERFACE

the output data register. It is reset high when the last bit of data

(either 16th bit or 24th bit) is read from the output register. If

data is not read from the output register, the DRDY line will

remain low. The output register will continue to be updated at

the output update rate, but DRDY will not indicate this. A read

from the device in this circumstance will access the most recent

word in the output register. If a new data-word becomes avail-

able to the output register while data is being read from the

output register, DRDY will not indicate this and the new data-

word will be lost to the user. DRDY is not affected by reading

from the control register or the calibration registers.

The AD7712’s serial communications port provides a flexible

arrangement to allow easy interfacing to industry-standard

microprocessors, microcontrollers, and digital signal processors.

A serial read to the AD7712 can access data from the output

register, the control register, or the calibration registers. A serial

write to the AD7712 can write data to the control register or the

calibration registers.

Two different modes of operation are available, optimized for

different types of interfaces where the AD7712 can act either as

master in the system (it provides the serial clock) or as slave (an

external serial clock can be provided to the AD7712). These

two modes, labeled self-clocking mode and external clocking

mode, are discussed in detail in the following sections.

Data can be accessed from the output data register only when

DRDY is low. If RFS goes low with DRDY high, no data trans-

fer will take place. DRDY does not have any effect on reading

data from the control register or from the calibration registers.

Self-Clocking Mode

The AD7712 is configured for its self-clocking mode by tying

the MODE pin high. In this mode, the AD7712 provides the

serial clock signal used for the transfer of data to and from the

AD7712. This self-clocking mode can be used with processors

that allow an external device to clock their serial port, including

most digital signal processors and microcontrollers such as the

68HC11 and 68HC05. It also allows easy interfacing to serial

parallel conversion circuits in systems with parallel data com-

munication, allowing interfacing to 74XX299 universal shift

registers without any additional decoding. In the case of shift

registers, the serial clock line should have a pull-down resistor

instead of the pull-up resistor shown in Figures 11 and 12.

Figure 11 shows a timing diagram for reading from the AD7712

in the self-clocking mode. This read operation shows a read

from the AD7712’s output data register. A read from the con-

trol register or calibration registers is similar, but, in these cases,

the DRDY line is not related to the read function. Depending

on the output update rate, it can go low at any stage in the

control/calibration register read cycle without affecting the read

and its status should be ignored. A read operation from either

the control or calibration registers must always read 24 bits of

data from the respective register.

Figure 11 shows a read operation from the AD7712. For the

timing diagram shown, it is assumed that there is a pull-up

resistor on the SCLK output. With DRDY low, the RFS input

is brought low. RFS going low enables the serial clock of the

AD7712 and also places the MSB of the word on the serial data

line. All subsequent data bits are clocked out on a high to low

transition of the serial clock and are valid prior to the following

rising edge of this clock. The final active falling edge of SCLK

clocks out the LSB, and this LSB is valid prior to the final active

rising edge of SCLK. Coincident with the next falling edge of

SCLK, DRDY is reset high. DRDY going high turns off the

SCLK and the SDATA outputs, which means that the data

hold time for the LSB is slightly shorter than for all other bits.

Read Operation

Data can be read from the output register, the control register, or

the calibration registers. A0 determines whether the data read

accesses data from the control register or from the output/calibra-

tion registers. This A0 signal must remain valid for the duration

of the serial read operation. With A0 high, data is accessed from

either the output register or from the calibration registers. With

A0 low, data is accessed from the control register.

The function of the DRDY line is dependent on only the output

update rate of the device and the reading of the output data

register. DRDY goes low when a new data-word is available in

DRDY (O)

t₃

t₂

A0 (I)

t₅

t₄

RFS (I)

t₉

t₆

SCLK (O)

t₈

MSB

t₁₀

t₇

THREE-STATE

LSB

SDATA (O)

Figure 11. Self-Clocking Mode, Output Data Read Operation

–20–

REV. F

AD7712

Write Operation

register or from the calibration registers. With A0 low, data is

accessed from the control register.

Data can be written to either the control register or calibration

registers. In either case, the write operation is not affected by

the DRDY line, and the write operation does not have any

effect on the status of DRDY. A write operation to the control

register or the calibration register must always write 24 bits to

the respective register.