AD8370-EVALZ [ADI]
LF to 750 MHz, Digitally Controlled VGA; LF至750兆赫，数字控制VGA
| 型号： | AD8370-EVALZ |
| 厂家： | 
ADI |
| 描述： | LF to 750 MHz, Digitally Controlled VGA |
| 文件： | 总28页 (文件大小：774K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LF to 750 MHz,
Digitally Controlled VGA
Data Sheet
AD8370
FEATURES
FUNCTIONAL BLOCK DIAGRAM
Programmable low and high gain (<2 dB resolution)
Low range: −11 dB to +17 dB
High range: 6 dB to 34 dB
Differential input and output
200 Ω differential input
VCCI
VCCO VCCO
3
11
6
4
2
5
7
PWUP
ICOM
BIAS CELL
VOCM
OCOM
100 Ω differential output
1
8
OPHI
INHI
PRE
AMP
OUTPUT
AMP
TRANSCONDUCTANCE
7 dB noise figure @ maximum gain
Two-tone IP3 of 35 dBm @ 70 MHz
−3 dB bandwidth of 750 MHz
40 dB precision gain range
Serial 8-bit digital interface
Wide input dynamic range
Power-down feature
9
16
15
INLO
ICOM
OPLO
OCOM
10
SHIFT REGISTER
AND LATCHES
AD8370
14
12
13
DATA CLCK LTCH
Figure 1.
Single 3 V to 5 V supply
70
40
CODE = LAST 7 BITS OF GAIN CODE
(NO MSB)
APPLICATIONS
60
50
40
30
20
10
0
30
20
10
0
Differential ADC drivers
IF sampling receivers
RF/IF gain stages
Cable and video applications
SAW filter interfacing
Single-ended-to-differential conversion
HIGH GAIN MODE
LOW GAIN MODE
HIGH GAIN MODE
GAIN
0.409
0.059
CODE
–10
–20
–30
GENERAL DESCRIPTION
GAIN
CODE
The AD8370 is a low cost, digitally controlled, variable gain
amplifier (VGA) that provides precision gain control, high IP3,
and low noise figure. The excellent distortion performance and
wide bandwidth make the AD8370 a suitable gain control
device for modern receiver designs.
LOW GAIN MODE
0
10 20 30 40 50 60 70 80 90 100 110 120 130
GAIN CODE
Figure 2. Gain vs. Gain Code at 70 MHz
For wide input, dynamic range applications, the AD8370 provides
two input ranges: high gain mode and low gain mode. A vernier,
7-bit, transconductance (gm) stage provides 28 dB of gain range
at better than 2 dB resolution and 22 dB of gain range at better than
1 dB resolution. A second gain range, 17 dB higher than the first,
can be selected to provide improved noise performance.
Gain control of the AD8370 is through a serial 8-bit gain control
word. The MSB selects between the two gain ranges, and the
remaining 7 bits adjust the overall gain in precise linear gain steps.
Fabricated on the ADI high speed XFCB process, the high
bandwidth of the AD8370 provides high frequency and low
distortion. The quiescent current of the AD8370 is 78 mA
typically. The AD8370 amplifier comes in a compact, thermally
enhanced 16-lead TSSOP package and operates over the
temperature range of −40°C to +85°C.
The AD8370 is powered on by applying the appropriate logic
level to the PWUP pin. When powered down, the AD8370
consumes less than 4 mA and offers excellent input to output
isolation. The gain setting is preserved when operating in a
power-down mode.
Rev. B
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registeredtrademarks arethe property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113 © 2005-2011 Analog Devices, Inc. All rights reserved.
AD8370
Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1
Basic Connections...................................................................... 15
Gain Codes.................................................................................. 15
Power-Up Feature....................................................................... 15
Choosing Between Gain Ranges .............................................. 16
Layout and Operating Considerations .................................... 16
Package Considerations............................................................. 17
Single-Ended-to-Differential Conversion............................... 17
DC-Coupled Operation............................................................. 18
ADC Interfacing......................................................................... 19
3 V Operation ............................................................................. 20
Evaluation Board and Software .................................................... 22
Appendix ......................................................................................... 25
Characterization Equipment..................................................... 25
Composite Waveform Assumption.......................................... 25
Definitions of Selected Parameters.......................................... 25
Outline Dimensions....................................................................... 28
Ordering Guide .......................................................................... 28
Applications....................................................................................... 1
General Description ......................................................................... 1
Functional Block Diagram .............................................................. 1
Revision History ............................................................................... 2
Specifications..................................................................................... 3
Absolute Maximum Ratings............................................................ 5
ESD Caution.................................................................................. 5
Pin Configuration and Function Descriptions............................. 6
Typical Performance Characteristics ............................................. 7
Theory of Operation ...................................................................... 13
Block Architecture...................................................................... 13
Preamplifier................................................................................. 13
Transconductance Stage ............................................................ 13
Output Amplifier........................................................................ 14
Digital Interface and Timing .................................................... 14
Applications..................................................................................... 15
REVISION HISTORY
12/11—Rev. A to Rev. B
Changes to Slew Rate Parameters, Table 1 .................................... 3
Updated Outline Dimensions....................................................... 28
Changes to Ordering Guide .......................................................... 28
7/05—Rev. 0 to Rev. A
Changes to Features.......................................................................... 1
Changes to Table 1............................................................................ 3
Changes to Figure 11 and Figure 15............................................... 8
Added Figure 12; Renumbered Sequentially ................................ 8
Added Figure 16; Renumbered Sequentially ................................ 9
Changes to Evaluation Board and Software Section.................. 22
Changes to Figure 60...................................................................... 23
Updated Outline Dimensions....................................................... 28
Changes to Ordering Guide .......................................................... 28
1/04—Revision 0: Initial Version
Rev. B | Page 2 of 28
Data Sheet
AD8370
SPECIFICATIONS
VS = 5 V, T = 25°C, ZS = 200 Ω, ZL = 100 Ω at gain code HG127, 70 MHz, 1 V p-p differential output, unless otherwise noted.
Table 1.
Parameter
Conditions
Min Typ
Max Unit
DYNAMIC PERFORMANCE
−3 dB Bandwidth
Slew Rate
VOUT < 1 V p-p
750
5.75
3.5
MHz
V/ns
V/ns
Gain Code HG127, RL = 1 kΩ, AD8370 in compression
Gain Code LG127, RL = 1 kΩ, VOUT = 2 V p-p
Pins INHI and IHLO
INPUT STAGE
Maximum Input
Input Resistance
Common-Mode Input Range
CMRR
Gain Code LG2, 1 dB compression
Differential
3.2
200
3.2
77
V p-p
Ω
V p-p
dB
Differential, f = 10 MHz, Gain Code LG127
Input Noise Spectral Density
GAIN
1.9
nV/√Hz
Maximum Voltage Gain
High Gain Mode
Gain Code = HG127
Gain Code = LG127
34
52
17
7.4
dB
V/V
dB
Low Gain Mode
V/V
Minimum Voltage Gain
High Gain Mode
Gain Code = HG1
Gain Code = LG1
−8
0.4
dB
V/V
dB
V/V
(V/V)/Code
(V/V)/Code
mdB/°C
ns
Low Gain Mode
Gain Step Size
−25
0.06
0.408
0.056
–2
High Gain Mode
Low Gain Mode
Gain Code = HG127
For 6 dB gain step, settled to 10% of final value
Pins OPHI and OPLO
RL ≥ 1 kΩ (1 dB compression)
Differential
VINHI = VINLO, over all gain codes
Gain Temperature Sensitivity
Step Response
20
OUTPUT INTERFACE
Output Voltage Swing
Output Resistance
Output Differential Offset
NOISE/HARMONIC PERFORMANCE
10 MHz
8.4
95
60
V p-p
Ω
mV
Gain Flatness
Noise Figure
Within 10 MHz of 10 MHz
0.01
7.2
dB
dB
Second Harmonic1
Third Harmonic1
Output IP3
Output 1 dB Compression Point
70 MHz
VOUT = 2 V p-p
VOUT = 2 V p-p
−77
−77
35
dBc
dBc
dBm
dBm
17
Gain Flatness
Noise Figure
Within 10 MHz of 70 MHz
0.02
7.2
dB
dB
Second Harmonic1
Third Harmonic1
Output IP3
VOUT = 2 V p-p
VOUT = 2 V p-p
−65
−62
35
dBc
dBc
dBm
dBm
Output 1 dB Compression Point
17
Rev. B | Page 3 of 28
AD8370
Data Sheet
Parameter
Conditions
Min Typ
0.03
Max Unit
140 MHz
Gain Flatness
Within 10 MHz of 140 MHz
dB
Noise Figure
Second Harmonic1
Third Harmonic1
Output IP3
Output 1 dB Compression Point
7.2
−54
−50
33
dB
VOUT = 2 V p-p
VOUT = 2 V p-p
dBc
dBc
dBm
dBm
17
190 MHz
Gain Flatness
Within 10 MHz of 240 MHz
0.03
dB
Noise Figure
7.2
−43
−43
33
dB
Second Harmonic1
Third Harmonic1
Output IP3
VOUT = 2 V p-p
VOUT = 2 V p-p
dBc
dBc
dBm
dBm
Output 1 dB Compression Point
17
240 MHz
Gain Flatness
Within 10 MHz of 240 MHz
0.04
dB
Noise Figure
7.4
–28
–33
32
dB
Second Harmonic1
Third Harmonic1
Output IP3
VOUT = 2 V p-p
VOUT = 2 V p-p
dBc
dBc
dBm
dBm
Output 1 dB Compression Point
17
380 MHz
Gain Flatness
Within 10 MHz of 240 MHz
0.04
dB
Noise Figure
Output IP3
Output 1 dB Compression Point
8.1
27
14
dB
dBm
dBm
POWER-INTERFACE
Supply Voltage
Quiescent Current3
3.02
72.5 79
5.5
V
PWUP High, GC = LG127, RL = ∞, 4 seconds after
power-on, thermal connection made to exposed
paddle under device
85.5
mA
vs. Temperature4
−40°C ≤ TA ≤ +85°C
105
mA
mA
Total Supply Current
PWUP High, VOUT = 1 V p-p, ZL = 100 Ω reactive,
GC = LG127 (includes load current)
82
Power-Down Current
vs. Temperature4
PWUP low
−40°C ≤TA ≤ +85°C
Pin PWUP
Voltage to enable the device
Voltage to disable the device
PWUP = 0 V
3.7
mA
mA
5
POWER-UP INTERFACE
Power-Up Threshold4
Power-Down Threshold4
PWUP Input Bias Current
GAIN CONTROL INTERFACE
1.8
V
V
nA
0.8
400
Pins CLCK, DATA, and LTCH
Voltage for a logic high
Voltage for a logic low
4
VIH
VIL
1.8
V
V
4
0.8
Input Bias Current
900
nA
1 Refer to Figure 22 for performance into a lighter load.
2 See the 3 V Operation section for more information.
3 Minimum and maximum specified limits for this parameter are guaranteed by production test.
4 Minimum or maximum specified limit for this parameter is a 6-sigma value and not guaranteed by production test.
Rev. B | Page 4 of 28
Data Sheet
AD8370
ABSOLUTE MAXIMUM RATINGS
Table 2.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those listed in the operational sections
of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Parameter
Rating
Supply Voltage, VS
PWUP, DATA, CLCK, LTCH
Differential Input Voltage,
5.5 V
VS + 500 mV
2 V
V
INHI – VINLO
Common-Mode Input Voltage,
VINHI or VINLO, with Respect to
ICOM or OCOM
VS + 500 mV (max),
VICOM – 500 mV,
VOCOM – 500 mV (min)
Internal Power Dissipation
575 mW
θJA (Exposed Paddle Soldered Down)
θJA (Exposed Paddle Not Soldered Down)
θJC (At Exposed Paddle)
30°C/W
95°C/W
9°C/W
Maximum Junction Temperature
Operating Temperature Range
Storage Temperature Range
Lead Temperature Range
(Soldering 60 sec)
150°C
−40°C to +85°C
−65°C to +150°C
235°C
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate
on the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. B | Page 5 of 28
AD8370
Data Sheet
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
INHI
ICOM
VCCI
INLO
ICOM
DATA
CLCK
LTCH
VCCO
OCOM
OPLO
AD8370
PWUP
VOCM
VCCO
OCOM
OPHI
TOP VIEW
(Not to Scale)
Figure 3.16-Lead TSSOP
Table 3. Pin Function Descriptions
Pin No.
Mnemonic Description
1
INHI Balanced Differential Input. Internally biased.
2, 15, PADDLE ICOM
Input Common. Connect to a low impedance ground. This node is also connected to the exposed pad
on the bottom of the device.
3
4
5
VCCI
PWUP
VOCM
Input Positive Supply. 3.0 V to 5.5 V. Should be properly bypassed.
Power Enable Pin. Device is operational when PWUP is pulled high.
Common-Mode Output Voltage Pin. The midsupply ((VVCCO − VOCOM)/2) common-mode voltage is delivered
to this pin for external bypassing for additional common-mode supply decoupling. This can be achieved
with a bypass capacitor to ground. This pin is an output only and is not to be driven externally.
6, 11
7, 10
8
VCCO
OCOM
OPHI
Output Positive Supply. 3.0 V to 5.5 V. Should be properly bypassed.
Output Common. Connect to a low impedance ground.
Balanced Differential Output. Biased to midsupply.
9
12
OPLO
LTCH
Balanced Differential Output. Biased to midsupply.
Serial Data Latch Pin. Serial data is clocked into the shift register via the DATA pin when LTCH is low. Data
in shift register is latched on the next high-going edge.
13
14
16
CLCK
DATA
INLO
Serial Clock Input Pin.
Serial Data Input Pin.
Balanced Differential Input. Internally biased.
Rev. B | Page 6 of 28
Data Sheet
AD8370
TYPICAL PERFORMANCE CHARACTERISTICS
VS = 5 V, Z S = 200 Ω, ZL = 100 Ω, T = 25°C, unless otherwise noted.
40
35
30
25
20
15
10
5
70
60
50
40
30
20
10
0
40
HIGH GAIN CODES SHOWN WITH DASHED LINES
HG127
CODE = LAST 7 BITS OF GAIN CODE
(NO MSB)
30
HG77
HIGH GAIN MODE
HG102
HG51
20
LG127
HG25
LOW GAIN MODE
HIGH GAIN MODE
10
∆ GAIN
∆ CODE
LG90
≅ 0.409
HG18
HG9
LG36
HG3
0
–10
–20
–30
0
∆ GAIN
∆ CODE
≅ 0.059
LG18
LG9
–5
–10
LOW GAIN MODE
LOW GAIN CODES SHOWN WITH SOLID LINES
10
100
1000
0
10 20 30 40 50 60 70 80 90 100 110 120 130
GAIN CODE
FREQUENCY (MHz)
Figure 4. Gain vs. Gain Code at 70 MHz
Figure 7. Frequency Response vs. Gain Code
40
35
30
25
20
15
10
50
40
35
30
25
20
15
10
5
30
25
20
15
10
5
+25°C
UNIT CONVERSION NOTE FOR
100Ω LOAD: dBVrms = dBm–10dB
HIGH GAIN MODE
45
40
35
30
25
20
LOW GAIN MODE
+85°C
–40°C
SHADING INDICATES ±3σ FROM THE
MEAN. DATA BASED ON 30 PARTS
FROM TWO BATCH LOTS.
SHADING INDICATES ±3σ FROM THE
MEAN. DATA BASED ON 30 PARTS
FROM TWO BATCH LOTS.
0
0
50
100
150
200
250
300
350
400
–5
140
0
20
40
60
80
100
120
FREQUENCY (MHz)
GAIN CODE
Figure 5. Output Third-Order Intercept vs. Gain Code at 70 MHz
Figure 8. Output Third-Order Intercept vs. Frequency at Maximum Gain
25
45
40
35
30
25
20
15
10
5
20
LG127
15
380MHz
LOW GAIN MODE
10
HG18
70MHz
HG127
5
380MHz
70MHz
20
HIGH GAIN MODE
0
0
100
200
300
400
500
600
0
40
60
80
100
120
140
FREQUENCY (MHz)
GAIN CODE
Figure 9. Noise Figure vs. Frequency at Various Gains
Figure 6. Noise Figure vs. Gain Code at 70 MHz
Rev. B | Page 7 of 28
AD8370
Data Sheet
20
16
12
8
2.0
1.5
LOW GAIN MODE
100Ω LOAD
1kΩ LOAD
HIGH GAIN MODE
LOW GAIN MODE
1.0
0.5
HIGH GAIN MODE
–40°C
+85°C
0
4
–0.5
–1.0
–1.5
–2.0
UNIT CONVERSION NOTE:
FOR 100Ω LOAD: dBV rms = dBm–10dB
FOR 1kΩ LOAD: dBV rms = dBm
0
ERROR AT –40°C AND +85°C WITH RESPECT TO +25°C.
SHADING INDICATES ±3σ FROM THE MEAN. DATA
BASED ON 30 PARTS FROM ONE BATCH LOT.
–4
SHADING INDICATES ±3σ FROM THE
MEAN. DATA BASED ON 30 PARTS
FROM TWO BATCH LOTS.
–8
0
10
100
1000
20
40
60
80
100
120
140
FREQUENCY (MHz)
GAIN CODE
Figure 10. Output P1dB vs. Gain Code at 70 MHz
Figure 13. Gain Error over Temperature vs. Frequency, RL = 100 Ω
0
–10
–20
–30
–40
–50
–60
–70
–80
–50
20
18
16
14
12
10
8
+25°C, 100Ω LOAD
–60
+85°C, 100Ω LOAD
18
16
14
12
10
8
–70
HIGH GAIN MODE
–80
UNIT CONVERSION NOTE:
–90
RE 100Ω LOAD: dBV rms = dBm – 10dB
RE 1kΩ LOAD: dBV rms = dBm
–100
–110
–120
–130
–140
–40°C, 100Ω LOAD
+25°C, 1kΩ LOAD
+85°C, 1kΩ LOAD
LOW GAIN MODE
6
SHADING INDICATES ±3σ FROM
THE MEAN. DATA BASED ON 30
PARTS FROM TWO BATCH LOTS.
–90
0
–40°C, 1kΩ LOAD
250 300 350
FREQUENCY (MHz)
20
40
60
80
100
120
140
4
400
6
0
50
100
150
200
GAIN CODE
Figure 11. Two-Tone Output IMD3 vs. Gain Code at 70 MHz,
RL = 1 kΩ, VOUT = 2 V p-p Composite Differential
Figure 14. Output P1dB vs. Frequency
35
25
20
15
10
5
–50
–52
–54
–56
–58
–60
–62
–64
–66
–68
–70
–72
–74
–76
–78
–80
–82
–84
30
25
20
15
10
5
HIGH GAIN
MODE
LOW GAIN
MODE
–40°C
0
+25°C
–5
–10
–15
+85°C
0
–5
0
50
100
150
200
250
300
350
400
0
20
40
60
80
100
120
140
GAIN CODE
FREQUENCY (MHz)
Figure 15. Two-Tone Output IMD3 vs. Frequency at Maximum Gain,
RL = 1 kΩ, VOUT = 2 V p-p Composite Differential
Figure 12. Output Third-Order Intercept vs. Gain Code at 70 MHz,
RL = 1 kΩ, VOUT = 2 V p-p Composite Differential
Rev. B | Page 8 of 28
Data Sheet
AD8370
90
34
32
30
28
26
24
22
20
18
16
14
24
22
20
18
16
14
12
10
8
120
60
1GHz
+85°C
150
30
–40°C
+25°C
S
22
5MHz
180
0
330
210
S
11
6
240
300
4
400
0
50
100
150
200
250
300
350
270
FREQUENCY (MHz)
Figure 16. Output Third-Order Intercept vs. Frequency at Maximum Gain,
RL = 1 kΩ, VOUT = 2 V p-p Composite Differential
Figure 19. Input and Output Reflection Coefficients, S11 and S22,
ZO = 100 Ω Differential
250
200
150
100
50
100
2.0
1.5
1.0
16 DIFFERENT GAIN
CODES REPRESENTED
R+jX FORMAT
50
0.5
0
–40°C
0
+85°C
–50
–100
–150
–0.5
–1.0
ERROR AT –40°C AND +85°C WITH RESPECT TO +25°C.
SHADING INDICATES ±3σ FROM THE MEAN. DATA
BASED ON 30 PARTS FROM ONE BATCH LOT.
–1.5
–2.0
0
0
100
200
300
400
500
600
700
10
100
1000
FREQUENCY (MHz)
FREQUENCY (MHz)
Figure 17. Gain Error over Temperature vs. Frequency, RL = 1 kΩ
Figure 20. Input Resistance and Reactance vs. Frequency
0
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
LOW GAIN R = 1kΩ
L
–10
HIGH GAIN R = 1kΩ
L
–20
–30
–40
–50
–60
–70
–80
–90
LOW GAIN, R = 1kΩ
HIGH GAIN R = 100Ω
L
L
LOW GAIN R = 100Ω
L
HIGH GAIN, R = 100Ω
LOW GAIN, R = 100Ω
L
L
HIGH GAIN, R = 1kΩ
L
0
20
40
60
80
100
120
140
0
20
40
60
80
100
120
140
GAIN CODE
GAIN CODE
Figure 18. Second-Order Harmonic Distortion vs. Gain Code at 70 MHz,
OUT = 2 V p-p Differential
Figure 21. Third-Order Harmonic Distortion vs. Gain Code at 70 MHz,
OUT = 2 V p-p Differential
V
V
Rev. B | Page 9 of 28
AD8370
Data Sheet
0
–10
–20
–30
–40
–50
–60
–70
–80
120
110
100
90
HD
R = 100Ω
L
2
HD
R
= 100Ω
3
L
80
70
60
HD
R = 1kΩ
L
3
50
40
HD
R
= 1kΩ
2
L
30
20
–90
0
50
100
150
200
250
300
350
400
1
10
100
1000
FREQUENCY (MHz)
FREQUENCY (MHz)
Figure 25. Power Supply Rejection Ratio vs. Frequency at Maximum Gain
Figure 22. Harmonic Distortion vs. Frequency at Maximum Gain,
OUT = 2 V p-p Composite Differential
V
0
120
100
80
60
40
20
0
80
60
40
20
0
FORWARD TRANSMISSION, HG0
–20
FORWARD TRANSMISSION, LG0
–40
–60
–80
16 DIFFERENT GAIN
CODES REPRESENTED
R+jX FORMAT
–100
–120
FORWARD TRANSMISSION, PWUP LOW
REVERSE TRANSMISSION, HG127
–20
–40
700
0
100
200
300
400
500
600
10
100
1000
FREQUENCY (MHz)
FREQUENCY (MHz)
Figure 23. Output Resistance and Reactance vs. Frequency
Figure 26. Various Forms of Isolation vs. Frequency
860
840
820
800
780
760
740
720
700
1400
1300
1200
1100
1000
900
R
= 1kΩ
L
HIGH GAIN MODE
R
= 100Ω
L
LOW GAIN MODE
800
700
600
0
10 20 30 40 50 60 70 80 90 100 110 120 130
0
100
200
300
400
500
600
700
800
900
GAIN CODE
FREQUENCY (MHz)
Figure 24. Group Delay vs. Gain Code at 70 MHz
Figure 27. Group Delay vs. Frequency at Maximum Gain
Rev. B | Page 10 of 28
Data Sheet
AD8370
80
70
60
DIFFERENTIAL OUTPUT (50mV/DIV)
LG32, LG127
ZERO
HG32, HG127
50
40
30
20
10
0
PWUP (2V/DIV)
GAIN CODE HG127
GND
INPUT = –30dBm, 70MHz 100 AVERAGES
10
100
FREQUENCY (MHz)
1000
TIME (40ns/DIV)
Figure 28. Common-Mode Rejection Ratio vs. Frequency
Figure 31. PWUP Time Domain Response
12
10
8
DIFFERENTIAL OUTPUT (10mV/DIV)
ZERO
6dB GAIN STEP (HG36 TO LG127)
LG127
6
LTCH (2V/DIV)
4
HG18
2
HG127
GND
INPUT = –30dBm, 70MHz
NO AVERAGING
0
10
110
210
310
410
510
610
FREQUENCY (MHz)
TIME (20ns/DIV)
Figure 32. Gain Step Time Domain Response
Figure 29. Input Referred Noise Spectral Density vs.
Frequency at Various Gains
V
DIFFERENTIAL
OUT
V
V
OPHI
OPLO
DIFFERENTIAL V
DIFFERENTIAL V
OUT
GND
IN
GND
TIME (2ns/DIV)
TIME (2ns/DIV)
Figure 30. DC-Coupled Large Signal Pulse Response
Figure 33. Overdrive Recovery
Rev. B | Page 11 of 28
AD8370
Data Sheet
85
80
75
70
65
60
55
2.75
2.70
2.65
2.60
2.55
2.50
2.45
2.40
+85°C
+25°C
LOW GAIN
HIGH GAIN
–40°C
LOW GAIN MODE
32 64 96
HIGH GAIN MODE
32 64 96
50
0
16
32
48
64
GAIN CODE
80
96
112
128
0
0
128
GAIN CODE
Figure 34. Supply Current vs. Gain Code
Figure 36. Common-Mode Output Voltage vs. Gain Code at
Various Temperatures
35
30
25
20
15
10
5
MEAN: 51.9
σ: 0.518
DATA FROM 136 PARTS
FROM ONE BATCH LOT
0
50
51
52
53
54
55
GAIN (V/V)
Figure 35. Distribution of Voltage Gain, HG127, 70 MHz, RL = 100 Ω
Rev. B | Page 12 of 28
Data Sheet
AD8370
THEORY OF OPERATION
set by passive components. See Figure 38 for a simplified
The AD8370 is a low cost, digitally controlled, fine adjustment
variable gain amplifier (VGA) that provides both high IP3 and
low noise figure. The AD8370 is fabricated on an ADI proprietary
high performance 25 GHz silicon bipolar process. The –3 dB
bandwidth is approximately 750 MHz throughout the variable
gain range. The typical quiescent current of the AD8370 is
78 mA. A power-down feature reduces the current to less than
4 mA. The input impedance is approximately 200 Ω differential,
and the output impedance is approximately 100 Ω differential
to be compatible with saw filters and matching networks used
in intermediate frequency (IF) radio applications. Because there
is no feedback between the input and output and stages within
the amplifier, the input amplifier is isolated from variations in
output loading and from subsequent impedance changes, and
excellent input to output isolation is realized. Excellent distortion
performance and wide bandwidth make the AD8370 a suitable
gain control device for modern differential receiver designs. The
AD8370 differential input and output configuration is ideally
suited to fully differential signal chain circuit designs, although
it can be adapted to single-ended system applications, if required.
schematic of the input interface.
1mA
INHI/INLO
2kΩ
VCC/2
1mA
Figure 38. INHI/INLO Simplified Schematic
TRANSCONDUCTANCE STAGE
The digitally controlled gm section has 42 dB of controllable
gain and makes gain adjustments within each gain range. The
step size resolution ranges from a fine ~ 0.07 dB up to a coarse
6 dB per bit, depending on the gain code. As shown in Figure 39,
of the 42 dB total range, 28 dB has resolution of better than
2 dB, and 22 dB has resolution of better than 1 dB.
BLOCK ARCHITECTURE
The three basic building blocks of the AD8370 are a high/low
gain selectable input preamplifier, a digitally controlled
transconductance (gm) block, and a fixed gain output stage.
Figure 39 shows typical input levels that can be applied to this
amplifier at different gain settings. The maximum input was
determined by finding the 1 dB compression or expansion point
of the VOUT/VSOURCE gain. Note that this is not VOUT/VIN. In this
way, the change in the input impedance of the device is also
taken into account.
VCCI
3
VCCO VCCO
11
6
4
2
5
7
PWUP
ICOM
BIAS CELL
VOCM
OCOM
1
8
OPHI
INHI
PRE
AMP
OUTPUT
AMP
TRANSCONDUCTANCE
16
9
3.2
INLO
OPLO
OCOM
ICOM 15
10
<0.5dB
RES
2.8
2.4
2.0
1.6
1.2
0.8
0.4
0
SHIFT REGISTER
AND LATCHES
LOW GAIN
<1dB
RES
AD8370
HIGH GAIN
34dB
GAIN
17dB
GAIN
14
12
13
DATA CLCK LTCH
<0.5dB
RESOLUTION
Figure 37. Functional Block Diagram
<2dB
RES
12dB
GAIN
PREAMPLIFIER
<1dB
RES
0.1dB GAIN
There are two selectable input preamplifiers. Selection is made
by the most significant bit (MSB) of the serial gain control data-
word. In the high gain mode, the overall device gain is 7.1 V/V
(17 dB) above the low gain setting. The two preamplifiers give
the AD8370 the ability to accommodate a wide range of input
amplitudes. The overlap between the two gain ranges allows the
user some flexibility based on noise and distortion demands.
See the Choosing Between Gain Ranges section for more
information.
6dB
GAIN
<2dB
RES
–5dB GAIN
–8dB GAIN
–11dB GAIN
–25dB GAIN
0
0.2
0.4
0.6
V
0.8
SOURCE
1.0
1.2
1.4
1.6
1.8
[V peak] (V)
Figure 39. Gain Resolution and Nominal Input and
Output Range over the Gain Range
The input impedance is approximately 200 Ω differential,
regardless of which preamplifier is selected. Note that the input
impedance is formed by using active circuit elements and is not
Rev. B | Page 13 of 28
AD8370
Data Sheet
Table 4. Serial Programming Timing Parameters
OUTPUT AMPLIFIER
Parameter
Min
Unit
ns
ns
ns
ns
The output impedance is approximately 100 Ω differential and,
like the input preamplifier, this impedance is formed using
active circuit elements. See Figure 40 for a simplified schematic
of the output interface.
Clock Pulse Width (TPW)
Clock Period (TCK)
25
50
Setup Time Data vs. Clock (TDS)
Setup Time Latch vs. Clock (TES)
Hold Time Latch vs. Clock (TEH)
10
20
10
ns
10µA
OPHI/OPLO
740Ω
VCC/2
CLCK/DATA/LTCH/PWUP
Figure 42. Simplified Circuit for Digital Inputs
Figure 40. OPHI/OPLO Simplified Circuit
The gain of the output amplifier, and thus the AD8370 as a
whole, is load dependent. The following equation can be used to
predict the gain deviation of the AD8370 from that at 100 Ω as
the load is varied.
VOCM
75Ω
1.98
98
VCC/2
GainDeviation =
1+
RLOAD
For example, if RLOAD is 1 kΩ, the gain is a factor of 1.80 (5.12 dB)
above that at 100 Ω, all other things being equal. If RLOAD is 50
Ω, the gain is a factor of 0.669 (3.49 dB) below that at 100 Ω.
Figure 43. Simplified Circuit for VOCM Output
DIGITAL INTERFACE AND TIMING
The digital control port uses a standard TTL interface. The 8-bit
control word is read in a serial fashion when the LTCH pin is
held low. The levels presented to the DATA pin are read on each
rising edge of the CLCK signal. Figure 41 illustrates the timing
diagram for the control interface. Minimum values for timing
parameters are presented in Table 4. Figure 42 is a simplified
schematic of the digital input pins.
T
DS
DATA
(PIN 14)
MSB MSB-1 MSB-2 MSB-3 LSB+3 LSB+2 LSB+1 LSB
T
CK
T
PW
CLCK
(PIN 13)
T
EH
T
ES
LTCH
(PIN 12)
Figure 41. Digital Timing Diagram
Rev. B | Page 14 of 28
Data Sheet
AD8370
APPLICATIONS
BASIC CONNECTIONS
GAIN CODES
Figure 44 shows the minimum connections required for basic
operation of the AD8370. Supply voltages between 3.0 V and
5.5 V are allowed. The supply to the VCCO and VCCI pins
should be decoupled with at least one low inductance, surface-
mount ceramic capacitor of 0.1 μF placed as close as possible to
the device.
The AD8370’s two gain ranges are referred to as high gain (HG)
and low gain (LG). Within each range, there are 128 possible
gain codes. Therefore, the minimum gain in the low gain range
is given by the nomenclature LG0 whereas the maximum gain
in that range is given by LG127. The same is true for the high
gain range. Both LG0 and HG0 essentially turn off the variable
transconductance stage, and thus no output is available with
these codes (see Figure 26).
SERIAL CONTROL
INTERFACE
1nF
1nF
The theoretical linear voltage gain can be expressed with respect
to the gain code as
R
S
16 15 14 13 12 11 10
9
2
AV = GainCode Vernier (1 + (PreGain − 1) MSB)
BALANCED
SOURCE
BALANCED
LOAD
R
L
AD8370
where:
R
S
AV is the linear voltage gain.
1
2
3
4
5
6
7
8
2
GainCode is the digital gain control word minus the MSB
1nF
1nF
1nF
(the final 7 bits).
100pF
100pF
0.1F
0.1F
Vernier = 0.055744 V/V
PreGain = 7.079458 V/V
FERRITE
BEAD
FERRITE
BEAD
+V (3.0V TO 5.0V)
S
MSB is the most significant bit of the 8-bit gain control word.
The MSB sets the device in either high gain mode (MSB = 1)
or low gain mode (MSB = 0).
Figure 44. Basic Connections
The AD8370 is designed to be used in differential signal chains.
Differential signaling allows improved even-order harmonic
cancellation and better common-mode immunity than can be
achieved using a single-ended design. To fully exploit these
benefits, it is necessary to drive and load the device in a
balanced manner. This requires some care to ensure that the
common-mode impedance values presented to each set of
inputs and outputs are balanced. Driving the device with an
unbalanced source can degrade the common-mode rejection
ratio. Loading the device with an unbalanced load can cause
degradation to even-order harmonic distortion and premature
output compression. In general, optimum designs are fully
balanced, although the AD8370 still provides impressive
performance when used in an unbalanced environment.
For example, a gain control word of HG45 (or 10101101 binary)
results in a theoretical linear voltage gain of 17.76 V/V,
calculated as
45 × 0.055744 × (1 + (7.079458 − 1) × 1)
Increments or decrements in gain within either gain range are
simply a matter of operating on the GainCode. Six –dB gain
steps, which are equivalent to doubling or halving the linear
voltage gain, are accomplished by doubling or halving the
GainCode.
When power is first applied to the AD8370, the device is
programmed to code LG0 to avoid overdriving the circuitry
following it.
The AD8370 is a fine adjustment, VGA. The gain control
transfer function is linear in voltage gain. On a decibel scale,
this results in the logarithmic transfer functions shown in
Figure 4. At the low end of the gain transfer function, the slope
is steep, providing a rather coarse control function. At the high
end of the gain control range, the decibel step size decreases,
allowing precise gain adjustment.
POWER-UP FEATURE
The power-up feature does not affect the GainCode, and the
gain setting is preserved when in power-down mode. Powering
down the AD8370 (bringing PWUP low while power is still
applied to the device) does not erase or change the GainCode
from the AD8370, and the same gain code is in place when the
device is powered up, that is, when PWUP is brought high
again. Removing power from the device all together and
reapplying, however, reprograms to LG0.
Rev. B | Page 15 of 28
AD8370
Data Sheet
gain is increased beyond this point, which explains the knee in
the OIP3 curve. The IIP3 curve has a knee for the same reason;
however, as the gain is increased beyond the knee, the IIP3
starts to decrease rather than increase. This is because in this
region OIP3 is constant, therefore the higher the gain, the lower
the IIP3. The two gain ranges have equal SFDR at
CHOOSING BETWEEN GAIN RANGES
There is some overlap between the two gain ranges; users can
choose which one is most appropriate for their needs. When
deciding which preamp to use, consider resolution, noise,
linearity, and spurious-free dynamic range (SFDR). The most
important points to keep in mind are
approximately 13 dB power gain.
•
•
The low gain range has better gain resolution.
The high gain range has a better noise figure.
LAYOUT AND OPERATING CONSIDERATIONS
Each input and output pin of the AD8370 presents either a
100 Ω or 50 Ω impedance relative to their respective ac grounds.
To ensure that signal integrity is not seriously impaired by the
printed circuit board, the relevant connection traces should
provide an appropriate characteristic impedance to the ground
plane. This can be achieved through proper layout.
•
•
The high gain range has better linearity and SFDR at
higher gains.
Conversely, the low gain range has higher SFDR at lower
gains.
Figure 45 provides a summary of noise, OIP3, IIP3, and SFDR
as a function of device power gain. SFDR is defined as
When laying out an RF trace with a controlled impedance,
consider the following:
2
SFDR =
(
IIP3 − NF − NS
)
•
•
Space the ground plane to either side of the signal trace at
least three line-widths away to ensure that a microstrip
(vertical dielectric) line is formed, rather than a coplanar
(lateral dielectric) waveguide.
3
where:
IIP3 is the input third-order intercept point, the output
intercept point in dBm minus the gain in dB.
Ensure that the width of the microstrip line is constant and
that there are as few discontinuities as possible, such as
component pads, along the length of the line. Width
variations cause impedance discontinuities in the line and
may result in unwanted reflections.
NF is the noise figure in dB.
NS is source resistor noise, –174 dBm for a 1 Hz bandwidth at
300°K (27°C).
•
Do not use silkscreen over the signal line because it alters
the line impedance.
In general, NS = 10 log10(kTB), where k = 1.374 ×10−23 , T is the
temperature in degrees Kelvin, and B is the noise bandwidth in
Hertz.
Keep the length of the input and output connection lines as
short as possible.
50
180
170
160
150
140
130
120
110
100
Figure 46 shows the cross section of a PC board, and Table 5
show the dimensions that provide a 100 Ω line impedance for
FR-4 board material with εr = 4.6.
NF LOW GAIN
OIP3 LOW GAIN
40
OIP3 HIGH GAIN
30
IIP3 LOW GAIN
Table 5.
20
IIP3 HIGH GAIN
100 Ω
22 mils
53 mils
2 mils
50 Ω
10
W
H
T
13 mils
8 mils
2 mils
NF HIGH GAIN
0
–10
–20
–30
SFDR LOW GAIN
SFDR HIGH GAIN
3W
W
3W
T
–30
–20
–10
0
10
20
30
40
POWER GAIN (dB)
E
H
R
Figure 45. OIP3, IIP3, NF, and SFDR Variation with Gain
As the gain increases, the input amplitude required to deliver
the same output amplitude is reduced. This results in less
distortion at the input stage, and therefore the OIP3 increases.
At some point, the distortion of the input stage becomes small
enough such that the nonlinearity of the output stage becomes
dominant. The OIP3 does not improve significantly because the
Figure 46. Cross-Sectional View of a PC Board
It possible to approximate a 100 Ω trace on a board designed
with the 50 Ω dimensions above by removing the ground plane
within 3 line-widths of the area directly below the trace.
Rev. B | Page 16 of 28
Data Sheet
AD8370
1 nF capacitors for CAC presents a capacitive reactance of
The AD8370 contains both digital and analog sections. Care
should be taken to ensure that the digital and analog sections
are adequately isolated on the PC board. The use of separate
ground planes for each section connected at only one point via
a ferrite bead inductor ensures that the digital pulses do not
adversely affect the analog section of the AD8370.
−j1.6 Ω on each input node at 100 MHz. This attenuates the
applied input voltage by 0.003 dB. If 10 pF capacitors had been
selected, the voltage delivered to the input would be reduced
by 2.1 dB when operating with a 200 Ω source impedance.
0.5
Due to the nature of the AD8370’s circuit design, care must be
taken to minimize parasitic capacitance on the input and output.
The AD8370 could become unstable with more than a few pF of
shunt capacitance on each input. Using resistors in series with
input pins is recommended under conditions of high source
capacitance.
HIGH GAIN MODE
(GAIN CODE HG255)
0
–0.5
LOW GAIN MODE
(GAIN CODE LG127)
High transient and noise levels on the power supply, ground,
and digital inputs can, under some circumstances, reprogram the
AD8370 to an unintended gain code. This further reinforces the
need for proper supply bypassing and decoupling. The user
should also be aware that probing the AD8370 and associated
circuitry during circuit debug may also induce the same effect.
–1.0
0
100
200
300
400
500
FREQUENCY (MHz)
Figure 48. Differential Output Balance for a Single-Ended Input Drive at
Maximum Gain (RL = 1 kΩ, CAC = 10 nF)
Figure 48 illustrates the differential balance at the output for a
single-ended input drive for multiple gain codes. The differential
balance is better than 0.5 dB for signal frequencies less than
250 MHz. Figure 49 depicts the differential balance over the
entire gain range at 10 MHz. The balance is degraded for lower
gain settings because the finite common gain allows some of the
input signal applied to INHI to pass directly through to the
OPLO pin. At higher gain settings, the differential gain dominates
and balance is restored.
PACKAGE CONSIDERATIONS
The package of the AD8370 is a compact, thermally enhanced
TSSOP 16-lead design. A large exposed paddle on the bottom of
the device provides both a thermal benefit and a low inductance
path to ground for the circuit. To make proper use of this pack-
aging feature, the PCB needs to make contact directly under the
device, connected to an ac/dc common ground reference with
as many vias as possible to lower the inductance and thermal
impedance.
0.6
LOW GAIN MODE
HIGH GAIN MODE
SINGLE-ENDED-TO-DIFFERENTIAL CONVERSION
0.5
0.4
0.3
0.2
0.1
0
SERIAL CONTROL
INTERFACE
C
C
AC
AC
R
S
16 15 14 13 12 11 10
9
SINGLE-
ENDED
SOURCE
R
L
AD8370
1
2
3
4
5
6
7
8
0
32
64
96
0
32
64
96
128
C
C
AC
AC
GAIN CODE
0.1µF
1nF
0.1µF
Figure 49. Differential Output Balance at 10 MHz for a Single-Ended Drive vs.
Gain Code (RL = 1 kΩ, CAC = 10 nF)
+V
S
Figure 47. Single-Ended-to-Differential Conversion
Even though the amplifier is no longer being driven in a balanced
manner, the distortion performance remains adequate for most
applications. Figure 50 illustrates the harmonic distortion
performance of the circuit in Figure 47 over the entire gain range.
The AD8370 is primarily designed for differential signal inter-
facing. The device can be used for single-ended-to-differential
conversion simply by terminating the unused input to ground
using a capacitor as depicted in Figure 47. The ac coupling
capacitors should be selected such that their reactance is
negligible at the frequency of operation. For example, using
Rev. B | Page 17 of 28
AD8370
Data Sheet
SERIAL CONTROL
INTERFACE
If the amplifier is driven in single-ended mode, the input
impedance varies depending on the value of the resistor used to
terminate the other input as
V
OCM
499Ω
+5V
100Ω
16 15 14 13 12 11 10
9
RinSE = RinDIFF + RTERM
499Ω
where RTERM is the termination resistor connected to the other
input.
R
T
R
L
AD8138
AD8370
–40
–50
–60
499Ω
1
2
3
4
5
6
7
8
R
S
R
T
2
499Ω
100Ω
V
OCM
1nF
0.1µF
1nF
+5V
HD2
HD2
SINGLE-ENDED GROUND
REFERENCED SOURCE
–70
–80
Figure 52. DC Coupling the AD8370. The AD8138 is used as a unity-gain level
shifting amplifier to lift the common-mode level of the source to midsupply.
HD3
HD3
–90
The AD8370 is also a dc accurate VGA. The common-mode dc
voltage present at the output pins is internally set to midsupply
using what is essentially a buffered resistive divider network
connected between the positive supply rail and the common
(ground) pins. The input pins are at a slightly higher dc potential,
typically 250 mV to 550 mV above the output pins, depending
on gain setting. In a typical single-supply application, it is
necessary to raise the common-mode reference level of the
source and load to roughly midsupply to maintain symmetric
swing and to avoid sinking or sourcing strong bias currents
from the input and output pins. It is possible to use balanced
dual supplies to allow ground referenced source and load, as
shown in Figure 51. By connecting the VOCM pin and unused
input to ground, the input and output common-mode potentials
are forced to virtual ground. This allows direct coupling of
ground referenced source and loads. The initial differential
input offset is typically only a few 100 µV. Over temperature,
the input offset could be as high as a few tens of mVs. If precise
dc accuracy is needed over temperature and time, it may be
necessary to periodically measure the input offset and to apply
the necessary opposing offset to the unused differential input,
canceling the resulting output offset.
LOW GAIN MODE
32 64 96
HIGH GAIN MODE
32 64 96
–100
0
0
128
GAIN CODE
Figure 50. Harmonic Distortion of the Circuit in Figure 47
DC-COUPLED OPERATION
–2.5V
SERIAL CONTROL
INTERFACE
0V
1nF
R
T
R
S
16 15 14 13 12 11 10
9
SINGLE-
ENDED
GROUND
REFERENCED
SOURCE
R
L
AD8370
1
2
3
4
5
6
7
8
0V
–2.5V
1nF
0.1µF
+2.5V
0.1µF
To address situations where dual supplies are not convenient, a
second option is presented in Figure 52. The AD8138 differential
amplifier is used to translate the common-mode level of the driving
source to midsupply, which allows dc accurate performance
with a ground-referenced source without the need for dual
supplies. The bandwidth of the solution in Figure 52 is limited
by the gain-bandwidth product of the AD8138. The normalized
frequency response of both implementations is shown in Figure 53.
Figure 51. DC Coupling the AD8370. Dual supplies are used to set the input
and output common-mode levels to 0 V.
Rev. B | Page 18 of 28
Data Sheet
AD8370
10
Often it is wise to include input and output parasitic suppression
resistors, RIP and ROP. Parasitic suppressing resistors help to
prevent resonant effects that occur as a result of internal bond-
wire inductance, pad to substrate capacitance, and stray
capacitance of the printed circuit board trace artwork. If
omitted, undesirable settling characteristics may be observed.
Typically, only 10 Ω to 25 Ω of series resistance is all that is
needed to help dampen resonant effects. Considering that most
ADCs present a relatively high input impedance, very little
signal is lost across the RIP and ROP series resistors.
8
6
AD8370 WITH
AD8138 SINGLE
+5V SUPPLY
4
2
0
–2
–4
–6
–8
–10
AD8370
USING DUAL
±2.5V SUPPLY
Depending on the input impedance presented by the input
system of the ADC, it may be desirable to terminate the ADC
input down to a lower impedance by using a terminating
resistor, RT. The high frequency response of the AD8370
exhibits greater peaking when driving very light loads. In
addition, the terminating resistor helps to better define the
input impedance at the ADC input. Any part-to-part variability
of ADC input impedance is reduced when shunting down the
ADC inputs by using a moderate tolerance terminating resistor
(typically a 1% value is acceptable).
1
10
100
1k
10k 100k
1M
10M 100M 1G
FREQUENCY (Hz)
Figure 53. Normalized Frequency Response of the Two Solutions in
Figure 51 and Figure 52
ADC INTERFACING
Although the AD8370 is designed to provide a 100 Ω output
source impedance, the device is capable of driving a variety
of loads while maintaining reasonable gain and distortion
performance. A common application for the AD8370 is ADC
driving in IF sampling receivers and broadband wide dynamic
range digitizers. The wide gain adjustment range allows the use
of lower resolution ADCs. Figure 54 illustrates a typical ADC
interface network.
After defining reasonable values for coupling capacitors,
suppressing resistors, and the terminating resistor, it is time to
design the intermediate filter network. The example in
Figure 54 suggests a second-order, low-pass filter network
comprised of series inductors and a shunt capacitor. The order
and type of filter network used depends on the desired high
frequency rejection required for the ADC interface, as well as
on pass-band ripple and group delay. In some situations, the
signal spectra may already be sufficiently band-limited such
that no additional filter network is necessary, in which case ZS
would simply be a short and ZP would be an open. In other
situations, it may be necessary to have a rather high-order
antialiasing filter to help minimize unwanted high frequency
spectra from being aliased down into the first Nyquist zone of
the ADC.
R
C
Z
S
R
IP
OP
AC
V
V
AD8370
IN
Z
R
T
P
ADC
Z
100Ω
IN
IN
R
C
Z
R
IP
V
OP
AC
S
OCM
Figure 54. Generic ADC Interface
Many factors need to be considered before defining component
values used in the interface network, such as the desired frequency
range of operation, the input swing, and input impedance of the
ADC. AC coupling capacitors, CAC, should be used to block any
potential dc offsets present at the AD8370 outputs, which would
otherwise consume the available low-end range of the ADC.
The CAC capacitors should be large enough so that they present
negligible reactance over the intended frequency range of
operation. The VOCM pin may serve as an external reference
for ADCs that do not include an on-board reference. In either
case, it is suggested that the VOCM pin be decoupled to ground
through a moderately large bypassing capacitor (1 nF to 10 nF)
to help minimize wideband noise pick-up.
To properly design the filter network, it is necessary to consider
the overall source and load impedance presented by the AD8370
and ADC input, including the additional resistive contribution
of suppression and terminating resistors. The filter design can
then be handled by using a single-ended equivalent circuit, as
shown in Figure 55. A variety of references that address filter
synthesis are available. Most provide tables for various filter
types and orders, indicating the normalized inductor and
capacitor values for a 1 Hz cutoff frequency and 1 Ω load. After
scaling the normalized prototype element values by the actual
desired cut-off frequency and load impedance, it is simply a
matter of splitting series element reactances in half to realize the
final balanced filter network component values.
Rev. B | Page 19 of 28
AD8370
Data Sheet
SOURCE
LOAD
A complete design example is shown in Figure 58. The AD8370
is configured for single-ended-to-differential conversion with
the input terminated down to present a single-ended 75 Ω input.
A sixth-order Chebyshev differential filter is used to interface
the output of the AD8370 to the input of the AD9430
R
Z
S
S
SINGLE-ENDED
EQUIVALENT
V
Z
Z
R
R
S
P
L
R
Z
S
S
170 MSPS, 12-bit ADC. The filter minimizes aliasing effects
and improves harmonic distortion performance.
2
2
L
2
BALANCED
CONFIGURATION
V
S
P
R
L
The input of the AD9430 is terminated with a 1.5 kΩ resistor so
that the overall load presented to the filter network is ~1 kΩ.
The variable gain of the AD8370 extends the useable dynamic
range of the ADC. The measured intermodulation distortion of
the combination is presented in Figure 57 at 42 MHz.
2
R
Z
S
2
S
2
Figure 55. Single-Ended-to-Differential Network Conversion
As an example, a second-order, Butterworth, low-pass filter
design is presented where the differential load impedance is
1200 Ω, and the padded source impedance of the AD8370 is
assumed to be 120 Ω. The normalized series inductor value for
the 10-to-1, load-to-source impedance ratio is 0.074 H, and the
normalized shunt capacitor is 14.814 F. For a 70 MHz cutoff
frequency, the single-ended equivalent circuit consists of a
200 nH series inductor followed by a 27 pF capacitor. To realize
the balanced equivalent, simply split the 200 nH inductor in
half to realize the network shown in Figure 56.
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
–110
–120
–130
R
R
S
L
R
=
= 0.1
S
L
= 0.074H
N
NORMALIZED
SINGLE-ENDED
EQUIVALENT
V
C
14.814F
R = 1Ω
0
10
20
30
40
50
60
70
S
N
L
FREQUENCY (MHz)
fC = 1Hz
Figure 57. FFT Plot of Two-Tone Intermodulation Distortion at
42 MHz for the Circuit in Figure 58
R
= 120Ω
200nH
S
In Figure 57, the intermodulation products are comparable to
the noise floor of the ADC. The spurious-free dynamic range of
the combination is better than 66 dB for a 70 MHz measurement
bandwidth.
DE-NORMALIZED
SINGLE-ENDED
EQUIVALENT
V
27pF
R = 1200Ω
S
L
fC = 70MHz
3 V OPERATION
R
2
S
= 60Ω
= 60Ω
100nH
It is possible to operate the AD8370 at voltages as low as 3 V
with only minor performance degradation. Table 6 gives typical
specifications for operation at 3 V.
R
2
L
= 600Ω
= 600Ω
BALANCED
CONFIGURATION
V
27pF
S
R
L
2
R
100nH
S
Table 6.
2
Parameter
Output IP3
P1dB
Typical (70 MHz, RL = 100 Ω)
+23.5 dBm
+12.7 dBm
Figure 56. Second-Order, Butterworth, Low-Pass Filter Design Example
−3 dB Bandwidth
IMD3
650 MHz (HG 127)
−82 dBc (RL = 1 kΩ)
Rev. B | Page 20 of 28
Data Sheet
AD8370
SERIAL CONTROL INTERFACE
FROM 75Ω
Tx-LINE
C
AC
C
100nF
120Ω
AC
68nH
180nH
220nH
25Ω
V
A
IN
R
S
100nF
16 15 14 13 12 11 10
9
27pF
39pF
27pF
1.5kΩ
AD8370
AD9430
1
2
3
4
5
6
7
8
C
AC
25Ω
68nH
180nH
220nH
V
B
IN
C
AC
100nF
1nF
100nF
0.1µF
0.1µF
+V
S
Figure 58. ADC Interface Example
Rev. B | Page 21 of 28
AD8370
Data Sheet
EVALUATION BOARD AND SOFTWARE
The evaluation board allows quick testing of the AD8370 by
using standard 50 Ω test equipment. The schematic is shown in
Figure 59. Transformers T1 and T2 are used to transform 50 Ω
source and load impedances to the desired input and output
reference levels. The top and bottom layers are shown in
Figure 63 and Figure 64. The ground plane was removed under
the traces between T1 and Pins INHI and INLO to approximate
a 100 Ω characteristic impedance.
The evaluation board comes with the AD8370 control software
that allows serial gain control from most computers. The
evaluation board is connected via a cable to the parallel port of
the computer. Adjusting the appropriate slider bar in the control
software automatically updates the gain code of the AD8370 in
either a linear or linear-in-dB fashion.
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
21
22
23
24
25
D-SUB 25 PIN MALE
C9 OPEN
L2*
R7 R6 R5
1kΩ 1kΩ 1kΩ
C1
C3
1nF
1nF
TC4-1W
JTX-2-10T
T1
T2
IN+
IN–
50Ω Tx LINE
50Ω Tx LINE
50Ω Tx LINE
50Ω Tx LINE
OUT+
OUT–
16 15 14 13 12 11 10
9
R2
0Ω
R4
0Ω
1:4
2:1
AD8370
R1
R3
0Ω
0Ω
1
2
3
4
5
6
7
8
C2
C4
1nF
1nF
C8
0.1µF
C5
0.1µF
C6
1µF
SW1
PWUP
L1*
VOCM
+V
S
R9
OPEN
C7
0.1µF
GND
R8 49.9Ω
C10 OPEN
P2
1
2
3
4
5
GND
*EMI SUPPRESSION FERRITE
HZ1206E601R-00
V
S
Figure 59. AD8370 Evaluation Board Schematic
Rev. B | Page 22 of 28
Data Sheet
AD8370
Figure 60. Evaluation Software
Table 7. AD8370 Evaluation Board Configuration Options
Component Function
Default Condition
VS, GND, VOCM Power Interface Vector Pins. Apply supply voltage between VS and GND. The VOCM
pin allows external monitoring of the common-mode input and output bias levels.
Not applicable
SW1, R8,
C10, PWUP
Device Enable. Set to Position B to power up the device. When in Position A, the PWUP SW1 = installed
pin is connected to the PWUP vector pin. The PWUP pin allows external power cycling R8 = 49.9 Ω (Size 0805)
of the device. R8 and C10 are provided to allow for proper cable termination.
C10 = open (Size 0805)
P1, R5, R6,
R7, C9
Serial Control Interfaces. The evaluation board can be controlled using most PCs.
Windows®-based control software is shipped with the evaluation kit. A 25-pin, D-sub
connector cable is required to connect the PC to the evaluation board. It may be
necessary to use a capacitor on the clock line, depending on the quality of the PC port
signals. A 1 nF capacitor for C9 is usually sufficient for reducing clock overshoot.
P1 = installed
R5, R6, R7 = 1 kΩ (Size 0603)
C9 = open (Size 0603)
J1, J2, J6, J7
Input and Output Signal Connectors. These SMA connectors provide a convenient way Not applicable
to interface the evaluation board with 50 Ω test equipment. Typically, the device is
evaluated using a single-ended source and load. The source should connect to
J1 (IN+), and the load should connect to J6 (OUT+).
C1, C2, C3, C4
T1, T2
AC Coupling Capacitors. Provide ac coupling of the input and output signals.
Impedance Transformers. T1 provides a 50 Ω to 200 Ω impedance transformation.
T2 provides a 100 Ω to 50 Ω impedance transformation.
C1, C2, C3, C4 = 1 nF (Size 0603)
T1 = TC4 −1W (Mini-Circuits)
T2 = JTX−2−10T (Mini-Circuits)
R1, R2, R3, R4
Single-Ended or Differential. R2 and R4 are used to ground the center tap of the
secondary windings on transformers T1 and T2. R1 and R3 should be used to ground
J2 and J7 when used in single-ended applications.
R1, R2, R3, R4 = 0 Ω (Size 0603)
C5, C6, C7,
C8 L1, L2
Power Supply Decoupling. Nominal supply decoupling consists of a ferrite bead
series inductor followed by a 1 µF capacitor to ground followed by a 0.1 µF capacitor
to ground positioned as close to the device as possible. C7 provides additional
decoupling of the input common-mode voltage. L1 provides high frequency
isolation between the input and output power supply. L2 provides high
frequency isolation between the analog and digital ground.
C6 = 1 µF (Size 0805)
C5, C7, C8 = 0.1 µF (Size 0603)
L1, L2 = HZ1206E601R-00
(Steward, Size 1206)
Rev. B | Page 23 of 28
AD8370
Data Sheet
Figure 61. Evaluation Board Top Silkscreen
Figure 63. Evaluation Board Top
Figure 62. Evaluation Board Bottom Silkscreen
Figure 64. Evaluation Board Bottom
Rev. B | Page 24 of 28
Data Sheet
AD8370
APPENDIX
CHARACTERIZATION EQUIPMENT
DEFINITIONS OF SELECTED PARAMETERS
An Agilent N4441A Balanced-Measurement System was used to
obtain the gain, phase, group delay, reverse isolation, CMRR,
and s-parameter information contained in this data sheet. With
the exception of the s-parameter information, T-attenuator pads
were used to match the 50 Ω impedance of this instrument’s ports
to the AD8370. An Agilent 4795A Spectrum Analyzer was used
to obtain nonlinear measurements IMD, IP3, and P1dB through
matching baluns and/or attenuator networks. Various other
measurements were taken with setups shown in this section.
Common-mode rejection ratio (Figure 28) has been defined for
this characterization effort as
Differential Mode Gain
Common Mode Gain
where the numerator is the gain into a differential load at the
output due to a differential source at the input, and the
denominator is the gain into a differential-mode load at the
output due to a common-mode source at the input. In terms of
mixed-mode s-parameters, this equates to
COMPOSITE WAVEFORM ASSUMPTION
The nonlinear two-tone measurements made for this data sheet,
that is, IMD and IP3, are based on the assumption of a fixed
value composite waveform at the output, generally 1 V p-p. The
frequencies of interest dictate the use of RF test equipment, and
because this equipment is generally not designed to work in
units of volts, but rather watts and dBm, an assumption was
made to facilitate equipment setup and operation. Two sinusoidal
tones can be represented as
SDD21
SDC21
More information on mixed-mode s-parameters can be
obtained in a reference by Bockelman, D.E. and Eisenstadt,
W.R., Combined Differential and Common-Mode Scattering
Parameters: Theory and Simulation. IEEE Transactions on
Microwave Theory and Techniques, v 43, n 7, 1530 (July 1995).
Reverse isolation (Figure 26) is defined as SDD12.
Power supply rejection ratio (PSRR) is defined as
V1 = V sin (2∏f1t)
V2 = V sin (2∏f2t)
Adm
As
The RMS average voltage of one tone is
2
1 T
1
(
)
V1 dt =
∫
where Adm is the differential mode forward gain (SDD21), and
As is the gain from the power supply pins (VCCI and VCCO,
taken together) to the output (OPLO and OPHI, taken
differentially), corrected for impedance mismatch. The
following reference provides more information: Gray, P.R.,
Hurst, P.J., Lewis, S.H. and Meyer, R.G., Analysis and Design of
Analog Integrated Circuits, 4th Edition, John Wiley & Sons, Inc.,
page 422.
T
2
0
where T is the period of the waveform. The RMS average
voltage of the two-tone composite signal is
2
1 T
(
V1 +V2 dt =1
)
∫
T
0
It can be shown that the average power of this composite
waveform is twice (3 dB) that of the single tone. This also
means that the composite peak-to-peak voltage is twice (6 dB)
that of a single tone. This principle can be used to set correct
input amplitudes from generators scaled in dBm and is correct
if the two tones are of equal amplitude and are reasonably close
in frequency.
Rev. B | Page 25 of 28
AD8370
Data Sheet
–22.5dB
PORT 1
PORT 1
SERIAL DATA
SOURCE
BIAS TEE
CONNECTION
TO PORT 1
SERIAL DATA
SOURCE
V
5.0V
S
1nF
1nF
T1
T2
PORT 2
MINI-
CIRCUITS
TC2-1T
1nF
1nF
MINI-
CIRCUITS
TC4-1W
16 15 14 13 12 11 10
9
PORT 2
MINI-
16 15 14 13 12 11 10
9
0Ω
CIRCUITS
TC2-1T
AD8370
200Ω
AD8370
1
2
3
4
5
6
7
8
1
2
3
4
5
6
7
8
1nF
1nF
1nF
1nF
1nF
V
5.0V
V 5.0V
S
S
1µF
1nF
1µF
1nF
1nF
Figure 65. PSRR Adm Test Setup
Figure 66. PSRR As Test Setup
TEKTRONIX TDS5104
DPO OSCILLOSCOPE
50Ω
AUX IN INPUT
50Ω
INPUT
50Ω
50Ω
HP8133A
3GHz PULSE
GENERATOR
INPUT INPUT
3dB
ATTEN
TRIG
6dB
SPLITTER
SERIAL DATA
SOURCE
OUT
V
5.0V
S
475Ω
3dB
2dB
ATTEN
ATTEN
52.3Ω
16 15 14 13 12 11 10
9
3dB
ATTEN
200Ω
AD8370
6dB
SPLITTER
OUT
1
2
3
4
5
6
7
8
475Ω
2dB
ATTEN
3dB
ATTEN
52.3Ω
V
5.0V
S
1µF
1nF
1µF
1nF
1nF
V
5.0V
S
Figure 67. DC Pulse Response and Overdrive Recovery Test Setup
Rev. B | Page 26 of 28
Data Sheet
AD8370
AGILENT 8648D
SIGNAL
GENERATOR
TEKTRONIX
TDS5104 DPO
OSCILLOSCOPE
SERIAL DATA
SOURCE
TEKTRONIX
P6205 ACTIVE
FET PROBE
RF OUT
50Ω INPUT
V
5.0V
S
475Ω
105Ω
1nF
T1
1nF
T2
50Ω INPUT
MINI-
CIRCUITS
JTX-2-10T
MINI-
CIRCUITS
TC4-1W
16 15 14 13 12 11 10
9
0Ω
AD8370
1
2
3
4
5
6
7
8
475Ω
1nF
1nF
V
5.0V
V 5.0V
S
S
1µF
1nF
1µF
1nF
1nF
Figure 68. Gain Step Time Domain Response Test Setup
AGILENT 8648D
SIGNAL
GENERATOR
TEKTRONIX
TDS5104 DPO
OSCILLOSCOPE
SERIAL DATA
SOURCE
RF OUT
10MHz REF OUT
V
5.0V
S
475Ω
105Ω
1nF
1nF
T1
T2
50Ω INPUT
MINI-
CIRCUITS
JTX-2-10T
MINI-
CIRCUITS
TC4-1W
16 15 14 13 12 11 10
9
0Ω
AD8370
1
2
3
4
5
6
7
8
475Ω
1nF
1nF
TEKTRONIX
P6205 ACTIVE
FET PROBE
10MHz IN
OUTPUT
50Ω INPUT
V
5.0V
V 5.0V
S
AGILENT 33250A
S
FUNCTION/ARBITRARY
WAVEFORM
1µF
1nF
1µF
1nF
GENERATOR
52.3Ω
1nF
Figure 69. PWUP Response Time Domain Test Setup
Rev. B | Page 27 of 28
AD8370
Data Sheet
OUTLINE DIMENSIONS
5.10
5.00
4.90
9
8
16
3.05
3.00 SQ
2.95
4.50
4.40
4.30
EXPOSED
PAD
(Pins Up)
TOP
VIEW
6.40
BSC
1
BOTTOM VIEW
0.65 BSC
1.05
1.00
0.80
1.20 MAX
8°
0°
0.20
0.09
0.15
0.05
0.30
0.19
0.75
0.60
0.45
SEATING
PLANE
COPLANARITY
0.10
COMPLIANT TO JEDEC STANDARDS MO-153-ABT
Figure 70. 16-Lead Thin Shrink Small Outline Package with Exposed Pad [TSSOP_EP]
(RE-16-2)
Dimensions shown in millimeters
ORDERING GUIDE
Model1
Temperature Range
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
Package Description
Package Option
AD8370ARE-REEL7
AD8370AREZ
16-lead TSSOP_EP, 7” Tape and Reel
16-lead TSSOP_EP, Tube
RE-16-2
RE-16-2
RE-16-2
AD8370AREZ-RL7
AD8370-EVALZ
16-lead TSSOP_EP, 7” Tape and Reel
Evaluation Board
1 Z = RoHS Compliant Part.
© 2005-2011 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D03692–0–12/11(B)
Rev. B | Page 28 of 28
相关型号：
AD8372ACPZ-R7
SPECIALTY ANALOG CIRCUIT, QCC32, 5 X 5 MM, ROHS COMPLIANT, MO-220VHHD-2, LFCSP-32
ROCHESTER
©2020 ICPDF网 联系我们和版权申明