AD8555ACP-R2 [ADI]
Zero-Drift, Digitally Programmable Sensor Signal Amplifier; 零漂移，数字式可编程传感器信号放大器
| 型号： | AD8555ACP-R2 |
| 厂家： | 
ADI |
| 描述： | Zero-Drift, Digitally Programmable Sensor Signal Amplifier |
| 文件： | 总28页 (文件大小：503K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Zero-Drift, Digitally Programmable
Sensor Signal Amplifier
AD8555
FUNCTIONAL BLOCK DIAGRAM
FEATURES
VDD
Very low offset voltage: 10 µV maximum over temperature
Very low input offset voltage drift: 60 nV/°C maximum
High CMRR: 96 dB minimum
Digitally programmable gain and output offset voltage
Single-wire serial interface
Open and short wire fault detection
Low-pass filtering
Stable with any capacitive load
Externally programmable output clamp voltage for driving
low voltage ADCs
LFCSP-16 and SOIC-8 packages
2.7 V to 5.5 V operation
VCLAMP
VDD
A5
P3
VNEG
R4
R1
R6
VDD
A3
A1
VSS
VDD
VSS
P1
R3
P2
R2
RF
VOUT
A4
VDD
A2
VSS
R7
FILT/
DIGOUT
VSS
R5
VPOS
P4
VDD
DAC
VSS
VSS
−40°C to +125°C operation
APPLICATIONS
Automotive sensors
Pressure and position sensors
Thermocouple amplifiers
Industrial weigh scales
Precision current sensing
Strain gages
Figure 1.
In addition to extremely low input offset voltage and input off-
set voltage drift and very high dc and ac CMRR, the AD8555
also includes a pull-up current source at the input pins and a
pull-down current source at the VCLAMP pin. This allows open
wire and shorted wire fault detection. A low-pass filter function
is implemented via a single low cost external capacitor. Output
clamping set via an external reference voltage allows the
AD8555 to drive lower voltage ADCs safely and accurately.
GENERAL DESCRIPTION
The AD8555 is a zero-drift, sensor signal amplifier with digi-
tally programmable gain and output offset. Designed to easily
and accurately convert variable pressure sensor and strain
bridge outputs to a well-defined output voltage range, the
AD8555 also accurately amplifies many other differential or
single-ended sensor outputs. The AD8555 uses the ADI pat-
ented low noise auto-zero and DigiTrim® technologies to create
an incredibly accurate and flexible signal processing solution in
a very compact footprint.
When used in conjunction with an ADC referenced to the same
supply, the system accuracy becomes immune to normal supply
voltage variations. Output offset voltage can be adjusted with a
resolution of better than 0.4% of the difference between VDD
and VSS. A lockout trim after gain and offset adjustment further
ensures field reliability.
Gain is digitally programmable in a wide range from 70 to 1,280
through a serial data interface. Gain adjustment can be fully
simulated in-circuit and then permanently programmed with
proven and reliable poly-fuse technology. Output offset voltage
is also digitally programmable and is ratiometric to the supply
voltage.
The AD8555AR is fully specified over the extended industrial
temperature range of −40°C to +125°C. Operating from
single-supply voltages of 2.7 V to 5.5 V, the AD8555 is offered in
the narrow 8-lead SOIC package and the 4 mm × 4 mm
16-lead LFCSP.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
Fax: 781.326.8703
www.analog.com
© 2004 Analog Devices, Inc. All rights reserved.
AD8555
TABLE OF CONTENTS
Electrical Specifications................................................................... 3
Device Programming................................................................. 19
Filtering Function....................................................................... 25
Driving Capacitive Loads.......................................................... 25
RF Interference ........................................................................... 26
Single-Supply Data Acquisition System .................................. 26
Using the AD8555 with Capacitive Sensors ........................... 27
Outline Dimensions....................................................................... 28
Ordering Guide .......................................................................... 28
Absolute Maximum Ratings............................................................ 7
Pin Configurations and Function Descriptions ........................... 8
Typical Performance Characteristics ............................................. 9
Theory of Operation ...................................................................... 17
Gain Values.................................................................................. 18
Open Wire Fault Detection....................................................... 19
Shorted Wire Fault Detection ................................................... 19
Floating VPOS, VNEG, or VCLAMP Fault Detection........... 19
REVISION HISTORY
4/04—Revision 0: Initial Version
Rev. 0 | Page 2 of 28
AD8555
ELECTRICAL SPECIFICATIONS
At VDD = 5.0 V, VSS = 0.0 V, VCM = 2.5 V, VO = 2.5 V, −40°C ≤ TA ≤ +125°C, unless otherwise specified.
Table 1.
Parameter
Symbol Conditions
Min Typ
Max Unit
INPUT STAGE
Input Offset Voltage
Input Offset Voltage Drift
Input Bias Current
VOS
TCVOS
IB
2
25
16
10
65
22
25
1
µV
nV/°C
nA
nA
nA
TA = 25°C
TA = 25°C
12
Input Offset Current
IOS
0.2
1.5
3.8
nA
V
dB
dB
ppm
ppm
%
Input Voltage Range
Common-Mode Rejection Ratio
0.6
80
96
CMRR
VCM = 0.9 V to 3.6 V, AV = 70
VCM = 0.9 V to 3.6 V, AV = 1,280
VO = 0.2 V to 3.4 V
92
112
20
1000
0.35
0.5
15
Linearity
VO = 0.2 V to 4.8 V
Differential Gain Accuracy
Differential Gain Temperature Coefficient
Second Stage Gain = 17.5 to 100
Second Stage Gain = 140 to 200
Second Stage Gain = 17.5 to 100
Second Stage Gain = 140 to 200
1.6
2.5
40
%
ppm/°C
ppm/°C
40
100
RF
14
18
22
kΩ
RF Temperature Coefficient
DAC
700
ppm/°C
Accuracy
AV = 70, Offset Codes = 8 to 248
AV = 70, Offset Codes = 8 to 248
AV = 70, Offset Codes = 8 to 248
0.7
50
5
0.8
%
ppm
mV
Ratiometricity
Output Offset
Temperature Coefficient
VCLAMP
35
15
3.3
ppm FS/°C
Input Bias Current
TA = 25°C, VCLAMP = 5 V
200
500
nA
nA
V
Input Voltage Range
OUTPUT BUFFER STAGE
Buffer Offset
Short-Circuit Current
Output Voltage, Low
Output Voltage, High
POWER SUPPLY
1.25
4.94
7
15
10
30
mV
mA
mV
V
ISC
VOL
VOH
5
RL = 10 kΩ to 5 V
RL = 10 kΩ to 0 V
4.94
Supply Current
ISY
VO = 2.5 V, VPOS = VNEG = 2.5V,
VDAC Code = 128
AV = 70
2.0
2.5
mA
dB
Power Supply Rejection Ratio
DYNAMIC PERFORMANCE
Gain Bandwidth Product
PSRR
109
125
GBP
First Gain Stage, TA = 25°C
Second Gain Stage, TA = 25°C
Output Buffer Stage
AV = 70, RL = 10 kΩ, C L = 100 pF
To 0.1%, AV = 70, 4 V Output Step
2
8
1.5
1.2
8
MHz
MHz
MHz
V/µs
µs
Output Buffer Slew Rate
Settling Time
SR
ts
NOISE PERFORMANCE
Input Referred Noise
Low Frequency Noise
Total Harmonic Distortion
TA = 25°C, f = 1 kHz
f = 0.1 Hz to 10 Hz
VIN = 16.75 mV rms, f = 1 kHz, AV = 100
32
0.5
−100
nV/√Hz
µV p-p
dB
en p-p
THD
Rev. 0 | Page 3 of 28
AD8555
Parameter
Symbol Conditions
Min Typ
Max Unit
DIGITAL INTERFACE
Input Current
DIGIN Pulse Width to Load 0
DIGIN Pulse Width to Load 1
Time between Pulses at DIGIN
DIGIN Low
DIGIN High
DIGOUT Logic 0
DIGOUT Logic 1
2
0.05
50
µA
tw0
tw1
tws
TA = 25°C
TA = 25°C
TA = 25°C
TA = 25°C
TA = 25°C
TA = 25°C
TA = 25°C
10
µs
µs
µs
V
V
V
10
1
1
4
4
V
Rev. 0 | Page 4 of 28
AD8555
At VDD = 2.7 V, VSS = 0.0 V, VCM = 1.35 V, VO = 1.35 V, −40°C ≤ TA ≤ +125°C, unless otherwise specified.
Table 2.
Parameter
Symbol
Conditions
Min
Typ
Max Unit
INPUT STAGE
Input Offset Voltage
Input Offset Voltage Drift
Input Bias Current
Input Offset Current
VOS
TCVOS
IB
2
10
60
µV
nV/°C
nA
25
16
0.2
TA = 25°C
TA = 25°C
12
IOS
1
nA
1.5
1.6
nA
V
dB
dB
ppm
ppm
%
Input Voltage Range
Common-Mode Rejection Ratio
0.5
80
96
CMRR
VCM = 0.9 V to 1.3 V, AV = 70
VCM = 0.9 V to 1.3 V, AV = 1,280
VO = 0.2 V to 3.4 V
92
112
20
1000
0.35
0.5
15
Linearity
VO = 0.2 V to 4.8 V
Differential Gain Accuracy
Differential Gain Temperature Coefficient
Second Stage Gain = 17.5 to 100
Second Stage Gain = 140 to 200
Second Stage Gain = 17.5 to 100
Second Stage Gain = 140 to 200
%
ppm/°C
40
ppm/°C
RF
14
18
22
kΩ
RF Temperature Coefficient
DAC
700
ppm/°C
Accuracy
AV = 70, Offset Codes = 8 to 248
AV = 70, Offset Codes = 8 to 248
AV = 70, Offset Codes = 8 to 248
0.7
50
5
%
ppm
mV
Ratiometricity
Output Offset
Temperature Coefficient
VCLAMP
35
3.3
ppm FS/°C
Input Bias Current
TA = 25°C, VCLAMP = 2.7 V
200
500
nA
nA
V
Input Voltage Range
OUTPUT BUFFER STAGE
Buffer Offset
Short-Circuit Current
Output Voltage, Low
Output Voltage, High
POWER SUPPLY
1.25
2.64
7
15
9.5
30
mV
mA
mV
V
ISC
VOL
VOH
4.5
RL = 10 kΩ to 5 V
RL = 10 kΩ to 0 V
2.64
Supply Current
ISY
VO = 1.35 V, VPOS = VNEG = 1.35 V,
VDAC Code = 128
AV = 70
2.0
mA
dB
Power Supply Rejection Ratio
DYNAMIC PERFORMANCE
Gain Bandwidth Product
PSRR
109
125
GBP
First Gain Stage, TA = 25°C
Second Gain Stage, TA = 25°C
Output Buffer Stage
AV = 70, RL = 10 kΩ, CL = 100 pF
To 0.1%, AV = 70, 4 V Output Step
2
8
1.5
1.2
8
MHz
MHz
MHz
V/µs
µs
Output Buffer Slew Rate
Settling Time
SR
ts
NOISE PERFORMANCE
Input Referred Noise
Low Frequency Noise
Total Harmonic Distortion
TA = 25°C, f = 1 kHz
f = 0.1 Hz to 10 Hz
VIN = 16.75 mV rms, f = 1 kHz, AV = 100
32
0.3
−100
nV/√Hz
µV p-p
dB
en p-p
THD
Rev. 0 | Page 5 of 28
AD8555
Parameter
Symbol
Conditions
Min
Typ
Max Unit
DIGITAL INTERFACE
Input Current
2
µA
DIGIN Pulse Width to Load 0
DIGIN Pulse Width to Load 1
Time between Pulses at DIGIN
tw0
tw1
tws
TA = 25°C
TA = 25°C
TA = 25°C
0.05
50
10
10
µs
µs
µs
Rev. 0 | Page 6 of 28
AD8555
ABSOLUTE MAXIMUM RATINGS
Table 3.
Parameter
Supply Voltage
Input Voltage
Differential Input Voltage1
Table 4.
2
Rating
Package Type
8-Lead SOIC (R)
16-Lead LFCSP (CP)
θJA
θJC
Unit
°C/W
°C/W
6 V
158
44
43
31.5
VSS − 0.3 V to VDD + 0.3 V
5.0 V
Output Short-Circuit
Indefinite
Duration to VSS or VDD
Storage Temperature Range
Operating Temperature Range
Junction Temperature Range
Lead Temperature Range
(Soldering, 10 sec)
−65°C to +150°C
−40°C to +125°C
−65°C to +150°C
300°C
1 Differential input voltage is limited to 5.0 V or the supply voltage, which-
ever is less.
2 θJA is specified for the worst-case conditions, i.e., θJA is specified for device
soldered in circuit board for SOIC and LFCSP packages.
Rev. 0 | Page 7 of 28
AD8555
PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS
VDD
FILT/DIGOUT
DIGIN
1
2
3
4
8
7
6
5
VSS
AD8555
TOP VIEW
(Not to Scale)
VOUT
VCLAMP
VPOS
VNEG
12 VOUT
11 NC
NC
FILT/DIGOUT
NC
1
2
3
4
PIN 1
INDICATOR
AD8555
Figure 2. 8-Lead SOIC (Not Drawn to Scale)
10 VCLAMP
TOP VIEW
DIGIN
9 NC
NC = NO CONNECT
Figure 3. 16-Lead LFCSP (Not Drawn to Scale)
Table 5. Pin Configuration
SOIC
LFCSP
Mnemonic
Pin No. Mnemonic
Pin No.
N/A
2
Description
1
2
VDD
FILT/DIGOUT
N/A
FILTDIGOUT
Positive Supply Voltage.
Unbuffered Amplifier Output In Series with a Resistor RF. Adding a capacitor
between FILT and VDD or VSS implements a low-pass filtering function. In
read mode, this pin functions as a digital output.
3
4
5
6
7
DIGIN
VNEG
VPOS
VCLAMP
VOUT
4
6
8
10
12
DIGIN
VNEG
VPOS
VCLAMP
VOUT
Digital Input.
Negative Amplifier Input (Inverting Input).
Positive Amplifier Input (Noninverting Input).
Set Clamp Voltage at Output.
Buffered Amplifier Output. Buffered version of the signal at the FILT/DIGOUT
pin. In read mode, VOUT is a buffered digital output.
8
VSS
N/A
N/A
N/A
N/A
13, 14
15, 16
1, 3, 5, 7, 9, 11
N/A
Negative Supply Voltage.
Negative Supply Voltage.
Positive Supply Voltage.
Do Not Connect.
N/A
N/A
N/A
DVSS, AVSS
DVDD, AVDD
NC
Rev. 0 | Page 8 of 28
AD8555
TYPICAL PERFORMANCE CHARACTERISTICS
40
35
30
25
20
15
10
5
V
= 5V
S
V
= 5V
S
40
30
20
10
0
0
–9
–6
–3
0
3
6
9
0
12.5
25.0
37.5
V (nV/°C)
OS
50.0
62.5
75.0
V
(µV)
OS
T
C
Figure 4. Input Offset Voltage Distribution
Figure 7. TCVOS @ VS = 5 V
1.0
50
45
40
35
30
25
20
15
10
5
0.5
0
–0.5
–1.0
–1.5
–2.0
–2.5
–3.0
–3.5
–4.0
0
0
0.5
1.0
1.5
2.0
V
2.5
(V)
3.0
3.5
4.0
4.5
0
12.5
25.0
37.5
V (nV/°C)
OS
50.0
62.5
75.0
= 5V
T
CM
C
Figure 5. Input Offset Voltage vs. Common-Mode Voltage
Figure 8. TCVOS @ VS = 2.7 V
10
10.0
7.5
V
S
8
6
5.0
4
2.5
VOUT = 0.3V
2
0
0
–2
–4
–6
–8
–10
–2.5
–5.0
–7.5
–10.0
VOUT = 4.7V
–50
–25
0
25
50
75
100
125
150
–50
–25
0
25
50
75
100
125
150
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 6. Input Offset Voltage vs. Temperature
Figure 9. Output Buffer Offset vs. Temperature
Rev. 0 | Page 9 of 28
AD8555
100
2.5
2.0
1.5
1.0
0.5
0
V
= 5V
V = 5.5V
S
S
10
1
–75
–25
25
75
125
175
0
1
2
3
4
5
6
TEMPERATURE (°C)
DIGITAL INPUT VOLTAGE(V)
Figure 10. Input Bias Current at VPOS, VNEG vs. Temperature
Figure 13. Digital Input Current vs. Digital Input Voltage (Pin 3)
100
10
1
1000
100
10
V
= 5V
V
= 5V
S
S
+125°C
+25°C
–40°C
0
1
2
3
4
5
0
1
2
3
4
5
6
V
(V)
VCLAMP VOLTAGE (V)
CM
Figure 11. Input Bias Current at VPOS, VNEG vs. Common-Mode Voltage
Figure 14. VCLAMP Current Over Temperature at VS = 5 V vs. VCLAMP Voltage
0.5
0.4
1000
V
= 2.7V
S
0.3
+125
0.2
+25
–40
0.1
0
100
–0.1
–0.2
–0.3
–0.4
–0.5
10
–50
–25
0
25
50
75
100
125
150
0
1
2
2.7
TEMPERATURE (°C)
VCLAMP VOLTAGE (V)
Figure 15. VCLAMP Current Over Temperature at VS = 2.7 vs. VCLAMP Voltage
Figure 12. Input Offset Current vs. Temperature
Rev. 0 | Page 10 of 28
AD8555
3
2
1
0
V
= ±2.5V
S
GAIN = 1280
120
80
0
100
1k
10k
100k
1M
0
1
2
3
4
5
6
FREQUENCY (Hz)
SUPPLY VOLTAGE (V)
Figure 16. Supply Current (ISY) vs. Supply Voltage
Figure 19. CMRR vs. Frequency
3.0
2.5
2.0
1.5
1.0
0.5
V
= 5V
S
145
135
125
115
5V
1280
2.7V
800
400
200
105 100
70
95
85
75
–75
–50
–25
0
25
50
75
100
125
150
–75
–50
–25
0
25
50
75
100
125
150
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 17. Supply Current (ISY) vs. Temperature
Figure 20. CMRR vs. Temperature at Different Gains
V
= ±2.5V
S
V
= ±2.5V
S
GAIN = 70
GAIN = 70
120
60
50
40
30
20
10
80
40
0
100
1k
10k
100k
1M
0
5
10
FREQUENCY (Hz)
FREQUENCY (kHz)
Figure 18. CMRR vs. Frequency
Figure 21. Input Voltage Noise Density vs. Frequency (0 Hz to 10 kHz)
Rev. 0 | Page 11 of 28
AD8555
V
C
= ±2.5V
= 40PF
S
GAIN = 1280
L
V
= ±2.5V
S
GAIN = 70
60
40
20
0
35
30
25
20
15
10
5
GAIN = 70
1k
10k
100k
1M
FREQUENCY (Hz)
0
250
500
FREQUENCY (kHz)
Figure 25. Closed-Loop Gain vs. Frequency Measured at Filter Pin
Figure 22. Input Voltage Noise Density vs. Frequency (0 Hz to 500 kHz)
V
= ±2.5V
S
V
= ±2.5V
GAIN = 1280
S
GAIN = 1000
0.6
0.4
60
40
20
0
0.2
GAIN = 70
0
–0.2
–0.4
–0.6
1k
10k
100k
1M
FREQUENCY (Hz)
TIME (1s/DIV)
Figure 26. Closed-Loop Gain vs. Frequency Measured at Output Pin
Figure 23. Low Frequency Input Voltage Noise (0.1 Hz to 10 Hz)
V
= ±2.5V
S
8
4
0
–4
–8
1k
10k
100k
1M
10M
FREQUENCY (Hz)
Figure 27. Output Buffer Gain vs. Frequency
Figure 24. Low Frequency Input Voltage Noise (0.1 Hz to 10 Hz)
Rev. 0 | Page 12 of 28
AD8555
60
50
40
30
20
10
0
15
12
9
V
= ±2.5V
S
R
R
= 0
S
S
SINK 5V
C
= 1nF
L
OUTPUT
BUFFER
SINK 2.7V
6
3
0
R
= 10
S
–3
–6
–9
–12
–15
R
= 20
S
R
= 50
S
SOURCE 5V
SOURCE 2.7V
R
= 100
1.0
S
0.1
10.0
100.0
–75 –50 –25
0
25
50
75
100 125 150 175
LOAD CAPACITANCE (nF)
TEMPERATURE (°C)
Figure 28. Output Buffer Positive Overshoot
Figure 31. Output Short Circuit vs. Temperature
60
50
40
30
20
10
0
V
= ±2.5V
S
SUPPLY VOLTAGE
R
S
R
= 0
C
S
L
4
2
0
3
2
1
0
R
= 10
S
R
= 20
S
V
OUT
R
= 50
S
R
= 100
S
0.1
1.0
10.0
100.0
TIME (100µs/DIV)
LOAD CAPACITANCE (nF)
Figure 29. Output Buffer Negative Overshoot
Figure 32. Power-On Response at 25°C
1.000
0.100
0.010
0.001
V
= ±2.5V
S
6
5
4
3
2
1
0
SUPPLY VOLTAGE
SOURCE
SINK
V
OUT
0.01
0.10
1.00
10.0
TIME (100µs/DIV)
LOAD CURRENT (mA)
Figure 30. Output Voltage to Supply Rail vs. Load Current
Figure 33. Power-On Response at 125°C
Rev. 0 | Page 13 of 28
AD8555
T
V
= ±2.5V
S
GAIN = 70
C
F
= 0.1µF
= 10kHz
L
6
5
4
3
2
1
0
IN
SUPPLY VOLTAGE
2
V
OUT
TIME (100µs/DIV)
TIME (100µs/DIV)
Figure 34. Power-On Response at −40°C
Figure 37. Small Signal Response
150
145
140
135
130
125
120
115
110
105
V
= 2.7V TO 5.5V
S
T
V
= ±2.5V
S
GAIN = 70
C
= 100pF
= 1kHz
L
F
IN
2
100
–75
–50
–25
0
25
50
75
100
125
150
TEMPERATURE (°C)
TIME (100µs/DIV)
Figure 35. PSRR vs. Temperature
Figure 38. Small Signal Response
140
120
100
80
T
V
= ±2.5V
S
GAIN = 70
C
= 100pF
L
2
60
40
20
0
0.01
0.1
1
10
100
FREQUENCY (kHz)
TIME (10µs/DIV)
Figure 36. PSRR vs. Frequency
Figure 39. Large Signal Response
Rev. 0 | Page 14 of 28
AD8555
T
V
= ±2.5V
S
GAIN = 70
V
IN
C
= 0.05µF
L
0V
2
V
OUT
0V
TIME (10µs/DIV)
Figure 40. Large Signal Response
Figure 43. Positive Overload Recovery (Gain = 70)
1k
V
A
= ±2.5V
SY
= 70
V
0V
V
100
10
1
IN
0V
–2.5V
0.1
1
10
100
1M
FREQUENCY (kHz)
Figure 41. Output Impedance vs. Frequency
Figure 44. Negative Overload Recovery (Gain = 1280)
0V
V
V
IN
0V
IN
0V
V
OUT
V
OUT
0V
Figure 42. Negative Overload Recovery (Gain = 70)
Figure 45. Positive Overload Recovery (Gain = 1280)
Rev. 0 | Page 15 of 28
AD8555
1.00
0.50
GAIN = 70
OFFSET = 128
V
= ±2.5V
S
V
= ±2.5V
S
0.20
0.10
0.05
+V
4V pp
0.1µF
1
20.5Ω
294Ω
4
5
6
7
AD8555
8
0.1µF
1kΩ 10kΩ
–V
0.02
0.01
10kΩ
OUT
20
50
100 200
500
1k
2k
5k
10k 20k
FREQUENCY (Hz)
Figure 46. Settling Time 0.1%
Figure 48. THD vs. Frequency
GAIN = 70
OFFSET = 128
V
= ±2.5V
S
+V
4V pp
0.1µF
1
20.5Ω
294Ω
4
5
6
7
AD8555
8
0.1µF
–V
10k
Ω
1k
Ω
10k
Ω
OUT
Figure 47. Settling Time 0.01%
Rev. 0 | Page 16 of 28
AD8555
THEORY OF OPERATION
VDD. The input to A5, VCLAMP, has a very high input resis-
A1, A2, R1, R2, R3, P1, and P2 form the first gain stage of the
differential amplifier. A1 and A2 are auto-zeroed op amps that
minimize input offset errors. P1 and P2 are digital potentiome-
ters, guaranteed to be monotonic. Programming P1 and P2
allows the first stage gain to be varied from 4.0 to 6.4 with 7-bit
resolution (see Table 6 and Equation 3), giving a fine gain
adjustment resolution of 0.37%. R1, R2, R3, P1, and P2 each
have a similar temperature coefficient, so the first stage gain
temperature coefficient is lower than 100 ppm/°C.
tance. It should be connected to a known voltage and not left
floating. However, the high input impedance allows the clamp
voltage to be set using a high impedance source, e.g., a potential
divider. If the maximum value of VOUT does not need to be
limited, VCLAMP should be connected to VDD.
A4 implements a rail-to-rail input and output unity-gain volt-
age buffer. The output stage of A4 is supplied from a buffered
version of VCLAMP instead of VDD, allowing the positive
swing to be limited. The maximum output current is limited
between 5 mA to 10 mA.
A3, R4, R5, R6, R7, P3, and P4 form the second gain stage of the
differential amplifier. A3 is also an auto-zeroed op amp that
minimize input offset errors. P3 and P4 are digital potentiome-
ters, allowing the second stage gain to be varied from 17.5 to
200 in eight steps (see Table 7); they allow the gain to be varied
over a wide range. R4, R5, R6, R7, P3, and P4 each have a similar
temperature coefficient, so the second stage gain temperature
coefficient is lower than 100 ppm/°C.
An 8-bit digital-to-analog converter (DAC) is used to generate a
variable offset for the amplifier output. This DAC is guaranteed
to be monotonic. To preserve the ratiometric nature of the input
signal, the DAC references are driven from VSS and VDD, and
the DAC output can swing from VSS (Code 0) to VDD (Code
255). The 8-bit resolution is equivalent to 0.39% of the differ-
ence between VDD and VSS, e.g., 19.5 mV with a 5 V supply.
The DAC output voltage (VDAC) is given approximately by
RF together with an external capacitor connected between
FILT/DIGOUT and VSS or VDD form a low-pass filter. The
filtered signal is buffered by A4 to give a low impedance output
at VOUT. RF is nominally 16 kΩ, allowing a 1 kHz low-pass
filter to be implemented by connecting a 10 nF external
capacitor between FILT/DIGOUT and VSS or between
FILT/DIGOUT and VDD. If low-pass filtering is not needed,
then the FILT/DIGOUT pin must be left floating.
Code + 0.5
⎛
⎜
⎞
⎟
VDAC ≈
(
VDD −VSS
)
+VSS
(1)
256
⎝
⎠
The temperature coefficient of VDAC is lower than
200 ppm/°C.
The amplifier output voltage (VOUT) is given by
A5 implements a voltage buffer, which provides the positive
supply to the amplifier output buffer A4. Its function is to limit
VOUT to a maximum value, useful for driving analog-to-digital
converters (ADC) operating on supply voltages lower than
VOUT = GAIN VPOS −VNEG +VDAC
( )
(2)
where GAIN is the product of the first and second stage gains.
VDD
VCLAMP
A5
VDD
P3
VNEG
R4
R1
R6
VDD
A3
A1
VSS
VDD
VSS
P1
R3
P2
R2
RF
VOUT
A4
VDD
A2
VSS
R7
FILT/
DIGOUT
VSS
R5
VPOS
P4
VDD
DAC
VSS
VSS
Figure 49. AD8555 Functional Schematic
Rev. 0 | Page 17 of 28
AD8555
GAIN VALUES
Table 6. First Stage Gain vs. Gain Code
First Stage
Gain Code
First Stage
Gain Code
First Stage
Gain Code
First Stage
Gain Code
First Stage Gain
4.000
4.015
4.030
4.045
4.060
4.075
4.090
4.105
4.120
4.135
4.151
4.166
4.182
4.197
4.213
4.228
4.244
4.260
4.276
4.291
4.307
4.323
4.339
4.355
4.372
4.388
4.404
4.420
4.437
4.453
4.470
4.486
First Stage Gain
4.503
4.520
4.536
4.553
4.570
4.587
4.604
4.621
4.638
4.655
4.673
4.690
4.707
4.725
4.742
4.760
4.778
4.795
4.813
4.831
4.849
4.867
4.885
4.903
4.921
4.939
4.958
4.976
4.995
5.013
5.032
5.050
First Stage Gain
5.069
5.088
5.107
5.126
5.145
5.164
5.183
5.202
5.221
5.241
5.260
5.280
5.299
5.319
5.339
5.358
5.378
5.398
5.418
5.438
5.458
5.479
5.499
5.519
5.540
5.560
5.581
5.602
5.622
5.643
5.664
5.685
First Stage Gain
5.706
5.727
5.749
5.770
5.791
5.813
5.834
5.856
5.878
5.900
5.921
5.943
5.965
5.988
6.010
6.032
6.054
6.077
6.099
6.122
6.145
6.167
6.190
6.213
6.236
6.259
6.283
6.306
6.329
6.353
6.376
6.400
0
1
2
3
4
5
6
7
32
33
34
35
36
37
38
39
40
41
42
43
44
45
46
47
48
49
50
51
52
53
54
55
56
57
58
59
60
61
62
63
64
65
66
67
68
69
70
71
72
73
74
75
76
77
78
79
80
81
82
83
84
85
86
87
88
89
90
91
92
93
94
95
96
97
98
99
100
101
102
103
104
105
106
107
108
109
110
111
112
113
114
115
116
117
118
119
120
121
122
123
124
125
126
127
8
9
10
11
12
13
14
15
16
17
18
19
20
21
22
23
24
25
26
27
28
29
30
31
Table 7. Second Stage Gain and Gain Ranges vs. Gain Code
Code
127
⎛
⎜
⎞
⎟
6.4
4
⎝
⎠
⎛
⎜
⎝
⎞
⎟
⎠
Second
Stage Gain
Code
Second
Stage
Gain
Minimum
Combined
Gain
Maximum
Combined
Gain
GAIN1 ≈ 4×
(3)
0
1
2
3
4
5
6
7
17.5
25
35
50
70
100
140
200
70
112
160
224
320
448
640
896
1280
100
140
200
280
400
560
800
Rev. 0 | Page 18 of 28
AD8555
OPEN WIRE FAULT DETECTION
FLOATING VPOS, VNEG, OR VCLAMP FAULT
DETECTION
The inputs to A1 and A2, VNEG and VPOS, each have a com-
parator to detect whether VNEG or VPOS exceeds a threshold
voltage, nominally VDD − 1.1 V. If (VNEG > VDD − 1.1 V) or
(VPOS > VDD − 1.1 V), VOUT is clamped to VSS. The output
current limit circuit is disabled in this mode, but the maximum
sink current is approximately 50 mA when VDD = 5 V. The
inputs to A1 and A2, VNEG and VPOS, are also pulled up to
VDD by currents IP1 and IP2. These are both nominally 18 nA
and matched to within 5 nA. If the inputs to A1 or A2 are acci-
dentally left floating, e.g., an open wire fault, IP1 and IP2 pull
them to VDD, which would cause VOUT to swing to VSS, al-
lowing this fault to be detected. It is not possible to disable IP1
and IP2, nor the clamping of VOUT to VSS, when VNEG or
VPOS approaches VDD.
A floating fault condition at the VPOS, VNEG, or VCLAMP
pins is detected by using a low current to pull a floating input
into an error voltage range, which is defined in the previous
section. In this way, the VOUT pin is shorted to VSS when a
floating input is detected. Table 9 lists the currents used.
Table 9. Floating Fault Detection at VPOS, VNEG, and
VCLAMP
Pin
Typical Current
16 nA pull-up
16 nA pull-up
Goal of Current
VPOS
VNEG
Pull VPOS above VINH
Pull VNEG above VINH
Pull VCLAMP below VCLL
VCLAMP 0.2 µA pull-down
DEVICE PROGRAMMING
Digital Interface
SHORTED WIRE FAULT DETECTION
The AD8555 provides fault detection, in the case where VPOS,
VNEG, or VCLAMP shorts to VDD and VSS. Figure 50 shows
the voltage regions at VPOS, VNEG, and VCLAMP that trigger
an error condition. When an error condition occurs, the VOUT
pin is shorted to VSS. Table 8 lists the voltage levels shown in
Figure 50.
The digital interface allows the first stage gain, second stage
gain, and output offset to be adjusted and allows desired values
for these parameters to be permanently stored by selectively
blowing polysilicon fuses. To minimize pin count and board
space, a single-wire digital interface is used. The digital input
pin, DIGIN, has hysteresis to minimize the possibility of inad-
vertent triggering with slow signals. It also has a pull-down
current sink to allow it to be left floating when programming is
not being performed. The pull-down ensures inactive status of
the digital input by forcing a dc low voltage on DIGIN.
VPOS
VNEG
VCLAMP
VDD
VDD
VDD
ERROR
ERROR
VINH
VINH
NORMAL
ERROR
NORMAL
ERROR
NORMAL
ERROR
A short pulse at DIGIN from low to high and back to low again,
e.g., between 50 ns and 10 µs long, loads a 0 into a shift register.
A long pulse at DIGIN, e.g., 50 µs or longer, loads a 1 into the
shift register. The time between pulses should be at least 10 µs.
Assuming VSS = 0 V, voltages at DIGIN between VSS and 0.2 ×
VDD are recognized as a low, and voltages at DIGIN between
0.8 × VDD and VDD are recognized as a high. A timing dia-
gram example showing the waveform for entering code 010011
into the shift register is shown in Figure 51.
VCLL
VSS
VINL
VSS
VINL
VSS
Figure 50. Voltage Regions at VPOS, VNEG, and VCLAMP
That Trigger a Fault Condition
Table 8. Typical VINL, VINH, and VCLL Values (VDD = 5 V)
Voltage Typical Min
Typical Max
Purpose
VINH
VINL
VCLL
3.9 V
0.195 V
1 V
4.2 V
Short to VDD
Fault Detection
Short to VSS
Fault Detection
0.55 V
1.2 V
Short to VSS
Fault Detection
Rev. 0 | Page 19 of 28
AD8555
tW1
tWS
tW1
tWS
tWS
tW0
tWS
tW0
tW0
tWS
tW1
WAVEFORM
CODE
0
1
0
0
1
1
Figure 51. Timing Diagram for Code 010011
Table 10. Timing Specifications
Timing Parameter
Description
Specification
Between 50 ns and 10 µs
≥50 µs
tw0
tw1
tws
Pulse Width for Loading 0 into Shift Register
Pulse Width for Loading 1 into Shift Register
Width between Pulses
≥10 µs
Table 11. 38-Bit Serial Word Format
Field No.
Bits
Description
Field 0
Bits 0 to 11
Bits 12 to 13
12-Bit Start of Packet 1000 0000 0001
2-Bit Function
Field 1
00: Change Sense Current
01: Simulate Parameter Value
10: Program Parameter Value
11: Read Parameter Value
2-Bit Parameter
Field 2
Bits 14 to 15
00: Second Stage Gain Code
01: First Stage Gain Code
10: Output Offset Code
11: Other Functions
Field 3
Field 4
Bits 16 to 17
Bits 18 to 25
2-Bit Dummy 10
8-Bit Value
Parameter 00 (Second Stage Gain Code): 3 LSBs Used
Parameter 01 (First Stage Gain Code): 7 LSBs Used
Parameter 10 (Output Offset Code): All 8 Bits Used
Parameter 11 (Other Functions)
Bit 0 (LSB): Master Fuse
Bit 1: Fuse for Production Test at Analog Devices
Bit 2: Parity Fuse
Field 5
Bits 26 to 37
12-Bit End of Packet 0111 1111 1110
A 38-bit serial word is used, divided into 6 fields. Assuming
each bit can be loaded in 60 µs, the 38-bit serial word transfers
in 2.3 ms. Table 11 summarizes the word format.
Fields 0 and 5 are the start of packet and end of packet field,
respectively. Matching the start of packet field with 1000 0000
0001 and the end of packet field with 0111 1111 1110 ensures
that the serial word is valid and enables decoding of the other
fields. Field 3 breaks up the data and ensures that no data com-
bination can inadvertently trigger the start of packet and end of
packet fields. Field 0 should be written first and Field 5 written
last. Within each field, the MSB must be written first and the
LSB written last. The shift register features power-on reset to
minimize the risk of inadvertent programming; power-on reset
occurs when VDD is between 0.7 V and 2.2 V.
Rev. 0 | Page 20 of 28
AD8555
Initial State
fuse to blow. There is no need to measure the supply current
during programming; the best way to verify correct program-
ming is to use the read mode to read back the programmed
values and to remeasure the gain and offset to verify these
values. Programmed fuses have no effect on the gain and output
offset until the master fuse is blown; after blowing the master
fuse, the gain and output offset are determined solely by the
blown fuses and the simulation mode is permanently deacti-
vated.
Initially, all the polysilicon fuses are intact. Each parameter has
the value 0 assigned (see Table 12).
Table 12. Initial State before Programming
Second Stage Gain Code = 0
First Stage Gain Code = 0
Output Offset Code = 0
Master Fuse = 0
Second Stage Gain = 17.5
First Stage Gain = 4.0
Output Offset = VSS
Master Fuse Not Blown
When power is applied to a device, parameter values are taken
either from internal registers if the master fuse is not blown or
from the polysilicon fuses if the master fuse is blown.
Programmed values have no effect until the master fuse is
blown. The internal registers feature power-on reset so that
unprogrammed devices enter a known state after power-up;
power-on reset occurs when VDD is between 0.7 V and 2.2 V.
Parameters are programmed by setting Field 1 to 10, selecting
the desired parameter in Field 2, and selecting a single bit with
the value 1 in Field 4.
As an example, suppose the user wants to permanently set the
second stage gain to 50. Parameter 00 needs to have the value
0000 0011 assigned. Two bits have the value 1, so two fuses need
to be blown. Since only one fuse can be blown at a time, the
code 1000 0000 0001 10 00 10 0000 0010 0111 1111 1110 can be
used to blow one fuse. The MOS switch that blows the fuse
closes when the complete packet is recognized and opens when
the start-of-packet, dummy, or end-of-packet fields are no
longer valid. After 1 ms, the second code 1000 0000 0001 10 00
10 0000 0001 0111 1111 1110 can be entered to blow the second
fuse.
Simulation Mode
The simulation mode allows any parameter to be changed tem-
porarily. These changes are retained until the simulated value is
reprogrammed, the power is removed, or the master fuse is
blown. Parameters are simulated by setting Field 1 to 01, select-
ing the desired parameter in Field 2, and the desired value for
the parameter in Field 4. Note that a value of 11 for Field 2 is
ignored during the simulation mode. Examples of temporary
settings follow:
To set the first stage gain permanently to a nominal value of
4.151, Parameter 01 needs to have the value 000 1011 assigned.
Three fuses need to be blown, and the following codes can be
used, with a 1 ms delay after each code:
• By setting the second stage gain code (Parameter 00) to 011
and the second stage gain to 50, 1000 0000 0001 01 00 10
0000 0011 0111 1111 1110 is the result.
1000 0000 0001 10 01 10 0000 1000 0111 1111 1110
1000 0000 0001 10 01 10 0000 0010 0111 1111 1110
1000 0000 0001 10 01 10 0000 0001 0111 1111 1110
• By setting the first stage gain code (Parameter 01) to 000 1011
and the first stage gain to 4.166, 1000 0000 0001 01 01 10
0000 1011 0111 1111 1110 is the result.
To set the output offset permanently to a nominal value of
1.260 V when VDD = 5 V and VSS = 0 V, Parameter 10 needs to
have the value 0100 0000 assigned. One fuse needs to be blown,
and the following code can be used: 1000 0000 0001 10 10 10
0100 0000 0111 1111 1110.
A first stage gain of 4.166 with a second stage gain of 50 gives
a total gain of 208.3. This gain has a maximum tolerance of
2.5%.
• Set the output offset code (Parameter 10) to 0100 0000 and
the output offset to 1.260 V when VDD = 5 V and VSS = 0 V.
This output offset has a maximum tolerance of 0.8%: 1000
0000 0001 01 10 10 0100 0000 0111 1111 1110.
Finally, to blow the master fuse to deactivate the simulation
mode and prevent further programming, the code 1000 0000
0001 10 11 10 0000 0001 0111 1111 1110 can be used.
Programming Mode
There are a total of 20 programmable fuses. Since each fuse
requires 1 ms to blow, and each serial word can be loaded in
2.3 ms, the maximum time needed to program the fuses can be as
low as 66 ms.
Intact fuses give a bit value of 0. Bits with a desired value of 1
need to have the associated fuse blown. Since a relatively large
current is needed to blow a fuse, only one fuse can be reliably
blown at a time. Thus, a given parameter value may need several
38-bit words to allow reliable programming. A 5.5 V supply is
required when blowing fuses to minimize the on resistance of
the internal MOS switches that blow the fuse. The power supply
must be able to deliver 250 mA of current, and at least 0.1 µF of
decoupling capacitance is needed across the power pins of the
device. A minimum period of 1 ms should be allowed for each
Parity Error Detection
A parity check is used to determine whether the programmed
data of an AD8555 is valid, or whether data corruption has
occurred in the nonvolatile memory. Figure 52 shows the sche-
matic implemented in the AD8555.
Rev. 0 | Page 21 of 28
AD8555
I0
VA0
VA1
VA2
VB0
VB1
VB2
VB3
VB4
VB5
VB6
VC0
VC1
VC2
VC3
VC4
VC5
VC6
VC7
IN01
IN02
IN03
IN04
IN05
IN06
IN07
IN08
IN09
IN10
IN11
IN12
IN13
IN14
IN15
IN16
IN17
IN18
DOT_SUM
PFUSE
EOR18 OUT
PAR_SUM
MFUSE
IN1
IN2
EOR2
I1
OUT
IN1
IN2
AND2
I2
OUT
PARITY_ERROR
Figure 52. Functional Circuit of AD8555 Parity Check
Table 13. Examples of DAT_SUM
Second Stage Gain Code
First Stage Gain Code
Output Offset Code
0000 0000
1000 0000
1000 0001
0000 0000
0000 0000
0000 0000
1000 0000
1111 1111
Number of Bits with 1
DAT_SUM
000
000
000
000
000
001
001
111
000 0000
000 0000
000 0000
000 0001
100 0001
000 0000
000 0001
111 1111
0
1
2
1
2
1
3
18
0
1
0
1
0
1
1
0
When PARITY_ERROR is 0, the AD8555 behaves as a pro-
grammed amplifier. When PARITY_ERROR is 1, a parity error
has been detected, and VOUT is connected to VSS.
VA0 to VA2 is the 3-bit control signal for the second stage gain,
VB0 to VB6 is the 7-bit control signal for the first stage gain,
and VC0 to VC7 is the 8-bit control signal for the output offset.
PFUSE is the signal from the parity fuse, and MFUSE is the
signal from the master fuse.
The 18-bit data signal (VA0 to VA2, VB0 to VB6, and VC0 to
VC7) is fed to an 18-input exclusive-OR gate (Cell EOR18). The
output of Cell EOR18 is the signal DAT_SUM. DAT_SUM = 0 if
there is an even number of 1s in the 18-bit word; DAT_SUM =
1 if there is an odd number of 1s in the 18-bit word. Examples
are given in Table 13.
The function of the 2-input AND gate (cell and2) is to ignore
the output of the parity circuit (signal PAR_SUM) when the
master fuse has not been blown. PARITY_ERROR is set to 0
when MFUSE = 0. In the simulation mode, for example, parity
check is disabled. After the master fuse has been blown, i.e.,
after the AD8555 has been programmed, the output from the
parity circuit (signal PAR_SUM) is fed to PARITY_ERROR.
Rev. 0 | Page 22 of 28
AD8555
Sense Current
After the second stage gain, first stage gain, and output offset
have been programmed, DAT_SUM should be computed and
the parity bit should be set equal to DAT_SUM. If DAT_SUM is
0, the parity fuse should not be blown in order for the PFUSE
signal to be 0. If DAT_SUM is 1, the parity fuse should be blown
to set the PFUSE signal to 1. The code to blow the parity fuse is
1000 0000 0001 10 11 10 0000 0100 0111 1111 1110.
A sense current is sent across each polysilicon fuse to determine
whether it has been blown or not. When the voltage across the
fuse is less than approximately 1.5 V, the fuse is considered not
blown and Logic 0 is output from the OTP cell. When the volt-
age across the fuse is greater than approximately 1.5 V, the fuse
is considered blown and Logic 1 is output.
When the AD8555 is manufactured, all fuses have a low resis-
tance. When a sense current is sent through the fuse, a voltage
less than 0.1 V is developed across the fuse. This is much lower
than 1.5 V, so Logic 0 is output from the OTP cell. When a fuse
is electrically blown, it should have a very high resistance. When
the sense current is applied to the blown fuse, the voltage across
the fuse should be larger than 1.5 V, so Logic 1 is output from
the OTP cell.
After setting the parity bit, the master fuse can be blown to pre-
vent further programming, using the code 1000 0000 0001 10 11
10 0000 0001 0111 1111 1110.
Signal PAR_SUM is the output of the 2-input exclusive-OR
gate (Cell EOR2). After the master fuse has been blown,
PARITY_ERROR is set to PAR_SUM. As mentioned earlier,
the AD8555 behaves as a programmed amplifier when
PARITY_ERROR = 0 (no parity error). On the other hand,
VOUT is connected to VSS when a parity error has been
detected, i.e., when PARITY_ERROR = 1.
It is theoretically possible (though very unlikely) for a fuse
to be incompletely blown during programming, assuming the
required conditions are met. In this situation, the fuse could
have a medium resistance (neither low nor high), and a voltage
of approximately 1.5 V could be developed across the fuse.
Thus, the OTP cell could sometimes output Logic 0 or a Logic 1,
depending on temperature, supply voltage, and other variables.
To detect this undesirable situation, the sense current can be
lowered by a factor of 4 using a special code. The voltage devel-
oped across the fuse would then change from 1.5 V to 0.38 V,
and the output of the OTP would be a Logic 0 instead of the
Logic 1 expected from a blown fuse. Correctly blown fuses
would still output a Logic 1. In this way, incorrectly blown fuses
can be detected. Another special code would return the sense
current to the normal (larger) value. The sense current cannot
be permanently programmed to the low value. When the
AD8555 is powered up, the sense current defaults to the high
value.
Read Mode
The values stored by the polysilicon fuses can be sent to the
FILT/DIGOUT pin to verify correct programming. Normally,
the FILT/DIGOUT pin is connected to only the second gain
stage output via RF. During read mode, however, the
FILT/DIGOUT pin is also connected to the output of a shift
register to allow the polysilicon fuse contents to be read. Since
VOUT is a buffered version of FILT/DIGOUT, VOUT also out-
puts a digital signal during read mode.
Read mode is entered by setting Field 1 to 11 and selecting the
desired parameter in Field 2; Field 4 is ignored. The parameter
value, stored in the polysilicon fuses, is loaded into an internal
shift register, and the MSB of the shift register is connected to
the FILT/DIGOUT pin. Pulses at DIGIN shift the shift register
contents out to the FILT/DIGOUT pin, allowing the 8‒bit
parameter value to be read after seven additional pulses; shift-
ing occurs on the falling edge of DIGIN. An eighth pulse at
DIGIN disconnects FILT/DIGOUT from the shift register and
terminates the read mode. If a parameter value is less than 8 bits
long, the MSBs of the shift register are padded with 0s.
The code to use the low sense current is 1000 0000 0001 00 00
10 XXXX XXX1 0111 1111 1110.
The code to use the normal (high) sense current is 1000 0000
0001 00 00 10 XXXX XXX0 0111 1111 1110.
For example, to read the second stage gain, the code 1000 0000
0001 11 00 10 0000 0000 0111 1111 1110 can be used. Since the
second stage gain parameter value is only three bits long, the
FILT/DIGOUT pin has a value of 0 when this code is entered
and remains 0 during four additional pulses at DIGIN. The
fifth, sixth, and seventh pulse at DIGIN returns the 3-bit value
at FILT/DIGOUT, the seventh pulse returning the LSB. An
eighth pulse at DIGIN terminates the read mode.
Rev. 0 | Page 23 of 28
AD8555
Suggested Programming Procedure
Suggested Algorithm to Determine
Optimal Gain and Offset Codes
1. Set VDD and VSS to the desired values in the application.
Use simulation mode to test and determine the desired
codes for the second stage gain, first stage gain, and output
offset. The nominal values for these parameters are shown
in Table 6, Table 7, Equation 1, and Equation 2; the codes
corresponding to these values can be used as a starting
point. However, since actual parameter values for given
codes vary from device to device, some fine tuning is nec-
essary for the best possible accuracy.
1.
Determine the desired gain, GA (e.g., using measure-
ments).
2a.
Use Table 7 to determine the second stage gain G2 such
that (4.00 × 1.04) < (GA/G2) < (6.4/1.04). This ensures
that the first and last codes for the first stage gain are not
used, thereby allowing enough first stage gain codes
within each second stage gain range to adjust for the 3%
accuracy.
2b.
3a.
Use simulation mode to set the second stage gain to G2.
Set the output offset to allow the AD8555 gain to be
measured, e.g., use Code 128 to set it to midsupply.
Use Table 6 or Equation 3 to set the first stage gain code
One way to choose these values is to set the output offset
to an approximate value, e.g., Code 128 for midsupply, to
allow the required gain to be determined. Then set the sec-
ond stage gain such that the minimum first stage gain
(Code 0) gives a lower gain than required, and the maxi-
mum first stage gain (Code 127) gives a higher gain than
required. After choosing the second stage gain, the first
stage gain can be chosen to fine tune the total gain. Finally,
the output offset can be adjusted to give the desired value.
After determining the desired codes for second stage gain,
first stage gain, and output offset, the device is ready for
permanent programming.
3b.
3c.
3d.
3e.
CG1 such that the first stage gain is nominally GA/G2.
Measure the resulting gain GB. GB should be within
3% of GA.
Calculate the first stage gain error (in relative terms)
E
G1 = GB/GA − 1.
Calculate the error (in the number of the first stage gain
codes) CEG1 = EG1/0.00370.
3f.
3g.
3h.
3i.
Set the first stage gain code to CG1 − CEG1.
Measure the gain GC. GC should be closer to GA than to GB.
Calculate the error (in relative terms) EG2 = GC/GA − 1.
Calculate the error (in the number of the first stage gain
codes) CEG2 = EG2/0.00370.
2. Set VSS to 0 V and VDD to 5.5 V. Use program mode to
permanently enter the desired codes for the second stage
gain, first stage gain, and output offset. Blow the master
fuse to allow the AD8555 to read data from the fuses and
to prevent further programming.
3j.
Set the first stage gain code to CG1 − CEG1 − CEG2. The
resulting gain should be within one code of GA.
Determine the desired output offset OA, e.g., using the
measurements.
Use Equation 1 to set the output offset code CO1 such
that the output offset is nominally OA.
4a.
4b.
4c.
3. Set VDD and VSS to the desired values in the application.
Use read mode with low sense current followed by high
sense current to verify programmed codes.
Measure the output offset OB. OB should be within
3% of OA.
4. Measure gain and offset to verify correct functionality.
4d.
4e.
Calculate the error (in relative terms) EO1 = OB/OA − 1.
Calculate the error (in the number of the output offset
codes) CEO1 = EO1/0.00392.
4f.
4g.
Set the output offset code to CO1 − CEO1.
Measure the output offset OC. OC should be closer to OA
than to OB.
4h.
4i.
Calculate the error (in relative terms) EO2 = OC/OA − 1.
Calculate the error (in the number of the output offset
codes) CEO2 = EO2/0.00392.
4j.
Set the output offset code to CO1 − CEO1 − CEO2. The
resulting offset should be within one code of OA.
Rev. 0 | Page 24 of 28
AD8555
FILTERING FUNCTION
DRIVING CAPACITIVE LOADS
The AD8555’s FILT/DIGOUT pin can be used to create a simple
low-pass filter. The AD8555’s internal 18 kΩ resistor can be
used with an external capacitor for this purpose. Typical
responses of the AD8555, configured for a gain of 70 and gain
of 1280, are shown in Figure 54 and Figure 55, respectively. This
filtering feature can be used to pass the signals within the filter’s
pass band while limiting the out-of-band signals bandwidth
and, therefore, reducing the noise of the overall solution.
The AD8555 can drive large capacitive loads. This feature is
useful when the amplifier, placed close to the sensor, has to
drive long cables. Most instrumentation amplifiers have diffi-
culty driving capacitance due to the degradation of the phase
margin caused by the additional phase lag from the capacitive
load. Higher capacitance at the output can increase the amount
of overshoot and ringing in the amplifier’s step response and
could even affect the stability of the device. Additionally, the
value of the capacitive load that an amplifier can drive before
oscillation varies with gain, supply voltage, input signal, and
temperature. Figure 57 and Figure 58 show the overshoot
response of AD8555 versus the capacitive load with a different
value isolation resistor (RS) in Figure 56. Similar to all amplifi-
ers, the AD8555 responds with overshoot when driving large CL,
but after a point (approximately 22 nF), the overshoot decreases.
This is because the pole created by CL dominates at first; how-
ever, at some point, the pole is farther in than the pole setting of
the buffer amplifier and is ignored by AD8555.
VDD
VSS
1
2
3
4
8
7
6
5
VDD
VSS
V
FILT/DIGOUT VOUT
OUT
C
FILTER
DIGIN
VNEG
VCLAMP
VPOS
VDD
AD8555
V
IN
Figure 53. AD8555 Configured to Filter Noise
VDD
VSS
1
2
3
4
8
7
6
5
VDD
VSS
C
= 0.010µF
C
FILTER
FILTER
R
S
C
= 0.001µF
FILT/DIGOUT VOUT
V
FILTER
OUT
40
20
0
C
L
DIGIN
VNEG
VCLAMP
VPOS
VDD
AD8555
C
= 0.100µF
FILTER
Figure 56. Test Circuit for Driving Capacitive Loads
60
50
40
30
20
10
0
V
= ±2.5V
S
R
R
= 0
S
S
C
= 1nF
L
10
100
1k
10k
50k
OUTPUT
BUFFER
Figure 54. Typical Response of the AD8555 at FILT/DIGOUT Pin (Gain = 70)
R
= 10
S
R = 20
S
C
= 0.001
µ
F
FILTER
R
= 50
S
60
40
20
0
R
= 100
1
S
0.1
10
100
C
= 0.010
µF
LOAD CAPACITANCE (nF)
FILTER
Figure 57. Positive Overshoot Graph vs. CL
C
= 0.100µF
FILTER
10
100
1k
10k
100k
Figure 55. Typical Response of the AD8555 at FILT/DIGOUT Pin (Gain = 1280)
Rev. 0 | Page 25 of 28
AD8555
60
needed to maintain common-mode rejection at low frequencies.
This introduces a second low-pass network, R1 + R2 and C3
that has a −3 dB frequency equal to 1/(2 π × (R1 + R2)(C3)).
This circuit’s −3 dB signal bandwidth is approximately 4 kHz
when a C3 value of 0.047 µF is used (see Figure 59).
V
= ±2.5V
S
R
S
50
40
30
20
10
0
R
= 0
C
S
L
VDD
VSS
R
= 10
S
1
2
3
4
8
7
6
5
VDD
VSS
FILT/DIGOUT VOUT
R
= 20
S
R2
4.02kΩ
DIGIN
VNEG
VCLAMP
VPOS
VDD
VNEG
VPOS
C2
1nF
C1
1nF
AD8555
C3
R
= 50
S
R
= 100
S
0.047µF
0.1
1.0
10.0
100.0
LOAD CAPACITANCE (nF)
R1
4.02kΩ
Figure 58. Negative Overshoot Graph vs. CL
Figure 59. RFI Suppression Method
RF INTERFERENCE
SINGLE-SUPPLY DATA ACQUISITION SYSTEM
All instrumentation amplifiers show dc offset as the result of
rectification of high frequency out-of-band signals that appear
at their inputs. The circuit in Figure 59 provides good RFI sup-
pression without reducing performance within the AD8555 pass
band. Resistor R1 and Capacitor C1, and likewise Resistor R2
and Capacitor C2, form a low-pass RC filter that has a −3 dB
bandwidth equal to f(−3 dB) = 1/2 π × R1 × C1. It can be seen that
R1, R2 and C1, C2 form a bridge circuit whose output appears
across the amplifier’s input pins. Any mismatch between C1, C2
unbalances the bridge and reduce the common-mode rejection.
Using the component values shown, this filter has a bandwidth
of approximately 40 kHz. To preserve common-mode rejection
in the AD8555’s pass band, capacitors need to be 5% (silver
mica) or better and should be placed as close to its inputs as
possible. Resistors should be 1% metal film. Capacitor C3 is
Interfacing bipolar signals to single-supply analog-to-digital
converters (ADCs) presents a challenge. The bipolar signal must
be mapped into the input range of the ADC. Figure 60 shows
how this translation can be achieved. The output offset can be
programmed to a desirable level to accommodate the input
voltage requirement of the ADC.
VDD
4
V
DD
1
2
3
4
8
7
6
5
VDD
VSS
10nF
2
FILT/DIGOUT VOUT
AIN
12 BIT
DIGIN
VNEG
VCLAMP
VPOS
VDD
100Ω
100Ω
100Ω
100Ω
AD7476
AD8555
0
S
DIGIN
Figure 60. A Single-Supply Data Acquisition Circuit Using the AD8555
Rev. 0 | Page 26 of 28
AD8555
tial offset voltage between VPOS and VNEG. This differential
offset voltage is amplified by the AD8555. The input bias cur-
rent at VNEG, on the other hand, flows into RP1 and create a
common-mode shift. This has little impact on VOUT. Despite
this weakness, the arrangement in Figure 61 should work if the
user wants to minimize the number of components around the
sensor, and if the error introduced by the input bias current at
VPOS is considered negligible.
The bridge circuit with a sensitivity of 2 mV/V is excited by a
5 V supply. The full-scale output voltage from the bridge
( 10 mV) therefore has a common-mode level of 2.5 V. The
AD8555 removes the common-mode component and amplifies
the input signal by a factor of 200 (G1 = 4, G2 = 50, Offset =
128). This results in an output signal of 2.0 V. In order to pre-
vent this signal from running into the AD8555’s ground rail, the
output offset voltage has to be raised to 2.5 V. This signal is
within the input voltage range of the ADC.
If greater accuracy is needed, the circuit in Figure 62 is recom-
mended. RP1, RP2, and CS are the same as in Figure 61; RP1 and
USING THE AD8555 WITH CAPACITIVE SENSORS
RP2 should be between 1 kΩ to 1 MΩ. RS in Figure 61 has been
Figure 61 shows a crude way of using the AD8555 with capaci-
tive sensors. RP1 and RP2 are resistors implementing a potential
divider to bias VNEG to VDD/2. Recommended values range
from 1 kΩ to 1 MΩ. CS is the capacitive sensor, and RS is a shunt
resistor used to prevent leakage currents from integrating on
the sensor. The value of RS is application specific.
split into two resistors, RS1 and RS2, in Figure 62. Again, the only
way for the capacitive sensor to discharge is through (RS1 + RS2).
The input bias current at VPOS flows through RS2 and RP1, and
the input bias current at VNEG flows through RS1 and RP1. If RS1
is made equal to RS2 and if the input bias currents are equal, the
input bias currents give a common-mode shift at VPOS and
VNEG with no differential offset. This common-mode shift is
attenuated by the AD8555 common-mode rejection. Further-
more, changes in input bias current, e.g., with temperature,
manifest as an input common-mode change, also rejected by the
AD8555.
Note that although VNEG is tied to a dc voltage, the only
impedance across the capacitive sensor is RS. Therefore, the only
way for charge to leak away from CS is through RS, assuming the
input bias currents at VPOS and VNEG are negligible.
R
P2
VPOS
VOUT
R
P2
C
R
VDD
S
S
R
S2
VPOS
CS
VOUT
VNEG
AD8555
VDD
R
P1
AD8555
VNEG
R
S1
R
P1
Figure 61. Crude Way of Using the AD8555 with Capacitive Sensors
Figure 62. Recommended Way of Using the AD8555 with Capacitive Sensors
The weakness of the circuit in Figure 61 is that the AD8555
input bias current at VPOS flows into RS and creates a differen-
Rev. 0 | Page 27 of 28
AD8555
OUTLINE DIMENSIONS
5.00 (0.1968)
4.80 (0.1890)
8
1
5
4
6.20 (0.2440)
5.80 (0.2284)
4.00 (0.1574)
3.80 (0.1497)
1.27 (0.0500)
BSC
0.50 (0.0196)
0.25 (0.0099)
×
45°
1.75 (0.0688)
1.35 (0.0532)
0.25 (0.0098)
0.10 (0.0040)
8°
0.51 (0.0201)
0.31 (0.0122)
0° 1.27 (0.0500)
COPLANARITY
0.10
0.25 (0.0098)
0.17 (0.0067)
SEATING
PLANE
0.40 (0.0157)
COMPLIANT TO JEDEC STANDARDS MS-012AA
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
Figure 63. 8-Lead Standard Small Outline Package [SOIC] Narrow Body
(R-8)
Dimensions shown in millimeters (inches)
4.0
BSC SQ
0.60 MAX
PIN 1
INDICATOR
0.60 MAX
0.65 BSC
13
12
16
1
PIN 1
INDICATOR
2.25
2.10 SQ
1.95
TOP
VIEW
3.75
BSC SQ
BOTTOM
VIEW
0.75
0.60
0.50
4
9
8
5
0.25 MIN
1.95 BSC
0.80 MAX
0.65 TYP
12° MAX
0.05 MAX
0.02 NOM
1.00
0.85
0.80
0.35
0.28
0.25
0.20 REF
COPLANARITY
0.08
SEATING
PLANE
COMPLIANT TO JEDEC STANDARDS MO-220-VGGC
Figure 64. 16-Lead Lead Frame Chip Scale Package [LFCSP] 4 mm × 4 mm Body
(CP-16)
Dimensions shown in millimeters
ORDERING GUIDE
Model
Temperature Range
Package Description
8-Lead SOIC
8-Lead SOIC
Package Option
AD8555AR
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
R-8
R-8
R-8
AD8555AR-REEL
AD8555AR-REEL7
AD8555AR-EVAL
AD8555ACP-R2
AD8555ACP-REEL
AD8555ACP-REEL7
8-Lead SOIC
Evaluation Board
16-Lead LFCSP
16-Lead LFCSP
16-Lead LFCSP
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
CP-16
CP-16
CP-16
© 2004 Analog Devices, Inc. All rights reserved. Trademarks and regis-
tered trademarks are the property of their respective owners.
D04598–0–4/04(0)
Rev. 0 | Page 28 of 28
相关型号：
©2020 ICPDF网 联系我们和版权申明