ADP1111AR-5 [ADI]
Micropower, Step-Up/Step-Down SW Regulator; Adjustable and Fixed 3.3 V, 5 V, 12 V; 微功耗，升压/降压型稳压器SW ;可调和固定3.3 V ， 5 V ， 12 V
| 型号： | ADP1111AR-5 |
| 厂家： | 
ADI |
| 描述： | Micropower, Step-Up/Step-Down SW Regulator; Adjustable and Fixed 3.3 V, 5 V, 12 V |
| 文件： | 总16页 (文件大小：409K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Micropower, Step-Up/Step-Down SW
a Regulator; Adjustable and Fixed 3.3 V, 5 V, 12 V
ADP1111
FEATURES
FUNCTIONAL BLOCK DIAGRAMS
Operates from 2 V to 30 V Input Voltage Range
72 kHz Frequency Operation
SET
Utilizes Surface Mount Inductors
Very Few External Components Required
Operates in Step-Up/Step-Down or Inverting Mode
Low Battery Detector
User Adjustable Current Limit
Internal 1 A Power Switch
ADP1111
A2
A0
V
IN
GAIN BLOCK/
ERROR AMP
I
LIM
SW1
1.25V
REFERENCE
Fixed or Adjustable Output Voltage
8-Pin DIP or SO-8 Package
OSCILLATOR
A1
DRIVER
APPLICATIONS
COMPARATOR
3 V to 5 V, 5 V to 12 V Step-Up Converters
9 V to 5 V, 12 V to 5 V Step-Down Converters
Laptop and Palmtop Computers
Cellular Telephones
GND
SW2
FB
SET
Flash Memory VPP Generators
Remote Controls
Peripherals and Add-On Cards
Battery Backup Supplies
ADP1111-5
ADP1111-12
A2
A0
V
IN
GAIN BLOCK/
ERROR AMP
I
LIM
Uninterruptible Supplies
SW1
Portable Instruments
1.25V
REFERENCE
OSCILLATOR
A1
DRIVER
COMPARATOR
R2 220k
SW2
R1
GENERAL DESCRIPTION
GND
SENSE
The ADP1111 is part of a family of step-up/step-down switch-
ing regulators that operates from an input voltage supply of 2 V
to 12 V in step-up mode and up to 30 V in step-down mode.
The ADP1111 can be programmed to operate in step-up/step-
down or inverting applications with only 3 external components.
Maximum switch current can be programmed with a single
resistor, and an open collector gain block can be arranged in
multiple configuration for low battery detection, as a post linear
regulator, undervoltage lockout, or as an error amplifier.
The fixed outputs are 3.3 V, 5 V and 12 V; and an adjustable
version is also available. The ADP1111 can deliver 100 mA at
5 V from a 3 V input in step-up mode, or it can deliver 200 mA
at 5 V from a 12 V input in step-down mode.
If input voltages are lower than 2 V, see the ADP1110.
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700
Fax: 617/326-8703
World Wide Web Site: http://www.analog.com
© Analog Devices, Inc., 1996
(0؇C ≤ T ≤ +70؇C, V = 3 V unless otherwise noted)
ADP1111–SPECIFICATIONS
A
IN
Parameter
Conditions
Switch Off
VS
IQ
Min
Typ
Max
Units
QUIESCENT CURRENT
INPUT VOLTAGE
300
500
µA
Step-Up Mode
VIN
2.0
12.6
30.0
V
V
Step-Down Mode
COMPARATOR TRIP POINT
VOLTAGE
ADP11111
1.20
1.25
1.30
V
OUTPUT SENSE VOLTAGE
ADP1111-3.3
ADP1111-52
ADP1111-122
VOUT
3.13
4.75
11.40
3.30
5.00
12.00
3.47
5.25
12.60
V
V
V
COMPARATOR HYSTERESIS
OUTPUT HYSTERESIS
ADP1111
8
12.5
mV
ADP1111-3.3
ADP1111-5
ADP1111-12
21
32
75
50
50
120
mV
mV
mV
OSCILLATOR FREQUENCY
DUTY CYCLE
fOSC
DC
tON
54
43
5
72
50
7
88
65
9
kHz
%
Full Load
SWITCH ON TIME
ILIM Tied to VIN
µs
SW SATURATION VOLTAGE
STEP-UP MODE
TA = +25°C
V
V
IN = 3.0 V, ISW = 650 mA
IN = 5.0 V, ISW = 1 A
VSAT
0.5
0.8
1.1
0.65
1.0
1.5
V
V
V
STEP-DOWN MODE
VIN = 12 V, ISW = 650 mA
ADP1111 VFB = 0 V
VSET = VREF
FEEDBACK PIN BIAS CURRENT
SET PIN BIAS CURRENT
GAIN BLOCK OUTPUT LOW
IFB
160
270
300
400
nA
nA
ISET
I
SINK = 300 µA
VSET = 1.00 V
VOL
0.15
0.4
V
REFERENCE LINE REGULATION
5 V ≤ VIN ≤ 30 V
2 V ≤ VIN ≤ 5 V
0.02
0.4
0.075
%/V
%/V
GAIN BLOCK GAIN
CURRENT LIMIT
RL = 100 kΩ3
AV
1000
6000
V/V
TA = +25°C
220 Ω from ILIM to VIN
ILIM
400
mA
CURRENT LIMIT TEMPERATURE
COEFFICIENT
–0.3
%/°C
SWITCH OFF LEAKAGE CURRENT
TA = +25°C
Measured at SW1 Pin
VSW1 = 12 V
1
10
µA
MAXIMUM EXCURSION BELOW GND
TA = +25°C
I
SW1 ≤ 10 µA, Switch Off
–400
–350
mV
NOTES
1This specification guarantees that both the high and low trip points of the comparator fall within the 1.20 V to 1.30 V range.
2The output voltage waveform will exhibit a sawtooth shape due to the comparator hysteresis. The output voltage on the fixed output versions will always be within
the specified range.
3100 kΩ resistor connected between a 5 V source and the AO pin.
4All limits at temperature extremes are guaranteed via correlation using standard statistical methods.
Specifications subject to change without notice.
REV. 0
–2–
ADP1111
PIN DESCRIPTIONS
Function
ABSOLUTE MAXIMUM RATINGS
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36 V
SW1 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50 V
SW2 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . –0.5 V to VIN
Feedback Pin Voltage (ADP1111) . . . . . . . . . . . . . . . . . 5.5 V
Switch Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.5 A
Maximum Power Dissipation . . . . . . . . . . . . . . . . . . 500 mW
Operating Temperature Range
ADP1111A . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C
Storage Temperature Range . . . . . . . . . . . . . –65°C to 150°C
Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . . 300°C
Mnemonic
ILIM
For normal conditions this pin is connected to
VIN. When lower current is required, a resistor
should be connected between ILIM and VIN.
Limiting the switch current to 400 mA is achieved
by connecting a 220 Ω resistor.
VIN
Input Voltage.
SW1
Collector Node of Power Transistor. For step-
down configuration, connect to VIN. For step-up
configuration, connect to an inductor/diode.
TYPICAL APPLICATION
SW2
Emitter Node of Power Transistor. For step-
down configuration, connect to inductor/diode.
For step-up configuration, connect to ground.
Do not allow this pin to go more than a diode
drop below ground.
SUMIDA
CD54-220K
22µH
MBRS120T3
5V
100mA
3V
INPUT
GND
AO
Ground.
I
V
IN
LIM
Auxiliary Gain (GB) Output. The open collector
SW1
can sink 300 µA. It can be left open if unused.
10µF
(OPTIONAL)
ADP1111AR-5
SENSE
33µF
SET
Gain Amplifier Input. The amplifier’s positive
input is connected to SET pin and its negative
input is connected to the 1.25 V reference. It can
be left open if unused.
GND SW2
FB/SENSE
On the ADP1111 (adjustable) version this pin
is connected to the comparator input. On the
ADP1111-3.3, ADP1111-5 and ADP1111-12,
the pin goes directly to the internal application
resistor that sets output voltage.
Figure 1. 3 V to 5 V Step-Up Converter
ORDERING GUIDE
Output Voltage
Model
Package*
PIN CONFIGURATIONS
ADP1111AN
ADP1111AR
ADJ
ADJ
3.3 V
3.3 V
5 V
5 V
12 V
12 V
N-8
SO-8
N-8
SO-8
N-8
SO-8
N-8
8-Lead Plastic DIP
8-Lead SOIC
(SO-8)
(N-8)
ADP1111AN-3.3
ADP1111AR-3.3
ADP1111AN-5
ADP1111AR-5
ADP1111AN-12
ADP1111AR-12
I
1
2
3
4
8
7
6
5
I
1
2
3
4
8
FB (SENSE)*
FB (SENSE)*
LIM
LIM
ADP1111
TOP VIEW
(Not to Scale)
ADP1111
TOP VIEW
(Not to Scale)
7
6
5
SET
A0
SET
A0
V
V
IN
IN
SW1
SW2
SW1
SW2
SO-8
GND
GND
*N = Plastic DIP, SO = Small Outline Package.
*FIXED VERSIONS
*FIXED VERSIONS
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the ADP1111 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
REV. 0
–3–
ADP1111–Typical Characteristics
76
75
1.4
1.2
1.0
74
73
72
71
70
69
OSCILLATOR FREQUENCY
0.8
V
= 3V
IN
V
= 5V
IN
0.6
0.4
0.2
0
V
= 2V
IN
68
67
2
4
6
8
10
12 15
18
21
24
27
30
0.1
0.2
0.4
I
0.6
0.8
1.0
1.2
CURRENT – A
INPUT VOLTAGE – V
SWITCH
Figure 2. Saturation Voltage vs. ISWITCH Current in
Step-Up Mode
Figure 5. Oscillator Frequency vs. Input Voltage
2.0
1.8
1.9
1.7
1.6
1.4
1.5
1.3
1.1
0.9
0.7
0.5
STEP-DOWN WITH
= 12V
V
IN
1.2
1.0
V
= 12V
IN
0.8
0.6
0.4
STEP-UP WITH
2V < V < 5V
IN
0.3
0.1
0.2
0
1
10
100
1000
0.1
0.2
0.4
0.6
0.8
0.9
R – Ω
LIM
I
CURRENT – A
SWITCH
Figure 6. Maximum Switch Current vs. RLIM
Figure 3. Switch ON Voltage vs. ISWITCH Current In
Step-Down Mode
80
78
76
1400
1200
1000
74
QUIESCENT CURRENT
72
70
68
OSCILLATOR FREQUENCY
800
600
400
66
64
200
0
62
60
–40
25
70
85
0
1.5
3
6
9
12
15
18
21
24
27
30
TEMPERATURE – ؇C
INPUT VOLTAGE – V
Figure 4. Quiescent Current vs. Input Voltage
Figure 7. Oscillator Frequency vs. Temperature
REV. 0
–4–
ADP1111
7.5
7.4
1.10
1.05
1.00
0.95
0.90
7.3
7.2
7.1
7.0
V
IN
= 12V
@
I
= 0.65A
SW
ON TIME
6.9
6.8
0.85
0.80
6.7
6.6
–40
0
25
70
85
–40
0
25
TEMPERATURE – ؇C
70
85
TEMPERATURE – ؇C
Figure 8. Switch ON Time vs. Temperature
Figure 11. Switch ON Voltage vs. Temperature in Step-
Down Mode
58
56
500
450
400
DUTY CYCLE
350
54
QUIESCENT CURRENT
300
250
200
150
100
52
50
48
46
50
0
–40
0
25
70
85
–40
0
25
70
85
TEMPERATURE – ؇C
TEMPERATURE – ؇C
Figure 9. Duty Cycle vs. Temperature
Figure 12. Quiescent Current vs. Temperature
250
200
0.6
0.5
0.4
V
IN
= 3V
@
I
= 0.65A
SW
BIAS CURRENT
150
0.3
0.2
0.1
0
100
50
0
–40
0
25
70
85
–40
0
25
TEMPERATURE – ؇C
70
85
TEMPERATURE – ؇C
Figure 10. Saturation Voltage vs. Temperature in Step-Up
Mode
Figure 13. Feedback Bias Current vs. Temperature
REV. 0
–5–
ADP1111
350
300
250
200
150
100
50
The ADP1111 provides external connections for both the
collector and emitter of its internal power switch that permit
both step-up and step-down modes of operation. For the step-
up mode, the emitter (Pin SW2) is connected to GND, and the
collector (Pin SW1) drives the inductor. For step-down mode,
the emitter drives the inductor while the collector is connected
to VIN.
BIAS CURRENT
The output voltage of the ADP1111 is set with two external
resistors. Three fixed-voltage models are also available:
ADP1111–3.3 (+3.3 V), ADP1111–5 (+5 V) and ADP1111–12
(+12 V). The fixed-voltage models are identical to the
ADP1111, except that laser-trimmed voltage-setting resistors
are included on the chip. On the fixed-voltage models of the
ADP1111, simply connect the feedback pin (Pin 8) directly to
the output voltage.
0
–40
0
25
TEMPERATURE – ؇C
70
85
Figure 14. Set Pin Bias Current vs. Temperature
COMPONENT SELECTION
General Notes on Inductor Selection
THEORY OF OPERATION
When the ADP1111 internal power switch turns on, current
begins to flow in the inductor. Energy is stored in the inductor
core while the switch is on, and this stored energy is transferred
to the load when the switch turns off. Since both the collector
and the emitter of the switch transistor are accessible on the
ADP1111, the output voltage can be higher, lower, or of
opposite polarity than the input voltage.
The ADP1111 is a flexible, low-power, switch-mode power
supply (SMPS) controller. The regulated output voltage can be
greater than the input voltage (boost or step-up mode) or less
than the input (buck or step-down mode). This device uses a
gated-oscillator technique to provide very high performance
with low quiescent current.
A functional block diagram of the ADP1111 is shown on
the first page of this data sheet. The internal 1.25 V reference is
connected to one input of the comparator, while the other input
is externally connected (via the FB pin) to a feedback network
connected to the regulated output. When the voltage at the FB
pin falls below 1.25 V, the 72 kHz oscillator turns on. A driver
amplifier provides base drive to the internal power switch, and
the switching action raises the output voltage. When the voltage
at the FB pin exceeds 1.25 V, the oscillator is shut off. While
the oscillator is off, the ADP1111 quiescent current is only
300 µA. The comparator includes a small amount of hysteresis,
which ensures loop stability without requiring external compo-
nents for frequency compensation.
To specify an inductor for the ADP1111, the proper values of
inductance, saturation current and dc resistance must be
determined. This process is not difficult, and specific equations
for each circuit configuration are provided in this data sheet. In
general terms, however, the inductance value must be low
enough to store the required amount of energy (when both
input voltage and switch ON time are at a minimum) but high
enough that the inductor will not saturate when both VIN and
switch ON time are at their maximum values. The inductor
must also store enough energy to supply the load, without
saturating. Finally, the dc resistance of the inductor should be
low so that excessive power will not be wasted by heating the
windings. For most ADP1111 applications, an inductor of
15 µH to 100 µH with a saturation current rating of 300 mA to
1 A and dc resistance <0.4 Ω is suitable. Ferrite-core inductors
that meet these specifications are available in small, surface-
mount packages.
The maximum current in the internal power switch can be set
by connecting a resistor between VIN and the ILIM pin. When the
maximum current is exceeded, the switch is turned OFF. The
current limit circuitry has a time delay of about 1 µs. If an
external resistor is not used, connect ILIM to VIN. Further
information on ILIM is included in the “APPLICATIONS”
section of this data sheet.
To minimize Electro-Magnetic Interference (EMI), a toroid or
pot-core type inductor is recommended. Rod-core inductors are
a lower-cost alternative if EMI is not a problem.
The ADP1111 internal oscillator provides 7 µs ON and 7 µs
OFF times that are ideal for applications where the ratio
between VIN and VOUT is roughly a factor of two (such as
converting +3 V to + 5 V). However, wider range conversions
(such as generating +12 V from a +5 V supply) can easily be
accomplished.
CALCULATING THE INDUCTOR VALUE
Selecting the proper inductor value is a simple three step
process:
1. Define the operating parameters: minimum input voltage,
maximum input voltage, output voltage and output current.
An uncommitted gain block on the ADP1111 can be connected
as a low-battery detector. The inverting input of the gain block
is internally connected to the 1.25 V reference. The noninverting
input is available at the SET pin. A resistor divider, connected
between VIN and GND with the junction connected to the SET
pin, causes the AO output to go LOW when the low battery set
point is exceeded. The AO output is an open collector NPN
transistor that can sink 300 µA.
2. Select the appropriate conversion topology (step-up, step-
down, or inverting).
3. Calculate the inductor value using the equations in the
following sections.
REV. 0
–6–
ADP1111
INDUCTOR SELECTION–STEP-UP CONVERTER
In a step-up or boost converter (Figure 18), the inductor must
store enough power to make up the difference between the input
voltage and the output voltage. The power that must be stored
is calculated from the equation:
Substituting a standard inductor value of 68 µH with 0.2 Ω dc
resistance will produce a peak switch current of:
−1.0 Ω• 7µs
68 µH
6V
1.0 Ω
IPEAK
=
1 − e
= 587 mA
PL = V
+VD −VIN(MIN) • I
(Equation 1)
(
)
(
)
OUT
OUT
Once the peak current is known, the inductor energy can be
calculated from Equation 5:
where VD is the diode forward voltage (0.5 V for a 1N5818
Schottky). Because energy is only stored in the inductor while
the ADP1111 switch is ON, the energy stored in the inductor
on each switching cycle must be equal to or greater than:
1
EL
=
68 µH • 587 mA 2 =11.7µJ
) (
(
)
2
PL
(Equation 2)
Since the inductor energy of 11.7 µJ is greater than the PL/fOSC
requirement of 3.6 µJ, the 68 µH inductor will work in this
application. By substituting other inductor values into the same
equations, the optimum inductor value can be selected.
fOSC
in order for the ADP1111 to regulate the output voltage.
When the internal power switch turns ON, current flow in the
inductor increases at the rate of:
When selecting an inductor, the peak current must not exceed
the maximum switch current of 1.5 A. If the equations shown
above result in peak currents > 1.5 A, the ADP1110 should be
considered. Since this device has a 70% duty cycle, more energy
is stored in the inductor on each cycle. This results is greater
output power.
−R't
L
VIN
R'
IL t =
( )
1−e
(Equation 3)
where L is in Henrys and R' is the sum of the switch equivalent
resistance (typically 0.8 Ω at +25°C) and the dc resistance of
the inductor. In most applications, the voltage drop across the
switch is small compared to VIN so a simpler equation can be
used:
The peak current must be evaluated for both minimum and
maximum values of input voltage. If the switch current is high
when VIN is at its minimum, the 1.5 A limit may be exceeded at
the maximum value of VIN. In this case, the ADP1111’s current
limit feature can be used to limit switch current. Simply select a
resistor (using Figure 6) that will limit the maximum switch
current to the IPEAK value calculated for the minimum value of
VIN. This will improve efficiency by producing a constant IPEAK
as VIN increases. See the “Limiting the Switch Current” section
of this data sheet for more information.
V
L
IL t = IN t
(Equation 4)
( )
Replacing ‘t’ in the above equation with the ON time of the
ADP1111 (7 µs, typical) will define the peak current for a given
inductor value and input voltage. At this point, the inductor
energy can be calculated as follows:
Note that the switch current limit feature does not protect the
circuit if the output is shorted to ground. In this case, current is
only limited by the dc resistance of the inductor and the forward
voltage of the diode.
1
2
EL
=
L • I2 PEAK
(Equation 5)
As previously mentioned, EL must be greater than PL/fOSC so
that the ADP1111 can deliver the necessary power to the load.
For best efficiency, peak current should be limited to 1 A or
less. Higher switch currents will reduce efficiency because of
increased saturation voltage in the switch. High peak current
also increases output ripple. As a general rule, keep peak current
as low as possible to minimize losses in the switch, inductor and
diode.
INDUCTOR SELECTION–STEP-DOWN CONVERTER
The step-down mode of operation is shown in Figure 19.
Unlike the step-up mode, the ADP1111’s power switch does not
saturate when operating in the step-down mode; therefore,
switch current should be limited to 650 mA in this mode. If the
input voltage will vary over a wide range, the ILIM pin can be
used to limit the maximum switch current. Higher switch
current is possible by adding an external switching transistor as
shown in Figure 21.
In practice, the inductor value is easily selected using the
equations above. For example, consider a supply that will
generate 12 V at 40 mA from a 9 V battery, assuming a 6 V
end-of-life voltage. The inductor power required is, from
Equation 1:
The first step in selecting the step-down inductor is to calculate
the peak switch current as follows:
PL = 12V +0.5V − 6V • 40 mA = 260 mW
(
) (
)
2 IOUT
DC
VOUT + VD
IN −VSW +VD
IPEAK
=
(Equation 6)
On each switching cycle, the inductor must supply:
V
PL 260 mW
fOSC 72 kHz
=
= 3.6 µJ
where DC = duty cycle (0.5 for the ADP1111)
VSW = voltage drop across the switch
VD = diode drop (0.5 V for a 1N5818)
IOUT = output current
Since the required inductor power is fairly low in this example,
the peak current can also be low. Assuming a peak current of
500 mA as a starting point, Equation 4 can be rearranged to
recommend an inductor value:
VOUT = the output voltage
VIN
6V
VIN = the minimum input voltage
L =
t =
7µs = 84 µH
IL(MAX) 500 mA
REV. 0
–7–
ADP1111
As previously mentioned, the switch voltage is higher in step-
down mode than in step-up mode. VSW is a function of switch
During each switching cycle, the inductor must supply the
following energy:
current and is therefore a function of VIN, L, time and VOUT
.
PL 275 mW
For most applications, a VSW value of 1.5 V is recommended.
=
= 3.8 µJ
fOSC 72 kHz
The inductor value can now be calculated:
Using a standard inductor value of 56 µH with 0.2 Ω dc
resistance will produce a peak switch current of:
VIN MIN −VSW −VOUT
(
)
L =
• tON
(Equation 7)
IPEAK
−0.85 Ω• 7µs
56 µH
4.5V − 0.75V
0.65 Ω+0.2 Ω
where tON = switch ON time (7 µs).
IPEAK
=
1− e
= 445 mA
If the input voltage will vary (such as an application that must
operate from a 9 V, 12 V or 15 V source), an RLIM resistor
should be selected from Figure 6. The RLIM resistor will keep
switch current constant as the input voltage rises. Note that
there are separate RLIM values for step-up and step-down modes
of operation.
Once the peak current is known, the inductor energy can be
calculated from (Equation 9):
1
2
EL
=
56 µH • 445 mA 2 = 5.54 µJ
) (
(
)
For example, assume that +5 V at 300 mA is required from a
+12 V to +24 V source. Deriving the peak current from
Equation 6 yields:
Since the inductor energy of 5.54 µJ is greater than the PL/fOSC
requirement of 3.82 µJ, the 56 µH inductor will work in this
application.
2• 300 mA
5 + 0.5
12 −1.5 + 0.5
IPEAK
=
= 600 mA
The input voltage only varies between 4.5 V and 5.5 V in this
application. Therefore, the peak current will not change enough
to require an RLIM resistor and the ILIM pin can be connected
directly to VIN. Care should be taken, of course, to ensure that
the peak current does not exceed 650 mA.
0.5
Then, the peak current can be inserted into Equation 7 to
calculate the inductor value:
12 −1.5 − 5
600 mA
L =
• 7µs = 64 µH
CAPACITOR SELECTION
For optimum performance, the ADP1111’s output capacitor
must be selected carefully. Choosing an inappropriate capacitor
can result in low efficiency and/or high output ripple.
Since 64 µH is not a standard value, the next lower standard
value of 56 µH would be specified.
To avoid exceeding the maximum switch current when the
input voltage is at +24 V, an RLIM resistor should be specified.
Using the step-down curve of Figure 6, a value of 560 Ω will
limit the switch current to 600 mA.
Ordinary aluminum electrolytic capacitors are inexpensive but
often have poor Equivalent Series Resistance (ESR) and
Equivalent Series Inductance (ESL). Low ESR aluminum
capacitors, specifically designed for switch mode converter
applications, are also available, and these are a better choice
than general purpose devices. Even better performance can be
achieved with tantalum capacitors, although their cost is higher.
Very low values of ESR can be achieved by using OS-CON
capacitors (Sanyo Corporation, San Diego, CA). These devices
are fairly small, available with tape-and-reel packaging and have
very low ESR.
INDUCTOR SELECTION–POSITIVE-TO-NEGATIVE
CONVERTER
The configuration for a positive-to-negative converter using the
ADP1111 is shown in Figure 22. As with the step-up converter,
all of the output power for the inverting circuit must be supplied
by the inductor. The required inductor power is derived from
the formula:
The effects of capacitor selection on output ripple are demon-
strated in Figures 15, 16 and 17. These figures show the output
of the same ADP1111 converter that was evaluated with three
different output capacitors. In each case, the peak switch
current is 500 mA, and the capacitor value is 100 µF. Figure 15
shows a Panasonic HF-series 16-volt radial cap. When the
switch turns off, the output voltage jumps by about 90 mV and
then decays as the inductor discharges into the capacitor. The
rise in voltage indicates an ESR of about 0.18 Ω. In Figure 16,
the aluminum electrolytic has been replaced by a Sprague 293D
series, a 6 V tantalum device. In this case the output jumps
about 30 mV, which indicates an ESR of 0.06 Ω. Figure 17
shows an OS-CON 16–volt capacitor in the same circuit, and
ESR is only 0.02 Ω.
PL
=
V
+ VD • I
(Equation 8)
(
)
(
)
OUT
OUT
The ADP1111 power switch does not saturate in positive-to-
negative mode. The voltage drop across the switch can be
modeled as a 0.75 V base-emitter diode in series with a 0.65 Ω
resistor. When the switch turns on, inductor current will rise at
a rate determined by:
−R't
VL
R'
L
IL t =
( )
1− e
(Equation 9)
where: R' = 0.65 Ω + RL(DC)
VL = VIN – 0.75 V
For example, assume that a –5 V output at 50 mA is to be
generated from a +4.5 V to +5.5 V source. The power in the
inductor is calculated from Equation 8:
PL = |−5V|+0.5V| • 50 mA = 275 mW
(
) (
)
REV. 0
–8–
ADP1111
For most circuits, the 1N5818 is a suitable companion to the
ADP1111. This diode has a VF of 0.5 V at 1 A, 4 µA to 10 µA
leakage, and fast turn-on and turn-off times. A surface mount
version, the MBRS130T3, is also available.
For switch currents of 100 mA or less, a Shottky diode such as
the BAT85 provides a VF of 0.8 V at 100 mA and leakage less
than 1 µA. A similar device, the BAT54, is available in a SOT23
package. Even lower leakage, in the 1 nA to 5 nA range, can be
obtained with a 1N4148 signal diode.
General purpose rectifiers, such as the 1N4001, are not suitable
for ADP1111 circuits. These devices, which have turn-on times
of 10 µs or more, are far too slow for switching power supply
applications. Using such a diode “just to get started” will result
in wasted time and effort. Even if an ADP1111 circuit appears
to function with a 1N4001, the resulting performance will not
be indicative of the circuit performance when the correct diode
is used.
Figure 15. Aluminum Electrolytic
Figure 16. Tantalum Electrolytic
Figure 17. OS-CON Capacitor
CIRCUIT OPERATION, STEP-UP (BOOST) MODE
In boost mode, the ADP1111 produces an output voltage that is
higher than the input voltage. For example, +12 V can be gener-
ated from a +5 V logic power supply or +5 V can be derived
from two alkaline cells (+3 V).
Figure 18 shows an ADP1111 configured for step-up operation.
The collector of the internal power switch is connected to the
output side of the inductor, while the emitter is connected to
GND. When the switch turns on, pin SW1 is pulled near
ground. This action forces a voltage across L1 equal to
V
IN – VCE(SAT), and current begins to flow through L1. This
current reaches a final value (ignoring second-order effects) of:
V
IN
− V
IPEAK
CE (SAT) • 7µs
L
where 7 µs is the ADP1111 switch’s “on” time.
D1
1N5818
L1
V
V
IN
OUT
R3
(OPTIONAL)
2
1
R2
R1
I
V
IN
LIM
3
8
SW1
FB
+
ADP1111
C1
GND SW2
5
4
If low output ripple is important, the user should consider the
ADP3000. Because this device switches at 400 kHz, lower peak
current can be used. Also, the higher switching frequency
simplifies the design of the output filter. Consult the ADP3000
data sheet for additional details.
Figure 18. Step-Up Mode Operation
When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes, current begins to flow
through D1 into the load, and the output voltage is driven above
the input voltage.
DIODE SELECTION
The output voltage is fed back to the ADP1111 via resistors R1
and R2. When the voltage at pin FB falls below 1.25 V, SW1
turns “on” again, and the cycle repeats. The output voltage is
therefore set by the formula:
In specifying a diode, consideration must be given to speed,
forward voltage drop and reverse leakage current. When the
ADP1111 switch turns off, the diode must turn on rapidly if
high efficiency is to be maintained. Shottky rectifiers, as well as
fast signal diodes such as the 1N4148, are appropriate. The
forward voltage of the diode represents power that is not
delivered to the load, so VF must also be minimized. Again,
Schottky diodes are recommended. Leakage current is especially
important in low-current applications where the leakage can be
a significant percentage of the total quiescent current.
R2
VOUT = 1. 25 V • 1 +
R1
The circuit of Figure 18 shows a direct current path from VIN to
OUT, via the inductor and D1. Therefore, the boost converter
is not protected if the output is short circuited to ground.
V
REV. 0
–9–
ADP1111
V
IN
CIRCUIT OPERATION, STEP DOWN (BUCK) MODE)
The ADP1111’s step down mode is used to produce an output
voltage that is lower than the input voltage. For example, the
output of four NiCd cells (+4.8 V) can be converted to a +3 V
logic supply.
+
R
3
C
2
2
3
1
V
OUT
I
V
SW1
LIM
IN
8
4
FB
D2
L1
R2
ADP1111
SW2
A typical configuration for step down operation of the ADP1111
is shown in Figure 19. In this case, the collector of the internal
power switch is connected to VIN and the emitter drives the
inductor. When the switch turns on, SW2 is pulled up towards
VIN. This forces a voltage across L1 equal to VIN – VCE – VOUT
and causes current to flow in L1. This current reaches a final
value of:
GND
+
5
C
1
D1
R1
D1, D2 = 1N5818 SCHOTTKY DIODES
Figure 20. Step-Down Model, VOUT > 6.2 V
If the input voltage to the ADP1111 varies over a wide range, a
current limiting resistor at Pin 1 may be required. If a particular
circuit requires high peak inductor current with minimum input
supply voltage, the peak current may exceed the switch maxi-
mum rating and/or saturate the inductor when the supply
voltage is at the maximum value. See the “Limiting the Switch
Current” section of this data sheet for specific recommendations.
V
−VCE −VOUT
IN
IPEAK
• 7µs
L
where 7 µs is the ADP1111 switch’s “on” time.
V
IN
+
R
LIM
100Ω
C
2
2
3
1
I
V
IN
SW1
LIM
8
4
FB
INCREASING OUTPUT CURRENT IN THE STEP-DOWN
REGULATOR
L1
ADP1111
V
OUT
SW2
AO SET GND
+
R2
R1
Unlike the boost configuration, the ADP1111’s internal power
switch is not saturated when operating in step-down mode. A
conservative value for the voltage across the switch in step-down
mode is 1.5 V. This results in high power dissipation within the
ADP1111 when high peak current is required. To increase the
output current, an external PNP switch can be added (Figure
21). In this circuit, the ADP1111 provides base drive to Q1
through R3, while R4 ensures that Q1 turns off rapidly. Because
the ADP1111’s internal current limiting function will not work
in this circuit, R5 is provided for this purpose. With the value
shown, R5 limits current to 2 A. In addition to reducing power
dissipation on the ADP1111, this circuit also reduces the switch
voltage. When selecting an inductor value for the circuit of
Figure 21, the switch voltage can be calculated from the
formula:
6
7
5
D1
1N5818
C
L
NC NC
Figure 19. Step-Down Mode Operation
When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes, and the switch side of the
inductor is driven below ground. Schottky diode D1 then turns
on, and current flows into the load. Notice that the Absolute
Maximum Rating for the ADP1111’s SW2 pin is 0.5 V below
ground. To avoid exceeding this limit, D1 must be a Schottky
diode. If a silicon diode is used for D1, Pin SW2 can go to
–0.8 V, which will cause potentially damaging power dissipation
within the ADP1111.
VSW = VR5 + VQ1(SAT) 0.6 V + 0.4 V
1 V
The output voltage of the buck regulator is fed back to the
ADP1111’s FB pin by resistors R1 and R2. When the voltage at
pin FB falls below 1.25 V, the internal power switch turns “on”
again, and the cycle repeats. The output voltage is set by the
formula:
INPUT
R5
0.3Ω
R4
220Ω
+
Q1
C
INPUT
MJE210
R3
330Ω
1
L1
I
LIM
3
8
SW1
OUTPUT
R2
VOUT = 1. 25 V • 1 +
R1
R2
R1
ADP1111
2
V
IN
FB
AO SET GND SW2
+
6
7
5
4
C
L
D1
1N5821
When operating the ADP1111 in step-down mode, the output
voltage is impressed across the internal power switch’s emitter-
base junction when the switch is off. To protect the switch, the
output voltage should be limited to 6.2 V or less. If a higher
output voltage is required, a Schottky diode should be placed in
series with SW2 as shown in Figure 20.
NC NC
Figure 21. High Current Step-Down Operation
REV. 0
–10–
ADP1111
Table I. Component Selection for Typical Converters
Input
Voltage
Output
Voltage
Output
Current (mA)
Circuit
Figure
Inductor
Value
Inductor
Part No.
Capacitor
Value
Notes
2 to 3.1
2 to 3.1
2 to 3.1
2 to 3.1
5
5
5
90 mA
10 mA
30 mA
10 mA
90 MA
30 mA
50 mA
300 mA
300 mA
7 mA
4
4
4
4
4
4
5
5
5
6
6
15 µH
47 µH
15 µH
47 µH
33 µH
47 µH
15 µH
56 µH
120 µH
56 µH
120 µH
CD75-150K
CTX50-1
CD75-150K
CTX50-1
CD75-330K
CTX50-1
33 µF
10 µF
22 µF
10 µF
22 µF
15 µF
47 µF
47 µF
47 µF
47 µF
100 µF
*
12
12
12
12
5
5
5
–5
–5
5
6.5 to 11
12 to 20
20 to 30
5
**
**
**
CTX50-4
CTX100-4
CTX50-4
CTX100-4
12
250 mA
**
NOTES
CD = Sumida.
CTX = Coiltronics.
**Add 47 Ω from ILIM to VIN
.
**Add 220 Ω from ILIM to VIN
.
also reduces the circuit’s output voltage sensitivity to tempera-
ture, which otherwise would be dominated by the –2 mV VBE
contribution of Q1. The output voltage for this circuit is
determined by the formula:
POSITIVE-TO-NEGATIVE CONVERSION
The ADP1111 can convert a positive input voltage to a negative
output voltage as shown in Figure 22. This circuit is essentially
identical to the step-down application of Figure 19, except that
the “output” side of the inductor is connected to power ground.
When the ADP1111’s internal power switch turns off, current
flowing in the inductor forces the output (–VOUT) to a negative
potential. The ADP1111 will continue to turn the switch on
until its FB pin is 1.25 V above its GND pin, so the output
voltage is determined by the formula:
R2
VOUT = 1. 25 V •
R1
Unlike the positive step-up converter, the negative-to-positive
converter’s output voltage can be either higher or lower than the
input voltage.
D1
1N5818
R2
VOUT = 1. 25 V • 1 +
R1
L1
POSITIVE
OUTPUT
R2
+
R
LIM
C
L
D2
Q1
MJE210
2N3906
INPUT
2
1
+
+
C2
I
V
IN
R
LIM
LIM
3
8
SW1
FB
C
INPUT
ADP1111
2
3
1
10kΩ
L1
OUTPUT
I
V
IN
SW1
LIM
AO SET GND SW2
4
8
SW2
6
7
5
4
R1
ADP1111
R2
R1
FB
AO SET GND
NC NC
NEGATIVE
INPUT
+
6
7
5
D1
1N5818
C
L
NC NC
NEGATIVE
OUTPUT
Figure 23. ADP1111 Negative-to-Positive Converter
LIMITING THE SWITCH CURRENT
Figure 22. Positive-to-Negative Converter
The ADP1111’s RLIM pin permits the switch current to be
limited with a single resistor. This current limiting action occurs
on a pulse by pulse basis. This feature allows the input voltage
to vary over a wide range without saturating the inductor or
exceeding the maximum switch rating. For example, a particular
design may require peak switch current of 800 mA with a 2.0 V
input. If VIN rises to 4 V, however, the switch current will
exceed 1.6 A. The ADP1111 limits switch current to 1.5 A and
thereby protects the switch, but the output ripple will increase.
Selecting the proper resistor will limit the switch current to
800 mA, even if VIN increases. The relationship between RLIM
and maximum switch current is shown in Figure 6.
The design criteria for the step-down application also apply to
the positive-to-negative converter. The output voltage should be
limited to |6.2 V| unless a diode is inserted in series with the
SW2 pin (see Figure 20.) Also, D1 must again be a Schottky
diode to prevent excessive power dissipation in the ADP1111.
NEGATIVE-TO-POSITIVE CONVERSION
The circuit of Figure 23 converts a negative input voltage to a
positive output voltage. Operation of this circuit configuration is
similar to the step-up topology of Figure 18, except the current
through feedback resistor R2 is level-shifted below ground by a
PNP transistor. The voltage across R2 is VOUT –VBEQ1. How-
ever, diode D2 level-shifts the base of Q1 about 0.6 V below
ground thereby cancelling the VBE of Q1. The addition of D2
The ILIM feature is also valuable for controlling inductor current
when the ADP1111 goes into continuous-conduction mode.
REV. 0
–11–
ADP1111
R
LIM
This occurs in the step-up mode when the following condition is
met:
(EXTERNAL)
V
IN
I
V
LIM
IN
VOUT + VDIODE
VIN − VSW
1
80Ω
(INTERNAL)
R1
<
1 − DC
Q3
I
Q1
ADP1111
where DC is the ADP1111’s duty cycle. When this relationship
exists, the inductor current does not go all the way to zero
during the time that the switch is OFF. When the switch turns
on for the next cycle, the inductor current begins to ramp up
from the residual level. If the switch ON time remains constant,
the inductor current will increase to a high level (see Figure 24).
This increases output ripple and can require a larger inductor
and capacitor. By controlling switch current with the ILIM
resistor, output ripple current can be maintained at the design
values. Figure 25 illustrates the action of the ILIM circuit.
200
SW1
SW2
Q1
DRIVER
72kHz
OSC
Q2
POWER
SWITCH
Figure 26. ADP1111 Current Limit Operation
The delay through the current limiting circuit is approximately
1 µs. If the switch ON time is reduced to less than 3 µs, accuracy
of the current trip-point is reduced. Attempting to program a
switch ON time of 1 µs or less will produce spurious responses
in the switch ON time; however, the ADP1111 will still provide
a properly regulated output voltage.
PROGRAMMING THE GAIN BLOCK
The gain block of the ADP1111 can be used as a low-battery
detector, error amplifier or linear post regulator. The gain block
consists of an op amp with PNP inputs and an open-collector
NPN output. The inverting input is internally connected to the
ADP1111’s 1.25 V reference, while the noninverting input is
available at the SET pin. The NPN output transistor will sink
about 300 µA.
200mA/div
Figure 27a shows the gain block configured as a low-battery
monitor. Resistors R1 and R2 should be set to high values to
reduce quiescent current, but not so high that bias current in
the SET input causes large errors. A value of 33 kΩ for R2 is a
good compromise. The value for R1 is then calculated from the
formula:
Figure 24.
VLOBATT − 1. 25 V
R1 =
1. 25 V
R2
where VLOBATT is the desired low battery trip point. Since the
gain block output is an open-collector NPN, a pull-up resistor
should be connected to the positive logic power supply.
200mA/div
5V
R
47k
L
V
IN
Figure 25.
ADP1111
The internal structure of the ILIM circuit is shown in Figure 26.
Q1 is the ADP1111’s internal power switch that is paralleled by
sense transistor Q2. The relative sizes of Q1 and Q2 are scaled
so that IQ2 is 0.5% of IQ1. Current flows to Q2 through an
internal 80 Ω resistor and through the RLIM resistor. These two
resistors parallel the base-emitter junction of the oscillator-
disable transistor, Q3. When the voltage across R1 and RLIM
exceeds 0.6 V, Q3 turns on and terminates the output pulse. If
only the 80 Ω internal resistor is used (i.e. the ILIM pin is
connected directly to VIN), the maximum switch current will be
1.5 A. Figure 6 gives RLIM values for lower current-limit values.
R1
R2
1.25V
REF
V
BAT
AO
TO
SET
PROCESSOR
GND
33k
V
–1.25V
LB
R1= –––––––––
35.1µA
V
LB
= BATTERY TRIP POINT
Figure 27a. Setting the Low Battery Detector Trip Point
REV. 0
–12–
ADP1111
The circuit of Figure 27b may produce multiple pulses when
approaching the trip point due to noise coupled into the SET
input. To prevent multiple interrupts to the digital logic,
hysteresis can be added to the circuit (Figure 27). Resistor
RHYS, with a value of 1 MΩ to 10 MΩ, provides the hysteresis.
The addition of RHYS will change the trip point slightly, so the
new value for R1 will be:
9 V to 5 V Step-Down Converter
This circuit uses a 9 V battery to generate a +5 V output. The
circuit will work down to 6.5 V, supplying 50 mA at this lower
limit. Switch current is limited to around 500 mA by the 100 Ω
resistor.
INPUT
R
LIM
100Ω
9V
VLOBATT − 1. 25 V
R1 =
L1
2
3
1
CTX15-4
1. 25 V
R2
V L − 1. 25 V
RL + RHYS
I
V
SW1
LIM
IN
−
4
8
OUTPUT
ADP1111-5SW2
15µH
(9V TO 5V @ 150mA,
IN
6.5V TO 5V @ 50mA)
IN
SENSE
AO SET GND
where VL is the logic power supply voltage, RL is the pull-up
resistor, and RHYS creates the hysteresis.
+
6
7
5
D1
1N5818
C
L
22µF
5V
NC NC
R
47k
L
V
IN
Figure 29. 9 V to 5 V Step-Down Converter
20 V to 5 V Step-Down Converter
This circuit is similar to Figure 29, except it supplies much
higher output current and operates over a much wider range of
input voltage. As in the previous examples, switch current is
limited to 500 mA.
ADP1111
R1
R2
1.25V
REF
V
BAT
AO
TO
SET
PROCESSOR
GND
1.6M
33k
R
HYS
12V TO 28V
INPUT
R
LIM
100Ω
Figure 27b.
APPLICATION CIRCUITS
All Surface Mount 3 V to 5 V Step-Up Converter
This is the most basic application (along with the basic step-
down configuration to follow) of the ADP1111. It takes full
advantage of surface mount packaging for all the devices used in
the design. The circuit can provide +5 V at 100 mA of output
current and can be operated off of battery power for use in
portable equipment.
L1
CTX68-4
2
3
1
I
V
IN
SW1
LIM
4
8
OUTPUT
(+5V @ 300mA)
SW2
68µH
ADP1111-5
SENSE
AO SET GND
+
6
7
5
D1
1N5818
C
L
47µF
NC NC
Figure 30. 20 V to 5 V Step-Down Converter
+5 V to –5 V Converter
This circuit is essentially identical to Figure 22, except it uses a
fixed-output version of the ADP1111 to simplify the design
somewhat.
D1
L1
INPUT +3V
OUTPUT
(5V @ 100mA)
20µH
CTX20-4
MBRS120T3
R3*
(OPTIONAL)
2
1
I
V
IN
LIM
3
8
SW1
12V TO 28V
INPUT
+
C
33µF
ADP1111-5
L
R
LIM
100Ω
SENSE
AO SET GND SW2
L1
CTX33-2
2
3
1
6
7
5
4
I
V
IN
SW1
LIM
4
8
SW2
33µH
ADP1111-5
SENSE
AO SET GND
NC NC
+
6
7
5
D1
1N5818
C
33µF
L
Figure 28. All Surface Mount +3 V to +5 V Step-Up Converter
NC NC
–5V
@ 75mA
Figure 31. +5 V to –5 V Converter
REV. 0
–13–
ADP1111
Voltage-Controlled Positive-to-Negative Converter
High Power, Low Quiescent Current Step-Down Converter
By making use of the fact that the feedback pin directly controls
the internal oscillator, this circuit achieves a shutdown-like state
by forcing the feedback pin above the 1.25 V comparator
threshold. The logic level at the 1N4148 diode anode needs to
be at least 2 V for reliable standby operation.
By including an op amp in the feedback path, a simple positive-
to-negative converter can be made to give an output that is a
linear multiple of a controlling voltage, Vc. The op amp, an
OP196, rail-to-rail input and output amplifier, sums the
currents from the output and controlling voltage and drives the
FB pin either high or low, thereby controlling the on-board
oscillator. The 0.22 Ω resistor limits the short-circuit current to
about 3 A and, along with the BAT54 Schottky diode, helps
limit the peak switch current over varying input voltages. The
external power switch features an active pull-up to speed up the
turn-off time of the switch. Although an IRF9530 was used in
the evaluation, almost any device that can handle at least 3 A of
peak current at a VDS of at least 50 V is suitable for use in this
application, provided that adequate attention is paid to power
dissipation. The circuit can deliver 2 W of output power with a
+6-volt input from a control voltage range of 0 V to 5 V.
The external switch driver circuit features an active pull-up
device, a 2N3904 transistor, to ensure that the power MOSFET
turns off quickly. Almost any power MOSFET will do as the
switch as long as the device can withstand the 18 volt VGS and is
reasonably robust. The 0.22 Ω resistor limits the short-circuit
current to about 3 A and, along with the BAT54 Schottky
diode, helps to limit the peak switch current over varying input
voltages.
+8V TO +18V
LI
20µH
+5V
500mA
IRF9540
S
D
R
INPUT
LIM
0.22
+
BAT54
G
C
L
+5V TO +12V
D1
2k
R
LIM
220µF
IRF9530
IN5821
2N3904
51Ω
INPUT
2
1
0.22Ω
V
IN
I
LIM
BAT54
3
8
SW1
FB
2kΩ
51Ω
1N4148
121kΩ
ADP1111
2N3904
1
2
L1
20µH
AO SET GND SW2
V
IN
I
LIM
3
SW1
FB
CTX20-4
6
7
5
4
1N4148
D1
IN5821
C
L
ADP1111
+
40.2kΩ
V
IN
8
47µF
35V
NC NC
OPERATE/STANDBY
200kΩ
AO SET GND SW2
2
3
7
1kΩ
6
6
7
5
4
OUTPUT
= –5.13 *V
2V ≤ V ≤ 5
IN
–V
OUT
2W MAXIMUM OUTPUT
C
4
1N5231
NC NC
LI = COILTRONICS CTX20-4
1N4148
39kΩ (0V TO +5V)
V
C
Figure 34. High Power, Low Quiescent Current Step-Down
Converter
Figure 32. Voltage Controlled Positive-to-Negative
Converter
NOTES
1. All inductors referenced are Coiltronics CTX-series except
where noted.
+3 V to –22 V LCD Bias Generator
This circuit uses an adjustable-output version of the ADP1111
to generate a +22.5 V reference output that is level-shifted to
give an output of –22 V. If operation from a +5 volt supply is
desired, change R1 to 47 ohms. The circuit will deliver 7 mA
with a 3 volt supply and 40 mA with a 5 volt supply.
2. If the source of power is more than an inch or so from the
converter, the input to the converter should be bypassed with
approximately 10 µF of capacitance. This capacitor should
be a good quality tantalum or aluminum electrolytic.
D1
1N4148
L1
OUTPUT
25µH
R
LIM
100Ω
+3V
2
1
732kΩ
42.2kΩ
I
V
IN
LIM
3
SW1
FB
C
0.1µF
L
ADP1111
2xAA
CELLS
8
4.7
+
AO SET GND SW2
µF
6
7
5
4
1N5818
NC NC
L1 = CTX25-4
+
22µF
1N5818
–22V OUTPUT
7mA @ 2V INPUT
Figure 33. 3 V to –22 V LCD Bias Generator
REV. 0
–14–
ADP1111
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
8-Lead Plastic DIP
(N-8)
0.430 (10.92)
0.348 (8.84)
8
5
4
0.280 (7.11)
0.240 (6.10)
1
0.325 (8.25)
0.300 (7.62)
0.060 (1.52)
0.015 (0.38)
PIN 1
0.195 (4.95)
0.115 (2.93)
0.210 (5.33)
MAX
0.130
(3.30)
MIN
0.160 (4.06)
0.115 (2.93)
0.015 (0.381)
0.008 (0.204)
SEATING
PLANE
0.100
(2.54)
BSC
0.022 (0.558)
0.014 (0.356)
0.070 (1.77)
0.045 (1.15)
8-Lead SOIC
(SO-8)
0.1968 (5.00)
0.1890 (4.80)
8
1
5
4
0.1574 (4.00)
0.1497 (3.80)
0.2440 (6.20)
0.2284 (5.80)
PIN 1
0.0688 (1.75)
0.0532 (1.35)
0.0196 (0.50)
x 45°
0.0098 (0.25)
0.0040 (0.10)
0.0099 (0.25)
8°
0°
0.0500
(1.27)
BSC
0.0192 (0.49)
0.0138 (0.35)
SEATING
PLANE
0.0098 (0.25)
0.0075 (0.19)
0.0500 (1.27)
0.0160 (0.41)
REV. 0
–15–
–16–
相关型号：
ADP1111ARZ-12-REEL
Micropower, Step-Up/Step-Down SW Regulator; Adjustable and Fixed 3.3 V, 5 V, 12 V
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ADP1111ARZ-REEL
Micropower, Step-Up/Step-Down SW Regulator; Adjustable and Fixed 3.3 V, 5 V, 12 V
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