AN-940_技术文档

AN-940

APPLICATION NOTE

One Technology Way • P. O. Box 9106 • Norwood, MA 02062-9106, U.S.A. • Tel: 781.329.4700 • Fax: 781.461.3113 • www.analog.com

Low Noise Amplifier Selection Guide for Optimal Noise Performance

by Paul Lee

Noise from surrounding circuit components must be accounted

for. At temperatures above absolute zero, all resistances act as

noise sources due to thermal movement of charge carriers called

Johnson noise or thermal noise. This noise increases with resis-

tance, temperature, and bandwidth. Voltage noise is shown in

Equation 1.

INTRODUCTION

When evaluating an amplifier’s performance for a low noise

application, both internal and external noise sources must be

considered. This application note briefly discusses the funda-

mentals of both internal and external noise and identifies the

tradeoffs associated in selecting the optimal amplifier for low

noise design.

V_n 4kTBR

where:

(1)

EXTERNAL NOISE SOURCES

External noise includes any type of external influences, such

as external components and electrical/electromagnetic interfer-

ence. Interference is defined as any unwanted signals arriving

as either voltage or current, at any of the amplifier’s terminals

or induced in its associated circuitry. It can appear as spikes,

steps, sine waves, or random noise. Interference can come from

anywhere: machinery, nearby power lines, RF transmitters or

receivers, computers, or even circuitry within the same equip-

ment (that is, digital circuits or switching-type power supplies).

If all interference is eliminated by careful design and/or layout

of the board, there can still be random noise associated with the

amplifier and its circuit components.

V_nis voltage noise.

k is Boltzmann’s constant (1.38 × 10⁻²³J/K).

T is the temperature in Kelvin (K).

B is the bandwidth in hertz (Hz).

R is the resistance in ohms (Ω).

Current noise (noise associated with current flow) is shown in

Equation 2

4kTB

R

I_n

where:

(2)

I_nis current noise.

k is Boltzmann’s constant (1.38 × 10⁻²³J/K).

T is the temperature in Kelvin (K).

B is the bandwidth in hertz (Hz).

R is the resistance in ohms (Ω).

Rev. D | Page 1 of 12

AN-940

Application Note

TABLE OF CONTENTS

Introduction ...................................................................................... 1

Popcorn Noise ...............................................................................5

Summing the Noise Sources ........................................................5

Noise Gain......................................................................................6

Selecting Low Noise Op Amp..........................................................7

Conclusion..........................................................................................9

References........................................................................................ 12

External Noise Sources .................................................................... 1

Internal Noise Sources ..................................................................... 3

Input-Referred Voltage Noise ..................................................... 4

Input-Referred Current Noise .................................................... 4

Flicker Noise.................................................................................. 5

Rev. D | Page 2 of 12

Application Note

AN-940

Resistors

INTERNAL NOISE SOURCES

For the purposes of this application note, the resistor noise is

limited to thermal (Johnson) noise. To keep a low level of this

type of noise, resistance values should be as low as possible

because RMS voltage of thermal (Johnson) noise is proportional

to the square root of the resistor value. For example, a 1 kΩ

resistor has a thermal noise of ~4 nV/√Hz at room temperature.

Noise appearing at the amplifier’s output is usually measured as

a voltage. However, it is generated by both voltage and current

sources. All internal sources are generally referred to the input,

that is, treated as uncorrelated or independent random noise

generators in series or in parallel with the inputs of an ideal

noise-free amplifier (see Figure 1). Because these noise sources

are considered random and/or exhibit Gaussian distribution

behavior, it is important to take proper care when summing the

noise sources as discussed in the Summing the Noise Sources

section.

For an in-depth analysis and low noise designs, other types of

resistor noise should be accounted for, such as contact noise

and shot noise. A few practical notes follow and they should

be considered when selecting a resistor.

If the same noise appears at two or more points in a circuit (that

is, input bias current cancellation circuitry), the two noise sources

are correlated noise sources and a correlation coefficient factor

should be included in the noise analysis. Further analysis of

correlated noise is limited in this application note as typical

correlation noise sources are less than 10% to 15% and they

can usually be disregarded.



Choose the largest practical wattage resistors, as the contact

noise is decreased with a larger volume of material.

Choose low noise resistive element material

Resistive elements composed of pure metals and/or

metal alloys in bulk exhibits low noise characteristics.

Such as Vishay Bulk Metal® foil technology resistors

(such as, S102C, Z201)



Wirewound technology resistors composed of metal

alloys have similar noise characteristics as Bulk Metal

foil technology, but are much more inductive.

Metal film technology resistors as thin film are noisier

than Bulk Metal foil or wirewound technology resistors

because of significant noise contributions from occlusions,

surface imperfections, and nonuniform depositions.

Thick film and carbon composition resistors are the

nosiest resistors.

Internal amplifier noise falls into four categories:



Input-referred voltage noise

Input-referred current noise

Flicker noise

Popcorn noise

Input-referred voltage noise and input-referred current noise

are the most common specifications used for amplifier noise

analysis. They are often specified as an input-referred spectral

density function or the rms noise contained in Δf bandwidth

and usually given in terms of nV/√Hz (for voltage noise) or

pA/√Hz (for current noise). The /√Hz is needed because the

noise power adds with (is cumulative over) bandwidth (Hz) or

the voltage and current noise density adds with square root of

the bandwidth (√Hz) (see Equation 1 and Equation 2).

e_n



Reactances

Reactances, such as capacitors and inductors, do not generate

noise, but the noise current through reactances develops noise

voltage as well as the associated parasitic.

Practical Tips

Output noise from a circuit can be reduced by lowering the

total component resistance or by limiting the circuit bandwidth.

Temperature reduction is generally not very helpful unless a

resistor can be made very cold, because noise power is propor-

tional to the absolute temperature,

–

+

i_n

R

S

–

i_n

T(x) in Kelvin = x°C + 273.15°

(3)

R

1

All resistors in a circuit generate noise. The effect of generated

noise must always be considered. In practice, only resistors in

the input and feedback paths (typically in high gain configu-

rations) are likely to have an appreciable effect on total circuit

noise. The noise can be considered as coming from either

current sources or voltage sources (whichever is more conve-

nient in a given circuit).

R

2

Figure 1. Op Amp Noise Model

Rev. D | Page 3 of 12

AN-940

Application Note

INPUT-REFERRED VOLTAGE NOISE

INPUT-REFERRED CURRENT NOISE

Input-referred voltage noise (e_n) is typically viewed as a noise

voltage source.

Input-referred current noise (i_n) is typically seen as two noise

current sources pumping currents through the two differential

input terminals.

Voltage noise is the noise specification that is usually empha-

sized; however, if input impedance levels are high, current noise

is often the limiting factor in system noise performance. It is

analogous to offsets, where the input offset voltage often bears

the blame for output offset, when in reality the bias current

causes the output offset where input impedances are high.

Shot noise (sometimes called Schottky noise) is current noise

due to random distribution of charge carriers in the current flow

through a potential barrier, such as a PN junction. The shot

noise current, i_n, is obtained from the formula

i_n 2I_BqB

(4)

Note the following points about input-referred voltage noise:

where:



Op amp voltage noise can be lower than 1 nV/√Hz for the

highest performance amplifiers.

I_Bis the bias current in ampere (A).

q is the electron charge in coulomb (1.6 × 10⁻¹⁹C).

B is the bandwidth in hertz (Hz).



Although bipolar op amps traditionally have less voltage

noise than FET op amps, they also have substantially

greater current noise.

The current noise of a simple bipolar and JFET op amp is typically

within 1 dB or 2 dB of the shot noise of the input bias current.

This specification is not always listed on data sheets.



Bipolar amplifier noise characteristics are dependent on

the quiescent current.

Present day FET op amps are capable of obtaining both low

current noise and voltage noise similar to bipolar amplifier

performance, though not as low as the best bipolar input

amplifiers.

Note the following points regarding input-referred noise:



The current noise of typical bipolar transistor op amps,

such as the OP27, is about 400 fA/√Hz, where I_Bis 10 nA,

and does not vary much with temperature except for bias,

current-compensated amplifiers.



The current noise of JFET input op amps (such as the

AD8610: 5 fA/√Hz at I_B= 10 pA) while lower, doubles

for every 20°C chip temperature increase, because JFET

op amp bias currents double for every 10°C increase.

Traditional voltage feedback op amps with balanced inputs

usually have equal (correlated and uncorrelated) current

noise on both their inverting and noninverting inputs.

Many amplifiers, especially those amps with input bias

current cancellation circuits, have considerably larger

correlated than uncorrelated noise components. Overall,

noise can be improved by adding an impedance-balancing

resistor (matching impedances on both positive and

negative input pins).



Rev. D | Page 4 of 12

Application Note

AN-940

FLICKER NOISE

POPCORN NOISE

The noise of op amps is Gaussian with constant spectral density

(white noise), over a wide range of frequencies. As frequency

decreases, the spectral density starts to rise because of the fabri-

cation process, the IC device layout, and the device type at a

rate of about 3 dB per octave for CMOS amplifiers, 3.5 dB to

4.5 dB per octave for bipolar amplifiers, or up to 5 dB per

octave for JFET amplifiers.

Popcorn noise (not specified or advertised) is an abrupt shift in

offset voltage or current lasting for several milliseconds with

amplitude from several microvolts to hundreds of microvolts.

This burst or pop is random. Low temperatures and high source

resistances usually produce the most favorable conditions for

popcorn noise. Although the root cause of popcorn noise is

not absolute, both metallic contamination and internal or

surface defects in the silicon lattice can cause popcorn noise

in ICs. Although considerable work has been done to reduce

the sources of popcorn noise in modern wafer fabrication, it

cannot be eliminated. Further analysis of popcorn noise is

beyond the scope of this application note.

This low frequency noise characteristic is known as flicker

noise or 1/f noise because the noise power spectral density

goes inversely with frequency (1/f). It has a −1 slope on a log

plot. The frequency at which an extrapolated −3 dB per octave

(for a CMOS-type amplifier) spectral density line intersects the

broadband constant spectral density value is known as the 1/f

corner frequency and is a figure of merit for the amplifier (see

Figure 2). Bipolar and JFET amplifiers typically have lower 1/f

corner frequency than CMOS amplifiers.

SUMMING THE NOISE SOURCES

If the noise sources are uncorrelated (that is, one noise signal

cannot be transformed into the other), the resulting noise is

not their arithmetic sum, but the square root of the sum of

their squares.

100

2

V_{ni, TOTAL}

where:



(e_n)² (R  i_n)²V_n(R_EX

)

(5)

S

EXTRAPOLATED 1/f

SPECTRAL NOISE DENSITY

10

V

_{ni, TOTAL}is the total noise referred-to-input (RTI).

e_nis input-referred voltage noise.

i_nis input-referred current noise.

R_Sis an equivalent source or input resistance to the amplifier.

V_n(R_EX) is voltage noise from external circuitry.

1

EXTRAPOLATED

CONSTANT SPECTRAL

NOISE DENSITY

Note the following:

1/f CORNER FREQUENCY

0.1



Any resistance in the noninverting input has Johnson noise

and converts current noise to a voltage noise.

Johnson noise in feedback resistors can be significant in

high resistance circuits.

0.1

1

10

100

1k

10k

FREQUENCY (Hz)



Figure 2. Spectral Noise Density

Figure 3 visually shows the Equation 5 as the summation of

vectors by using the Pythagorean Theorem.

V

ni, TOTAL

V

(R )

EX

n

R

× i_n

S

e_n

Figure 3. Vector Summation of Noise Sources

Rev. D | Page 5 of 12

AN-940

Application Note

OP AMP

NOISE MODEL

COMBINED RTI NOISE

(V

)

ni, TOTAL

e_n

–

+

–

+

–

i_n

RESISTOR

NOISE

R

S

i_n

RESISTOR

NOISE

R

1

RESISTOR

NOISE

R

2

Figure 4. Simplifying the Amplifier Noise Circuit

In some cases, the noise gain and the signal gain are not equiv-

alent (see Figure 5). Note that the closed-loop bandwidth is

determined by dividing the gain bandwidth product (or unity

gain frequency) by the noise gain of the amplifier circuit.

NOISE GAIN

The noises previously discussed can be grouped into referred-

to-input (RTI) noise of the amplifier circuit. To calculate the

total output noise of the amplifier circuit, the total combined

noise on the input must be multiplied by the amplifier circuit’s

noise gain. Noise gain is the gain of the amplifier’s circuit for

referred-to-input noise and it is typically used to determine the

stability of the amplifier circuit.

NOISE

SOURCE

+

CASE 1:

SIGNAL

SOURCE

–

To simplify the noise gain calculation, the noise sources in the

simple amplifier circuit in Figure 1 can be reduced to a single

total RTI noise source (V_{ni, TOTAL}), as shown in Figure 4. It is a

common practice to lump the total combined RTI noise to the

noninverting input of the amplifier.

R

1

R

2

CASE 2:

SIGNAL

SOURCE

V_{no, TOTAL} G_NV_{ni, TOTAL}

Figure 5. Signal Gain vs. Noise Gain

where:

Case 1: In a noninverting configuration, both the signal gain

and the noise gain are equal to 1 + R₁/R₂.

V_{no, TOTAL}is the total referred-to-output (RTO) noise.

V_{ni, TOTAL}is the total referred-to-input (RTI) noise

Case 2: In an inverting configuration, signal gain is equal to

−(R₁/R₂), but the noise gain is still equal to 1 + R₁/R₂.

R₁

G_N1

R₂

where:

G_Nis the noise gain.

R₁is the feedback equivalent impedance.

R₂is the gain setting equivalent impedance.

Rev. D | Page 6 of 12

Application Note

AN-940

SELECTING LOW NOISE OP AMP

If an op amp is driven with a source resistance, the equivalent

noise input becomes the square root of the sum of the squares of

the amplifier’s voltage noise, the voltage generated by the source

resistance, and the voltage caused by the amplifiers current

noise flowing through the source impedance.

An amplifier can be selected where its noise contribution is

negligible compared to the source resistance by using a figure

of merit, R_{S, OP}, of an op amp. It can be calculated by using an

amplifier’s noise specification.

e_n

i_n

(7)

R_{S, OP}



For very low source resistances, the noise generated by the

source resistance and amplifier current noise contribute

insignificantly to the total. In this case, the noise at the

input is effectively only the voltage noise of the op amp.

where:

e_nis input-referred voltage noise.

i_nis input-referred current noise.

If the source resistance is high, the Johnson noise of the source

resistance may dominate both the op amp voltage noise and the

voltage due to the current noise. However, note that, because the

Johnson noise only increases with the square root of the resis-

tance, while the noise voltage due to the current noise is directly

proportional to the input impedance, the amplifier’s current noise

always dominates for a high enough value of input impedance.

When an amplifier’s voltage and current noise are high enough,

there may be no value of input resistance for which Johnson

noise dominates.

Figure 6 shows a comparison of the voltage noise density of

a number of high voltage (up to 44 V) op amps from Analog

Devices, Inc., vs. R_{S, OP}at 1 kHz. The diagonal line plots the

Johnson noise associated with resistance.

100

f = 1kHz

JOHNSON NOISE LINE

OF SOURCE RESISTANCE

AD8622/AD8624

OP285

OP467

AD8610/

AD8620

OP271

10

OPx177

OP275

OP213

OPx84

OP27/OP37

OP270

AD743/AD745

ADA4004

ADA4075-2

AD8597/AD8599

1

AD797

0.1

10

100

1k

10k

100k

1M

SOURCE RESISTANCE (Ω)

Figure 6. Analog Devices Op Amp Noise Plot

Rev. D | Page 7 of 12

AN-940

Application Note

Similar types of graph can be constructed for a chosen frequency

from the data in the op amp data sheet (see Figure 8). For example,

the AD8599 has an input-referred voltage noise of 1.07 nV/√Hz

and an input-referred current noise of 2.3 pA/√Hz at 1 kHz. The

2. Locate the given source resistance, such as 1 kꢀ, on the

Johnson noise line.

3. Create a horizontal line from the point located in Step 2 to

the right of the plot.

R

_{S, OP}is about ~465 ꢀ at 1 kHz. In addition, note the following:

4. Create a line down and to the left from the point located in

Step 2) by decreasing one decade of voltage noise per one

decade of resistance.



The Johnson noise associated with this device is equivalent

to a source resistor of about 69.6 ꢀsee Figure 6).

For a source resistance above ~465 ꢀ, the noise voltage

produced by the amplifier’s current noise exceeds that

contributed by the source resistance; the amplifier’s

current noise becomes the dominant noise source.

Any amplifiers below and to the right of the lines are good low

noise op amps for the design as highlighted in the shade of gray

in Figure 7.



For the example shown in Figure 7, the following devices are

good candidates for the design: AD8597, AD8599, AD797,

ADA4075-2, ADA4004, OP270, OP27/OP37, AD743/AD745,

and OP184.

To use the graph (see Figure 7), follow Step 1 through Step 4.

1. Typically, the source resistances are known (such as sensor

impedances). If the resistances are not known, calculate them

from the surrounding or preceding circuit components.

100

f = 1kHz

JOHNSON NOISE LINE

LOW NOISE BOUNDRY

IDEAL OP AMPS FOR A

LOW NOISE APPLICATION

AD8622/AD8624

OP285

AD8610/

AD8620

OP467

OP271

10

OPx177

STEP 2

STEP 3

OP275

OP213

OPx84

OP270

OP27/OP37

AD743/AD745

ADA4004

ADA4075-2

AD8597/AD8599

1

AD797

STEP 4

0.1

10

100

1k

10k

100k

1M

SOURCE RESISTANCE (Ω)

Figure 7. Selecting Op Amp for Low Noise Design

Rev. D | Page 8 of 12

Application Note

AN-940

CONCLUSION

Consider all potential noise sources when evaluating an

amplifier’s noise performance for low noise design.

For resistive noise sources, use the following rules:



Restrict bandwidth to only what is necessary.

Reduce resistor value where possible.

Use low noise resistors, such as bulk metal foil, wirewound,

and metal film technology resistors.

The key noise contribution of an op amp is dependent on

source resistance as follows:



R_S>> R_{S, OP}; input-referred current noise dominates.

R_S= R_{S, OP}; amplifier noise and resistor noise are equal

R_S<< R_{S, OP}; input-referred voltage noise dominates.



Reduce the number of resistive noise sources where

possible.

Use Figure 8 and Figure 9 to assist with the selection of

an Analog Devices low noise amplifier using the criteria

described in this application note.

In summary, reduce or eliminate interference signals by



Proper layout techniques to reduce parasitics.

Proper ground techniques, such as isolating digital

and analog ground.

For more information on noise, see the article, “Noise Opti-

mization in Sensor Signal Conditioning Circuit” available at

http://www.analog.com/noiseoptimization.



Proper shielding.

Rev. D | Page 9 of 12

AN-940

Application Note

V

MAX

(µV)

SLEW

RATE

(V/µs)

I

/AMP

MAX

(mA)

e_n

@

1kHz

i_{n @}

R

@

1/f

CORNER

(Hz)

I

B

MAX

(nA)

CMRR PSRR NUMBER

OS

SY

S, OP

1kHz

(Ω)

PART

V

TCV

(µV/°C)

GBP

(MHz)

1kHz

I

SC

(mA)

MIN

(dB)

MIN

(dB)

OF

AMPS

SY

OS

NUMBER

(V)

(nV/√Hz) (pA/√Hz)

AD797

10 TO 36

9 TO 36

40

0.2

0.8

8

20

15

10.5

5.7

0.9

2

450

465

60

9

900

200

80

52

120

1

AD8597/

AD8599

120

10

1.07

2.3

1/

2

ADA4004-1/

ADA4004-2/

ADA4004-4

10 TO 36

125

0.7

12

2.7

2.2

1.8

1.2

1500

5

90

25

110

1/

2/

4

AD8676

AD8675

10 TO 36

50

75

0.2

0.3

10

2.5

4

3.4

2.9

3.5

2.8

0.3*

—

10

2

40

30

111

114

100

106

120

110

2

1

AD8671/

AD8672/

AD8674

12

1/

2/

4

ADA4075-2 ±4.5 TO ±18

1000

100

75

0.3

0.2

6.5

8

12

2.8

17

2.25

5.7

2.8

3.2

1.2

0.4

0.6

2333

8000

5333

5

100

80

40

30

15

110

100

106

140

110

2

1

OP27

OP37

8 TO 44

9 TO 36

2.7

5

40

5

4.7

75

OP270

OP470

2.4

3.25

20

2/

4

AD743

AD745

9.6 TO 36

3 TO 36

1000

500

2

4.5

20

2.8

12.5

4

10

2

3.2

3.9

0.0069

0.4

463,768

9750

50

10

0.4

0.25

450

40

10

80

90

86

90

100

90

1

OP184/

OP284/

OP484

100

0.2

4.25

1/

2/

4

AD8655/

AD8656

2.7 TO 5.5

4 TO 36

250

150

0.4

0.2

28

11

4.5

3

4

—

3000

10

0.01

600

220

40

85

96

88

1/

2

OP113 /

OP213/

OP413

3.4

1.2

4.7

0.4

11,750

100

1/

2/

4

SSM2135

ADA4528-1

OP285

4 TO 36

2.2 TO 5.5

9 TO 36

2000

2.5

—

0.002

1

3.5

4

0.9

0.45

22

3

5.2

5.6*

6

0.5

0.7*

0.9

10,400

8000

3

750

0.4

30

65

87

135

80

90

130

85

2

1

2

1.7

2.5

3.5

NONE

125

250

100

9

6667

350

0.01

AD8610/

AD8620

10 TO 27

0.5

25

60

6

0.005

1,200,000

1000

90

100

1/

2

OP275

9 TO 44

9 TO 36

5 TO 36

1000

500

200

1800

60

2

3.5

1

9

22

170

84

2.5

6

1.5

0.8

4000

7500

2.24

8

350

600

5

14

40

45

10

25

80

85

96

2

4

1

4

OP467

28

19

6.5

1.3

6

ADA4627-1

OP471

7.500

2.75

0.5

6.1

6.5

7.9

0.0016

0.4

3,812,500

16,250

39,500

250

5

106

95

106

95

4

8

60

2

OP1177/

OP2177/

OP4177

0.2

0.7

0.2

10

120

1/

2/

4

AD8510/

AD8512/

AD8513

9 TO 36

400

1

8

20

2.5

8

—

100

0.08

70

86

1/

2/

4

AD8651/

AD8652

2.7 TO 5.5

350

4

50

24

41

11

14

8

0.025

—

320,000

—

10000

1000

0.01

80

67

76

63

1/

2

AD8646/

AD8647/

AD8648

2500

1.8

1.5

0.001

120

1/

2(SD)/

4

AD8605/

AD8606/

AD8608

2.7 TO 5.5

2.7 TO 6

300

2000

325

1

1.3

1

10

15

5

1.2

1.05

0.8

8

0.01

0.05

0.4

800,000

160,000

23,750

500

3000

10

0.001

600

80

30

85

70

80

60

1/

2/

4

AD8691/

AD8692/

AD8694

1(SD)/

2(SD)/

4(SD)

OP162/

OP262/

OP462

2.7 TO 12

13

9.5

1/

2/

4

OP07

6 TO 36

8 TO 36

75

0.3

0.5

1.5

0.6

24

0.3

0.2

12

4

1.3

2

9.6

10

0.12

0.074

0.05

80,000

135,135

200,000

100

8

4

1

30

106

120

80

94

115

70

1

OP07D

AD8677

150

130

500

8 TO 36

8

1

30

AD8615/

AD8616/

AD8618

2.7 TO 5.5

1000

0.001

150

1/

2/

4

AD8519/

AD8529

2.7 TO 12

5 TO 16

1100

2500

2

3

8

4

2.9

3.5

1.2

10

0.4

0.1

25,000

80

300

70

63

90

60

98

1/

2

AD8665/

AD8666/

AD8668

1.55

100,000

1000

0.001

140

1/

2/

4

AD8622/

AD8624

4 TO 36

5 TO 16

125

160

0.5

4

0.56

4

0.48

3.5

0.250

1.55

11

12

0.15

0.1

73,333

20

200

40

125

90

125

95

2/

4

AD8661/

AD8662/

AD8664

120,000

1000

0.001

140

1/

2/

4

OP97

4 TO 40

3 TO 36

75

0.3

0.9

0.7

0.2

0.38

0.35

14

15

0.02*

0.13

1,166,667

115,384

200

20

0.15

11

10

30

110

120

1/

2/

4

OP297

OP497

OP777/

OP727/

OP747

100

1/

2/

4

*REFER TO DEVICE DATA SHEET FOR SPECIFICATION CONDITIONS.

Figure 8. Analog Devices Low Input Voltage Noise Amplifier Selection Table

Rev. D | Page 10 of 12

Application Note

AN-940

V

MAX

(µV)

SLEW

RATE

(V/µs)

I

/AMP

MAX

(mA)

e_n

@

1kHz

i_{n @}

1kHz

(fA/√Hz)

R

@

1/f

CORNER

(Hz)

I

B

MAX

(pA)

CMRR PSRR NUMBER

OS

SY

S, OP

1kHz

(Ω)

PART

V

TCV

(µV/°C)

GBP

(MHz)

I

OUT

(mA)

MIN

(dB)

MIN

(dB)

OF

AMPS

SY

OS

NUMBER

(V)

(nV/√Hz)

AD549

10 TO 36

10 TO 26

500

750

10

5

3

5

0.7

35

16

0.22

0.5

159,090,909

35,000,000

100

-

0.06

1

20

90

76

90

80

1

AD8627/

AD8626/

AD8625

2.5

0.850

15*

1/

2/

4

AD8641/

AD8642

AD8643

5 TO 26

5 TO 36

750

2.5

2

3.5

1.9

3

0.290

0.900

27.5

16

0.5

0.8

57,000,000

20,000,000

250

90

1

12*

15*

90

70

1/

2/

4

AD820/

AD822/

AD824

1000

3*

10

74*

1/

2/

4

ADA4627-1

AD548K/B

8 TO 36

9 TO 36

10 TO 26

200

500

100

1

5

19

1

84

1.8

60

7.500

0.2

6.1

30

6

1.6

1.8

5

3,812,500

16,666,666

1,200,000

250

700

5

45

15

45

106

82

106

86

1

10

AD8610/

AD8620

0.5

25

3.500

1000

90

100

1/

2

ADA4062-2

ADA4062-4

8 TO 36

2500

4

1.4

3.3

0.220

36

5

7,200,000

30

50

20

73

74

2/

4

AD743

AD745

AD711C

9.6 TO 36

9 TO 36

1000

500

250

300

2

5

1

4.5

20

4

2.8

12.5

20

10

3.2

18

8

6.9

10

463,768

1,800,000

800,000

50

0.4

0.25

25

40

25

80

90

86

85

90

100

86

1

2.8

1.2

200

500

AD8605/

AD8606/

AD8608

2.7 TO 6

10

5

10

1

80

1/

2/

4

OP282/

OP482

9 TO 36

8 TO 36

3000

1000

1700

10

2

4

3.5

5

9

0.250

1.650

36

16

10

3,600,000

1,600,000

40

100

20

10

28

70

80

110

92

2/

4

AD8682

AD8684

2/

4

ADA4000-1

ADA4000-2

ADA4000-4

20

100

40

82

1/

2/

4

OP97/

OP297/

OP497

4 TO 40

75*

0.3*

0.9*

0.2*

0.38*

14*

20*

1,166,667*

200*

150

10

110*

1/

2/

4

AD8651/

AD8652

2.7 TO 5.5

2.7 TO 6

350

500

4

50

24

41

12

14

8

25

50

320,000

200,000

10,000

1000

10

1

80

76

70

1/

2

AD8615/

AD8616/

AD8618

1.5

1.3

10

150

1/

2/

4

AD8691/

AD8692/

AD8694

2.7 TO 6

5 TO 6

2000

160

1.3

4

10

4

5

1.05

1.55

8

50

160,000

120,000

3000

1000

1

80

70

90

80

95

1(SD)/

2(SD)/

4(SD)

AD8661/

AD8662/

AD8664

3.5

12

100

140

1/

2/

4

OP07

6 TO 36

5 TO 36

75

0.3

0.5

0.6

0.3

4

9.6

11

120

150

80,000

73,333

100

20

4000

200

30

40

106

125

94

1

AD8622/

AD8624

125

0.56

0.48

0.250

125

2/

4

*REFER TO DEVICE DATA SHEET FOR SPECIFICATION CONDITIONS.

Figure 9. Analog Devices Low Input Current Noise Amplifier Selection Table

Rev. D | Page 11 of 12

AN-940

Application Note

REFERENCES

Analog Devices, Inc., AN-280 Application Note Mixed Signal Circuit Techniques.

Barrow, J., and Paul Brokaw. 1989. “Grounding for Low- and High-Frequency Circuits.” Analog Dialogue. Analog Devices, Inc. (23-3).

Bennett, W. R. 1960. Electrical Noise. New York: McGraw-Hill.

Bowers, Derek F. 1989. “Minimizing Noise in Analog Bipolar Circuit Design.” IEEE Press.

Brockman, Don and Arnold Williams. AN-214 Application Note Ground Rules for High-Speed Circuits. Analog Devices, Inc.

Brokaw, Paul. 2000. AN-202 Application Note An IC Amplifier User’s Guide to Decoupling, Grounding, and Making Things Go Right for a

Change. Analog Devices, Inc. (February).

Brokaw, Paul and Jeff Barrow. AN-345 Application Note Grounding for Low- and High-Frequency Circuits. Analog Devices, Inc.

Bryant, James Bryant and Lew Counts. 1990. “Op Amp Issues–Noise ” Analog Dialogue. Analog Devices Inc. (24–2).

Freeman, J. J. 1958. Principles of Noise. New York: John Wiley & Sons, Inc.

Gupta, Madhu S., ed., 1977. Electrical Noise: Fundamentals & Sources. New York: IEEE Press. Collection of classical reprints.

Hernik, Yuval and Belman, Michael. Linearity and Noise Capabilities of Ultra High Precision Bulk Metal® Foil Resistors. Vishay Intertechnology,

Inc. (February 2010).

Johnson, J. B. 1928. “Thermal Agitation of Electricity in Conductors” (Physical Review 32): 97–109.

Motchenbacher, C. D., and J. A. Connelly. 1993. Low-Noise Electronic Design. New York: John Wiley & Sons, Inc.

Nyquist, H. 1928. “Thermal Agitation of Electric Charge in Conductors” (Physical Review 32): 110–113.

Rice, S.O. 1944. “Math Analysis for Random Noise” Bell System Technical Journal (July): 282–332.

Rich, Alan. 1982. “Understanding Interference-Type Noise.” Analog Dialogue. Analog Devices Inc., (16–3).

Rich, Alan. 1983. “Shielding and Guarding.” Analog Dialogue. Analog Devices Inc. (17–1).

Ryan, Al and Tim Scranton. 1984. “DC Amplifier Noise Revisited.” Analog Dialogue. Analog Devices, Inc., (18–1).

Schottky, W. 1926. “Small-Shot Effect and Flicker Effect.” (Phys. Rev. 28): 74–103.

Van Der Ziel, A. 1954. Noise. Englewood Cliffs, NJ: Prentice-Hall, Inc.

Vishay Intertechnology, Inc. AN0003 Application Note Audio Noise Reduction Through the Use of Bulk Metal® Foil Resistors-“Hear the

Difference”.

registered trademarks are the property of their respective owners.

AN07053-0-7/11(D)

Rev. D | Page 12 of 12


型号：	AN-940
厂家：	ADI
描述：	Low Noise Amplifier Selection Guide for Optimal Noise Performance 低噪声放大器选型指南最佳噪声性能放大器
文件：	总12页 (文件大小：169K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

AN-940 [ADI]

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