OP162HRU [ADI]
15 MHz Rail-to-Rail Operational Amplifiers; 15 MHz的轨至轨运算放大器
| 型号： | OP162HRU |
| 厂家： | 
ADI |
| 描述： | 15 MHz Rail-to-Rail Operational Amplifiers |
| 文件： | 总16页 (文件大小：557K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
15 MHz Rail-to-Rail
Operational Amplifiers
a
OP162/OP262/OP462
FEATURES
PIN CONFIGURATIONS
Wide Bandwidth: 15 MHz
Low Offset Voltage: 325 V max
Low Noise: 9.5 nV/÷Hz @ 1 kHz
Single-Supply Operation: +2.7 V to +12 V
Rail-to-Rail Output Swing
Low TCVOS: 1 V/؇C typ
High Slew Rate: 13 V/s
No Phase Inversion
8-Lead Narrow-Body SO
(S Suffix)
1
8
NULL
V+
OUT A
NC
NULL
–IN A
+IN A
V–
OP162
4
5
NC = NO CONNECT
Unity Gain Stable
8-Lead TSSOP
(RU Suffix)
APPLICATIONS
Portable Instrumentation
Sampling ADC Amplifier
Wireless LANs
Direct Access Arrangement
Office Automation
NULL
NULL
8
5
1
–IN A
+IN A
V–
V+
OP162
OUT A
NC
4
NC = NO CONNECT
8-Lead Narrow-Body SO
(S Suffix)
GENERAL DESCRIPTION
1
8
V+
OUT A
–IN A
+IN A
V–
The OP162 (single), OP262 (dual), OP462 (quad) rail-to-rail
15 MHz amplifiers feature the extra speed new designs require,
with the benefits of precision and low power operation. With
their incredibly low offset voltage of 45 mV (typ) and low noise,
they are perfectly suited for precision filter applications and
instrumentation. The low supply current of 500 mA (typ) is
critical for portable or densely packed designs. In addition, the
rail-to-rail output swing provides greater dynamic range and
control than standard video amplifiers provide.
OUT B
–IN B
+IN B
OP262
4
5
8-Lead TSSOP
(RU Suffix)
OUT A
–IN A
+IN A
V–
V+
8
5
1
OUT B
–IN B
+IN B
OP262
4
These products operate from single supplies as low as +2.7 V to
dual supplies of ± 6 V. The fast settling times and wide output
swings recommend them for buffers to sampling A/D converters.
The output drive of 30 mA (sink and source) is needed for
many audio and display applications; more output current can
be supplied for limited durations.
14-Lead Narrow-Body SO
(S Suffix)
1
14
OUT D
OUT A
–IN A
+IN A
V+
+IN B
–IN B
–IN D
+IN D
V–
+IN C
–IN C
OUT C
The OP162 family is specified over the extended industrial
temperature range (–40∞C to +125∞C). The single OP162
and dual OP262 are available in 8-lead SOIC and TSSOP
packages. The quad OP462 is available in 14-lead narrow-body
SOIC and TSSOP packages.
OP462
8
OUT B
7
14-Lead TSSOP
(RU Suffix)
OUT A
OUT D
1
14
OP462
–IN A
+IN A
V+
–IN D
+IN D
V–
+IN B
–IN B
OUT B
+IN C
–IN C
OUT C
7
8
REV. D
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, norforanyinfringementsofpatentsorotherrightsofthirdpartiesthat
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
Fax: 781/326-8703
www.analog.com
© Analog Devices, Inc., 2002
OP162/OP262/OP462–SPECIFICATIONS
(@ V = +5.0 V, VCM = 0 V, TA = +25؇C, unless otherwise noted)
ELECTRICAL CHARACTERISTICS
S
Parameter
Symbol
Conditions
Min
Typ
Max
Units
INPUT CHARACTERISTICS
Offset Voltage
VOS
OP162G, OP262G, OP462G,
–40∞C £ TA £ +125∞C
45
325
800
1
3
5
600
650
± 25
± 40
+4
mV
mV
mV
mV
mV
nA
nA
nA
nA
V
H Grade, –40∞C £ TA £ +125∞C
D Grade, –40∞C £ TA £ +125∞C
0.8
Input Bias Current
Input Offset Current
IB
360
± 2.5
–40∞C £ TA £ +125∞C
–40∞C £ TA £ +125∞C
IOS
Input Voltage Range
VCM
0
Common-Mode Rejection
CMRR
0 V £ VCM £ +4.0 V,
–40∞C £ TA £ +125∞C
RL = 2 kW, 0.5 £ VOUT £ 4.5 V
RL = 10 kW, 0.5 £ VOUT £ 4.5 V
RL = 10 kW, –40∞C £ TA £ +125∞C
G Grade1
70
110
30
88
dB
Large Signal Voltage Gain
AVO
V/mV
V/mV
V/mV
mV
mV/∞C
pA/∞C
65
40
Long-Term Offset Voltage
Offset Voltage Drift
Bias Current Drift
VOS
DVOS/DT
DIB/DT
600
Note 2
1
250
OUTPUT CHARACTERISTICS
Output Voltage Swing High
VOH
VOL
IL = 250 mA, –40∞C £ TA £ +125∞C
IL = 5 mA
IL = 250 mA, –40∞C £ TA £ +125∞C
IL = 5 mA
4.95
4.85
4.99
4.94
14
65
± 80
± 30
V
V
mV
mV
mA
mA
Output Voltage Swing Low
50
150
Short Circuit Current
Maximum Output Current
ISC
IOUT
Short to Ground
POWER SUPPLY
Power Supply Rejection Ratio
PSRR
ISY
VS = +2.7 V to +7 V
120
600
500
dB
dB
mA
mA
mA
mA
–40∞C £ TA £ +125∞C
OP162, VOUT = 2.5 V
–40∞C £ TA £ +125∞C
OP262, OP462, VOUT = 2.5 V
–40∞C £ TA £ +125∞C
90
Supply Current/Amplifier
750
1
700
850
DYNAMIC PERFORMANCE
Slew Rate
Settling Time
SR
tS
GBP
1 V < VOUT < 4 V, RL = 10 kW
To 0.1%, AV = –1, VO = 2 V Step
10
540
15
V/ms
ns
Gain Bandwidth Product
MHz
Phase Margin
f
61
Degrees
m
NOISE PERFORMANCE
Voltage Noise
Voltage Noise Density
Current Noise Density
e p-p
n
0.1 Hz to 10 Hz
f = 1 kHz
f = 1 kHz
0.5
9.5
0.4
mV p-p
nV/÷Hz
pA/÷Hz
e
n
i
n
NOTES
1Long-term offset voltage is guaranteed by a 1000 hour life test performed on three independent lots at +125∞C, with an LTPD of 1.3.
2Offset voltage drift is the average of the –40∞C to +25∞C delta and the +25∞C to +125∞C delta.
Specifications subj]ect to change without notice.
REV. D
–2–
OP162/OP262/OP462
(@ V = +3.0 V, VCM = 0 V, TA = +25؇C, unless otherwise noted)
OP162/OP262/OP462–SPECIFICATIONS
ELECTRICAL CHARACTERISTICS
S
Parameter
Symbol
Conditions
Min
Typ
Max
Units
INPUT CHARACTERISTICS
Offset Voltage
VOS
OP162G, OP262G, OP462G
H Grade, –40∞C £ TA £ +125∞C
D Grade, –40∞C £ TA £ +125∞C
50
325
1
3
mV
mV
mV
mV
nA
nA
V
0.8
5
Input Bias Current
IB
IOS
VCM
CMRR
360
± 2.5
600
± 25
+2
Input Offset Current
Input Voltage Range
Common-Mode Rejection
0
0 V £ VCM £ +2.0 V,
–40∞C £ TA £ +125∞C
RL = 2 kW, 0.5 V £ VOUT £ 2.5 V
RL = 10 kW, 0.5 V £ VOUT £ 2.5 V
G Grade1
70
20
110
20
30
dB
Large Signal Voltage Gain
Long-Term Offset Voltage
AVO
VOS
V/mV
V/mV
mV
600
OUTPUT CHARACTERISTICS
Output Voltage Swing High
VOH
VOL
IL = 250 mA
IL = 5 mA
IL = 250 mA
IL = 5 mA
2.95
2.85
2.99
2.93
14
V
V
mV
mV
Output Voltage Swing Low
50
150
66
POWER SUPPLY
Power Supply Rejection Ratio
PSRR
ISY
VS = +2.7 V to +7 V,
–40∞C £ TA £ +125∞C
OP162, VOUT = 1.5 V
–40∞C £ TA £ +125∞C
OP262, OP462, VOUT = 1.5 V
–40∞C £ TA £ +125∞C
60
110
600
dB
mA
mA
mA
mA
Supply Current/Amplifier
700
1
650
850
500
DYNAMIC PERFORMANCE
Slew Rate
Settling Time
SR
tS
RL = 10 kW
To 0.1%, AV = –1, VO = 2 V Step
10
575
V/ms
ns
Gain Bandwidth Product
Phase Margin
GBP
15
59
MHz
Degrees
f
m
NOISE PERFORMANCE
Voltage Noise
Voltage Noise Density
e p-p
n
0.1 Hz to 10 Hz
f = 1 kHz
f = 1 kHz
0.5
9.5
0.4
mV p-p
nV/÷Hz
pA/÷Hz
e
n
Current Noise Density
i
n
NOTES
1Long-term offset voltage is guaranteed by a 1000 hour life test performed on three independent lots at +125∞C, with an LTPD of 1.3.
Specifications subject to change without notice.
REV. D
–3–
OP162/OP262/OP462–SPECIFICATIONS
(@ V = ؎5.0 V, VCM = 0 V, TA = +25؇C, unless otherwise noted)
ELECTRICAL CHARACTERISTICS
S
Parameter
Symbol
Conditions
Min
Typ
Max
Units
INPUT CHARACTERISTICS
Offset Voltage
VOS
OP162G, OP262G, OP462G
–40∞C £ TA £ +125∞C
25
325
800
1
3
5
500
650
± 25
± 40
+4
mV
mV
mV
mV
mV
nA
nA
nA
nA
V
H Grade, –40∞C £ TA £ +125∞C
D Grade, –40∞C £ TA £ +125∞C
0.8
Input Bias Current
Input Offset Current
IB
260
± 2.5
–40∞C £ TA £ +125∞C
–40∞C £ TA £ +125∞C
IOS
Input Voltage Range
Common-Mode Rejection
VCM
CMRR
–5
70
–4.9 V £ VCM £ +4.0 V,
–40∞C £ TA £ +125∞C
RL = 2 kW, –4.5 V £ VOUT £ 4.5 V
RL = 10 kW, –4.5 V £ VOUT £ 4.5 V
–40∞C £ TA £ +125∞C
G Grade1
110
35
120
dB
Large Signal Voltage Gain
AVO
V/mV
V/mV
V/mV
mV
mV/∞C
pA/∞C
75
25
Long-Term Offset Voltage
Offset Voltage Drift
Bias Current Drift
VOS
DVOS/DT
DIB/DT
600
Note 2
1
250
OUTPUT CHARACTERISTICS
Output Voltage Swing High
VOH
VOL
IL = 250 mA, –40∞C £ TA £ +125∞C 4.95
4.99
4.94
–4.99
–4.94
± 80
V
V
V
V
mA
mA
IL = 5 mA
4.85
Output Voltage Swing Low
IL = 250 mA, –40∞C £ TA £ +125∞C
IL = 5 mA
Short to Ground
–4.95
–4.85
Short Circuit Current
Maximum Output Current
ISC
IOUT
± 30
POWER SUPPLY
Power Supply Rejection Ratio
PSRR
ISY
VS = ± 1.35 V to ± 6 V,
–40∞C £ TA £ +125∞C
OP162, VOUT = 0 V
60
110
650
dB
mA
mA
mA
mA
V
Supply Current/Amplifier
800
1.15
775
–40∞C £ TA £ +125∞C
OP262, OP462, VOUT = 0 V
–40∞C £ TA £ +125∞C
550
1
Supply Voltage Range
VS
+3.0 (± 1.5)
+12 (± 6)
DYNAMIC PERFORMANCE
Slew Rate
Settling Time
SR
tS
–4 V < VOUT < 4 V, RL = 10 kW
To 0.1%, AV = –1, VO = 2 V Step
13
475
V/ms
ns
Gain Bandwidth Product
Phase Margin
GBP
15
64
MHz
Degrees
f
m
NOISE PERFORMANCE
Voltage Noise
Voltage Noise Density
e p-p
n
0.1 Hz to 10 Hz
f = 1 kHz
f = 1 kHz
0.5
9.5
0.4
mV p-p
nV/÷Hz
pA/÷Hz
e
n
Current Noise Density
i
n
NOTES
1Long-term offset voltage is guaranteed by a 1000 hour life test performed on three independent lots at +125∞C, with an LTPD of 1.3.
2Offset voltage drift is the average of the –40∞C to +25∞C delta and the +25∞C to +125∞C delta.
Specifications subject to change without notice.
–4–
REV. D
OP162/OP262/OP462
ABSOLUTE MAXIMUM RATINGS
ORDERING GUIDE
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 6 V
Input Voltage1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 6 V
Differential Input Voltage2 . . . . . . . . . . . . . . . . . . . . . ± 0.6 V
Internal Power Dissipation
SOIC (S) . . . . . . . . . . . . . . . . . . . Observe Derating Curves
TSSOP (RU) . . . . . . . . . . . . . . . . Observe Derating Curves
Output Short-Circuit Duration . . . . Observe Derating Curves
Storage Temperature Range . . . . . . . . . . . . –65∞C to +150∞C
Operating Temperature Range . . . . . . . . . . –40∞C to +125∞C
Junction Temperature Range . . . . . . . . . . . . –65∞C to +150∞C
Lead Temperature Range (Soldering, 10 sec) . . . . . . . +300∞C
Temperature
Range
Package
Description
Package
Option
Model
OP162GS
–40∞C to +125∞C 8-Lead SOIC
RN-8
RU-8
RU-8
RU-8
RN-8
RU-8
RU-14
RN-14
RN-14
RU-14
OP162DRU –40∞C to +125∞C 8-Lead TSSOP
OP162HRU –40∞C to +125∞C 8-Lead TSSOP
OP262DRU –40∞C to +125∞C 8-Lead TSSOP
–40∞C to +125∞C 8-Lead SOIC
OP262HRU –40∞C to +125∞C 8-Lead TSSOP
OP462DRU –40∞C to +125∞C 14-Lead TSSOP
–40∞C to +125∞C 14-Lead SOIC
–40∞C to +125∞C 14-Lead SOIC
OP462HRU –40∞C to +125∞C 14-Lead TSSOP
OP262GS
OP462DS
OP462GS
3
Package Type
JA
Units
JC
8-Lead SOIC (S)
158
240
120
180
43
43
36
35
∞C/W
∞C/W
∞C/W
∞C/W
8-Lead TSSOP (RU)
14-Lead SOIC (S)
14-Lead TSSOP (RU)
NOTES
1For supply voltages greater than 6 volts, the input voltage is limited to less than or
equal to the supply voltage.
2For differential input voltages greater than 0.6 volts the input current should be
limited to less than 5 mA to prevent degradation or destruction of the input devices.
3qJA is specified for the worst case conditions, i.e., qJA is specified for device soldered
in circuit board for SOIC and TSSOP packages.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the OP162/OP262/OP462 features proprietary ESD protection circuitry, permanent
damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper
ESD precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
REV. D
–5–
OP162/OP262/OP462–Typical Performance Characteristics
420
340
260
180
100
250
200
150
100
100
V
= 5V
V
= 5V
= 25؇C
S
S
V
= 5V
= 25؇C
S
T
A
T
A
80
COUNT =
COUNT =
720 OP AMPS
360 OP AMPS
60
40
50
0
20
0
0
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0
COMMON-MODE VOLTAGE – Volts
–200 –140 –80 –20
40
100 160
0.2 0.3 0.5 0.7 0.9 1.1 1.3 1.5
INPUT OFFSET VOLTAGE – V
INPUT OFFSET DRIFT, TCV – V/؇C
OS
Figure 3. OP462 Input Bias Current
vs. Common-Mode Voltage
Figure 1. OP462 Input Offset Voltage
Distribution
Figure 2. OP462 Input Offset Voltage
Drift (TCVOS
)
125
0
–100
–200
–300
–400
15
V
= 5V
V = 5V
S
S
V
= 5V
S
100
75
10
5
50
25
0
–500
0
–75 –50 –25
0
25 50 75 100 125 150
TEMPERATURE –
–50 –25
0
25 50 75 100 125 150
–75 –50 –25
0
25 50 75 100 125 150
TEMPERATURE – ؇C
؇
C
TEMPERATURE – ؇C
Figure 4. OP462 Input Offset Voltage
vs. Temperature
Figure 6. OP462 Input Offset Current
vs. Temperature
Figure 5. OP462 Input Bias Current
vs. Temperature
5.12
100
0.100
V
= 5V
S
V
= 5V
S
R
= 10k⍀
L
5.06
5.00
4.94
4.88
4.82
80
60
40
20
0.080
0.060
0.040
0.020
0.000
I
= 5mA
OUT
V
= 5V
S
I
= 250A
OUT
R
= 2k⍀
I
= 5mA
L
OUT
R
= 600⍀
L
I
= 250A
OUT
0
–75 –50 –25
0
25 50 75 100 125 150
TEMPERATURE – ؇C
–75 –50 –25
0
25 50 75 100 125 150
–75 –50 –25
0
25 50 75 100 125 150
TEMPERATURE – ؇C
TEMPERATURE – ؇C
Figure 9. OP462 Open-Loop Gain
vs. Temperature
Figure 7. OP462 Output High Voltage
vs. Temperature
Figure 8. OP462 Output Low Voltage
vs. Temperature
–6–
REV. D
OP162/OP262/OP462
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
100
80
60
40
20
0
0.7
T
= 25؇C
A
V
= 10V
S
V
= 5V
S
0.6
0.5
0.4
V
= 3V
S
V
= 10V
S
V
= 3V
S
0
2
4
6
8
10
12
0
1
2
3
4
5
6
7
–75 –50 –25
0
25 50 75 100 125 150
LOAD CURRENT – mA
SUPPLY VOLTAGE – Volts
TEMPERATURE – ؇C
Figure 10. Output Low Voltage to
Supply Rail vs. Load Current
Figure 12. OP462 Supply Current/
Amplifier vs. Supply Voltage
Figure 11. Supply Current/Amplifier
vs. Temperature
50
5
4
3
60
V
T
= 5V
= 25؇C
S
A
40
30
V
T
R
= 5V
= +25؇C
GAIN
S
40
20
A
45
= 830⍀
≤ 5pF
L
L
C
20
10
0
90
PHASE
135
180
225
270
V
= 5V
S
2
0
A
R
C
= 1
= 10k⍀
= 15pF
VCL
L
L
–10
–20
–30
1
0
–20
–40
T
= 25°C
A
DISTORTION < 1%
100k
1M
10M
100M
10k
100k
1M
10M
10k
100k
1M
10M
100M
FREQUENCY – Hz
FREQUENCY – Hz
FREQUENCY – Hz
Figure 13. Open-Loop Gain and
Phase vs. Frequency (No Load)
Figure 15. Maximum Output Swing
vs. Frequency
Figure 14. Closed-Loop Gain vs.
Frequency
4
3
70
60
50
V
T
= 5V
= 25؇C
S
A
60
50
40
30
20
10
0
0.1%
0.01%
V
T
= 5V
= 25؇C
S
A
V
= 5V
2
1
S
T
= 25؇C
= ؎50mV
= 10k⍀
A
40
30
20
10
0
V
IN
R
+OS
L
0
–OS
–1
–2
–3
–4
0.1%
0.01%
800
0
200
400
600
1000
1
10
100
1k
10
100
CAPACITANCE – pF
1000
SETTLING TIME – ns
FREQUENCY – Hz
Figure 16. Settling Time vs. Step Size
Figure 17. Small-Signal Overshoot
vs. Capacitance
Figure 18. Voltage Noise Density vs.
Frequency
REV. D
–7–
OP162/OP262/OP462
300
250
200
150
100
7
90
80
70
60
50
40
30
20
V
T
= 5V
= 25؇C
V
T
= 5V
= 25؇C
V
= 5V
S
= 25؇C
S
S
6
5
4
3
2
T
A
A
A
A
= 10
VCL
A
= 1
VCL
50
0
1
0
100k
1M
FREQUENCY – Hz
10M
1
10
100
1k
1k
10k
100k
1M
10M
FREQUENCY – Hz
FREQUENCY – Hz
Figure 20. Output Impedance vs.
Frequency
Figure 19. Current Noise Density vs.
Frequency
Figure 21. CMRR vs. Frequency
90
2V
V
T
= 5V
= 25؇C
20mV
2s
S
A
V
V
A
= 12V p-p
= ؎5V
= 1
IN
S
80
70
60
100
90
100
90
V
+PSRR
–PSRR
50
10
0%
10
0%
V
A
= 5V
= 100k⍀
= 0.5V p-p
40
30
20
S
V
2V
20s
e
n
1k
10k
100k
1M
10M
FREQUENCY – Hz
Figure 22. PSRR vs. Frequency
Figure 23. 0.1 Hz to 10 Hz Noise
Figure 24. No Phase Reversal; [VIN =
12 V p-p, VS = ±5 V, AV = 1]
V
= 5V
= 1
= 25؇C
= 100pF
S
V
= 5V
= 1
= 25؇C
= 100pF
S
A
V
A
100
90
100
90
A
V
A
T
T
C
L
C
L
10
10
0%
0%
500mV
100s
20mV
200ns
Figure 26. Large Signal Transient
Response
Figure 25. Small Signal Transient
Response
–8–
REV. D
OP162/OP262/OP462
VCC. It is important to avoid accidentally connecting the wiper
to VEE, as this will damage the device. The recommended value
for the potentiometer is 20 kW.
APPLICATIONS SECTION
Functional Description
The OPx62 family is fabricated using Analog Devices’ high
speed complementary bipolar process, also called XFCB. The
process includes trench isolating each transistor to lower para-
sitic capacitances thereby allowing high speed performance.
This high speed process has been implemented without trading
off the excellent transistor matching and overall dc performance
characteristic of Analog Devices’ complementary bipolar pro-
cess. This makes the OPx62 family an excellent choice as an
extremely fast and accurate low voltage op amp.
+5V
20k⍀
1
8
3
7
6
OP162
V
OS
4
2
Figure 27 shows a simplified equivalent schematic for the OP162.
A PNP differential pair is used at the input of the device. The
cross connecting of the emitters is used to lower the transcon-
ductance of the input stage, which improves the slew rate of the
device. Lowering the transconductance through cross connect-
ing the emitters has another advantage in that it provides a
lower noise factor than if emitter degeneration resistors were
used. The input stage can function with the base voltages taken
all the way to the negative power supply, or up to within 1 V of
the positive power supply.
–5V
Figure 28. Schematic Showing Offset Adjustment
Rail-to-Rail Output
The OP162/OP262/OP462 has a wide output voltage range that
extends to within 60 mV of each supply rail with a load current
of 5 mA. Decreasing the load current will extend the output
voltage range even closer to the supply rails. The common-
mode input range extends from ground to within 1 V of the
positive supply. It is recommended that there be some minimal
amount of gain when a rail-to-rail output swing is desired. The
minimum gain required is based on the supply voltage and can
be found as:
V
CC
VS
VS –1
AV,min
=
where VS is the positive supply voltage. With a single supply
voltage of +5 V, the minimum gain to achieve rail-to-rail output
should be 1.25.
+IN
–IN
V
OUT
Output Short-Circuit Protection
To achieve a wide bandwidth and high slew rate, the output of
the OP162/OP262/OP462 is not short-circuit protected. Short-
ing the output directly to ground or to a supply rail may destroy
the device. The typical maximum safe output current is ± 30 mA.
Steps should be taken to ensure the output of the device will not
be forced to source or sink more than 30 mA.
V
EE
Figure 27. Simplified Schematic
In applications where some output current protection is needed,
but not at the expense of reduced output voltage headroom, a
low value resistor in series with the output can be used. This is
shown in Figure 29. The resistor is connected within the feed-
back loop of the amplifier so that if VOUT is shorted to ground
and VIN swings up to +5 V, the output current will not exceed
30 mA.
Two complementary transistors in a common-emitter configura-
tion are used for the output stage. This allows the output of the
device to swing to within 50 mV of either supply rail at load
currents less than 1 mA. As load current increases, the maxi-
mum voltage swing of the output will decrease. This is due to
the collector-to-emitter saturation voltages of the output transis-
tors increasing. The gain of the output stage, and consequently
the open-loop gain of the amplifier, is dependent on the load
resistance connected at the output. And because the dominant
pole frequency is inversely proportional to the open-loop gain,
the unity-gain bandwidth of the device is not affected by the
load resistance. This is typically the case in rail-to-rail output
devices.
For single +5 V supply applications, resistors less than 169 W
are not recommended.
+5V
V
IN
169⍀
V
OPx62
OUT
Offset Adjustment
Because the OP162/OP262/OP462 has such an exceptionally
low typical offset voltage, adjustment to correct offset voltage
may not be needed. However, the OP162 does have pinouts
where a nulling resistor can be attached. Figure 28 shows how
the OP162 offset voltage can be adjusted by connecting a poten-
tiometer between Pins 1 and 8, and connecting the wiper to
Figure 29. Output Short-Circuit Protection
REV. D
–9–
OP162/OP262/OP462
Figures 30 and 31 provide a convenient way to see if the device
is being overheated. The maximum safe power dissipation can
be found graphically, based on the package type and the ambi-
ent temperature around the package. By using the previous
equation, it is a simple matter to see if PDISS exceeds the device’s
power derating curve. To ensure proper operation, it is impor-
tant to observe the recommended derating curves shown in
Figures 30 and 31.
Input Overvoltage Protection
The input voltage should be limited to ± 6 V or damage to the
device can occur. Electrostatic protection diodes placed in the
input stage of the device help protect the amplifier from static
discharge. Diodes are connected between each input as well as
from each input to both supply pins as shown in the simplified
equivalent circuit in Figure 27. If an input voltage exceeds
either supply voltage by more than 0.6 V, or if the differential
input voltage is greater than 0.6 V, these diodes begin to ener-
gize and overvoltage damage could occur. The input current
should be limited to less than 5 mA to prevent degradation or
destruction of the device.
2.0
1.5
This can be done by placing an external resistor in series with
the input that could be overdriven. The size of the resistor can
be calculated by dividing the maximum input voltage by 5 mA.
For example, if the differential input voltage could reach 5 V,
the external resistor should be 5 V/5 mA = 1 kW. In practice,
this resistance should be placed in series with both inputs to
balance any offset voltages created by the input bias current.
1.0
0.5
0
8-PIN SOIC
PACKAGE
8-PIN TSSOP
PACKAGE
Output Phase Reversal
The OP162/OP262/OP462 is immune to phase reversal as long
as the input voltage is limited to ± 6 V. Figure 24 shows a photo
of the output of the device with the input voltage driven beyond
the supply voltages. Although the device’s output will not
change phase, large currents due to input overvoltage could
result, damaging the device. In applications where the possibility
of an input voltage exceeding the supply voltage exists, over-
voltage protection should be used, as described in the previous
section.
–40
–20
0
20
40
60
80
100
120
AMBIENT TEMPERATURE – ؇C
Figure 30. Maximum Power Dissipation vs. Temperature
for 8-Pin Package Types
2.0
1.5
Power Dissipation
14-PIN SOIC
PACKAGE
The maximum power that can be safely dissipated by the
OP162/OP262/OP462 is limited by the associated rise in junc-
tion temperature. The maximum safe junction temperature is
150∞C, and should not be exceeded or device performance
could suffer. If this maximum is momentarily exceeded, proper
circuit operation will be restored as soon as the die temperature
is reduced. Leaving the device in an “overheated” condition for
an extended period can result in permanent damage to the device.
1.0
14-PIN TSSOP
PACKAGE
0.5
0
–40
–20
0
20
40
60
80
100
120
To calculate the internal junction temperature of the OPx62,
the following formula can be used:
AMBIENT TEMPERATURE –
؇C
Figure 31. Maximum Power Dissipation vs. Temperature
for 14-Pin Package Types
TJ = PDISS ¥ qJA + TA
where: TJ = OPx62 junction temperature;
Unused Amplifiers
P
DISS = OPx62 power dissipation;
It is recommended that any unused amplifiers in a dual or a
quad package be configured as a unity gain follower with a 1 kW
feedback resistor connected from the inverting input to the
output and the noninverting input tied to the ground plane.
q
JA = OPx62 package thermal resistance, junction-to-
ambient; and
TA = Ambient temperature of the circuit.
Power On Settling Time
The power dissipated by the device can be calculated as:
The time it takes for the output of an op amp to settle after a
supply voltage is delivered can be an important consideration in
some power-up sensitive applications. An example of this
would be in an A/D converter where the time until valid data
can be produced after power-up is important.
P
DISS = ILOAD ¥ (VS – VOUT
LOAD is the OPx62 output load current;
VS is the OPx62 supply voltage; and
OUT is the OPx62 output voltage.
)
where:
I
V
The OPx62 family has a rapid settling time after power-up.
Figure 32 shows the OP462 output settling times for a single
supply voltage of VS = +5 V. The test circuit in Figure 33 was
used to find the power on settling times for the device.
–10–
REV. D
OP162/OP262/OP462
2V
500ns
V
= 5V
S
A
C
R
= 1
= 300pF
= 10k⍀
100
90
V
L
L
100
90
WITH SNUBBER:
R
C
= 140⍀
= 10nF
X
X
V
A
R
= 5V
= 1
= 10k⍀
S
10
0%
V
L
10
0%
50mV
50mV
1s
Figure 32. Oscilloscope Photo of VS and VOUT
Figure 36. A Photo of a Nice Square Wave at the Output
The network operates in parallel with the load capacitor, CL,
and provides compensation for the added phase lag. The actual
values of the network resistor and capacitor are determined
empirically to minimize overshoot while maximizing unity-gain
bandwidth. Table I shows a few sample snubber networks for
large load capacitors:
+1
+
0 TO +5V
SQUARE
–
V
OUT
OP462
10k⍀
Table I. Snubber Networks for Large Capacitive Loads
CLOAD
RX
CX
Figure 33. Test Circuit for Power On Settling Time
Capacitive Load Drive
<300 pF
500 pF
1 nF
140 W
100 W
80 W
10 nF
10 nF
10 nF
47 nF
The OP162/OP262/OP462 is a high speed, extremely accurate
device and can tolerate some capacitive loading at its output.
As load capacitance increases, however, the unity-gain band-
width of the device will decrease. There will also be an increase
in overshoot and settling time for the output. Figure 35 shows
an example of this with the device configured for unity gain and
driving a 10 kW resistor and 300 pF capacitor placed in parallel.
10 nF
10 W
Obviously, higher load capacitance will also reduce the unity-
gain bandwidth of the device. Figure 37 shows a plot of unity-
gain bandwidth versus capacitive load. The snubber network
will not provide any increase in bandwidth, but it will substan-
tially reduce ringing and overshoot, as shown in the difference
between Figures 35 and 36.
By connecting a series R-C network, commonly called a “snub-
ber” network, from the output of the device to ground, this
ringing can be eliminated and overshoot can be significantly
reduced. Figure 34 shows how to set up the snubber network,
and Figure 36 shows the improvement in output response with
the network added.
10
9
8
+5V
7
6
5
V
OPx62
OUT
4
V
C
R
IN
L
X
3
C
X
2
Figure 34. Snubber Network Compensation for Capacitive
Loads
1
0
10pF
100pF
1nF
10nF
C
Figure 37. Unity GainLBOAaDndwidth vs. CLOAD
V
= 5V
S
A
C
R
= 1
Total Harmonic Distortion and Crosstalk
V
L
L
100
90
= 300pF
= 10k⍀
The OPx62 device family offers low total harmonic distortion.
This makes it an excellent device choice for audio applications.
Figure 38 shows a graph of THD plus noise figures at 0.001%
for the OP462.
10
Figure 39 shows a graph of the worst case crosstalk between two
amplifiers in the OP462 device. A 1 V rms signal is applied to
one amplifier while measuring the output of an adjacent ampli-
fier. Both amplifiers are configured for unity gain and supplied
with ± 2.5 V.
0%
50mV
1s
Figure 35. A Photo of a Ringing Square Wave
REV. D
–11–
OP162/OP262/OP462
The audio signal is ac coupled to each noninverting input
through a 10 mF capacitor. The gain of the amplifier is con-
trolled by the feedback resistors and is: (R2/R1) + 1. For this
example, the gain is 6. By removing R1 altogether, the amplifier
would have unity gain. A 169 W resistor is placed at the output
in the feedback network to short-circuit protect the output of
the device. This would prevent any damage to the device from
occurring if the headphone output became shorted. A 270 mF
capacitor is used at the output to couple the amplifier to the
headphone. This value is much larger than that used for the
input because of the low impedance of headphones, which can
range from 32 W to 600 W or more.
0.010
V
A
= ؎2.5V
= 1
= 1.0V rms
= 10k⍀
S
V
V
IN
R
L
BANDWIDTH:
<10Hz TO 22kHz
0.001
0.0001
R1 = 10k⍀
10F
R2 = 50k⍀
20
100
1k
10k 20k
FREQUENCY – Hz
5V
Figure 38. THD+N vs. Frequency Graph
LEFT IN
10F
10k⍀
270F
47k⍀
L VOLUME
CONTROL
169⍀
HEADPHONE
LEFT
OP262-A
–40
–50
–60
–70
A
= 1
V
V
= 1.0V rms
5V
IN
(0dBV)
100k⍀
100k⍀
10F
R
= 10k⍀
= ؎2.5V
L
S
V
–80
5V
–90
–100
–110
10k⍀
10F
270F
47k⍀
169⍀
HEADPHONE
RIGHT
OP262-B
R VOLUME
CONTROL
RIGHT IN
–120
R2 = 50k⍀
–130
–140
R1 = 10k⍀
20
100
1k
10k 20k
FREQUENCY – Hz
10F
Figure 39. Crosstalk vs. Frequency Graph
Figure 40. Headphone Output Amplifier
PCB Layout Considerations
Because the OP162/OP262/OP462 can provide gain at high
frequency, careful attention to board layout and component
selection is recommended. As with any high speed application,
a good ground plane is essential to achieve the optimum perfor-
mance. This can significantly reduce the undesirable effects of
ground loops and I¥R losses by providing a low impedance refer-
ence point. Best results are obtained with a multilayer board
design with one layer assigned to ground plane.
Instrumentation Amplifier
Because of its high speed, low offset voltages and low noise
characteristics, the OP162/OP262/OP462 can be used in a wide
variety of high speed applications, including a precision instru-
mentation amplifier. Figure 41 shows an example of such an
application.
–V
IN
Chip capacitors should be used for supply bypassing, with one
end of the capacitor connected to the ground plane and the
other end connected within 1/8 inch of each power pin. An
additional large tantalum electrolytic capacitor (4.7 mF–10 mF)
should be connected in parallel. This capacitor does not need to
be placed as close to the supply pins, as it is to provide current
for fast large-signal changes at the device’s output.
OP462-A
2k⍀
1k⍀
2k⍀
10k⍀
2k⍀
OP462-D
OP462-C
OUTPUT
R
G
1k⍀
1.9k⍀
10k⍀
OP462-B
200⍀
APPLICATION CIRCUITS
Single Supply Stereo Headphone Driver
10 TURN
(OPTIONAL)
+V
IN
Figure 40 shows a stereo headphone output amplifier that can
be run from a single +5 V supply. The reference voltage is
derived by dividing the supply voltage down with two 100 kW
resistors. A 10 mF capacitor prevents power supply noise from
contaminating the audio signal and establishes an ac ground for
the volume control potentiometers.
Figure 41. A High Speed Instrumentation Amplifier
–12–
REV. D
OP162/OP262/OP462
Direct Access Arrangement
The differential gain of the circuit is determined by RG, where:
Figure 44 shows a schematic for a +5 V single supply transmit/
receive telephone line interface for 600 W transmission systems.
It allows full duplex transmission of signals on a transformer
coupled 600 W line. Amplifier A1 provides gain that can be
adjusted to meet the modem output drive requirements. Both
A1 and A2 are configured so as to apply the largest possible
differential signal to the transformer. The largest signal available
on a single +5 V supply is approximately 4.0 V p-p into a 600 W
transmission system. Amplifier A3 is configured as a difference
amplifier to extract the receive information from the transmis-
sion line for amplification by A4. A3 also prevents the transmit
signal from interfering with the receive signal. The gain of A4
can be adjusted in the same manner as A1’s to meet the modem’s
input signal requirements. Standard resistor values permit the
use of SIP (Single In-line Package) format resistor arrays. Couple
this with the OP462 14-lead SOIC or TSSOP package and this
circuit can offer a compact solution.
2
ADIFF = 1+
RG
with the RG resistor value in kW. Removing RG will set the cir-
cuit gain to unity.
The fourth op amp, OP462-D, is optional and is used to im-
prove CMRR by reducing any input capacitance to the ampli-
fier. By shielding the input signal leads and driving the shield
with the common-mode voltage, input capacitance is eliminated
at common-mode voltages. This voltage is derived from the
midpoint of the outputs of OP462-A and OP462-B by using two
10 kW resistors followed by OP462-D as a unity gain buffer.
It is important to use 1% or better tolerance components for the
2 kW resistors, as the common-mode rejection is dependent on
their ratios being exact. A potentiometer should also be con-
nected in series with the OP462-C noninverting input resistor to
ground to optimize common-mode rejection.
P1
The circuit in Figure 41 was implemented to test its settling
time. The instrumentation amp was powered with ± 5 V, so the
input step voltage went from –5 V to +4 V to keep the OP462
within its input range. Therefore, the 0.05% settling range is
when the output is within 4.5 mV. Figure 42 shows the positive
slope settling time to be 1.8 ms, and Figure 43 shows a settling
time of 3.9 ms for the negative slope.
TX GAIN
ADJUST
R2
9.09k⍀
C1
R1
TRANSMIT
TXA
TO TELEPHONE
LINE
0.1F
2k⍀
10k⍀
R3
2
360⍀
1
7
1:1
A1
3
R5
6.2V
6.2V
Z
O
10k⍀
600⍀
5V DC
T1
R6
6
5
R7
10k⍀
MIDCOM
10k⍀
5mV
2V
671-8005
A2
100
90
R8
10F
10k⍀
R9
R10
10k⍀
10k⍀
P2
RX GAIN
ADJUST
R13
R14
RECEIVE
RXA
2
3
10k⍀ 14.3k⍀
R11
10k⍀
1
10
A3
0%
2k⍀
C2
0.1F
6
R12
7
1s
A4
10k⍀
5
A1, A2 = 1/2 AD8532
A3, A4 = 1/2 AD8532
Figure 42. Positive Slope Settling Time
Figure 44. A Single-Supply Direct Access Arrangement for
Modems
5mV
2V
100
90
10
0%
1s
Figure 43. Negative Slope Settling Time
REV. D
–13–
OP162/OP262/OP462
Spice Macro-Model
* OP162/OP262/OP462 SPICE Macro-model
* 7/96, Ver. 1
* Troy Murphy / ADSC
E1 23
R6 23
R7 24
C5 23
*
98 (21, 98)
24 53E+3
98 53E+3
24 1E-12
2
*
* Copyright 1996 by Analog Devices
*
* SECOND GAIN STAGE
*
* Refer to “README.DOC” file for License Statement. Use of this model
* indicates your acceptance of the terms and provisions in the License
* Statement
*
G3 25
R8 25
D3 25
D4 50
*
98 (24, 98) 40E-6
98 1.65E+6
99 DX
25 DX
* Node Assignments
*
*
*
*
*
*
*
noninverting input
* OUTPUT STAGE
*
|
|
|
|
|
|
1
inverting input
|
|
|
|
|
2
positive supply
GSY 99
R9 99
R10 20
Q3 45
Q4 45
EB1 99
EB2 42
RB1 40
RB2 42
CF 45
D5 46
D6 47
V3 46
V4 47
.
50 POLY (1)
(99, 50) 277.5E-6 7.5E-6
|
|
|
negative supply
|
|
20 100E3
50 100E3
41 99 POUT
43 50 NOUT
40 POLY (1)
50 POLY (1)
41 500
43 500
25 11E-12
99 DX
output
|
|
4
2
|
|
.SUBCKT OP162
*
*INPUT STAGE
*
Q1
Q2
Ios
I1
EOS 7
RC1 5
RC2 6
RE1 3
RE2 4
99
50
45
(98, 25) 0.70366
(25, 98) 0.73419
1
1
5
6
1
7
2
2
3
4
PIX 5
PIX 5
1.25E-9
43 DX
41 0.7
50 0.7
99 15 85E-6
POLY(1) (14, 20) 45E-6 1
1
50 3.035E+3
50 3.035E+3
15 607
MODEL PIX
PNP (Bf=117.7)
.MODEL POUT PNP (BF=119, IS=2.782E-17, VAF=28, KF=3E-7)
.MODEL NOUT NPN (BF=110, IS=1.786E-17, VAF=90, KF=3E-7)
15 607
C1
D1
D2
V1 99
V2 99
*
5
3
4
6
8
9
8
9
600E-15
DX
DX
DC 1
DC 1
.MODEL DX
.ENDS
D()
* 1st GAIN STAGE
*
EREF 98
0
(20, 0)
98 10 (5, 6)
10 98
10 98 3.3E-9
1
10.5
G1
R1
C2
*
1
* COMMON-MODE STAGE WITH ZERO AT 4kHz
*
ECM 13 98 POLY (2) (1, 98) (2, 98)
0
0.5 0.5
R2
R3
C3
*
13 14 1E+6
14 98 70
13 14 80E-12
* POLE AT 1.5MHz, ZERO AT 3MHz
*
G2 21 98 (10, 98) .588E-6
R4 21 98 1.7E6
R5 21 22 1.7E6
C4 22 98 31.21E-15
*
* POLE AT 6MHz, ZERO AT 3MHz
*
–14–
REV. D
OP162/OP262/OP462
OUTLINE DIMENSIONS
14-Lead Standard Small Outline Package [SOIC]
Narrow Body
8-Lead Standard Small Outline Package [SOIC]
Narrow Body
(RN-14)
(RN-8)
Dimensions shown in millimeters and (inches)
Dimensions shown in millimeters and (inches)
8.75 (0.3445)
8.55 (0.3366)
5.00 (0.1968)
4.80 (0.1890)
8
1
5
4
14
1
8
7
4.00 (0.1575)
3.80 (0.1496)
6.20 (0.2441)
5.80 (0.2283)
6.20 (0.2440)
5.80 (0.2284)
4.00 (0.1574)
3.80 (0.1497)
1.75 (0.0689)
1.35 (0.0531)
1.27 (0.0500)
BSC
0.50 (0.0197)
0.25 (0.0098)
0.50 (0.0196)
0.25 (0.0099)
1.27 (0.0500)
BSC
؋
45؇ ؋
45؇ 0.25 (0.0098)
0.10 (0.0039)
1.75 (0.0688)
1.35 (0.0532)
0.25 (0.0098)
8؇
0؇
0.10 (0.0040)
0.51 (0.0201)
0.33 (0.0130)
SEATING
PLANE
1.27 (0.0500)
0.40 (0.0157)
8؇
0.25 (0.0098)
0.19 (0.0075)
COPLANARITY
0.10
0.51 (0.0201)
0.33 (0.0130)
0؇ 1.27 (0.0500)
COPLANARITY
0.10
0.25 (0.0098)
0.19 (0.0075)
SEATING
PLANE
0.41 (0.0160)
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MS-012AA
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
8-Lead Thin Shrink Small Outline Package [TSSOP]
(RU-8)
14-Lead Thin Shrink Small Outline Package [TSSOP]
(RU-14)
Dimensions shown in millimeters
Dimensions shown in millimeters
3.10
3.00
2.90
5.10
5.00
4.90
8
5
4
14
8
7
4.50
4.40
4.30
4.50
4.40
4.30
6.40 BSC
6.40
BSC
1
1
PIN 1
PIN 1
0.65
BSC
1.05
1.00
0.80
0.65
BSC
0.15
0.05
0.20
0.09
1.20
1.20
0.75
0.60
0.45
MAX
MAX
8؇
0؇
8؇
0؇
0.15
0.05
0.30
0.19
0.75
0.60
0.45
0.30
0.19
SEATING
PLANE
0.20
0.09
SEATING
PLANE
COPLANARITY
0.10
COPLANARITY
0.10
COMPLIANT TO JEDEC STANDARDS MO-153AA
COMPLIANT TO JEDEC STANDARDS MO-153AB-1
REV. D
–15–
OP162/OP262/OP462
Revision History
Location
Page
10/02—Data Sheet changed from REV. C to REV. D.
Deleted 8-Lead Plastic DIP (N-8) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Universal
Deleted 14-Lead Plastic DIP (N-14) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Universal
Edits to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Edits to Figure 30 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Edits to Figure 31 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Updated OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
–16–
REV. D
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