OP176GBC [ADI]
Bipolar/JFET, Audio Operational Amplifier; 双极性/ JFET ，音频运算放大器
| 型号： | OP176GBC |
| 厂家： | 
ADI |
| 描述： | Bipolar/JFET, Audio Operational Amplifier |
| 文件： | 总21页 (文件大小：683K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Bipolar/JFET,
Audio Operational Amplifier
a
OP176*
FEATURES
PIN CONNECTIONS
Low Noise: 6 nV/√Hz
High Slew Rate: 25 V/µs
8-Lead Narrow-Body SO
(S Suffix)
8-Lead Epoxy DIP
(P Suffix)
Wide Bandwidth: 10 MHz
Low Supply Current: 2.5 mA
Low Offset Voltage: 1 mV
Unity Gain Stable
NULL
–IN
NULL
–IN
NC
1
8
7
6
5
NC
1
2
3
4
8
7
6
5
OP176
2
3
4
V+
V+
SO-8 Package
OP176
+IN
OUT
NULL
+IN
OUT
NULL
APPLICATIONS
Line Driver
V–
V–
Active Filters
Fast Amplifiers
Integrators
200 µV. This allows the OP176 to be used in many dc coupled
or summing applications without the need for special selections
or the added noise of additional offset adjustment circuitry.
GENERAL DESCRIPTION
The OP176 is a low noise, high output drive op amp that
features the Butler Amplifier front-end. This new front-end
design combines both bipolar and JFET transistors to attain
amplifiers with the accuracy and low noise performance of
bipolar transistors, and the speed and sound quality of JFETs.
Total Harmonic Distortion plus Noise equals previous audio
amplifiers, but at much lower supply currents.
The output is capable of driving 600 Ω loads to 10 V rms while
maintaining low distortion. THD + Noise at 3 V rms is a low
0.0006%.
The OP176 is specified over the extended industrial (–40°C to
+85°C) temperature range. OP176s are available in both plastic
DIP and SO-8 packages. SO-8 packages are available in 2500
piece reels. Many audio amplifiers are not offered in SO-8
surface mount packages for a variety of reasons, however, the
OP176 was designed so that it would offer full performance in
surface mount packaging.
Improved dc performance is also provided with bias and offset
currents greatly reduced over purely bipolar designs. Input
offset voltage is guaranteed at 1 mV and is typically less than
*Protected by U.S. Patent No. 5101126.
7
RB4
RB6
RB7
RB5
QB6
RB2
RB3
QB5
QB4
QB7
R4
CB1
QB3
Q10
RS1
J1
J2
Q1
Z2
Q2
QS1
2
3
Q9
R5
6
JB1
CCB
CF
RS2
QS2
CC2
Q6
QB2
Q5
Q4
QS3
Q3
Q8
Q11
RB1
R3
Q7
R2L
R1L
R2P1
R2P2
R1P1
QB8
QB9
QB1
Z1
CC1
R1P2
R2S
R1S
R2A
R1A
1
5
4
Simplified Schematic
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood. MA 02062-9106, U.S.A.
Tel: 617/329-4700 Fax: 617/326-8703
OP176–SPECIFICATIONS
(@ VS = ±15.0 V, TA = +25°C unless otherwise noted)
ELECTRICAL CHARACTERISTICS
Parameter
Symbol
Conditions
Min
Typ
Max
Units
INPUT CHARACTERISTICS
Offset Voltage
Offset Voltage
VOS
VOS
IB
1
mV
mV
nA
nA
nA
nA
V
–40°C ≤ TA ≤ +85°C
VCM = 0 V
VCM = 0 V, –40°C ≤ TA ≤ +85°C
VCM = 0 V
1.25
350
400
±50
±100
+10.5
Input Bias Current
Input Offset Current
IOS
VCM = 0 V, –40°C ≤ TA ≤ +85°C
Input Voltage Range
VCM
–10.5
Common-Mode Rejection
CMRR
VCM = ±10.5 V,
–40°C ≤ TA ≤ +85°C
RL = 2 kΩ
RL = 2 kΩ, –40°C ≤ TA ≤ +85°C
RL = 600 Ω
80
250
175
106
dB
Large Signal Voltage Gain
Offset Voltage Drift
AVO
V/mV
V/mV
V/mV
µV/°C
200
5
∆VOS/∆T
OUTPUT CHARACTERISTICS
Output Voltage Swing
VO
ISC
RL = 2 kΩ, –40°C ≤ TA ≤ +85°C
RL = 600 Ω, VS = ±18 V
–13.5
–14.8
±25
+13.5
+14.8
V
V
mA
Output Short Circuit Current
±50
POWER SUPPLY
Power Supply Rejection Ratio
PSRR
ISY
VS = ±4.5 V to ±18 V
–40°C ≤ TA ≤ +85°C
VS = ±4.5 V to ±18 V, VO = 0 V,
RL = ∞, –40°C ≤ TA ≤ +85°C
VS = ±22 V, VO = 0 V, RL = ∞,
–40°C ≤ TA ≤ +85°C
86
80
108
dB
dB
Supply Current
2.5
mA
Supply Current
ISY
2.75
±22
mA
V
Supply Voltage Range
VS
±4.5
DYNAMIC PERFORMANCE
Slew Rate
Gain Bandwidth Product
SR
GBP
RL = 2 kΩ
15
25
10
V/µs
MHz
AUDIO PERFORMANCE
THD + Noise
VIN = 3 V rms,
RL = 2 kΩ, f = 1 kHz
f = 1 kHz
0.001
6
0.5
%
nV/√Hz
pA/√Hz
Voltage Noise Density
Current Noise Density
en
in
f = 1 kHz
Specifications subject to change without notice.
–2–
REV. 0
OP176
(@ V = ±15.0 V, T = +25°C unless otherwise noted)
WAFER TEST LIMITS
Parameter
S
A
Symbol
Conditions
Limit
Units
Offset Voltage
Input Bias Current
VOS
IB
IOS
VCM
CMRR
PSRR
AVO
VO
1
mV max
nA max
nA max
V min
dB min
dB min
V/mV min
V min
VCM = 0 V
VCM = 0 V
350
±50
±10.5
80
Input Offset Current
Input Voltage Range1
Common-Mode Rejection
Power Supply Rejection Ratio
Large Signal Voltage Gain
Output Voltage Range
VCM = ±10.5 V
V = ±4.5 V to ±18 V
RL = 2 kΩ
86
250
13.5
14.8
2.75
2.5
RL = 2 kΩ
VS = ±18.0 V, RL = 600 Ω
VS = ±22.0 V, VO = 0 V, RL = ∞
VS = ±4.5 V to ±18 V,
VO = 0 V, RL = ∞
V min
mA max
mA max
Supply Current
ISY
NOTES
Electrical tests and wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed for standard
product dice. Consult factory to negotiate specifications based on dice lot qualifications through sample lot assembly and testing.
1Guaranteed by CMR test.
ABSOLUTE MAXIMUM RATINGS1
DICE CHARACTERISTICS
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±22 V
Input Voltage2. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±18 V
NULL
Differential Input Voltage2 . . . . . . . . . . . . . . . . . . . . . . . ±7.5 V
Output Short-Circuit Duration to GND . . . . . . . . . . Indefinite
Storage Temperature Range
OUT
V+
P, S Package . . . . . . . . . . . . . . . . . . . . . . . .–65°C to +150°C
Operating Temperature Range
OP176G . . . . . . . . . . . . . . . . . . . . . . . . . . . .–40°C to +85°C
Junction Temperature Range
P, S Package . . . . . . . . . . . . . . . . . . . . . . . .–65°C to +150°C
Lead Temperature Range (Soldering, 60 sec) . . . . . . . +300°C
V
–
3
NULL
Package Type
θJA
θJC
Units
–IN
+IN
8-Pin Plastic DIP (P)
8-Pin SOIC (S)
103
158
43
43
°C/W
°C/W
OP176 Die Size 0.069 × 0.067 Inch, 4,623 Sq. Mils.
Substrate (Die Backside) Is Connected to V–.
Transistor Count, 26.
NOTES
1Absolute maximum ratings apply to both DICE and packaged parts, unless
otherwise noted.
2For input voltages greater than ±7.5 V limit input current to less than 5 mA.
3θJA is specified for the worst case conditions, i.e., θJA is specified for device in socket
for P-DIP packages; θJA is specified for device soldered in circuit board for SOIC
package.
ORDERING GUIDE
Temperature Range Package Description
Model
Package Option
OP176GP
OP176GS
–40°C to +85°C
–40°C to +85°C
8-Pin Plastic DIP
8-Pin SOIC
N-8
SO-8
OP176GSR –40°C to +85°C
OP176GBC +25°C
SO-8 Reel, 2500 Pieces
DICE
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the OP176 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
REV. 0
–3–
OP176–Typical Characteristics
120
30
25
20
15
10
5
±V = ±15V
±V = ±15V
S
S
100
ΩT = +25°C
≤–40°C ≤ T ≤ +85°C
A
A
R
= 2k
L
80
60
40
20
0
BASED ON 300 OP AMPS
0
0
1
2
3
4
5
6
7
8
1k
10k
100k
1M
10M
FREQUENCY – Hz
µ
tC
V
– µV/°C
OS
Figure 1. Input Offset Voltage Drift Distribution @ ±15 V
Figure 4. Maximum Output Swing vs. Frequency
16
16
V
T
= ±15V
= +25°C
S
ΩV = ±18V, +V , R = 600Ω
S
OM
L
A
14
12
10
8
V = ±18V, –V
, R = 600Ω
S
OM
L
POSITIVE SWING
15
14
13
12
NEGATIVE SWING
V = ±15V, +V , R = 2kΩ
S
OM
L
6
V = ±15V, –V
, R = 2kΩ
S
OM
L
V = ±15V, +V , R = 600Ω
4
2
0
S
OM
L
V = ±15V, –V
, R = 600Ω
S
OM
L
10
100
1k
10k
–50
–25
0
25
50
75
100
ΩLOAD RESISTANCE – Ω
TEMPERATURE – °C
Figure 2. Output Swing vs. Temperature
Figure 5. Maximum Output Swing vs. Load Resistance
2.50
300
±V = ±15V
S
V
= 0V
250
200
150
100
50
CM
2.25
T
= +85°C
= +25°C
A
2.00
1.75
1.50
T
A
T
= –40°C
A
0
–50
±5±
±10±
±15±
±20±
±25±
–25
0
25
50
75
100
0
SUPPLY VOLTAGE – V
TEMPERATURE – °C
Figure 3. Input Bias Current vs. Temperature
Figure 6. Supply Current per Amplifier vs. Supply Voltage
–4–
REV. 0
OP176
80
70
60
50
40
30
20
10
0
120
100
80
60
40
20
0
T
= +25°C
±V = ±15V
A
S
+PSRR
±V = ±15V
S
SINK
SOURCE
–PSRR
100
1k
10k
100k
1M
–50
–25
0
25
50
75
100
FREQUENCY – Hz
TEMPERATURE – °C
Figure 10. Power Supply Rejection vs. Frequency
Figure 7. Short Circuit Current vs. Temperature @ ±15 V
120
2000
±V = ±15V
T
= +25°C
= ±15V
Ω
= >600
S
A
100
80
60
40
20
0
1750
±V = ±10V
V
R
O
S
L
1500
GAIN
Ω–GAIN, R = 2kΩ
L
1250
1000
90
PHASE
+GAIN, R = 2kΩ
L
135
180
225
750
–GAIN, R = 600Ω
PHASE MARGIN = 60°
L
500
250
–20
–40
–60
+GAIN, R = 600Ω
L
0
1k
10k
100k
1M
10M
100M
–50
–25
0
25
50
75
100
FREQUENCY – Hz
TEMPERATURE – °C
Figure 8. Open-Loop Gain & Phase vs. Frequency
Figure 11. Open-Loop Gain vs. Temperature
50
40
T
= +25°C
= ±15V
T
V
= +25°C
= ±15V
A
A
40
30
V
S
S
30
20
10
0
20
10
A
= +100
V
0
–10
–20
–30
A
= +10
V
A
= +1
V
100
1k
10k
100k
1M
1k
10k
100k
1M
10M
100M
FREQUENCY – Hz
FREQUENCY – Hz
Figure 9. Closed-Loop Gain vs. Frequency
Figure 12. Closed-Loop Output Impedance vs. Frequency
REV. 0
–5–
OP176
140
65
60
55
50
45
14
12
10
8
T
= +25°C
= ±15V
A
±V = ±15V
S
V
120
100
80
60
40
20
0
S
PHASE
GAIN
6
125
100
1k
10k
100k
1M
–75
–50
–25
0
25
50
75
100
FRERQUENCY – Hz
TEMPERATURE – °C
Figure 16. Gain Bandwidth Product & Phase Margin vs.
Temperature
Figure 13. Common-Mode Rejection vs. Frequency
100
50
±V = ±15V
S
90
ΩR = 2kΩ
L
NEGATIVE SWING
POSITIVE SWING
V
= 100mVp-p
80
70
60
50
40
30
20
10
0
40
IN
AV = 1
CL
NEGATIVE SLEW RATE
30
POSITIVE SLEW RATE
20
V
= ±15V
S
R
= 2kΩ
L
SWING = ±10V
10
0
SLEW WINDOW = ±5V
T
= +25°C
A
0
100 200 300 400 500 600 700 800 900 1000
LOAD CAPACITANCE – pF
0
200 400 600 800 1000 1200 1400 1600 1800 2000
LOAD CAPACITANCE – pF
Figure 14. Small Signal Overshoot vs. Load Capacitance
Figure 17. Slew Rate vs. Load Capacitance
40
35
30
25
20
15
10
5
35
±V = ±15V
S
ΩR = 2kΩ
30
25
20
15
10
5
ΩV = ±15V
S
SR–
L
R
= 2kΩ
L
T
= +25°C
A
SR+
SR+ AND SR–
0
0
–50
–25
0
25
50
75
100
0
0.4
0.8
1.2
1.6
2.0
TEMPERATURE – °C
DIFFERENTIAL INPUT VOLTAGE – V
Figure 15. Slew Rate vs. Differential Input Voltage
Figure 18. Slew Rate vs. Temperature
–6–
REV. 0
OP176
2.5
2.0
25
20
15
10
5
V
T
= ±15V
S
±V = ±15V
S
= +25°C
A
ΩT = +25°C
A
1.5
1.0
0.5
0
0
10
100
1k
FREQUENCY – Hz
10k
10
100
FREQUENCY – Hz
1k
10k
Figure 19. Voltage Noise Density vs. Frequency
Figure 21. Current Noise Density vs. Frequency
100
90
100
90
V
V
OUT
OUT
(5V/DIV)
(50mV/DIV)
10
10
0%
0%
50mV
100nS
500nS
5V
TIME –100ns/DIV
TIME – 500ns/DIV
Figure 20. Small Signal Transient Response
Figure 22. Large Signal Transient Response
REV. 0
–7–
OP176
APPLICATIONS
0.1
Short Circuit Protection
±V = ±18V
S
The OP176 has been designed with output short circuit
protection. The typical output drive current is ±50 mA. This
high output current and wide output swing combine to yield an
excellent audio amplifier, even when driving large signals, at low
power and in a small package.
ΩR = 600Ω
L
0.010
Total Harmonic Distortion
Total Harmonic Distortion + Noise (THD + N) of the OP176
is well below 0.001% with any load down to 600 Ω. However,
this is dependent upon the peak output swing. In Figure 23 it is
seen that the THD + Noise with 3 V rms output is below
0.001%. In the following Figure 24, THD + Noise is below
0.001% for the 10 kΩ and 2 kΩ loads but increases to above
0.01% for the 600 Ω load condition. This is a result of the
output swing capability of the OP176. Notice the results in
Figure 25, showing THD vs. VIN (V rms).
10Vrms
Ω5Vrms
Ω3Vrms
0.001
Ω1Vrms
.0001
20
100
1k
10k
20k
0.1
Figure 25. THD + Noise vs. Output Amplitude (V rms)
±V = ±15V
S
V
= 3Vrms
O
The output of the OP176 is designed to maintain low harmonic
distortion while driving 600 Ω loads. However, driving 600 Ω
loads with very high output swings results in higher distortion if
clipping occurs.
0.010
To attain low harmonic distortion with large output swings,
supply voltages may be increased. Figure 26 shows the perfor-
mance of the OP176 driving 600 Ω loads with supply voltages
varying from ±18 volts to ±20 volts. Notice that with ±18 volt
supplies the distortion is fairly high, while with ±20 volt supplies
it is a very low 0.0007%.
Ω600Ω
0.001
0.1
.0001
20
100
1k
10k
20k
ΩR = 600Ω
L
FIGURE 23. THD + Noise vs. Frequency
0.010
0.1
V
= ±18V
±22V
O
±V = ±18V
S
V
= 10Vrms
O
0.001
0.010
±19V
±20V
0.0001
20
100
1k
10k
20k
0.001
Ω600Ω
Figure 26. THD + Noise vs. Supply Voltage
10kΩ
2kΩ
0.0001
20
100
1k
10k
20k
Figure 24. THD + Noise vs. RLOAD
–8–
REV. 0
OP176
Noise
If the original 5534 socket includes offset nulling circuitry, one
would find a 10 kΩ to 100 kΩ potentiometer connected between
Pins 1 and 8 with said potentiometer’s wiper arm connected to
V+. In order to upgrade the socket to the OP176, this circuit
should be removed before inserting the OP176 for its offset
nulling scheme uses Pins 1 and 5. Whereas the wiper arm of the
5534 trimming potentiometer is connected to the positive
supply, the OP176’s wiper arm is connected to the negative
supply. Directly substituting the OP176 into the original socket
would inject a large current imbalance into its input stage. In
this case, the potentiometer should be removed altogether, or, if
nulling is still required, the trimming potentiometer should be
rewired to match the nulling circuit as illustrated in Figure 29.
The voltage noise density of the OP176 is below 6 nV/√Hz from
30 Hz. This enables low noise designs to have good perfor-
mance throughout the full audio range. Figure 27 shows a
typical OP176 with a 1/f corner at 6 Hz.
CH A: 80.0 µV FS
10.0 µV /DIV
MKR: 15.9 µV/ Hz
+V
S
7
\
0
Hz
50Hz /
300 mHz
2
3
5.4 Hz
BW:
MKR:
V
OUT
6
OP176
5
Figure 27. 1/f Noise Corner
P1
1
Noise Testing
4
For audio applications the noise density is usually the most
important noise parameter. For characterization the OP176 is
tested using an Audio Precision, System One. The input signal
to the Audio Precision must be amplified enough to measure
accurately. For the OP176 the noise is gained by approximately
1020 using the circuit shown in Figure 28. Any readings on the
Audio Precision must then be divided by the gain. In imple-
menting this test fixture, good supply bypassing is essential.
ΩP1 = 10kΩ
TRIM RANGE = ±2mV
V
OS
–V
S
Figure 29. Offset Voltage Nulling Scheme
Input Overcurrent Protection
The maximum input differential voltage that can be applied to
the OP176 is determined by a pair of internal Zener diodes
connected across its inputs. They limit the maximum differen-
tial input voltage to ±7.5 V. This is to prevent emitter-base
junction breakdown from occurring in the input stage of the
OP176 when very large differential voltages are applied.
However, in order to preserve the OP176’s low input noise
voltage, internal resistances in series with the inputs were not
used to limit the current in the clamp diodes. In small signal
applications, this is not an issue; however, in applications where
large differential voltages can be inadvertently applied to the
device, large transient currents can flow through these diodes.
Although these diodes have been designed to carry a current of
±5 mA, external resistors as shown in Figure 30 should be used
in the event that the OP176’s differential voltage were to exceed
±7.5 V.
OP176
OP37
OUTPUT
OP37
909Ω
100Ω
909Ω
100Ω
4.42kΩ
490Ω
Figure 28. Noise Test
Upgrading “5534‘’ Sockets
The OP176 is a superior amplifier for upgrading existing
designs using the industry standard 5534. In most application
circuits, the OP176 can directly replace the 5534 without any
modifications to the surrounding circuitry. Like the 5534, the
OP176 follows the industry standard, single op amp pinout. The
difference between these two devices is the location of the null
pins and the 5534’s compensation capacitor.
1.4k
2
–
6
OP176
1.4k
3
+
The 5534 normally requires a 22 pF capacitor between Pins 5
and 8 for stable operation. Since the OP176 is internally
compensated for unity gain operation, it does not require
external compensation. Nevertheless, if the 5534 socket already
includes a capacitor, the OP176 can be inserted without
removing it. Since the OP176’s Pin 8 is a “NO CONNECT’’
pin, there is no internal connection to that pin. Thus, the 22 pF
capacitor would be electrically connected through Pin 5 to the
internal nulling circuitry. With the other end left open, the
capacitor should have no effect on the circuit. However, to
avoid altogether any possibility for noise injection, it is recom-
mended that the 22 pF capacitor be cut out of the circuit
entirely.
Figure 30. Input Overcurrent Protection
REV. 0
–9–
OP176
Output Voltage Phase Reversal
+15V
Since the OP176’s input stage combines bipolar transistors for
low noise and p-channel JFETs for high speed performance, the
output voltage of the OP176 may exhibit phase reversal if either
of its inputs exceeds the specified negative common-mode input
voltage. This might occur in some applications where a trans-
ducer, or a system, fault might apply very large voltages upon
the inputs of the OP176. Even though the input voltage range
of the OP176 is ±10.5 V, an input voltage of approximately
–13.5 V will cause output voltage phase reversal. In inverting
amplifier configurations, the OP176’s internal 7.5 V clamping
diodes will prevent phase reversal; however, they will not
prevent this effect from occurring in noninverting applications.
For these applications, the fix is a 3.92 kΩ resistor in series
with the noninverting input of the device and is illustrated in
Figure 31.
10µF
+
0.1µF
2
3
7
V
6
OUT
OP176
V
IN
R
2kΩ
L
4
0.1µF
10µF
R
*
FB
–15V
Figure 33. Unity Gain Follower
V
OUT
6
2
3
V
OP176
IN
+15V
ΩR
L
10µF
+
ΩR
S
2kΩ
3.92kΩ
*R IS OPTIONAL
0.1µF
FB
10pF
Figure 31. Output Voltage Phase Reversal Fix
Overdrive Recovery
4.99kΩ
V
IN
4.99kΩ
2
3
7
The overdrive recovery time of an operational amplifier is the
time required for the output voltage to recover to a rated output
level from a saturated condition. This recovery time is impor-
tant in applications where the amplifier must recover quickly
after a large abnormal transient event. The circuit shown in
Figure 32 was used to evaluate the OP176’s overload recovery
V
OUT
6
OP176
2kΩ
4
2.49kΩ
0.1µF
time. The OP176 takes approximately 1 µs to recover to VOUT
+10 V and approximately 900 ns to recover to VOUT = –10 V.
=
10µF
+
–15V
R1
R2
Ω1kΩ
Ω10kΩ
Figure 34. Unity Gain Inverter
2
3
V
OUT
In inverting and noninverting applications, the feedback
resistance forms a pole with the source resistance and capaci-
tance (RS and CS) and the OP176’s input capacitance (CIN), as
shown in Figure 35. With RS and RF in the kΩ range, this pole
can create excess phase shift and even oscillation. A small
capacitor, CFB, in parallel with RFB eliminates this problem. By
setting RS (CS + CIN) = RFB CFB, the effect of the feedback pole is
completely removed.
6
OP176
R
V
L
IN
R
909Ω
S
4Vp-p
@100Hz
Ω2.43kΩ
Figure 32. Overload Recovery Time Test Circuit
High Speed Operation
C
FB
As with most high speed amplifiers, care should be taken with
supply decoupling, lead dress, and component placement.
Recommended circuit configurations for inverting and
R
FB
noninverting applications are shown in Figure 33 and Figure 34.
V
OUT
R
C
C
IN
S
S
Figure 35. Compensating the Feedback Pole
–10–
REV. 0
OP176
R
R
F
Attention to Source Impedances Minimizes Distortion
Since the OP176 is a very low distortion amplifier, careful
attention should be given to source impedances seen by both
inputs. As with many FET-type amplifiers, the p-channel
JFETs in the OP176’s input stage exhibit a gate-to-source
capacitance that varies with the applied input voltage. In an
inverting configuration, the inverting input is held at a virtual
ground and, as such, does not vary with input voltage. Thus,
since the gate-to-source voltage is constant, there is no distor-
tion due to input capacitance modulation. In noninverting
applications, however, the gate-to-source voltage is not
constant. The resulting capacitance modulation can cause
distortion above 1 kHz if the input impedance is > 2 kΩ and
unbalanced.
G
V
IN
C
F
R
X
V
OP176
OUT
C
L
R
= R
R
WHERE R = OPEN-LOOP OUTPUT RESISTANCE
X
O
G O
R
F
I
+
R
R
F
G
=
I +
C
C
R
(
)
(
R
F
)
[
]
F
L O
|A
|
CL
Figure 37. In-the-Loop Compensation Technique for
Driving Capacitive Loads
Figure 36 shows some guidelines for maximizing the distortion
performance of the OP176 in noninverting applications. The
best way to prevent unwanted distortion is to ensure that the
parallel combination of the feedback and gain setting resistors
(RF and RG) is less than 2 kΩ. Keeping the values of these
resistors small has the added benefits of reducing the thermal
noise of the circuit and dc offset errors. If the parallel combina-
tion of RF and RG is larger than 2 kΩ, then an additional
resistor, RS, should be used in series with the noninverting
APPLICATIONS USING THE OP176
A High Speed, Low Noise Differential Line Driver
The circuit of Figure 38 is a unique line driver widely used in
many applications. With ±18 V supplies, this line driver can
deliver a differential signal of 30 V p-p into a 2.5 kΩ load. The
high slew rate and wide bandwidth of the OP176 combine to
yield a full power bandwidth of 130 kHz while the low noise
front end produces a referred-to-input noise voltage spectral
density of 15 nV/√Hz. The circuit is capable of driving lower
impedance loads as well. For example, with a reduced output
level of 5 V rms (14 V p-p), the circuit exhibits a full-power
bandwidth of 190 kHz while driving a differential load of 249 Ω!
The design is a transformerless, balanced transmission system
where output common-mode rejection of noise is of paramount
importance. Like the transformer-based design, either output
can be shorted to ground for unbalanced line driver applications
without changing the circuit gain of 1. Other circuit gains can
be set according to the equation in the diagram. This allows the
design to be easily set for noninverting, inverting, or differential
operation.
R
R
G
F
V
OP176
OUT
R *
S
V
IN
* R = R //R IF R //R > 2kΩ
S
G
F
G
F
FOR MINIMUM DISTORTION
Figure 36. Balanced Input Impedance to Mininize
Distortion in Noninverting Amplifier Circuits
input. The value of RS is determined by the parallel combina-
tion of RF and RG to maintain the low distortion performance of
the OP176. For a more generalized treatment on circuit
impedances and their effects on circuit distortion, please review
the section on Active Filters at the end of the Applications
section.
R3
2kΩ
R9
2
50Ω
6
A2
V
O1
3
R7
R11
2kΩ
1kΩ
R1
2kΩ
Driving Capacitive Loads
R4
As with any high speed amplifier, care must be taken when
driving capacitive loads. The graph in Figure 14 shows the
OP176’s overshoot versus capacitive load. The test circuit is a
standard noninverting voltage follower; it is this configuration
that places the most demand on an amplifier’s stability. For
capacitive loads greater than 400 pF, overshoot exceeds 40%
and is roughly equivalent to a 45° phase margin. If the applica-
tion requires the OP176 to drive loads larger than 400 pF, then
external compensation should be used.
V
IN
2kΩ
V
– V = V
O1
3
2
O2
IN
6
P1
10kΩ
A1
R5
2kΩ
R6
2kΩ
R2
2kΩ
R12
1kΩ
R10
50Ω
2
6
A3
V
O2
3
R8
2kΩ
Figure 37 shows a simple circuit which uses an in-the-loop
compensation technique that allows the OP176 to drive any
capacitive load. The equations in the figure allow optimization
of the output resistor, RX, and the feedback capacitor, CF, for
optimal circuit stability. One important note is that the circuit
bandwidth is reduced by the feedback capacitor, CF, and is
given by:
A1, A2, A3 = OP176
R3
R1
GAIN =
SET R2, R4, R5 = R1 AND R6, R7, R8 = R3
Figure 38. A High Speed, Low Noise Differential Line
Driver
1
BW =
2 π RF CF
REV. 0
–11–
OP176
A Low Noise Microphone Preamplifier with a Phantom
1.0
Power Option
Figure 39 is an example of a circuit that combines the strengths
of the SSM2017 and the OP176 into a variable gain micro-
phone preamplifier with an optional phantom power feature.
The SSM2017’s strengths lie in its low noise and distortion, and
gain flexibility/simplicity. However, rated only for 2 kΩ or
higher loads, this makes driving 600 Ω loads somewhat limited
with the SSM2017 alone. A pair of OP176s are used in the
circuit as a high current output buffer (U2) and a DC servo
stage (U3). The OP176’s high output current drive capability
provides a high level drive into 600 Ω loads when operating
from ±18 V supplies. For a complete treatment of the circuit
design details, the interested reader should consult application
note AN-242, available from Analog Devices.
±V = ±18V
S
80kHz LPF
0.1
G = 2000
0.010
G = 200
G = 4
G = 20
This amplifier’s performance is quite good over programmed
gain ranges of 2 to 2000. For a typical audio load of 600 Ω,
THD + N at various gains and an output level of 10 V rms is
illustrated in Figure 40. For all but the very highest gain, the
THD + N is consistent and well below 0.01%, while the gain of
2000 becomes more limited by noise. The noise performance of
the circuit is exceptional with a referred-to-input noise voltage
spectral density of 1 nV/√Hz at a circuit gain of 1000.
0.001
20
100
1k
10k
20k
Figure 40. Low Noise Microphone Preamplifier THD + N
Performance at Various Gains (VOUT = 10 V rms and
RL = 600 Ω)
+48V
+18V
+V
S
+
C8
47µF/
63V
+
+
ΩR10
100Ω
C6
C3
PHANTOM POWER SUPPLY CONNECTIONS,
INTERLOCKED WITH +/–V (SEE NOTE 5).
S
0.1µF
100µF/25V
R9
6.81kΩ
R8
6.81kΩ
C7
0.1µF
C4
100µF/25V
–18V
–V
S
Z1
Z2
C5
33pF
1)
R2
+
–IN
20kΩ
+V
2200µF/ 10V
+
U1
SSM2017P
S
7
4
ΩR 1
C
1
P
IN
C
2
R
1
RF
B
TO MICROPHONE
COMMON
49.9Ω
47µF/
63V
–V
100pF
U2
10kΩ
S
C
1
R1
G
OP176
3
C
N
ΩR3
49.9Ω
10kΩ
4
7
6
2
8
1
4.7nF/
FILM
3)
2
R
G
6
5
OUTPUT
3
2
C
R7
1kΩ
G
R
2
B
+
C
1
RF
10kΩ
100pF
R4
2200µF/ 10V
Z4
221kΩ
+
–V
+V
S
S
+IN
OUT COMMON
R6
10kΩ
ΩR 2
49.9Ω
C
47µF/
63V
2
C1
1µF FILM
P
IN
Z3
D1
U3
OP176
1N458
D2
+V
S
1N458
R5
NOTES:
1) Z1–Z4 1N752 (SEE TEXT).
2) C , C LOW LEAKAGE ELECTROLYTIC TYPES (SEE TEXT).
7
2
3
6
221kΩ
INX
GX
3) GAIN = G = 2 x ((10k/R ) +1) (SEE TEXT).
G
4
4) ALL RESISTORS 1% METAL FILM.
5) DOTTED PHANTOM POWER RELATED COMPONENTS OPTIONAL (SEE TEXT).
–V
S
C2
1µF FILM
Figure 39. A Low Noise Microphone Preamplifier
–12–
REV. 0
OP176
A Low Noise, +5 V/+10 V Reference
A Differential ADC Driver
In many high resolution applications, voltage reference noise
can be a major contributor to overall system error. Monolithic
voltage references often exhibit too much wide band noise to be
used alone in these systems. Only through careful filtering and
buffering of these monolithic references can one realize wide-
band microvolt noise levels. The circuit illustrated in Figure 41
is an example of a low noise precision reference optimized for
both ac and dc performance around the OP176. With a +10 V
reference (the AD587), the circuit exhibits a 1 kHz spot output
noise spectral density < 10 nV/√Hz. The reference output
voltage is selectable between 5 V and 10 V, depending only on
the selection of the monolithic reference. The output table
illustrated in the figure provides a selection of monolithic
references compatible with this circuit.
High performance audio sigma-delta ADCs, such as the stereo
16-bit AD1878 and the 18-bit AD1879, present challenging
design problems with regards to input interfacing. Because of
an internal switched capacitor input circuit, the ADC input
structure presents a difficult dynamic load to the drive amplifier
with fast transient input currents due to their 3 MHz ADC
sampling rate. Also, these ADCs inputs are differential with a
rated full-scale range of ±6.3 V, or about 4.4 V rms. Hence, the
ADC interface circuit of Figure 42 is designed to accept a
balanced input signal to drive the low dynamic impedances seen
at the inputs of these ADCs. The circuit uses two OP176
+12V
+V
S
ANALOG
0.1µF
0.1µF
100µ/25V
TO
U1, U2
COM
OUTPUT TABLE
TOLERANCE
100µ/25V
C1
100pF
–12V
ANALOG
–V
V
(+/–mV)
U1
S
OUT
10V
10V
10V
10V
5V
5V
5V
5V
5 TO 10
AD587
REF01
REF10
AD581
REF195
AD586
REF02
REF05
R2
R1
ΩR5
51Ω
30 TO 100
30 TO 50
5 TO 30
5.62kΩ
5.76kΩ
BALANCED
C4
0.01µF
TO
INPUTS
2 TO 10
V
V
–
+
AD1878/
AD1879
SIGMA-
DELTA
ADC
(+)
U1
IN
R4
2.5 TO 20
15 TO 50
15 TO 25
100Ω
V
OUT
U1, U2 = OP176
C2 100pF
(–)
C3
0.0047µF
C3
100µF/25V
U2
6
L & R
OP176
IN
+15V
R3
5.49kΩ
U2
ΩR6
51Ω
R4
INPUTS
4
2
7
R1
R3
100Ω
5.62kΩ
C5
0.01µF
R6
3.3Ω
2
1kΩ
3
6
5
8
= AG, PIN 10 OR 18
U1
C1
100µF/25V
R2
10kΩ
(+)
R5
1.1kΩ
4
NOTES
C4
0.1µF
C1–C5 = NPO CERAMIC, NON-INDUCTIVE,
C3-C5 CLOSE TO ADC
R1–R6 = 1% METAL FILM
R
5kΩ
TRIM
C5
10µF/25V
5kΩ
C2
100µF/25V
USE
FOR
SINGLE-ENDED
INPUTS
10kΩ
(OPTIONAL)
REF
COMMON
Figure 41. A Low Noise, +5 V/+10 V Reference
Figure 42. A Balanced Driver Circuit for Sigma-Delta ADCs
In operation, the basic reference voltage is set by U1, either a
5 V or 10 V 3-terminal reference chosen from the table. In this
case, the reference used is a 10 V buried Zener reference, but
all U1 IC types shown can plug into the pinout and can be
optionally trimmed. The stable 10 V from the reference is then
applied to the R1-C1-C2noise filter, which uses electrolytic
capacitors for a low corner frequency. When electrolytic
capacitors are used for filtering, one must be cognizant of their
dc leakage current errors. Here, however, a dc bootstrap of C1
is used, so this capacitor sees only the small R2 dc drop as bias,
effectively lowering its leakage current to negligible levels. The
resulting low noise, dc-accurate output of the filter is then
buffered by a low noise, unity gain op amp using an OP176.
With the OP176’s low VOS and control of the source resistances,
the dc performance of this circuit is quite good and will not
compromise voltage reference accuracy and/or drift. Also, the
OP176 has a typical current limit of 50 mA, so it can provide
higher output currents when compared to a typical IC reference
alone.
amplifiers as inverting low-pass filters for their speed and high
output current drive. The outputs of the OP176s then drive the
differential ADC inputs through an RC network. This RC
network buffers the amplifiers against step changes at the ADC
sampling inputs using one differential (C3) and two common-
mode connected capacitors (C4 and C5). The 51 Ω series
resistors isolate the OP176s from the heavily capacitive loads,
while the capacitors absorb the transient currents. Operating on
±12 V supplies, this circuit exhibits a very low THD + N of
0.001% at 5 V rms outputs. For single-ended drive sources, a
third op amp unity gain inverter can be added between R2’s (+)
input terminal and R4. For best results, short-lead, noninduc-
tive capacitors are suggested for C3, C4, and C5 (which are
placed close to the ADC), and 1% metal-film types for R1
through R6. For surface mount PCBs, these components can
be NPO ceramic chip capacitors and thin-film chip resistors.
REV. 0
–13–
OP176
parasitics. One percent metal-film resistors and two percent
film capacitors of polystyrene or polypropylene are recom-
mended. Using the suggested values, the frequency response
relative to the ideal RIAA characteristic is within ±0.2 dB over
20 Hz–20 kHz. Even tighter response can be achieved by using
the alternate values, shown in brackets “[ ],” with the trade-off
of a non off-the-shelf part.
An RIAA Phono Preamp
Figure 43 illustrates a simple phono preamplifier using RIAA
equalization. The OP176 is used here to provide gain and is
chosen for its low input voltage noise and high speed perfor-
mance. The feedback equalization network (R1, R2, C1, and
C2) forms a three time constant network, providing reasonably
accurate equalization with standard component values. The
input components terminate a moving magnet phono cartridge
as recommended by the manufacturer, the element values
shown being typical. When this ac coupled circuit is built with a
low noise bipolar input device such as the OP176, amplifier bias
current makes direct cartridge coupling difficult. This circuit
uses input and output capacitor coupling to minimize biasing
interactions.
As previously mentioned, the OP176 was chosen for three
reasons: (1) For optimal circuit noise performance, the
amplifier used should exhibit voltage and current noise densities
of 5 nV/√Hz and 1 pA/√Hz, respectively. (2) For high gain
accuracy, especially at high stage gains, the amplifier should
exhibit a gain bandwidth product in excess of 5 MHz. (3)
Equally important because of the 100% feedback through the
network at high frequencies, the amplifier must be unity gain
stable. With the OP176, the circuit exhibits low distortion over
the entire range, generally well below 0.01% at outputs levels of
5 V rms using ±18 V supplies. To achieve maximum perfor-
mance from this high gain, low level circuit, power supplies
should be well regulated and noise free, and care should be
taken with shielding and conductor layout.
Input ac coupling to the amplifier is provided via C5, and the
low frequency termination resistance, RT, is the parallel equiva-
lent of R6 and R7. R3 of the feedback network is ac grounded
via C4, a large value electrolytic. Additionally, this resistor is
set to a low value to minimize circuit noise from nonamplifier
sources. These design measures reduce the dc offset at the
output of the OP176 to a few millivolts. The output coupling
network of C3 and R4 is shown as suitable for wide band
response, but it can be set to a 7950 µs time constant for use as
a 20 Hz rumble filter.
Active Filter Circuits Using the OP176
A general active filter topology that lends itself to both high-pass
(HP) and low-pass (LP) filters is the well known Sallen-Key
(SK) VCVS (Voltage-Controlled, Voltage Source) architecture.
This filter type uses the op amp as a fixed gain voltage follower
at either unity or a higher gain. Discussed here are simplified 2-
pole, unity gain forms of these filters, which are attractive for
several reasons: One, at audio frequencies, using an amplifier
with a 10 MHz bandwidth such as the OP176, these filters
exhibit reasonably low sensitivities for unity gain and high
damping (low Q). Second, as voltage followers, they are also
inherently gain accurate within their pass band; hence, no gain
resistor scaling errors are generated. Third, they can also be
made “dc accurate,” with output dc errors of only a few
millivolts. The specific filter response in terms of HP, LP and
damping is determined by the RC network around the op amp,
as shown in Figure 44a.
The 1 kHz gain (“G”) of this circuit, controlled by R3, is
calculated as:
R1
R3
G (@ 1 kHz) = 0.101 × 1 +
For an R3 of 200 Ω, the circuit gain is just under 50 × (≈ 34 dB),
and higher gains are possible by decreasing R3. For any value
of R3, the R5-C6 time constant should be equal to R3 and the
series equivalent of C1 and C2.
Using readily available standard values for network elements
(R1, R2, C1, and C2) makes the design easily reproducible and
inexpensive. These components are ideally high quality
precision types, for low equalization errors and minimum
+V
+18V
S
0.1µF
0.1µF
100µF
100µF
–V
–18V
MOVING
MAGNET
PICKUP
S
C5
100µF/25V
+V
U1
S
C3
100µF/25V
3
2
7
6
ΩR5
499Ω
OP176
R6
Ω100kΩ
ΩR7
100kΩ
C
t
C1
V
OUT
4
0.03µF
2%
150pF
R1
100kΩ
1%
[97.6kΩ
]
–V
S
R2
8.25kΩ
1%
R
= R6| |R7
~ 50kΩ
C2
0.01µF
2%
t
–
[7.87kΩ]
ΩR4
100kΩ
C6
3nF
Ω200Ω (34dB)
Ω100Ω (40dB)
R3
C4
1000µF/16V
Figure 43. An RIAA Phono Preamplifier Circuit
–14–
REV. 0
OP176
High Pass Sections
Low Pass Sections
Figure 44a illustrates the high-pass form of a 2-pole SK filter
using an OP176. For simplicity and practicality, capacitors C1
and C2 are set equal (“C”), and resistors R2 and R1are
adjusted to a ratio, N, which provides the filter damping
coefficient, α, as per the design expressions. This high pass
design is begun with selection of standard capacitor values for
C1 and C2 and a calculation of N. The values for R1 and R2
are then determined from the following expressions:
In the LP SK arrangement of Figure 44b, the R and C elements
are interchanged where the resistors are made equal. Here, the
ratio of C2/C1 (“M”) is used to set the filter α, as noted.
Otherwise, this filter is similar to the HP section, and the
resulting 1 kHz LP response is shown in Figure 45. The design
begins with a choice of a standard capacitor value for C1 and a
calculation of M. This then forces a value of “M × C1” for C2.
Then, the value for R1 and R2 (“R”) is calculated according to
the following equation:
1
R1 =
1
R =
2π × FREQ × C ×
N
2π × FREQ × C1×
M
and
R2 = N × R1
IN
OUT
R1
11k
(11.254k)
C1
0.02µF
IN
OUT
R1
11k
(11.254k)
C1
GIVEN: α, FREQ
R2
11k
(11.254k)
0.01µF
+V
GIVEN: α, FREQ
SET C1 = C2 = C
2
M
4
α2
1
Q
S
OP176
3
α
=
=
C2
0.01µF
7
6
C2
C1
+V
S
1
Q
2
N
4
α2
M =
=
OP176
2
α =
=
C2
0.01µF
3
7
4
6
CHOOSE C1
R2
R1
N =
=
–V
2
S
R2
22k
(22.508k)
C2 = M x C1
4
1
R1 =
1
R =
2 π FREQ x C x
N
–V
S
2 π FREQ x C1 x
M
Z
COMP
R2 = N x R1
1 kHz BW SHOWN
1 kHz BW SHOWN
Z
Z
(LOW PASS)
OUTPUT
COMP
COMP
IN (–)
Z
(HIGH PASS)
OUTPUT
COMP
C2
IN (–)
R2
R1
C1
R2
C1
C2
R1
Figures 44b. Two-Pole Unity Gain HP/LP Active Filters
Figures 44a. Two-Pole Unity Gain HP/LP Active Filters
10.000
LP
In this examples, circuit α (or 1/Q) is set equal to √2, providing
a Butterworth (maximally flat) characteristic. The filter corner
frequency is normalized to 1 kHz, with resistor values shown in
both rounded and (exact) form. Various other 2-pole response
shapes are possible with appropriate selection of α, and fre-
quency can be easily scaled, using inversely proportional R or
C values for a given α. The 22 V/µs slew rate of the OP176 will
support 20 V p-p outputs above 100 kHz with low distortion.
The frequency response resulting with this filter is shown as the
dotted HP portion of Figure 45.
0.0
HP
–10.00
–20.00
–30.00
–40.00
–50.00
–60.00
–70.00
20
100
1k
10k
50k
FREQUENCY – Hz
Figure 45. Relative Frequency Response of 2-Pole, 1 kHz
Butterworth LP (Left) and HP (Right) Active Filters
REV. 0
–15–
OP176
1
0.1
Passive Component Selection for Active Filters
The passive components suitable for active filters deserve more
than casual attention. Resistors should be 1%, low TC, metal-
film types of the RN55 or RN60 style. Capacitors should be 1%
or 2% film types preferably, such as polypropylene or polysty-
rene, or NPO (COG) ceramic for smaller values.
B1
Active Filter Circuit Subtleties
0.010
0.001
0.0001
In designing active filter circuits with the OP176, moderately
low values (10 kΩ or less) for R1 and R2 can be used to
minimize the effects of Johnson noise when critical. The
practical tradeoff is, of course, capacitor size and expense. DC
errors will result for larger values of resistance, unless compen-
sation for amplifier input bias current is used. To add bias
compensation in the HP filter section of Figure 42a, a feedback
compensation resistor equal to R2 can be used. This will
minimize bias current induced offset to the product of the
OP176’s IOS and R2. For an R2 of 25 kΩ, this produces a typical
compensated offset voltage of 50 µV. Similar compensation is
applied to Figure 42b, using a resistance equal to R1+ R2.
Using dc compensation, filter output dc errors using the OP176
will be dominated by its VOS, which is typically 1 mV or less. A
caveat here is that the additional resistors can increase noise
substantially. For example, a 10 kΩ resistor generates ~ 12 nV/
√Hz of noise and is about twice that of the OP176. These
resistors can be ac bypassed to eliminate their noise using a
simple shunt capacitor chosen such that its reactance (XC) is
much less than R at the lowest frequency of interest.
A1
B2
A2
20
100
1k
10 k
20k
FREQUENCY – Hz
Figure 46. THD + N (%) vs. Frequency for Various 1 kHz HP
Active Filters Illustrating the Effects of the ZCOMP Network
Curves A1 and B1 show performance with ZCOMP shorted,
while curves A2 and B2 illustrate operation with ZCOMP active.
For the “A” example values, distortion in the pass band of
1 kHz–20 kHz is below 0.001% compensated, and slightly
higher uncompensated. With the higher impedance “B” net-
work, there is a much greater difference between compensated
and uncompensated responses, underscoring the sensitivity to
higher impedances. Although the positive effect of ZCOMP is seen
for both “A” and “B” cases, there is a buffering effect which
takes place with lower impedances. As case “A” shows, when
using larger capacitance values in the source, the amplifier’s
nonlinear C-V input characteristics have less effect on the
signal.
A more subtle form of ac degradation is also possible in these
filters, namely nonlinear input capacitance modulation. This
issue was previously covered for general cases in the section on
minimizing distortion. In active filter circuits, a fully compen-
sating network (for both dc and ac performance) can be used to
minimize this distortion. To be most effective, this network
(ZCOMP) should include R1through C2 as noted for either filter
type, of the same style and value as their counterparts in the
forward path. The effects of a ZCOMP network on the THD + N
performance of two 1 kHz HP filters is illustrated in Figure 46.
One filter (A) is the example shown in Figure 44a (Curves A1
and A2), while the second (B) uses RC values scaled 10 times
upward in impedance (Curves B1 and B2). Both filters operate
with a 2 V rms input, ±18 V supplies, 100 kΩ loading, and
analyzer bandwidth of 80 kHz.
Thus, to minimize the necessity for the complete ZCOMP com-
pensation, effective filter designs should use the lowest capaci-
tive impedances practical, with an 0.01 µF lower value limit as a
goal for lowest distortion (while lower values can certainly be
used, they may suffer higher distortion without the use of full
compensation). Since most designs are likely to use low relative
impedances for reasons of low noise and offset, the effects of
CM distortion may or may not actually be apparent to a given
application.
–16–
REV. 0
OP176
97
E
P
I
1
98
4
R
R
7
6
5
CM2
8
9
35
D
12
D
15
D
–IN
+IN
Q1
Q2
2
1
V
V
V
N1
V
N3
V
N5
V
C
D1
IN
N1
N3
N5
I
36
D2
10
N2
13
N4
16
N6
C
C
1
OS
N1
D
D
D
E
E
OS
N
3
N2
N4
N6
14
17
11
CM1
5
6
C
2
98
R
R
4
3
E
M
97
V
2
C7
19
D
3
21
23
24
25
26
R
R
R
12
9
G
R
G
R
R
C
G
R
C
E
2
C
G
22
C
1
7
2
8
10
5
4
11
6
13
3
3
4
98
98
D
E
4
REF
20
V
3
51
99
D7
30
D8
G8
R
R
15
17
I
SY
V
D5
4
28
L
F
27
2
29
34
1
V
5
31
G5
C8
R
R
14
D6
D9
F
98
2
33
32
16
R
18
G9
C
9
D10
G6
G7
50
Figure 47. OP176 Spice Model Schematic
REV. 0
–17–
OP176
OP176 SPICE Model
*
* POLE/ZERO PAIR AT 1.5 MHz/2.7 MHz
*
* Node Assignments
*
*
*
*
*
*
*
.SUBCKT OP176
*
* INPUT STAGE & POLE AT 100 MHz
*
R8
R9
C4
G2 98
*
* POLE AT 100 MHz
*
R10 23
C5 23
G3 98
*
* POLE AT 100 MHz
*
R11 24
C6 24
G4 98
*
21
21
22
98
22
98
21
1E3
1.25E3
47.2E-12
(18,28)
1E-3
Output
|
|
98
98
23
1
1.59E-9
(21,28)
1
R3
R4
CIN
CM1 1
CM2 2
C2
I1
IOS
5
6
1
51
51
2
98
98
6
2.487
2.487
3.7E-12
7.5E-12
7.5E-12
320E-12
100E-3
98
98
24
1
1.59E-9
(23,28)
1
5
97
1
* COMMON-MODE GAIN NETWORK WITH ZERO AT
1 kHz
*
4
2
1E-9
EOS 9
3
2
9
4
POLY(1) (26,28)
0.2E-3
1
R12 25
C7 25
R13 26
26
26
98
98
Q1
Q2
R5
R6
D1
D2
EN
5
6
7
8
2
1
3
7
8
QX
QX
1.970
1.970
DZ
DZ
(10,0)
(13,0)
(16,0)
E2
*
25
4
36
36
1
2
1
* POLE AT 100 MHz
*
R14 27
C8 27
G5 98
*
1
1E-3
1E-3
98
98
27
GN1 0
GN2 0
*
1
EREF98
0
0
0
(28,0)
(99,0)
(50,0)
1
1
1
EP
97
EM 51
*
* VOLTAGE NOISE SOURCE
*
DN1 35
DN2 10
VN1 35
VN2 0
*
* CURRENT NOISE SOURCE
*
DN3 12
DN4 13
VN3 12
VN4 0
*
* CURRENT NOISE SOURCE
*
DN5 15
DN6 16
VN5 15
VN6 0
*
10
11
0
DEN
DEN
DC 2
DC 2
11
10E-3
10E-3
10E-3
10E-3
13
14
0
DIN
DIN
DC 2
DC 2
1
1
14
16
17
0
DIN
DIN
DC 2
DC 2
17
* GAIN STAGE & DOMINANT POLE AT 32 Hz
*
R7
C3
18
18
98
98
18
19
51
19
1.243E6
4E-9
(5,6) 4.021E-1
1.35
1.35
DX
G1 98
V2
V3
97
20
D3 18
D4 20 18 DX
*
–18–
REV. 0
OP176
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
8-Lead Plastic DIP (N-8)
8
5
0.280 (7.11)
0.240 (6.10)
PIN 1
1
4
0.325 (8.25)
0.300 (7.62)
0.430 (10.92)
0.348 (8.84)
0.060 (1.52)
0.015 (0.38)
0.195 (4.95)
0.115 (2.93)
0.210
(5.33)
MAX
0.130
(3.30)
MIN
0.160 (4.06)
0.115 (2.93)
0.015 (0.381)
0.008 (0.204)
SEATING
PLANE
0.100
(2.54)
0.070 (1.77)
0.045 (1.15)
0.022 (0.558)
0.014 (0.356)
BSC
8-Lead Narrow-Body SO (SO-8)
8
1
5
4
0.1574 (4.00)
0.1497 (3.80)
PIN 1
0.2440 (6.20)
0.2284 (5.80)
0.1968 (5.00)
0.1890 (4.80)
0.0196 (0.50)
x 45°
0.0099 (0.25)
0.0688 (1.75)
0.0532 (1.35)
0.0098 (0.25)
0.0040 (0.10)
8°
0°
0.0500 (1.27)
0.0160 (0.41)
0.0192 (0.49)
0.0138 (0.35)
0.0500
(1.27)
BSC
0.0098 (0.25)
0.0075 (0.19)
REV. 0
–19–
–20–
OP176
FOR CATALOG
ORDERING GUIDE
Model
Temperature Range Package Description
Package Option*
OP176GP
OP176GS
–40°C to +85°C
–40°C to +85°C
8-Pin Plastic DIP
8-Pin SOIC
N-8
SO-8
OP176GSR –40°C to +85°C
OP176GBC +25°C
SO-8 Reel, 2500 Pieces
DICE
*For outline information see Package Information section.
REV. 0
–21–
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