TMP01_技术文档

Low Power, Programmable

Temperature Controller

a

TMP01*

FUNCTIO NAL BLO CK D IAGRAM

FEATURES

–55؇C to +125؇C (–67؇F to +257؇F) Operation

؎1.0؇C Accuracy Over Tem perature (typ)

Tem perature-Proportional Voltage Output

User Program m able Tem perature Trip Points

User Program m able Hysteresis

20 m A Open Collector Trip Point Outputs

TTL/ CMOS Com patible

Single-Supply Operation (4.5 V to 13.2 V)

Low Cost 8-Pin DIP and SO Packages

TEMPERATURE

SENSOR &

VOLTAGE

2.5V

SENSOR

VREF

1

2

V+

8

7

REFERENCE

R1

R2

R3

SET

HIGH

OVER

WINDOW

COMPARATOR

SET

LOW

3

4

6

5

UNDER

VPTAT

APPLICATIONS

GND

HYSTERESIS

GENERATOR

Over/ Under Tem perature Sensor and Alarm

Board Level Tem perature Sensing

Tem perature Controllers

Electronic Therm ostats

Therm al Protection

TMP01

HVAC System s

Industrial Process Control

Rem ote Sensors

Hysteresis is also programmed by the external resistor chain and

is determined by the total current drawn out of the 2.5 V refer-

ence. T his current is mirrored and used to generate a hysteresis

offset voltage of the appropriate polarity after a comparator has

been tripped. T he comparators are connected in parallel, which

guarantees that there is no hysteresis overlap and eliminates

erratic transitions between adjacent trip zones.

GENERAL D ESCRIP TIO N

T he T MP01 is a temperature sensor which generates a voltage

output proportional to absolute temperature and a control signal

from one of two outputs when the device is either above or

below a specific temperature range. Both the high/low tempera-

ture trip points and hysteresis (overshoot) band are determined

by user-selected external resistors. For high volume production,

these resistors are available on-board.

T he T MP01 utilizes proprietary thin-film resistors in conjunc-

tion with production laser trimming to maintain a temperature

accuracy of ±1°C (typ) over the rated temperature range, with

excellent linearity. T he open-collector outputs are capable of

sinking 20 mA, enabling the T MP01 to drive control relays di-

rectly. Operating from a +5 V supply, quiescent current is only

500 µA (max).

T he T MP01 consists of a bandgap voltage reference combined

with a pair of matched comparators. T he reference provides

both a constant 2.5 V output and a voltage proportional to abso-

lute temperature (VPT AT ) which has a precise temperature co-

efficient of 5 mV/K and is 1.49 V (nominal) at +25°C. T he

comparators compare VPT AT with the externally set tempera-

ture trip points and generate an open-collector output signal

when one of their respective thresholds has been exceeded.

T he T MP01 is available in the low cost 8-pin epoxy mini-DIP

and SO (small outline) packages, and in die form.

*P r otected by U.S. P atent No. 5,195,827.

REV. C

Inform ation furnished by Analog Devices is believed to be accurate and

reliable. However, no responsibility is assum ed by Analog Devices for its

use, nor for any infringem ents of patents or other rights of third parties

which m ay result from its use. No license is granted by im plication or

otherwise under any patent or patent rights of Analog Devices.

One Technology Way, P.O. Box 9106, Norw ood. MA 02062-9106, U.S.A.

Tel: 617/ 329-4700

Fax: 617/ 326-8703

Plastic DIP and Surface Mount Packages

TMP01EP/FP, TMP01ES/FS–SPECIFICATIONS

(V+ = +5 V, GND = O V, –40؇C ≤ T ≤ +85؇C unless otherwise noted)

A

P aram eter

Sym bol

Conditions

Min

Typ

Max

Units

INPUT S SET HIGH, SET LOW

Offset Voltage

Offset Voltage Drift

Input Bias Current, “E”

Input Bias Current, “F”

V_OS

T CV_OS

I_B

0.25

3

25

mV

µV/°C

nA

50

100

I_B

25

nA

OUT PUT VPT AT¹

Output Voltage

Scale Factor

VPT AT

T C_{VPT AT}

T _A= +25°C, No Load

1.49

5

V

mV/K

°C

T emperature Accuracy, “E”

T emperature Accuracy, “F”

T emperature Accuracy, “E”

T emperature Accuracy, “F”

T emperature Accuracy, “E”

T emperature Accuracy, “F”

T emperature Accuracy, “E”

T emperature Accuracy, “F”

Repeatability Error

T_A= +25°C, No Load

–1.5

–3

±0.5

±1.0

±0.75

±1.5

±1

1.5

3

10°C < T_A< 40°C, No Load

–40°C < T _A< 85°C, No Load

–55°C < T _A< 125°C, No Load

Note 4

–3.0

–5.0

3.0

5.0

±2

±1.5

±2.5

0.25

±0.02

°C

∆VPT AT

Degree

%/V

Long T erm Drift Error

Power Supply Rejection Ratio

Notes 2 and 6

T _A= +25°C, 4.5 V ≤ V+ ≤ 13.2 V

0.5

±0.1

PSRR

OUT PUT VREF

Output Voltage, “E”

Output Voltage, “F”

Output Voltage, “E”

Output Voltage, “F”

Output Voltage, “E”

Output Voltage, “F”

Drift

Line Regulation

Load Regulation

Output Current, Zero Hysteresis

Hysteresis Current Scale Factor

T urn-On Settling T ime

VREF

T C_VREF

T _A= +25°C, No Load

–40°C < T _A< 85°C, No Load

–55°C < T _A< 125°C, No Load

2.495

2.490

2.485

2.500

2.5 ± 0.01

2.5 ± 0.015

–10

±0.01

±0.1

7

2.505

2.510

2.515

V

ppm/°C

%/V

%/mA

µA

µA/°C

µs

4.5 V ≤ V+ ≤ 13.2 V

10 µA ≤ I_VREF≤ 500 µA

±0.05

±0.25

I_VREF

SF_HYS

(Note 1)

T o Rated Accuracy

5.0

25

OPEN-COLLECT OR OUT PUT S OVER, UNDER

Output Low Voltage

Output Leakage Current

Fall T ime

V_OL

I_OH

t_HL

I_SINK= 1.6 mA

I_SINK= 20 mA

V+ = 12 V

0.25

0.6

1

0.4

V

µA

ns

100

See T est Load

40

POWER SUPPLY

Supply Range

Supply Current

Power Dissipation

V+

I_SY

P_DISS

4.5

13.2

500

800

2.5

V

Unloaded, +V = 5 V

Unloaded, +V = 13.2 V

+V = 5 V

400

450

2.0

µA

mW

NOT ES

¹K = °C + 273.15.

²Guaranteed but not tested.

³Does not consider errors caused by heating due to dissipation of output load currents.

⁴Maximum deviation between +25°C readings after temperature cycling between –55°C and +125°C.

⁵T ypical values indicate performance measured at T _A= +25°C.

⁶Observed in a group sample over an accelerated life test of 500 hours at 150°C.

Specifications subject to change without notice.

Test Load

V+

1kΩ

20pF

–2–

REV. C

TMP01

TO-99 Metal Can Package (V+ = +5 V, GND = O V, –40؇C ≤ T ≤ +85؇C

A

unless otherwise noted)

TMP01FJ–SPECIFICATIONS

P aram eter

Sym bol

Conditions

Min

Typ

Max

Units

INPUT S SET HIGH, SET LOW

Offset Voltage

Offset Voltage Drift

V_OS

T CV_OS

0.25

3

mV

µV/°C

Input Bias Current, “F”

I_B

25

100

nA

OUT PUT VPT AT¹

Output Voltage

Scale Factor

VPT AT

T C_{VPT AT}

T _A= +25°C, No Load

1.49

5

V

mV/K

T emperature Accuracy, “F”

Repeatability Error

T_A= +25°C, No Load

10°C < T_A< 40°C, No Load

–40°C < T _A< 85°C, No Load

–55°C < T _A< 125°C, No Load

Note 4

–3

±1.0

±1.5

±2

±2.5

0.25

±0.02

3

°C

Degree

%/V

–5.0

5.0

∆VPT AT

Long T erm Drift Error

Power Supply Rejection Ratio

Notes 2 and 6

T _A= +25°C, 4.5 V ≤ V+ ≤ 13.2 V

0.5

±0.1

PSRR

OUT PUT VREF

Output Voltage, “F”

Drift

Line Regulation

Load Regulation

Output Current, Zero Hysteresis

Hysteresis Current Scale Factor

T urn-On Settling T ime

VREF

T C_VREF

T _A= +25°C, No Load

–40°C < T _A< 85°C, No Load

–55°C < T _A< 125°C, No Load

2.490

2.480

2.500

2.5 ± 0.015

–10

±0.01

±0.1

7

2.510

2.520

V

ppm/°C

%/V

%/mA

µA

µA/°C

µs

4.5 V ≤ V+ ≤ 13.2 V

10 µA ≤ I_VREF≤ 500 µA

±0.05

±0.25

I_VREF

SF_HYS

(Note 1)

T o Rated Accuracy

5.0

25

OPEN-COLLECT OR OUT PUT S OVER, UNDER

Output Low Voltage

Output Leakage Current

Fall T ime

V_OL

I_OH

t_HL

I_SINK= 1.6 mA

I_SINK= 20 mA

V+ = 12 V

0.25

0.6

1

0.4

V

µA

ns

100

See T est Load, Note 2

40

POWER SUPPLY

Supply Range

Supply Current

Power Dissipation

V+

I_SY

P_DISS

4.5

13.2

500

800

2.5

V

Unloaded, +V = 5 V

Unloaded, +V = 13.2 V

+V = 5 V

400

450

2.0

µA

mW

NOT ES

¹K = °C + 273.15.

²Guaranteed but not tested.

³Does not consider errors caused by heating due to dissipation of output load currents.

⁴Maximum deviation between +25°C readings after temperature cycling between –55°C and +125°C.

⁵T ypical values indicate performance measured at T _A= +25°C.

⁶Observed in a group sample over an accelerated life test of 500 hours at 150°C.

Specifications subject to change without notice.

REV. C

–3–

TMP01

(V = +5.0 V, GND = 0 V, T = +25؇C, unless otherwise noted)

WAFER TEST LIMITS DD

P aram eter

A

Sym bol

Conditions

Min

Typ

Max

100

1.5

Units

nA

INPUT S SET HIGH, SET LOW

Input Bias Current

I_B

OUT PUT VPT AT

T emperature Accuracy

T_A= +25°C, No Load

°C

OUT PUT VREF

Nominal Value

Line Regulation

Load Regulation

VREF

T _A= +25°C, No Load

4.5 V ≤ V+ ≤ 13.2 V

10 µA ≤ I_VREF≤ 500 µA

2.490

2.510

±0.05

±0.25

V

%/V

%/mA

OPEN-COLLECT OR OUT PUT S OVER, UNDER

Output Low Voltage

Output Leakage Current

V_OL

I_OH

I_SINK= 1.6 mA

I_SINK= 20 mA

0.4

1.0

100

mV

V

µA

POWER SUPPLY

Supply Range

Supply Current

V+

I_SY

4.5

13.2

600

V

µA

Unloaded

NOT ES

Electrical tests are performed at wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed

for standard product dice. Consult factory to negotiate specifications based on dice lot qualification through sample lot assembly and testing.

D ICE CH ARACTERISTICS

D ie Size 0.078 × 0.071 inch, 5,538 sq. mils

(1.98 × 1.80 mm, 3.57 sq. mm)

T ransistor Count: 105

8

7

6

5

1. VREF

2. SETHIGH

3. SETLOW

4. GND (TWO PLACES)

(CONNECTED TO SUBSTRATE)

5. VPTAT

4

6. UNDER

7. OVER

8. V+

2

3

4

1

For additional DICE ordering information, refer to databook.

–4–

REV. C

TMP01

ABSO LUTE MAXIMUM RATINGS

GENERAL D ESCRIP TIO N

Maximum Supply Voltage . . . . . . . . . . . . . . . . –0.3 V to +15 V

Maximum Input Voltage

T he T MP01 is a very linear voltage-output temperature sensor,

with a window comparator that can be programmed by the user

to activate one of two open-collector outputs when a predeter-

mined temperature setpoint voltage has been exceeded. A low

drift voltage reference is available for setpoint programming.

(SET HIGH, SET LOW) . . . . . . . . . –0.3 V to [(V+) +0.3 V]

Maximum Output Current (VREF, VPT AT ) . . . . . . . . . 2 mA

Maximum Output Current (Open Collector Outputs) . . 50 mA

Maximum Output Voltage (Open Collector Outputs) . . . . 15 V

Operating T emperature Range . . . . . . . . . . . . –55°C to +150°C

Dice Junction T emperature . . . . . . . . . . . . . . . . . . . . . +150°C

Storage T emperature Range . . . . . . . . . . . . – 65°C to +150°C

Lead T emperature (Soldering, 60 sec) . . . . . . . . . . . . . +300°C

T he temperature sensor is basically a very accurately tempera-

ture compensated, bandgap-type voltage reference with a buff-

ered output voltage proportional to absolute temperature

(VPT AT ), accurately trimmed to a scale factor of 5 mV/K. See

the Applications Information following.

T he low drift 2.5 V reference output VREF is easily divided ex-

ternally with fixed resistors or potentiometers to accurately es-

tablish the programmed heat/cool setpoints, independent of

temperature. Alternatively, the setpoint voltages can be supplied

by other ground referenced voltage sources such as user-

programmed DACs or controllers. T he high and low setpoint

voltages are compared to the temperature sensor voltage, thus

creating a two-temperature thermostat function. In addition,

the total output current of the reference (I_VREF) determines the

magnitude of the temperature hysteresis band. T he open collec-

tor outputs of the comparators can be used to control a wide va-

riety of devices.

P ackage Type

θ_JA

θ_JC

Units

8-Pin Plastic DIP (P)

8-Lead SOIC (S)

8-Lead T O-99 Can (J)

103¹

158²

150¹

43

18

°C/W

NOT ES

¹θ_JAis specified for device in socket (worst case conditions).

²θ_JAis specified for device mounted on PCB.

CAUTIO N

1. Stresses above those listed under “Absolute Maximum Rat-

ings” may cause permanent damage to the device. T his is a

stress rating only and functional operation at or above this

specification is not implied. Exposure to the above maximum

rating conditions for extended periods may affect device

reliability.

HYSTERESIS

CURRENT

ENABLE

V+

1

8

7

6

5

VREF

CURRENT

MIRROR

I

HYS

2. Digital inputs and outputs are protected, however, permanent

damage may occur on unprotected units from high energy

electrostatic fields. Keep units in conductive foam or packag-

ing at all times until ready to use. Use proper antistatic han-

dling procedures.

SET

HIGH

OVER

UNDER

VPTAT

2

3

WINDOW

COMPARATOR

SET

LOW

3. Remove power before inserting or removing units from their

sockets.

HYSTERESIS

VOLTAGE

1kΩ

VOLTAGE

REFERENCE

AND

SENSOR

4

GND

O RD ERING GUID E

TEMPERATURE

OUTPUT

Tem perature P ackage

P ackage

O ption

Range^l

D escription

TMP01

Model/Grade

T MP01EP

T MP01FP

T MP01ES

T MP01FS

T MP01FJ²

T MP01GBC

XIND

+25°C

Plastic DIP

SOIC

T O-99 Can

Die

N-8

SO-8

H-08A

Figure 1. Detailed Block Diagram

NOT ES

¹XIND = –40°C to +85°C.

²Consult factory for availability of MIL/883 version in T O-99 can.

REV. C

–5–

TMP01

Tem per atur e H yster esis

T he hysteresis current is readily calculated, as shown. For

example, for 2 degrees of hysteresis, I_VREF= 17 µA. Next, the

setpoint voltages V_{SET HIGH}and V_{SET LOW}are determined using

the VPT AT scale factor of 5 mV/K = 5 mV/(°C + 273.15),

which is 1.49 V for +25°C. We then calculate the divider resis-

tors, based on those setpoints. T he equations used to calculate

the resistors are:

T he temperature hysteresis is the number of degrees beyond the

original setpoint temperature that must be sensed by the T MP01

before the setpoint comparator will be reset and the output dis-

abled. Figure 2 shows the hysteresis profile. T he hysteresis is

programmed by the user by setting a specific load on the refer-

ence voltage output VREF. T his output current I_VREFis also

called the hysteresis current, which is mirrored internally and

fed to a buffer with an analog switch.

V_SETHIGH= (T_SETHIGH+ 273.15)(5 mV/°C)

V_SETLOW= (T_SETLOW+ 273.15) (5 mV/°C)

HYSTERESIS

HIGH

HYSTERESIS

LOW

R1 (kΩ) = (V_VREF– V_SETHIGH)/I_VREF

= (2.5 V – V_SETHIGH)/I_VREF

=

HI

HYSTERESIS HIGH =

HYSTERESIS LOW

R2 (kΩ) = (V_SETHIGH– V_SETLOW)/I_VREF

R3 (kΩ) = V_SETLOW/I_VREF

OUTPUT

VOLTAGE

OVER, UNDER

LO

V

= 2.5V

= R1

8

7

6

5

V+

1

VREF

)/I

I

(V

– V

VREF

SETHIGH VREF

V

TEMPERATURE

TMP01

2

3

4

OVER

UNDER

VPTAT

SETHIGH

T_SETLOW

T_SETHIGH

(V

– V

)/I

= R2

SETHIGH

SETLOW VREF

V

SETLOW

= R3

Figure 2. TMP01 Hysteresis Profile

V

/I

SETLOW VREF

After a temperature setpoint has been exceeded and a compara-

tor tripped, the buffer output is enabled. T he output is a cur-

rent of the appropriate polarity which generates a hysteresis

offset voltage across an internal 1000 Ω resistor at the compara-

tor input. T he comparator output remains “on” until the volt-

age at the comparator input, now equal to the temperature

sensor voltage VPT AT summed with the hysteresis offset, has

returned to the programmed setpoint voltage. T he comparator

then returns LOW, deactivating the open-collector output and

disabling the hysteresis current buffer output. T he scale factor

for the programmed hysteresis current is:

GND

Figure 3. TMP01 Setpoint Program m ing

T he total R1 + R2 + R3 is equal to the load resistance needed

to draw the desired hysteresis current from the reference, or

I_VREF

.

T he formulas shown above are also helpful in understanding the

calculation of temperature setpoint voltages in circuits other

than the standard two-temperature thermostat. If a setpoint

function is not needed, the appropriate comparator should be

disabled. SET HIGH can be disabled by tying it to V+, SET -

LOW by tying it to GND. Either output can be left unconnected.

I_HYS= I_VREF= 5 µA/°C + 7 µA

T hus since VREF = 2.5 V, with a reference load resistance of

357 kΩ or greater (output current 7 µA or less), the temperature

setpoint hysteresis will be zero degrees. See the temperature

programming discussion below. Larger values of load resistance

will only decrease the output current below 7 µA and will have

no effect on the operation of the device. T he amount of hyster-

esis is determined by selecting a value of load resistance for

VREF, as shown below.

218

248

273

298

323

348

373

398

K

–18

0

–55

–25

0

25

50

75

100

125

°C

–67

–25

32 50

1.365

77 100

1.49

150

200 212

1.865

257

°F

1.09

1.24

1.615

1.74

1.99

P r ogr am m ing the TMP 01

VPTAT

In the basic fixed-setpoint application utilizing a simple resistor

ladder voltage divider, the desired temperature setpoints are

programmed in the following sequence:

Figure 4. Tem perature—VPTAT Scale

1. Select the desired hysteresis temperature.

2. Calculate the hysteresis current I_VREF

.

3. Select the desired setpoint temperatures.

4. Calculate the individual resistor divider ladder values needed

to develop the desired comparator setpoint voltages at

SET HIGH and SET LOW.

–6–

REV. C

TMP01

Under standing Er r or Sour ces

in practice. Comparator input offset directly impacts the pro-

grammed setpoint voltage and thus the resulting hysteresis

band, and must be included in error calculations.

T he accuracy of the VPT AT sensor output is well characterized

and specified, however preserving this accuracy in a heating or

cooling control system requires some attention to minimizing

the various potential error sources. T he internal sources of

setpoint programming error include the initial tolerances and

temperature drifts of the reference voltage VREF, the setpoint

comparator input offset voltage and bias current, and the hys-

teresis current scale factor. When evaluating setpoint program-

ming errors, remember that any VREF error contribution at the

comparator inputs is reduced by the resistor divider ratios. T he

comparator input bias current (inputs SET HIGH, SET LOW)

drops to less than 1 nA (typ) when the comparator is tripped.

T his can account for some setpoint voltage error, equal to the

change in bias current times the effective setpoint divider ladder

resistance to ground.

External error sources to consider are the accuracy of the pro-

gramming resistors, grounding error voltages, and the overall

problem of thermal gradients. T he accuracy of the external

programming resistors directly impacts the resulting setpoint

accuracy. T hus in fixed-temperature applications the user

should select resistor tolerances appropriate to the desired

programming accuracy. Resistor temperature drift must be

taken into account also. T his effect can be minimized by select-

ing good quality components, and by keeping all components in

close thermal proximity. Applications requiring high measure-

ment accuracy require great attention to detail regarding

thermal gradients. Careful circuit board layout, component

placement, and protection from stray air currents are necessary

to minimize common thermal error sources.

T he thermal mass of the T MP01 package and the degree of

thermal coupling to the surrounding circuitry are the largest

factors in determining the rate of thermal settling, which ulti-

mately determines the rate at which the desired temperature

measurement accuracy may be reached. T hus, one must allow

sufficient time for the device to reach the final temperature.

T he typical thermal time constant for the plastic package is

approximately 140 seconds in still air! T herefore, to reach the

final temperature accuracy within 1%, for a temperature change

of 60 degrees, a settling time of 5 time constants, or 12 min-

utes, is necessary.

Also, the user should take care to keep the bottom of the

setpoint programming divider ladder as close to GND (Pin 4)

as possible to minimize errors due to IR voltage drops and cou-

pling of external noise sources. In any case, a 0.1 µF capacitor

for power supply bypassing is always recommended at the chip.

Safety Considerations In Heating And Cooling System Design

Designers should anticipate potential system fault conditions

which may result in significant safety hazards which are outside

the control of and cannot be corrected by the T MP01-based

circuit. Governmental and industrial regulations regarding

safety requirements and standards for such designs should be

observed where applicable.

T he setpoint comparator input offset voltage and zero hyster-

esis current affect setpoint error. While the 7 µA zero hysteresis

current allows the user to program the T MP01 with moderate

resistor divider values, it does vary somewhat from device to de-

vice, causing slight variations in the actual hysteresis obtained

550

525

500

5.0

4.5

4.0

3.5

3.0

475

+125°C

450

+85°C

425

–55°C

400

375

350

+25°C

–40°C

–75

–50

–25

0

25

50

75

100

125

0

5

10

15

20

TEMPERATURE – °C

SUPPLY VOLTAGE – Volts

Figure 5. Supply Current vs. Supply Voltage

Figure 6. Minim um Supply Voltage vs. Tem perature

REV. C

–7–

TMP01

+2.0

2.510

2.508

2.506

2.504

2.502

2.500

2.498

2.496

2.494

2.492

2.490

X + 3σ

+1.5

+1.0

+0.5

0

V+ = +5V

CURVES NOT NORMALIZED

EXTRAPOLATED FROM OPERATING LIFE DATA

X

–0.5

–1.0

–1.5

–3.0

X – 3σ

–75

–50

–25

0

25

50

75

100

125

0

200

400

600

800

1000

TEMPERATURE –

°C

T = HOURS OF OPERATION AT 125°C; V+ = +5V

Figure 7. VPTAT Accuracy vs. Tem perature

Figure 10. VREF Long Term Drift Accelerated by Burn-In

2.508

100

V+ = +5V

= 10µA

80

2.506

I

VREF

2.504

2.502

60

40

20

2.500

0

2.498

2.496

–20

100

1k

10k

100k

1M

–75

–50

–25

0

25

50

75

100

125

FREQUENCY – Hz

TEMPERATURE – °C

Figure 11. VREF Power Supply Rejection vs. Frequency

Figure 8. VREF Accuracy vs. Tem perature

1.0

6.0

5.0

4.0

3.0

2.0

1.0

0

V

= +15V

C

V+ = +5V

= +25°C

T

A

0.1

V+ = +5V

I

= 7.5µA

VREF

0.01

–75

0

10

20

30

40

50

–50

–25

0

25

50

75

100

125

I

– mA

TEMPERATURE – °C

C

Figure 12. Set High, Set Low Input Offset Voltage vs.

Tem perature

Figure 9. Open-Collector Output (OVER, UNDER) Satura-

tion Voltage vs. Output Current

–8–

REV. C

TMP01

8

7

6

5

4

3

2

1

10

9

V+ = +5V

T

= +25°C

A

I

= 5µA

V+ = +5V

_A= +25°C

VREF

8

T

7

6

5

4

3

2

1

0

–0.4

0

–0.32

–0.24 –0.16 –0.08

OFFSET – mV

0

0.08

0.16

6.2

6.4

6.6

7

7.2

7.4 7.6

7.8

8

6.8

REFERENCE CURRENT – µA

Figure 13. Com parator Input Offset Distribution

Figure 14. Zero Hysteresis Current Distribution

AP P LICATIO NS INFO RMATIO N

Self-H eating Effects

In some applications the user should consider the effects of self-

heating due to the power dissipated by the open-collector out-

puts, which are capable of sinking 20 mA continuously. Under full

load, the T MP01 open-collector output device is dissipating

With excellent drift and noise characteristics, VREF offers a

good voltage reference for data acquisition and transducer exci-

tation applications as well. Output drift is typically better than

–10 ppm/°C, with 315 nV/√Hz (typ) noise spectral density at

1 kHz.

P r eser ving Accur acy O ver Wide Tem per atur e Range

O per ation

P_DISS= 0.6 V × .020A = 12 mW

T he TMP01 is unique in offering both a wide-range temperature

sensor and the associated detection circuitry needed to imple-

ment a complete thermostatic control function in one mono-

lithic device. While the voltage reference, setpoint comparators,

and output buffer amplifiers have been carefully compensated to

maintain accuracy over the specified temperature range, the user

has an additional task in maintaining the accuracy over wide op-

erating temperature ranges in this application. Since the T MP01

is both sensor and control circuit, in many applications it is pos-

sible that the external components used to program and inter-

face the device may be subjected to the same temperature

extremes. T hus it may be necessary to locate components in

close thermal proximity to minimize large temperature differen-

tials, and to account for thermal drift errors where appropriate,

such as resistor matching tempcos, amplifier error drift, and

the like. Circuit design with the T MP01 requires a slightly dif-

ferent perspective regarding the thermal behavior of electronic

components.

which in a surface-mount SO package accounts for a tempera-

ture increase due to self-heating of

∆T = P_DISS× θ_JA= .012 W × 158°C/W = 1.9°C.

T his will of course directly affect the accuracy of the T MP01

and will for example cause the device to switch the heating out-

put “OFF” 2 degrees early. Alternatively, bonding the same

package to a moderate heatsink limits the self-heating effect to

approximately

∆T = P_DISS× θ_JC= .012 W × 43°C/W = 0.52°C.

which is a much more tolerable error in most systems. T he

VREF and VPT AT outputs are also capable of delivering suffi-

cient current to contribute heating effects and should not be

ignored.

Buffer ing the Voltage Refer ence

As mentioned before, the reference output VREF is used to gen-

erate the temperature setpoint programming voltages for the

T MP01 and also is used to determine the hysteresis temperature

band by the reference load current I_VREF. T he on-board output

buffer amplifier is typically capable of 500 µA output drive into

as much as 50 pF load (max). Exceeding this load will affect the

accuracy of the reference voltage, could cause thermal sensing

errors due to dissipation, and may induce oscillations. Selection

of a low drift buffer functioning as a voltage follower with high

input impedance will ensure optimal reference accuracy, and

will not affect the programmed hysteresis current. Amplifiers

which offer the low drift, low power consumption, and low cost

appropriate to this application include the OP295, and members

of the OP90, OP97, OP177 families, and others as shown in the

following applications circuits.

Ther m al Response Tim e

T he time required for a temperature sensor to settle to a speci-

fied accuracy is a function of the thermal mass of the sensor,

and the thermal conductivity between the sensor and the object

being sensed. T hermal mass is often considered equivalent to

capacitance. T hermal conductivity is commonly specified using

the symbol Q, and can be thought of as the reciprocal of thermal

resistance. It is commonly specified in units of degrees per watt

of power transferred across the thermal joint. T hus, the time re-

quired for the T MP01 to settle to the desired accuracy is depen-

dent on the package selected, the thermal contact established in

that particular application, and the equivalent power of the heat

source. In most applications, the settling time is probably best

determined empirically.

REV. C

–9–

TMP01

Switching Loads With The O pen-Collector O utputs

TEMPERATURE

SENSOR &

VOLTAGE

In many temperature sensing and control applications some type

of switching is required. Whether it be to turn on a heater when

the temperature goes below a minimum value or to turn off a

motor that is overheating, the open-collector outputs Over and

Under can be used. For the majority of applications, the switches

used need to handle large currents on the order of 1 amp and

above. Because the T MP01 is accurately measuring tempera-

ture, the open-collector outputs should handle less than 20 mA

of current to minimize self-heating. Clearly, the Over-temp and

Under-temp outputs should not drive the equipment directly.

Instead, an external switching device is required to handle the

large currents. Some examples of these are relays, power

MOSFET s, thyristors, IGBT s, and Darlingtons.

VREF

VPTAT

V+

1

2

8

7

2.4kΩ (12V)

1.2kΩ (6V)

5%

REFERENCE

R1

R2

R3

NC

IRFR9024

OR EQUIV.

WINDOW

COMPARATOR

3

4

6

5

HEATING

ELEMENT

NC

HYSTERESIS

GENERATOR

TMP01

NC = NO CONNECT

Figure 15b. Driving a P-Channel MOSFET

Figure 15 shows a variety of circuits where the T MP01 controls

a switch. T he main consideration in these circuits, such as the

relay in Figure 15a, is the current required to activate the

switch.

TEMPERATURE

SENSOR &

VOLTAGE

VREF

VPTAT

V+

1

2

8

7

REFERENCE

HEATING

ELEMENT

R1

4.7kΩ

+12V

NC

TEMPERATURE

SENSOR &

VOLTAGE

IRF130

R2

R3

VREF

WINDOW

COMPARATOR

VPTAT

1

2

8

2N1711

MOTOR

SHUTDOWN

3

4

6

5

IN4001

REFERENCE

R1

R2

R3

OR EQUIV.

7

NC

HYSTERESIS

GENERATOR

2604-12-311

COTO

WINDOW

COMPARATOR

TMP01

3

4

6

5

NC = NO CONNECT

HYSTERESIS

GENERATOR

Figure 15c. Driving a N-Channel MOSFET

TMP01

Isolated Gate Bipolar T ransistors (IGBT ) combine many of the

benefits of power MOSFET s with bipolar transistors, and are

used for a variety of high power applications. Because IGBT s

have a gate similar to MOSFET s, turning on and off the devices

is relatively simple as shown in Figure 15d. T he turn on voltage

for the IGBT shown (IRGBC40S) is between 3.0 and 5.5 volts.

T his part has a continuous collector current rating of 50 A and a

maximum collector to emitter voltage of 600 V, enabling it to

work in very demanding applications.

Figure 15a. Reed Relay Drive

It is important to check the particular relay you choose to ensure

that the current needed to activate the coil does not exceed the

T MP01’s recommended output current of 20 mA. T his is easily

determined by dividing the relay coil voltage by the specified

coil resistance. Keep in mind that the inductance of the relay

will create large voltage spikes that can damage the T MP01 out-

put unless protected by a commutation diode across the coil, as

shown. T he relay shown has a contact rating of 10 watts maxi-

mum. If a relay capable of handling more power is desired, the

larger contacts will probably require a commensurately larger

coil, with lower coil resistance and thus higher trigger current.

As the contact power handling capability increases, so does the

current needed for the coil. In some cases an external driving

transistor should be used to remove the current load on the

T MP01 as explained in the next section.

TEMPERATURE

SENSOR &

VOLTAGE

VREF

VPTAT

V+

1

2

8

7

MOTOR

CONTROL

REFERENCE

R1

R2

R3

4.7kΩ

NC

IRGBC40S

WINDOW

COMPARATOR

2N1711

3

4

6

5

NC

HYSTERESIS

Power FET s are popular for handling a variety of high current

DC loads. Figure 15b shows the T MP01 driving a p-channel

MOSFET transistor for a simple heater circuit. When the out-

put transistor turns on, the gate of the MOSFET is pulled down

to approximately 0.6 V, turning it on. For most MOSFET s a

gate-to-source voltage or Vgs on the order of –2 V to –5 V is suf-

ficient to turn the device on. Figure 15c shows a similar circuit

for turning on an n-channel MOSFET , except that now the gate

to source voltage is positive. Because of this reason an external

transistor must be used as an inverter so that the MOSFET will

turn on when the “Under T emp” output pulls down.

GENERATOR

TMP01

NC = NO CONNECT

Figure 15d. Driving an IGBT

–10–

REV. C

TMP01

T he last class of high power devices discussed here are T hyris-

tors, which includes SCRs and T riacs. T riacs are a useful alter-

native to relays for switching ac line voltages. T he 2N6073A

shown in Figure 15e is rated to handle 4A (rms). T he

optoisolated MOC3011. T riac shown features excellent electri-

cal isolation from the noisy ac line and complete control over

the high power T riac with only a few additional components.

TEMPERATURE

SENSOR &

VOLTAGE

VREF

VPTAT

V+

4.7kΩ

1

2

8

7

I

C

REFERENCE

R1

R2

R3

2N1711

WINDOW

COMPARATOR

Q1

3

4

6

5

TEMPERATURE

SENSOR &

VOLTAGE

VREF

VPTAT

V+ = 5V

HYSTERESIS

GENERATOR

AC

1

2

8

7

LOAD

TMP01

REFERENCE

R1

R2

R3

300Ω

NC

150Ω

Figure 16a. An External Resistor Minim izes Self-Heating

1

2

3

6

5

4

WINDOW

COMPARATOR

MOC3011

3

4

6

5

TEMPERATURE

SENSOR &

VOLTAGE

VREF

VPTAT

V+

2N6073A

1

2

8

7

I

C

4.7kΩ

HYSTERESIS

REFERENCE

4.7kΩ

R1

R2

R3

GENERATOR

2N1711

TMP01

2N1711

NC = NO CONNECT

WINDOW

COMPARATOR

Q2

Q1

3

4

6

5

Figure 15e. Controlling the 2N6073A Triac

H igh Cur r ent Switching

HYSTERESIS

GENERATOR

As mentioned above, internal dissipation due to large loads on

the T MP01 outputs will cause some temperature error due to

self-heating. External transistors remove the load from the

T MP01, so that virtually no power is dissipated in the internal

transistors and no self-heating occurs. Figure 16 shows a few ex-

amples using external transistors. T he simplest case, using a

single transistor on the output to invert the output signal is

shown in Figure 16a. When the open-collector of the T MP01

turns “ON” and pulls the output down, the external transistor

Q1’s base will be pulled low, turning off the transistor. Another

transistor can be added to reinvert the signal as shown in Figure

16b. Now, when the output of the T MP01 is pulled down, the

first transistor, Q1, turns off and its collector goes high, which

turns Q2 on, pulling its collector low. T hus, the output taken

from the collector of Q2 is identical to the output of the

TMP01

Figure 16b. Second Transistor Maintains Polarity of

TMP01 Output

An example of a higher power transistor is a standard Darling-

ton configuration as shown in Figure 16c. T he part chosen,

T IP-110, can handle 2A continuous which is more than enough

to control many high power relays. In fact the Darlington itself

can be used as the switch, similar to MOSFET s and IGBT s.

T MP01. By picking a transistor that can accommodate large

amounts of current, many high power devices can be switched.

+12V

RELAY

MOTOR

SWITCH

TEMPERATURE

SENSOR &

VOLTAGE

I_C

VREF

VPTAT

V+

1

2

8

7

TIP-110

4.7kΩ

REFERENCE

4.7kΩ

R1

R2

R3

2N1711

WINDOW

COMPARATOR

3

4

6

5

HYSTERESIS

GENERATOR

TMP01

Figure 16c. Darlington Transistor Can Handle Large Currents

REV. C

–11–

TMP01

Buffer ing the Tem per atur e O utput P in

V+

T he VPT AT sensor output is a low impedance dc output volt-

age with a 5 mV/K temperature coefficient, and is useful in a

number of measurement and control applications. In many ap-

plications, this voltage needs to be transmitted to a central loca-

tion for processing. T he buffered VPT AT voltage output is

capable of 500 µA drive into 50 pF (max). As mentioned in the

discussion above regarding buffering circuits for the VREF out-

put, it is useful to consider external amplifiers for interfacing

VPT AT to external circuitry to ensure accuracy, and to mini-

mize loading which could create dissipation-induced tempera-

ture sensing errors. An excellent general-purpose buffer circuit

using the OP177 is shown in Figure 17 which is capable of driv-

ing over 10 mA, and will remain stable under capacitive loads of

up to 0.1 µF. Other interfacing ideas are shown below.

TEMPERATURE

SENSOR &

VOLTAGE

VREF

VPTAT

1

2

8

7

10kΩ

0.1µF

REFERENCE

R1

R2

R3

WINDOW

COMPARATOR

V+

3

4

6

5

V_OUT

100Ω

OP177

VPTAT

HYSTERESIS

GENERATOR

C_L

V–

TMP01

Figure 17. Buffer VPTAT to Handle Difficult Loads

receiving end. Figure 18 shows two amplifiers being used to

send the signal differentially, and an excellent differential

D iffer ential Tr ansm itter

In noisy industrial environments, it is difficult to send an accu-

rate analog signal over a significant distance. However, by send-

ing the signal differentially on a wire pair, these errors can be

significantly reduced. Since the noise will be picked up equally

on both wires, a receiver with high common-mode input rejec-

tion can be used to cancel out the noise very effectively at the

receiver, the AMP03, which features a common-mode rejection

ratio of 95 dB at dc and very low input and drift errors.

V+

TEMPERATURE

SENSOR &

VOLTAGE

VREF

VPTAT

1

2

8

7

REFERENCE

R1

R2

R3

WINDOW

COMPARATOR

10kΩ

3

4

6

5

50Ω

VPTAT

V+

V–

1/2

HYSTERESIS

GENERATOR

OP297

10kΩ

V_OUT

TMP01

AMP03

1/2

OP297

Figure 18. Send the Signal Differentially for Noise Im m unity

–12–

REV. C

TMP01

4 m A-20 m A Cur r ent Loop

high accuracy. For initial accuracy, a 10 kΩ trim potentiometer

can be included in series with R3, and the value of R3 lowered

to 95 kΩ. T he potentiometer should be adjusted to produce an

output current of 12.3 mA at 25°C.

Another, very common method of transmitting a signal over

long distances is to use a 4 mA-20 mA Loop, as shown in Fig-

ure 19. An advantage of using a 4 mA-20 mA loop is that the

accuracy of a current loop is not compromised by voltage drops

across the line. One requirement of 4 mA-20 mA circuits is that

the remote end must receive all of its power from the loop,

meaning that the circuit must consume less than 4 mA. Operat-

ing from +5 V, the quiescent current of the T MP01 is 500 µA

max, and the OP90s is 20 µA max, totaling less than 4 mA.

Although not shown, the open collector outputs and tempera-

ture setting pins can be connected to do any local control of

switching.

Tem per atur e-to-Fr equency Conver ter

Another common method of transmitting analog information is

to convert a voltage to the frequency domain. T his is easily

done with any of the low cost monolithic Voltage-to-Frequency

Converters (VFCs) available, which feature a robust, open-col-

lector digital output. A digital signal is very immune to noise

and voltage drops because the only important information is the

frequency. As long as the conversions between temperature and

frequency are done accurately, the temperature data can be suc-

cessfully transmitted.

T he current is proportional to the voltage on the VPT AT out-

put, and is calibrated to 4 mA at a temperature of –40°C, to

20 mA for +85°C. T he main equation governing the operation

of this circuit gives the current as a function of VPT AT :

A simple circuit to do this combines the T MP01 with an

AD654 VFC, as shown in Figure 20. T he AD654 outputs a

square wave that is proportional to the dc input voltage accord-

ing to the following equation:

1

VPTAT × R5 VREF × R3

R5

R2

I_OUT

=

–

1 +

R6

R2

R3 + R1

V_IN

F_OUT

=

T he resulting temperature coefficient of the output current is

128 µA/°C.

10 (R1 + R2) C_T

By simply connecting the VPT AT output to the input of the

AD654, the 5 mV/°C temperature coefficient gives a sensitivity

of 25 Hz/°C, centered around 7.5 kHz at 25°C. T he trimming

resistor R2 is needed to calibrate the absolute accuracy of the

AD654. For more information on that part, please consult the

AD654 data sheet. Finally, the AD650 can be used to accu-

rately convert the frequency back to a dc voltage on the receiv-

ing end.

1

4

8

5

V+

+5V TO +13.2V

VREF

GND

TMP01

VPTAT

R1

243kΩ

7

2

R2

39.2kΩ

2N1711

6

OP90

R3

100kΩ

4

V+

3

TEMPERATURE

SENSOR &

VOLTAGE

VREF

VPTAT

1

8

7

REFERENCE

R1

R6

100Ω

V+

4–20mA

2

C

T

0.1µF

R2

R5

WINDOW

COMPARATOR

V+

5kΩ

100kΩ

R

L

6

8

7

3

6

1

R3

AD654

VPTAT

5

F

OUT

OSC

4

HYSTERESIS

GENERATOR

Figure 19. 4-20 m A Current Loop

3

TMP01

T o determine the resistor values in this circuit, first note that

VREF remains constant over temperature. T hus the ratio of R5

over R2 must give a variation of I_OUTfrom 4 mA to 20 mA as

VPT AT varies from 1.165 V at –40°C to 1.79 V at +85°C. T he

absolute value of the resistors is not important, only the ratio.

For convenience, 100 kΩ is chosen for R5. Once R2 is calcu-

lated, the value of R3 and R1 is determined by substituting

4 mA for I_OUTand 1.165 V for VPT AT and solving. T he final

values are shown in the circuit. T he OP90 is chosen for this cir-

cuit because of its ability to operate on a single supply and its

R1

2

5

1.8kΩ

R2

500Ω

Figure 20. Tem perature-to-Frequency Converter

REV. C

–13–

TMP01

OP290

V+

TEMPERATURE

SENSOR &

VOLTAGE

VREF

VPTAT

V+

IL300XC

1

2

8

7

1

2

4

REFERENCE

V+

7

R1

R2

R3

6

REF43

3

2

100Ω

2

3

6

WINDOW

COMPARATOR

OP290

4

V+

7

2.5V

6

5

3

4

6

5

3

1.16V TO 1.7V

680pF

IN4148

6

OP90

4

I

1

2

4

2

HYSTERESIS

GENERATOR

R1

470kΩ

TMP01

100kΩ

ISOLATION

BARRIER

604kΩ

680pF

Figure 21. Isolation Am plifier

Isolation Am plifier

example, at room temperature, VPT AT = 1.49 V, so adjust R2

until V_OUT= 1.49 V as well. Both the REF43 and the OP90

operate from a single supply, and contribute no significant error

due to drift.

In many industrial applications the sensor is located in an envi-

ronment that needs to be electrically isolated from the central

processing area. Figure 21 shows a simple circuit that uses an

8-pin optoisolator (IL300XC) that can operate across a 5,000 V

barrier. IC1 (an OP290 single-supply amplifier) is used to drive

the LED connected between Pins 1 to 2. T he feedback actually

comes from the photodiode connected from Pins 3 to 4. T he

OP290 drives the LED such that there is enough current gener-

ated in the photodiode to exactly equal the current derived from

the VPT AT voltage across the 470 kΩ resistor. On the receiving

end, an OP90 converts the current from the second photodiode

to a voltage through its feedback resistor R2. Note that the other

amplifier in the dual OP290 is used to buffer the 2.5 V reference

voltage of the T MP01 for an accurate, low drift LED bias level

without affecting the programmed hysteresis current. A REF43

(a precision 2.5 V reference) provides an accurate bias level at

the receiving end.

In order to avoid the accuracy trim, and to reduce board space,

complete isolation amplifiers are available, such as the high

accuracy AD202.

O ut-of-Range War ning

By connecting the two open collector outputs of the T MP01

together into a “wired-OR” configuration, a temperature “out-

of-range” warning signal is generated. T his can be useful in sen-

sitive equipment calibrated to work over a limited temperature

range. R1, R2, and R3 in Figure 22 are chosen to give a tem-

perature range of 10°C around room temperature (25°C). T hus,

if the temperature in the equipment falls below +15°C or rises

above +35°C, the Undertemp Output or Overtemp Output re-

spectively will go low and turn the LED on. T he LED may be

replaced with a simple pull-up resistor to give a logic output for

controlling the instrument, or any of the switching devices dis-

cussed above can be used.

T o understand this circuit, it helps to examine the overall equa-

tion for the output voltage. First, the current (I1) in the photo-

diode is set by:

V+

TEMPERATURE

SENSOR &

VOLTAGE

2.5 V – VPTAT

I₁=

LED

VREF

VPTAT

1

2

8

7

470 kΩ

R1

47.5kΩ

REFERENCE

200Ω

Note that the IL300XC has a gain of 0.73 (typical) with a min

and max of 0.693 and 0.769 respectively. Since this is less than

1.0, R2 must be larger than R1 to achieve overall unity gain. T o

show this the full equation is:

R2

4.99kΩ

WINDOW

COMPARATOR

3

4

6

5

R3

71.5kΩ

VPTAT

HYSTERESIS

GENERATOR

2. 5 V – VPTAT

V

= 2. 5 V – I

R = 2. 5 V – 0. 7

2 2

644 kΩ = VPTAT

OUT

470 kΩ

TMP01

A trim is included for R2 to correct for the initial gain accuracy

of the IL300XC. T o perform this trim, simply adjust for an out-

put voltage equal to VPT AT at any particular temperature. For

Figure 22. Out-of-Range Warning

–14–

REV. C

TMP01

Tr anslating 5 m V/K to 10 m V/°C

However, the gain from VPT AT to the output is two, so that

5 mV/K becomes 10 mV/°C. T hus, for a temperature of +80°C,

the output voltage is 800 mV. Circuit errors will be due prima-

rily to the inaccuracies of the resistor values. Using 1% resistors

the observed error was less than 10 mV, or 1°C. T he 10 pF

feedback capacitor helps to ensure against oscillations. For bet-

ter accuracy, a adjustment potentiometer can be added in series

with either 100 kΩ resistor.

A useful circuit is shown in Figure 23 that translates the VPT AT

output voltage, which is calibrated in Kelvins, into an output

that can be read directly in degrees Celsius on a voltmeter

display. T o accomplish this, an external amplifier is configured

as a differential amplifier. T he resistors are scaled so the VREF

voltage will exactly cancel the VPT AT voltage at 0.0°C.

10pF

Tr anslating VP TAT to the Fahr enheit Scale

105kΩ

+15V

4.22kΩ

A very similar circuit to the one shown in Figure 23 can be used

to translate VPT AT into an output that can be read directly in

degrees Fahrenheit, with a scaling of 10 mV/°F. Only unity gain

or less is available from the first stage differentiating circuit, so

the second amplifier provides a gain of two to complete the con-

version to the Fahrenheit scale. Using the circuit in Figure 24, a

temperature of 0.0°F gives an output of 0.00 V. At room tem-

perature (70°F) the output voltage is 700 mV. A –40°C to

+85°C operating range translates into –40°F to +185°F. T he

errors are essentially the same as for the circuit in Figure 23.

100kΩ

1

5

7

2

3

VREF

TMP01

VPTAT

V_OUT(10mV/°C)

(V_OUT= 0.0V @ T = 0.0°C)

6

OP177

4

4.12kΩ

487Ω

100kΩ

–15V

Figure 23. Translating 5 m V/K to 10 m V/°C

10pF

100kΩ

90.9kΩ

1.0kΩ

+15V

100kΩ

6

5

100kΩ

1

5

V

= 0.0V @ T = 0.0°F

OUT

2

3

7

4

VREF

TMP01

VPTAT

7

(10mV/°F)

6

6.49kΩ

121Ω

1/2

OP297

1/2

OP297

100kΩ

–15V

Figure 24. Translating 5 m V/K to 10 m V/°F

REV. C

–15–

TMP01

O UTLINE D IMENSIO NS

D imensions shown in inches and (mm).

8-P in Epoxy D IP

8

1

5

0.280 (7.11)

0.240 (6.10)

4

0.070 (1.77)

0.045 (1.15)

0.325 (8.25)

0.300 (7.62)

0.430 (10.92)

0.348 (8.84)

0.015

0.210

(5.33)

MAX

0.195 (4.95)

0.115 (2.93)

(0.381) TYP

0.130

(3.30)

MIN

0.015 (0.381)

0.008 (0.204)

0.160 (4.06)

0.115 (2.93)

SEATING

0°- 15°

0.022 (0.558)

0.014 (0.356)

0.100

(2.54)

BSC

PLANE

8-P in SO IC

8

5

4

0.2440 (6.20)

0.2284 (5.80)

0.1574 (4.00)

0.1497 (3.80)

1

0.1968 (5.00)

0.1890 (4.80)

0.0196 (0.50)

× 45°

0.102 (2.59)

0.094 (2.39)

0.0099 (0.25)

0.0098 (0.25)

0.0040 (0.10)

0°-8°

0.0500 (1.27)

0.0160 (0.41)

0.0098 (0.25)

0.0075 (0.19)

0.0192 (0.49)

0.0138 (0.35)

0.0500 (1.27) BSC

SEATING

PLANE

8-P in TO -99

REFERENCE PLANE

0.750 (19.05)

0.500 (12.70)

0.185 (4.70)

0.165 (4.19)

0.250 (6.35)

MIN

0.050

(1.27)

MAX

0.115

(2.92)

BSC

0.160 (4.06)

0.110 (2.79)

5

6

8

4

2

0.335 (8.51)

0.305 (7.75)

0.045 (1.14)

0.027 (0.69)

0.230

(5.84)

BSC

7

3

0.370 (9.40)

0.335 (8.51)

1

0.115

(2.92)

BSC

0.019 (0.48)

0.016 (0.41)

0.040 (1.02) MAX

0.034 (0.86)

0.027 (0.69)

0.045 (1.14)

0.010 (0.25)

0.021 (0.53)

0.016 (0.41)

45

°

BSC

BASE & SEATING PLANE

–16–

REV. C


型号：	TMP01
厂家：	ADI
描述：	Low Power, Programmable Temperature Controller 低功耗，可编程温度控制器控制器
文件：	总16页 (文件大小：379K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

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