A4935_12 [ALLEGRO]
The A4935 is a 3-phase controller for use with N-channel external power MOSFETs and is specifically designed for automotive applications.; 的A4935是用于与N沟道外部功率MOSFET使用3相控制器，是专为汽车应用而设计的。
| 型号： | A4935_12 |
| 厂家： | 
ALLEGRO MICROSYSTEMS |
| 描述： | The A4935 is a 3-phase controller for use with N-channel external power MOSFETs and is specifically designed for automotive applications. |
| 文件： | 总25页 (文件大小：452K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
A4935
Automotive 3-Phase MOSFET Driver
Features and Benefits
Description
▪ High current 3-phase gate drive for N-channel MOSFETs
▪ Cross-conduction protection with adjustable dead time
▪ Top-off charge pump for 100% PWM
▪ Charge pump for low supply voltage operation
▪ Uncommitted current sense amplifier
▪ 5.5 to 50 V supply voltage range
The A4935 is a 3-phase controller for use with N-channel
external power MOSFETs and is specifically designed for
automotive applications.
A unique charge pump regulator provides full (>10 V) gate
drive for battery voltages down to 7 V and allows the A4935
to operate with a reduced gate drive, down to 5.5 V.
▪ Compatible with 3.3 V and 5 V logic
▪ Extensive diagnostic outputs
▪ Low-current sleep mode
A bootstrap capacitor is used to provide the above-battery
supply voltage required for N-channel MOSFETs.An internal
charge pump for the high-side drive allows DC (100% duty
cycle) operation.
Full control over all six power FETs in the 3-phase bridge is
provided,allowingmotorstobedrivenwithblockcommutation
or sinusoidal excitation. The power FETs are protected from
shoot-through by integrated crossover control and resistor-
adjustable dead time.
Package: 48-pin LQFP with exposed
thermal pad (suffix JP)
Bridge current can be measured using an integrated current
sense amplifier. This is an uncommitted differential amplifier
with a below-ground common mode range allowing it to be
used in low-side current sense applications. Gain and offset
are defined by external resistors.
Continued on the next page…
Not to scale
Typical Application
VBAT
VBAT
A8450
Regulator
3-Phase
BLDC
Motor
Control
A4935
DSP
or
Micro-
controller
Diagnostics
Current Sense
4935-DS, Rev. 4
A4935
Automotive 3-Phase MOSFET Driver
Description (continued)
Integrated diagnostics provide indication of undervoltage,
overtemperature, and power bridge faults. They can be configured
to protect the power FETs under most short circuit conditions.
Detailed diagnostics are available as a serial data word.
TheA4935 is supplied in a 48-pin LQFPwith exposed thermal pad,
(suffix JP). This is a small footprint (81 mm2) power package. It is
lead (Pb) free with 100% matte tin leadframe plating.
Selection Guide
Part Number
Packing
A4935KJPTR-T
1500 pieces per reel
Absolute Maximum Ratings*
Characteristic
Symbol
VBB
Notes
Rating
–0.3 to 50
–0.3 to 7
–0.3 to 16
–0.3 to 16
–0.3 to 6.5
–4 to 6.5
–4 to 6.5
–0.3 to 6.5
–5 to 55
Units
V
Load Supply Voltage
Logic Supply Voltage
VREG
VDD
V
V
CP1 and CP2
V
Logic Inputs and Outputs
CSP and CSN
V
V
LSS
V
CSO and VDSTH
SA, SB, and SC
V
V
RDEAD
–0.3 to 6.5
–5 to 55
V
VDRAIN
V
GHA, GHB, and GHC
GLA, GLB, and GLC
CA, CB, and CC
Operating Temperature Range
Junction Temperature
Sx to Sx+15
–5 to 16
V
V
–0.3 to Sx+15
–40 to 150
150
V
TA
Range K
ºC
ºC
TJ(max)
Overtemperature event not exceeding
10 s, lifetime duration not exceeding 10 hr,
guaranteed by design characterization
Transient Junction Temperature
TtJ
175
ºC
ºC
Storage Temperature Range
*With respect to AGND.
Tstg
–55 to 150
THERMAL CHARACTERISTICS may require derating at maximum conditions
Characteristic
Symbol
Test Conditions*
Value
Units
4-layer PCB based on JEDEC standard
23
ºC/W
RθJA
Package Thermal Resistance
2-layer PCB with 3 in.2 of copper area each side
44
2
ºC/W
ºC/W
RθJP
*Additional thermal information available on Allegro website.
Allegro MicroSystems, Inc.
115 Northeast Cutoff
2
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A4935
Automotive 3-Phase MOSFET Driver
Functional Block Diagram
Battery +
CP
VBB
CP1
CP2
Logic
Supply
VDD
VBAT
Charge
Pump
VREG
Regulator
CREG
COAST
PWMH
PWML
Charge
Pump
VDRAIN
CA
Bootstrap
Monitor
CBOOTA
GHA
SA
High Side
Drive
AHI
ALO
BHI
RGATE
Control
Logic
BLO
CHI
VREG
GLA
Low Side
Drive
Phase C
Phase B
RGATE
CLO
Phase A
(repeated
for B & C)
CCEN
RESET
RDEAD
LSS
Diagnostics and Protection
UVLO, OTF
ESF
CSP
CSN
+
Short to Supply
Short to Gnd
Short Load
FF1
FF2
–
AGND
CSOUT
VDSTH
Allegro MicroSystems, Inc.
115 Northeast Cutoff
3
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A4935
Automotive 3-Phase MOSFET Driver
ELECTRICAL CHARACTERISTICS valid at TJ = –40°C to 150°C, VDD = 3 to 5.5 V, VBB = 7 to 50 V, unless noted otherwise
Characteristics
Supply and Reference
Symbol
Test Conditions
Min.
Typ.
Max.
Units
Load Supply Voltage Functional
Operating Range1
VBB
5.5
–
50
V
IBBQ
IBBS
VDD
IDDQ
IDDS
RESET = high, outputs = low, VBB = 12 V
RESET = low, Sleep mode, VBB = 12 V
–
–
10
–
14
10
mA
μA
V
Load Supply Quiescent Current
Logic Supply Voltage
3.0
–
–
5.5
RESET = high, outputs = low
RESET = low
4
6
mA
μA
V
Logic Supply Quiescent Current
–
–
10
VBB > 9 V, IREG = 0 to 15 mA
7.5 V < VBB ≤ 9 V, IREG = 0 to 10 mA
12.5
12.5
13
13
13.80
13.80
V
VREG Output Voltage
VREG
2×VBB
– 2.5
6 V < VBB ≤ 7.5 V, IREG = 0 to 9 mA
–
–
V
5.5 V < VBB ≤ 6 V, IREG < 8 mA
ID = 10 mA
8.5
0.4
1.5
9.5
0.7
2.2
–
V
V
V
1.0
2.8
Bootstrap Diode Forward Voltage
Bootstrap Diode Resistance
VfBOOT
ID = 100 mA
rD(100mA)
=
rD
6
10
20
Ω
(VfBOOT(150mA) – VfBOOT(50mA)) / 100 mA
Bootstrap Diode Current Limit
Top-off Charge Pump Current Limit
High-Side Gate Drive Static Load Resistance
Gate Output Drive
IDBOOT
ITOCPM
RGSH
250
–
500
400
–
750
–
mA
μA
kΩ
250
–
Turn-On Time
tr
tf
CLOAD = 1 nF, 20% to 80%
CLOAD = 1 nF, 80% to 20%
TJ = 25°C, IGHx = –150 mA
TJ = 150°C, IGHx = –150 mA
TJ = 25°C, IGLx = 150 mA
TJ = 150°C, IGLx = 150 mA
–
–
35
20
8
–
–
ns
ns
Ω
Turn-Off Time
6
12
16
4
Pullup On Resistance
RDS(on)UP
10
2
13
3
Ω
Ω
Pulldown On Resistance
RDS(on)DN
3
4.5
6
Ω
VCx
– 0.2
GHx Output Voltage
VGHX
VGLX
tP(off)
tP(on)
Bootstrap capacitor fully charged
–
–
–
V
V
VREG
– 0.2
GLx Output Voltage
–
Input change to unloaded gate output
change
Turn-Off Propagation Delay2
Turn-On Propagation Delay2
60
60
90
90
150
150
ns
ns
Input change to unloaded gate output
change
Measured between corresponding
transition points on two sequential phases
Propagation Delay Matching, Phase-to-Phase
Propagation Delay Matching, On-to-Off
∆tPP
∆tOO
–
–
10
10
–
–
ns
ns
Measured across one phase
Continued on the next page…
Allegro MicroSystems, Inc.
115 Northeast Cutoff
4
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A4935
Automotive 3-Phase MOSFET Driver
ELECTRICAL CHARACTERISTICS (continued) valid at TJ = –40°C to 150°C, VDD = 3 to 5.5 V, VBB = 7 to 50 V,
unless noted otherwise
Characteristics
Symbol
Test Conditions
Min.
Typ.
0
Max.
Units
ns
RDEAD tied to GND
–
–
–
–
R
DEAD = 3 kΩ
DEAD = 30 kΩ
180
960
3.5
6
ns
Dead Time2
tDEAD
R
815
–
1110
–
ns
RDEAD = 240 kΩ
μs
RDEAD tied to VDD
–
–
μs
Logic Inputs and Outputs
FF1 and FF2 Fault Output
VFF(L)
IFF(H)
VDEAD(L)
IDEAD
IFF = 1 mA, fault not present
VFF = 5 V, fault present
–
–1
–
–
–
–
0.4
1
V
FF1 and FF2 Fault Output Leakage Current3
RDEAD Input Low Voltage
RDEAD Current3
μA
V
–
0.2
–70
RDEAD = GND
–200
μA
0.3 ×
VDD
Input Low Voltage
Input High Voltage
VIN(L)
VIN(H)
–
–
–
V
V
0.7 ×
VDD
–
Input Hysteresis
VINhys
IIN
300
–1
–
500
–
–
1
mV
μA
kΩ
μs
Input Current (Except RESET and CCEN)3
Input Pulldown Resistor (RESET and CCEN)
RESET Pulse Time4
0 V < VIN < VDD
RPD
tRES
tDR
50
–
–
0.1
–
3.5
200
RESET Delay4
–
ns
0.3 ×
VDD
FF2 Clock Input High Voltage
FF2 Clock Input Low Voltage
VILC
VIHC
–
–
–
V
V
0.7 ×
VDD
–
FF2 Clock Input Hysteresis
FF2 Clock Low to Valid Data Delay4
FF2 Clock Low to Fault Reset Delay4
FF2 Clock High Time4
VIChys
tDF
300
–
500
–
–
100
100
–
mV
ns
ns
ns
ns
tRF
–
–
tHF
500
500
–
FF2 Clock Low Time4
tLF
–
–
Current Sense Differential Amplifier
Differential Input Voltage
Input Bias Current3
VID
IBIAS
IOS
VID = CSP – CSN, –1.3 V < VCM < 4 V
CSP = CSN = 0 V
–VDD
–100
–100
–10
–
–
0
VDD
100
100
+10
–
mV
nA
Input Offset Current3
CSP = CSN = 0 V
0
nA
Input Offset Voltage
VIOS
∆VIOS
CMR
AVopn
AVclos
CSP = CSN = 0 V
–
mV
μV/°C
V
Input Offset Voltage Drift
Input Common Mode Range
Open Loop Gain
CSP = CSN = 0 V
10
–
CSP = CSN
–1.5
100
5
4
40 mV< VID < 175 mV, VCM in range
40 mV< VID < 175 mV, VCM in range
–
–
dB
Closed Loop Gain
–
–
V/V
Continued on the next page…
Allegro MicroSystems, Inc.
115 Northeast Cutoff
5
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A4935
Automotive 3-Phase MOSFET Driver
ELECTRICAL CHARACTERISTICS (continued) valid at TJ = –40°C to 150°C, VDD = 3 to 5.5 V, VBB = 7 to 50 V,
unless noted otherwise
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Units
Small Signal –3 dB Frequency Bandwidth
BWG
VID = 10 mVpp, G = 5 V/V
–
1.6
–
MHz
To within 10% of steady state,
VCSOUT = 1 Vpp square wave
Settling Time
tSETTLE
VCSOUT
–
400
–
–
ns
V
VDD
– 0.3
Output Dynamic Range
–100 μA < ICSOUT < 100 μA
0.1
Output Current Sink
ICSsink
ICSsource
PSRR
VID = –400 mV, VCSOUT = 1.5 V
VID = 400 mV, VCSOUT = 1.5 V
CSP = CSN = AGND, 0 to 300 kHz
CSP = CSN = 0 to 200 mV step
1
–
–
–
–
–
–
–1
–
mA
mA
dB
dB
dB
Output Current Source
–
VREG Supply Ripple Rejection
DC Common Mode Rejection
AC Common Mode Rejection
45
38
28
CMRR
CMRR
–
V
CM = 200 mVpp, 0 to 1 MHz
–
To within 100 mV of steady state,
VCM = +4 V step within CMR
Common Mode Recovery Time
Output Slew Rate
tCMrec
SR
–
–
–
1
–
–
–
μs
V/μs
ns
10% to 90%, VID = 0 to 175 mV step
20
To within 10% of steady state,
VID = 250 mV to 0 V step
Input Overload Recovery
Protection
tIDrec
500
VREGUVon
VREGUVoff
VBOOTUV
VBOOTUVhys
VDDUV
VREG rising
7.5
6.75
59
–
8
7.25
–
8.5
7.75
69
V
V
VREG Undervoltage Lockout Threshold
VREG falling
Bootstrap Undervoltage Threshold
Bootstrap Undervoltage Hysteresis
VDD Undervoltage Turn-Off Threshold
VDD Undervoltage Hysteresis
VDSTH Input Range
Cx with respect to Sx
%VREG
%VREG
V
13
–
VDD falling
2.45
50
0.1
–
2.7
100
–
2.85
150
2
VDDUVhys
VDSTH
mV
V
VDSTH Input Current
IDSTH
0 V < VDSTH < 2 V
10
30
μA
VDRAIN Input Voltage
VDRAIN
7
VBB
50
V
VDSTH = 2 V,
VBB = 12 V, 0 V < VDRAIN < VBB
VDRAIN Input Current
IDRAIN
–
–
250
μA
High-side on, VDSTH ≥ 1 V
High-side on, VDSTH < 1 V
Low-side on, VDSTH ≥ 1 V
Low-side on, VDSTH < 1 V
Temperature increasing
Recovery = TJF – TJFhys
–
–150
–
±100
±50
±100
±50
170
15
–
150
–
mV
mV
mV
mV
ºC
Short-to-Ground Threshold Offset5
VSTGO
Short-to-Battery Threshold Offset6
VSTBO
–150
150
–
150
–
Overtemperature Fault Flag Threshold
Overtemperature Fault Hysteresis
TJF
TJFhys
–
ºC
1Functions correctly, but parameters are not guaranteed, below the general limits (7 V).
2See Gate Drive Timing diagrams.
3For input and output current specifications, negative current is defined as coming out of (sourcing) the specified device pin.
4See Fault Output Timing diagram.
5As VSx decreases, fault occurs if VBAT –VSx > VSTG. STG threshold, VSTG = VDSTH + VSTGO
6As VSx increases, fault occurs if VSx – VLSS > VSTB. STB threshold, VSTB = VDSTH+VSTBO
.
.
Allegro MicroSystems, Inc.
115 Northeast Cutoff
6
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A4935
Automotive 3-Phase MOSFET Driver
Timing Diagrams
xHI
xHI
xHI
xLO
xLO
xLO
tDEAD
tP(off)
tP(on)
tP(off)
GHx
GLx
GHx
GLx
GHx
GLx
tP(off)
tDEAD
tP(on)
tP(off)
Synchronous Rectification
High-Side PWM
Low-Side PWM
(A) Gate Drive Timing, Phase Control Inputs
PWMH
PWML
tP(off)
tDEAD
tDEAD
tP(off)
GHx
GLx
GHx
GLx
tDEAD
tP(off)
tP(off)
tDEAD
xHI = 0, xLO = 1, PWMH = 1
xHI = 1, xLO = 0, PWML = 1
(B) Gate Drive Timing, PWM Inputs
COAST
COAST
tP(off)
tP(on)
tP(off)
tP(on)
GHx
GLx
GHx
GLx
xHI = 1, xLO = 0, PWMH = PWML = 1
xHI = 0, xLO = 1, PWMH = PWML = 1
(C) Gate Drive Timing, COAST Inputs
tDF
FF1
FF2
FF1
FF2
tLF
tHF
tRF
tDR
Gate Drive
RESET
Gate Drive
RESET
Disabled
Enabled
Disabled
Enabled
tRES
Fault Register Read
Simple Fault Reset
(D) Fault Output Timing
Allegro MicroSystems, Inc.
115 Northeast Cutoff
7
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A4935
Automotive 3-Phase MOSFET Driver
Functional Description
The A4935 is a three-phase MOSFET driver (pre-driver) with
separate supplies for the logic, and for the analog and drive sec-
tions. This permits operation with a regulated logic supply from
3 to 5.5 V and with an unregulated main supply of 7 to 50 V.
5.5 V. This provides a very rugged solution for use in the harsh
automotive environment.
Gate Drives
The six high current gate drives are capable of driving a wide
range of N-channel power MOSFETs, and are configured as three
high-side drives and three low-side drives. Each drive can be con-
trolled with a logic level input compatible with 3.3 or 5 V logic.
The A4935 provides all the necessary circuits to ensure that the
gate-source voltage of both high-side and low-side external FETs
are above 10 V, at supply voltages down to 7 V. For extreme
battery voltage drop conditions, correct functional operation is
guaranteed at supply voltages down to 5.5 V, but with a reduced
gate drive voltage.
The A4935 is designed to drive external, low on-resistance,
power N-channel MOSFETs. It supplies the large transient cur-
rents necessary to quickly charge and discharge the external FET
gate capacitance in order to reduce dissipation in the external
FET during switching. The charge and discharge rate can be
controlled using an external resistor in series with the connection
to the gate of the FET.
Gate Drive Voltage Regulation The gate drives are powered
by an internal regulator which limits the supply to the drives
and therefore the maximum gate voltage. When the VBB supply
greater than about 16 V, the regulator is a simple linear regulator.
Below 16 V, the regulated supply is maintained by a charge pump
boost converter, which requires a pump capacitor connected
between the CP1 and CP2 pins. This capacitor must have a mini-
mum value of 220 nF, and is typically 470 nF.
The control inputs to the A4935 provide a very flexible solution
for many motor control applications. For full sinusoidal excita-
tion, each phase can be driven with an independent PWM signal.
For less complex drive solutions, the two PWM inputs, PWML
and PWMH, allow simple high-side, low-side, or fast-decay con-
trol with a single PWM signal.
A current sense amplifier allows motor current to be sensed by a
low-value sense resistor in the ground connection to the power
bridge.
The regulated voltage, nominally 13 V, is available on the VREG
pin. A sufficiently large storage capacitor must be connected to
this pin to provide the transient charging current to the low-side
drives and the bootstrap capacitors.
The A4935 includes a number of protection features against
undervoltage, overtemperature, and power bridge faults. Fault
states enable responses by the device or by the external controller,
depending on the fault condition and logic settings. Two fault flag
outputs, FF1 and FF2, are provided to signal detected faults to an
external controller. Diagnostics include an internal fault register,
which can be accessed by serial read out using the fault flag pins.
Top-off Charge Pump An additional top-off charge pump is
provided for each phase. The charge pumps allow the high-side
drives to maintain the gate voltage on the external FETs indefi-
nitely, ensuring so-called 100% PWM if required. This is a low
current trickle charge pump, and is operated only after a high-side
FET has been signaled to turn on. The floating high-side gate
drive requires a small bias current (<20 μA) to maintain the high-
level output. Without the top-off charge pump, this bias current
would be drawn from the bootstrap capacitor through the Cx pin.
The charge pump provides sufficient current to ensure that the
bootstrap voltage and thereby the gate-source voltage is main-
tained at the necessary level.
Power Supplies
Two power supply voltages are required, one for the logic
interface and one for the analog and output drive sections. Both
supplies should be decoupled with ceramic capacitors connected
close to the supply and ground pins.
The logic supply, connected to VDD, allows the flexibility of a
3.3 or 5 V logic interface. The main power supply should be con-
nected to VBB through a reverse voltage protection circuit. The
A4935 operates within specified parameters with a VBB supply
from 7 to 50 V and functions correctly with a supply down to
Note that the charge required for initial turn-on of the high-side
gate is always supplied by the bootstrap capacitor. If the bootstrap
capacitor becomes discharged, the top-off charge pump will not
provide sufficient current to allow the FET to turn on.
Allegro MicroSystems, Inc.
115 Northeast Cutoff
8
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A4935
Automotive 3-Phase MOSFET Driver
In some applications a safety resistor is added between the gate
and source of each FET in the bridge. When a high-side FET is
held in the on-state, the current through the associated high-side
LSS Pin This is the low-side return path for discharge of the
capacitance on the FET gates. It should be tied directly to the
common sources of the low-side external FETs through an inde-
gate-source resistor (RGSH) is provided by the high-side drive and pendent low impedance connection.
therefore appears as a static resistive load on the top-off charge
RDEAD Pin This pin controls internal generation of dead time
during FET switching.
pump. The minimum value of RGSH for which the top-off charge
pump can provide current is shown in the Electrical Characteris-
tics table.
• When a resistor greater than 3 kΩ is connected between
RDEAD and AGND, cross-conduction is prevented by the gate
drive circuits, which introduce a dead time, tDEAD , between
switching one FET off and the complementary FET on. The
dead time is derived from the resistor value connected between
the RDEAD and AGND pins.
• When RDEAD is connected directly to VDD, cross-conduction
is prevented by the gate drive circuits. In this case, tDEAD
defaults to a value of 6 μs typical.
• When RDEAD is connected directly to AGND, internal dead
time generation is disabled. This allows dead times of any
duration to be determined by the external controller through the
relative timing of the phase logic control inputs, xHI and xLO.
Note that when using an external controller to determine the
dead time, care must be taken to ensure that unintentional shorts
across the supply are avoided.
GLA, GLB, and GLC Pins These are the low-side gate drive
outputs for the external N-channel MOSFETs. External resistors
between the gate drive output and the gate connection to the FET
(as close as possible to the FET) can be used to control the slew
rate seen at the gate, thereby providing some control of the di/dt
and dv/dt of the SA, SB, and SC outputs. GLx going high turns on
the upper half of the drive, sourcing current to the gate of the low-
side FET in the external power bridge, turning it on. GLx going
low turns on the lower half of the drive, sinking current from the
external FET gate circuit to the LSS pin, turning off the FET.
SA, SB, and SC Pins Directly connected to the motor, these
terminals sense the voltages switched across the load. These
terminals are also connected to the negative side of the bootstrap
capacitors and are the negative supply connections for the floating
high-side drives. The discharge current from the high-side FET
gate capacitance flows through these connections, which should
have low impedance circuit connections to the FET bridge.
Logic Control Inputs
Low voltage-level digital inputs provide control for the gate
drives. The input logic is shown in table 1.
GHA, GHB, and GHC Pins These terminals are the high-side gate
drive outputs for the external N-channel FETs. External resistors
between the gate drive output and the gate connection to the FET
(as close as possible to the FET) can be used to control the slew
rate seen at the gate, thereby controlling the di/dt and dv/dt of the
SA, SB, and SC outputs. GHx going high turns on the upper half of
the drive, sourcing current to the gate of the high-side FET in the
external motor-driving bridge, turning it on. GHx going low turns
on the lower half of the drive, sinking current from the external
FET gate circuit to the corresponding Sx pin, turning off the FET.
These logic inputs can be driven from either 3.3 or 5 V logic. All
have a nominal hysteresis of 500 mV to improve noise perfor-
mance.
AHI, BHI, CHI, ALO, BLO, and CLO Pins These are the phase
control inputs. The xHI inputs control the high-side drives and
the xLO inputs control the low-side drives. Internal lockout
logic ensures that the high-side output drive and low-side output
drive cannot be active simultaneously, except when RDEAD is
connected to AGND and at the same time CCEN is set high, as
described in the CCEN pin section.
CA, CB, and CC Pins These are the high-side connections
for the bootstrap capacitors and are the positive supply for the
high-side gate drives. The bootstrap capacitors are charged to
approximately VREG when the associated output Sx terminal is
low. When the Sx output swings high, the charge on the bootstrap
capacitor causes the voltage at the corresponding Cx terminal to
rise with the output to provide the boosted gate voltage needed
for the high-side FETs.
PWMH and PWML Pins These inputs can be used to externally
control motor torque and speed.
• Setting PWMH low turns off active high-side drives and turns
on the complementary low-side drives. This provides high-
side–chopped slow-decay PWM with synchronous rectification.
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A4935
Automotive 3-Phase MOSFET Driver
• Setting PWML low turns off active low-side drives and turns on correct operating condition. The charge pump stabilizes in
the complementary high-side drives. This provides low-side–
chopped slow-decay PWM with synchronous rectification.
• PWMH and PWML may also be connected together and driven
with a single PWM signal. This provides fast-decay PWM with
synchronous rectification.
approximately 3 ms under nominal conditions.
RESET can be used also to clear latched fault flags without
entering sleep mode. To do so, hold RESET low for less then the
reset pulse time, tRES. This clears any latched fault that disables
the outputs, such as short circuit detection or bootstrap capacitor
undervoltage, and also clears the fault register.
COAST Pin An active-low input, which forces low all gate drive
outputs, GHx and GLx, and turns off all external FETs. This can
be used to protect the FETs and the motor in the case of a short
circuit. Using COAST does not clear any faults, so the fault flags
can still be decoded and the fault register data word can be read.
Because COAST turns off all the external FETs, it can also be used
to provide fast-decay PWM without synchronous rectification.
Note that the A4935 can be configured to start without any exter-
nal logic input. To do so, pull up the RESET pin to VBB by means
of an external resistor. The resistor value should be between
20 and 33 kΩ.
CCEN Pin This input provides an override to allow both the high-
side and the low-side external FETs of any phase to be active at
the same time, enabling cross-conduction. As an extra level of
safety, cross-conduction can only occur when RDEAD is tied to
AGND and CCEN is set high. If the CCEN input is inadvertently
disconnected from the controller, an internal pull-down resistor
ensures that the outputs revert to a safe condition.
RESET Pin This is an active-low input, and when active it
allows the A4935 to enter sleep mode. When RESET is held low
for longer than the reset pulse time, tRES, the regulator and all
internal circuitry are disabled and the A4935 enters sleep mode.
During sleep mode, current consumption from the VBB and VDD
supplies is reduced to a minimal level. In addition, latched faults
and the corresponding fault flags are cleared. When the A4935 is
coming out of sleep mode, the protection logic ensures that the
gate drive outputs are off until the charge pump reaches its
ESF Pin This is the Enable Stop on Fault input. It determines the
action that is taken when certain faults are detected. See the Fault
Protection and Diagnostics section for details.
Table 1. Phase Control Truth Table
Inputs
Outputs
Comment
RDEAD RESET
CCEN COAST PWMH PWML
xHI
0
0
1
1
0
1
0
1
0
1
1
1
x
xLO
0
GHx
L
GLx
L
Sx
Z
x
1
1
1
1
1
1
1
1
1
1
1
1
0
1
1
1
x
x
x
x
x
x
x
x
x
x
x
x
x
x
0
1
x
1
1
x
1
1
1
1
1
1
1
1
x
0
1
1
x
1
1
x
0
0
1
1
0
0
1
0
x
x
1
1
x
1
1
x
1
1
0
0
0
0
0
1
x
x
1
1
Phase disabled
x
1
L
H
L
LS
HS
Z
Phase sinking
x
0
H
L
Phase sourcing
>0.2 V
1
L
Phase disabled
x
1
L
H
H
L
LS
LS
HS
HS
HS
LS
HS
LS
Z
Sink; high-side PWM on other phases
Slow decay, SR; low-side recirculation
Slow decay, SR; high-side recirculation
Source; low-side PWM on other phases
Fast decay, SR
x
0
L
x
1
H
H
H
L
x
0
L
x
x
1
L
0
H
L
Fast decay, SR
AGND
AGND
x
1
H
L
Slow decay, SR; high-side recirculation
Slow decay, SR; low-side recirculation
Low power shutdown
1
H
Z
x
Z
x
x
x
L
L
Z
Coast
AGND
AGND
1
1
1
L
L
Z
Phase disabled
1
H
H
U
Cross-conduction
x = don’t care, HS = high-side FET active, LS = low-side FET active, Z = high impedance, both FETs off, U = undefined, SR = synchronous rectification
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A4935
Automotive 3-Phase MOSFET Driver
damage to components, the external controller can take low the
COAST input or all of the xHi and xLO phase control inputs.
Current Sense Amplifier
An uncommitted differential sense amplifier is provided to allow
the use of either low value sense resistors or a current shunt as the VDSTH Pin Faults on the external FETs are determined by
current sensing element. The input common mode range, CMR,
allows the below-ground current sensing typically required in
PWM motor control during switching transients.
measuring the drain-source voltage, VDS , of each active FET
and comparing it to the threshold voltage applied to the VDSTH
input, VDSTH. To avoid false fault detection during switching tran-
sients, the comparison is delayed by an internal blanking timer.
Input is on the CSN and CSP pins. The output of the sense ampli-
fier is available at CSOUT and can be used in a peak current
control system.
VDRAIN This is a low-current sense input from the top of the
external FET bridge. This input allows accurate measurement of
the voltage at the drain of the high-side FETs. It should be con-
nected directly to the common connection point for the drains of
the power bridge FETs at the positive supply connection point.
The input current to the VDRAIN pin is proportional to the volt-
age on the VDSTH pin and can be approximated by:
The gain of the sense amplifier is set using external input and
feedback resistors. The gain must be set to be greater than the
specified minimum to ensure stability. Typically the gain will be
set between 5 and 50 V/V. Output offset can also be added using
external resistors. Examples of setting the sense amplifier gain
and offset are provided in the Applications Information section.
IVDRAIN = 72 × VDSTH + 52 ,
where IVDRAIN is the current into the VDRAIN pin, in μA, and
VDSTH is the voltage on the VDSTH pin, in V.
Diagnostics
FF1 and FF2 Pins are open drain output fault flags, which
indicate fault conditions by their state, as shown in table 2. In
the event that two or more faults are detected simultaneously, the
state of the fault flags will be determined by a logical OR of the
flag states for all detected faults.
Several diagnostic features are integrated into the A4935 to
provide indication of fault conditions and, if required, take action
to prevent permanent damage. In addition to system wide faults
such as undervoltage and overtemperature, the A4935 integrates
individual drain-source monitors for each external FET, to pro-
vide short circuit detection. When a short or undervoltage fault
is being reported, detailed fault information can be read from the
fault outputs as a serial data word.
Table 2. Fault Definitions
Flag State
Disable Outputs*
Flag
Latched
Fault Description
ESF
Low
ESF
High
FF1 FF2
0
0
0
0
1
1
1
1
0
1
1
1
0
1
1
1
No fault
No
No
No
–
If ESF high
If ESF high
If ESF high
No
Diagnostic Management Pins
Short-to-ground
Yes
Yes
Yes
Yes
Yes
Yes
Yes
ESF Pin This pin (Enable Stop on Fault) determines the action
taken when a short circuit or overtemperature fault is detected. It
does not affect undervoltage fault condition actions.
Short-to-supply
No
Shorted load
No
Overtemperature
VDD undervoltage
VREG undervoltage
Bootstrap undervoltage
No
When ESF is set to logic high, any short circuit or overtem-
perature fault condition will pull all the gate drive outputs low
and coast the motor. For short faults, this disabled state will be
latched until RESET goes low or a serial read is completed.
Yes
Yes
Yes
No
No
Yes
*Yes indicates all gate drives low, and all FETs off.
When ESF is set to logic low, under most conditions the A4935
will not disrupt normal operation and therefore will not protect
the drive circuit or motor from damage. This is the case even
though the fault flags are set. This allows the actions taken to be
controlled externally by the system control circuits. To prevent
When ESF is high, short faults will always cause the fault flags
to be latched. When ESF is low, a short fault will only be flagged
when the fault is present, and the flag state will not be latched.
This provides additional diagnostics flexibility during FET
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A4935
Automotive 3-Phase MOSFET Driver
switching. Any short faults detected will always be latched in the
fault register.
Any time the A4935 enters the VREG undervoltage fault state,
bit 7 in the fault register will be set and will remain set until cleared
by a register reset (see the Fault Register Serial Access section).
When a short or undervoltage fault is present, a clock can be
applied to FF2 and detailed fault information can be read from
FF1 as a serial word. This can be used to determine on which of
the six external FETs a short is being detected, or which of the
monitored voltages have gone below their undervoltage threshold
level. Fault register serial access operation is detailed in the Fault
Register Serial Access section.
Bootstrap Capacitor Undervoltage The A4935 monitors the
voltage across the individual bootstrap capacitors to ensure they
have sufficient charge to supply the current pulse for the high-
side drive. Before a high-side drive can be turned on, the voltage
across the associated bootstrap capacitor must be higher than the
turn-on voltage limit. If this is not the case, then the A4935 will
start a bootstrap charge cycle by activating the complementary
low-side drive. Under normal circumstances, this will charge the
bootstrap capacitor above the turn-on voltage in a few microsec-
onds and the high-side drive will then be enabled.
Fault States
Overtemperature If the junction temperature exceeds the over-
temperature threshold, typically 165°C, the A4935 will enter the
overtemperature fault state and FF1 will go high. The overtem-
perature fault state, and FF1, will only be cleared when the tem-
The bootstrap voltage monitor remains active while the high-side
drive is active and if the voltage drops below the turn-off voltage
a charge cycle is initiated.
In either case, if there is a fault that prevents the bootstrap capaci-
tor charging, then the charge cycle will timeout, the fault flags
(indicating an undervoltage) will be set, and the outputs will be
disabled. In addition, the appropriate bit in the fault register will
be set. This allows the specific phase giving the bootstrap under-
voltage to be determined by reading the serial data word.
perature drops below the recovery level defined by TJF – TJFhys
.
Note that an overtemperature fault does not permit access to the
fault register because FF2 is pulled low.
If ESF is set high when an overtemperature is detected, the out-
puts will be disabled automatically while the fault state is present.
If ESF is set low, then no circuitry will be disabled. In this case
external control circuits must take action to limit the power dis-
sipation in some way so as to prevent overtemperature damage to
the chip and unpredictable device operation.
The bootstrap undervoltage fault state remains latched until
RESET is set low or a serial read of the fault register is com-
pleted.
VDD Undervoltage The logic supply voltage at VDD is moni-
tored to ensure correct logical operation. If an undervoltage
on VDD is detected, the outputs will be disabled. In addition,
because the state of other reported faults cannot be guaranteed,
all fault states, fault flags, and the fault register are reset and
replaced by the fault flags corresponding to a VDD undervoltage
fault state. For example, a VDD undervoltage will reset an exist-
ing short circuit fault condition and replace it with a VDD under-
voltage fault. When the VDD undervoltage condition is removed,
all flags will be cleared and the outputs enabled.
VREG Undervoltage VREG supplies the low-side gate driver
and the bootstrap charge current. It is critical to ensure that the
voltages are sufficiently high before enabling any of the outputs.
If the voltage at VREG, VREG, drops below the falling VREG
undervoltage lockout threshold, VREGUVoff, then the A4935 will
enter the VREG undervoltage fault state. In this fault state, both
FF1 and FF2 will be high, and the outputs will be disabled. The
VREG undervoltage fault state and the fault flags will be cleared
when VREG rises above the rising VREG undervoltage lockout
Short Fault Operation Shorts in the power bridge are determined
by monitoring the drain-souce voltage, VDS, of each active FET
and comparing it to the fault threshold voltage at the VDSTH pin.
Because power MOSFETs take a finite time to reach the rated on-
resistance, the measured drain-source voltages will show a fault
as the phase switches. To avoid such false short fault detections,
threshold, VREGUVon
.
The VREG undervoltage monitor circuit is active during pow-
er-up, and the A4935 remains in the VREG undervoltage fault
state until VREG is greater than the rising VREG undervoltage
lockout threshold, VREGUVon
.
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A4935
Automotive 3-Phase MOSFET Driver
the output from the comparators are ignored under two condi-
tions:
Short to Ground A short from any of the motor phase connec-
tions to ground is detected by monitoring the voltage across the
high-side FETs in each phase, using the appropriate Sx pin and
the voltage at VDRAIN. This drain-source voltage, VDS, is con-
tinuously compared to the voltage on the VDSTH pin. The result
of this comparison is ignored if the FET is not active. It is ignored
also for one fault blank time interval after the FET is turned on.
If, when the comparator is not being ignored, its output indicates
that VDS exceeds the voltage at the VDSTH pin, FF2 will be high.
If also ESF is set high, FF2 will be latched high and the outputs
will be disabled. Alternatively, if also ESF is set low, the outputs
will not be disabled and FF2 will only be high while the output
of the comparator indicates that VDS exceeds the voltage at the
VDSTH pin.
▪ while the external FET is off, and
▪ until the end of the period, referred to as the fault blank time,
after the FET is turned on.
When the FET is turned on, if the drain-source voltage exceeds
the voltage at the VDSTH pin at any time after the fault blank
time, then a short fault will be detected. If also ESF is set high,
then this fault will be latched and the FET disabled until reset.
In some applications, the fault blank time may be insufficient to
avoid detecting false faults during the switching time of the exter-
nal FET. In these cases, the external controller driving the A4935
may be used to determine the correct fault condition by setting
ESF low. This will prevent latching of the fault flag when a short
fault is detected, and will not disable the FET. With ESF low, FF2
will remain high only while the measured VDS exceeds the fault
threshold. The external controller can then monitor the fault flags
and use its own timers to validate a fault condition.
Shorted Load The short-to-ground and short-to-supply monitor
circuits will also detect a short across a motor phase winding. In
most cases, a shorted winding will be indicated by a high-side
and low-side fault being detected at the same time. In some cases
the relative impedances may permit only one of the shorts to be
detected.
Note that any fault thus detected by the A4935 will still be
latched in the fault register and remain there until cleared.
When ESF is set low, the external FETs are not disabled by the
A4935 when a short fault is detected. To avoid permanent dam-
age to the external FETs or to the motor under this condition, the
A4935 can either be fully disabled by the RESET input or all
FETs can be switched off by pulling low the COAST input or all
the phase control inputs.
Differentiating Short Fault Conditions
The distinction between short-to-ground, short-to-supply, and
shorted load can only be made by examining the contents of the
fault register. It is not possible to determine where a short fault
has occurred when using the state of the fault flags, FF1 and
FF2, alone. The flag combination FF1 low and FF2 high simply
indicates the presence of a probable short circuit.
Short to Supply A short from any of the motor phase connec-
tions to the battery or VBB connection is detected by monitoring
the voltage across the low-side FETs in each phase, using the
appropriate Sx pin and the LSS pin. This drain-source voltage,
VDS, is continuously compared to the voltage on the VDSTH pin.
The result of this comparison is ignored if the FET is not active.
It is ignored also for one fault blank time interval after the FET is
turned on. If, when the comparator is not being ignored, its output
indicates that VDS exceeds the voltage at the VDSTH pin, then
FF2 will be high. If also ESF is set high, then FF2 will be latched
high and the outputs will be disabled. Alternatively, if also ESF
As described above, shorts are detected by monitoring the drain-
source voltage, VDS , of each of the six FETs in the power bridge.
The different short fault conditions are defined as follows:
▪ A short-to-ground is likely to be present if the VDS of any active
high-side FET is greater than the threshold defined by VDSTH
(fault bits AH, BH, or CH = 1)
▪ A short-to-supply is likely to be present if the VDS of any active
low-side FET is greater than the threshold defined by VDSTH
(fault bits AL, BL, or CL = 1)
is set low, then the outputs will not be disabled and FF2 will only ▪ A shorted load or phase is likely to be present if, at the same
be high while the output of the comparator indicates that VDS
exceeds the voltage at the VDSTH pin.
time, the VDS of an active high-side and an active low-side are
both greater than the threshold defined by VDSTH.
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A4935
Automotive 3-Phase MOSFET Driver
then an under voltage has been detected. In either case, the
sequence for reading the contents of the fault register is:
Fault Register
All undervoltage and short faults are recorded in a 10-bit fault
register as defined in table 3. The fault register accumulates all
detected faults until cleared by setting RESET low, by cycling the
power off and on, or by reading the contents. The contents will
also be cleared if a VDD undervoltage fault is detected. During a
VDD undervoltage fault condition, both fault flags will be high
but all the bits in the fault register will be reset.
1. The external controller takes any necessary additional action to
protect the FETs.
2. The external controller pulls FF2 low.
3. The A4935 outputs on FF1 the fault register first bit, AH.
4. The external controller reads the fault bit, and then cycles FF2
high then low for the next bit, BH.
Table 3. Fault Register Bit Definitions
5. Steps 3 and 4 alternate until all of the 10 bits in the fault regis-
ter have been read out.
Bit
AH
BH
CH
AL
BL
CL
VR
VA
VB
VC
Position
Function
First
V
DS exceeded on A phase high-side FET
6. After the final bit, VC, is output, the external controller cycles
FF2 high then low.
2
3
VDS exceeded on B phase high-side FET
VDS exceeded on C phase high-side FET
VDS exceeded on A phase low-side FET
VDS exceeded on B phase low-side FET
4
7. The A4935 resets the fault register and pulls FF1 and FF2 low
to indicate no fault present.
5
6
VDS exceeded on C phase low-side FET
8. The external controller releases FF2.
7
Undervoltage detected on VREG
The basic sequence for the three possible states of FF1 and FF2
are shown in figure 1.
8
Bootstrap undervoltage detected on phase A
Bootstrap undervoltage detected on phase B
Bootstrap undervoltage detected on phase C
9
At the end of the serial transfer, on the last high-to-low transition
input to FF2, the fault register and the fault flags are reset. How-
ever, it is possible that one of the three unlatched fault conditions,
VREG undervoltage, VDD undervoltage, or overtemperature, is
still present. In this case the fault flags will immediately show the
fault status.
Last
The contents of the fault register can be read serially from the
FF1 pin by applying a clock signal to the FF2 pin during an
undervoltage or short fault state.
The fault flag pins, FF1 and FF2, are open drain outputs and pas-
sively pulled high when a fault is present. This makes it possible
to drive one or both of these fault pins from an external source
during a fault condition, when the A4935 is not pulling the pin
low. FF2 can thus be used as a clock input to shift out the fault
status register, bit-by-bit, on the other fault flag, FF1.
Resetting the VR Bit
At power-up, on coming out of reset, or after a VDD or VREG
undervoltage fault, it is possible that the fault flags and fault reg-
ister will have cleared but the VR bit in the fault register remains
set. This would happen if, when a power-on-reset occurred,
VREG had not yet risen beyond the undervoltage threshold level,
VREGUVon. Although VREG undervoltage fault state is not latched
and the fault flags are cleared when the fault is removed, the
VR bit in the fault register is latched and may remain set after
the power-on-reset. For this reason it is recommended, when the
serial fault register is to be used, to perform a reset by taking the
RESET pin low for less than the reset pulse time, tRES, after the
A4935 is powered-up and all fault flags are clear (FF1 and FF2
are low).
When FF2 is being pulled low by the A4935, either when no fault
is present or when an overtemperature fault is present, then no
serial access is possible. The fault status register can be accessed
only when FF2 goes high. This occurs when either a short or an
undervoltage fault has been detected.
Faut Register Serial Access
To access the fault register, FF1 and FF2 must be monitored by
an external controller. If FF2 goes high and FF1 remains low,
then a short has been detected. If FF1 and FF2 go high together,
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A4935
Automotive 3-Phase MOSFET Driver
FF2
FF1
Overtemperature Condition Present
(A) Overtemperature Fault
FF2 pulled up by resistor
VDS fault detected
External controller pulls FF2 low
FF2
FF1
AH
BH
CH
AL
BL
CL
VR
VA
VB
VC
Phase A short-to-supply
A4935 outputs fault register on FF1
1 bit on each falling edge of FF2
A4935 pulls FF1 and FF2 low
and resets fault register
(B) VDS Fault Register Read
FF1 and FF2 pulled up by resistor
UV fault detected
External controller pulls FF2 low
FF2
FF1
AH
BH
CH
AL
BL
CL
VR
VA
VB
VC
VREG undervoltage
A4935 pulls FF1 and FF2 low
and resets fault register
A 4935 outputs fault register on FF1
1 bit on each falling edge of FF2
(C) Undervoltage Fault Register Read
Figure 1. Fault flag sequence diagrams
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A4935
Automotive 3-Phase MOSFET Driver
Applications Information
Power Bridge Management Using PWM Control
The A4935 provides individual high-side and low-side controls
for each phase, plus two PWM control signals and a coast con-
trol. This allows a wide variety of 3-phase bridge control schemes
to be implemented.
A
B
C
A
B
C
For advanced schemes using sinusoidal current control, each FET
in the 3-phase bridge can be controlled individually without using
the PWM and COAST inputs. This requires a higher perfor-
mance external controller, with a PWM output for each phase. If
full external control over dead time is required, then six PWM
outputs will be required, one for each FET in the bridge. In this
type of system, the external controller has full control over the
current-decay method, load current recirculation paths, braking,
and coasting.
Drive
A
1
0
H
L
Recirculate
B
1
0
H
L
C
0
1
L
H
A
0
B
0
1
C
0
1
L
H
Phase
xHI
xLO
GHx
GLx
Phase
xHI
xLO
GHx
GLx
1
L
L
H
H
(A) Slow decay, synchronous rectification, high-side PWM using phase inputs
Figure 2A shows an example of the paths of the bridge and load
currents when each phase is controlled directly. The PWM inputs
PWMH and PWML are both tied high and COAST is tied low. In
this case the high-side FETs are switched off during the current
decay time (PWM off-time) and load current recirculates through
the low-side FETs. This is commonly referred to as high-side
chopping or high-side PWM. During the PWM off-time, the
complementary FETs are turned on, to short the body diode and
provide synchronous rectification.
A
B
C
A
B
C
Drive
Recirculate
A
1
0
B
1
0
H
L
C
0
1
L
H
A
1
0
H
L
B
1
0
H
L
C
1
0
H
L
Phase
xHI
xLO
GHx
GLx
Phase
xHI
xLO
GHx
GLx
Figure 2A shows one combination of phase states, and the same
principal applies to any of the possible phase states. This princi-
pal also applies when the low-side FETs are turned off during the
PWM off-time and the load current recirculates through the high
side FETs, as shown in figure 2B.
H
L
(B) Slow decay, synchronous rectification, low-side PWM using phase inputs
For less complex control schemes, for example where simple
block commutation is used, it is possible to control the bridge
with three logic signals (one for each phase) and a single PWM
signal. Figure 2C shows an example of 2-phase excitation with
high-side PWM, as commonly used in a block commutation
scheme. The PWMH input is used to modulate the phase currents
and PWML is held high. During the PWM off-time, the active
high-side FET is turned off and the complementary low-side FET
is turned on. Note that the phase control signals in this case do
not change and all PWM switching, for any phase combination, is
managed by a single PWM signal. For low-side PWM, PWMH is
held high and the PWM signal is applied to PWML.
A
B
C
A
B
C
Drive
Recirculate
A
1
0
H
L
B
0
0
L
L
1
1
C
0
1
L
H
A
1
0
L
H
B
0
0
L
L
0
1
C
0
1
L
H
Phase
xHI
xLO
GHx
GLx
Phase
xHI
xLO
GHx
GLx
PWMH
PWML
PWMH
PWML
By tying PWMH and PWML together and applying a PWM
signal to them, the load current can be controlled using fast decay
by effectively reversing the supply polarity. This feature operates
(C) Slow decay, synchronous rectification, high-side PWM using PWMx inputs
Figure 2. Power bridge current paths
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A4935
Automotive 3-Phase MOSFET Driver
with either 2-phase or 3-phase excitation. When using fast decay,
a PWM duty cycle of 50% results in zero effective motor torque.
A duty cycle of less than 50% causes negative effective torque,
and greater than 50% causes positive effective torque.
The choice of power FET and external series gate resistance
determine the selection of the dead-time resistor, RDEAD. The
dead time should be long enough to ensure that one FET in a
phase has stopped conducting before the complementary FET
starts conducting. This should also take into account the tolerance
and variation of the FET gate capacitance, the series gate resis-
tance, and the on-resistance of the A4935 internal drives.
To reduce power dissipation in the external FETs, the A4935 can
be instructed to turn on the appropriate low-side and high-side
drives during the load current recirculation PWM off-cycle. This
synchronous rectification allows current to flow through the
selected FETs, rather than the source-drain body diode, during
the decay time. The body diodes of the recirculating power FETs
conduct only during the dead time that occurs at each PWM
transition.
Internally-generated dead time will be present only if the on-com-
mand for one FET occurs within tDEAD after the off-command
for its complementary FET. In the case where one side of a phase
drive is permanently off, for example when using diode rectifica-
tion with slow decay, then the dead time will not occur. In this
case the gate drive will turn on within the specified propagation
delay after the corresponding phase input goes high. (Refer to the
Gate Drive Timing diagrams.)
Dead Time
To prevent cross-conduction (shoot through) in any phase of
the power FET bridge, it is necessary to have a dead time delay,
Fault Blank Time
t
DEAD , between a high- or low-side turn-off and the next comple-
To avoid false short fault detection, the output from the VDS
monitor for any FET is ignored when that FET is off and for a
period of time after it is turned on. This period of time is the fault
blank time. Its length is the dead time, tDEAD , plus an additional
period of time that compensates for the delay in the VDS moni-
tors. This additional delay is typically 300 to 600 ns. When tDEAD
mentary turn-on event. The potential for cross-conduction occurs
when any complementary high-side and low-side pair of FETs are
switched at the same time; for example, when using synchronous
rectification or after a bootstrap capacitor charging cycle. In the
A4935, the dead time for all three phases is set by a single dead-
time resistor (RDEAD) between the RDEAD and AGND pins.
For RDEAD values between 3 kΩ and 240 kΩ, at 25°C the nomi-
nal value of tDEAD in ns can be approximated by:
7200
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0.0
,
tDEAD(nom)
50 +
=
(1)
1.2 + (200 / RDEAD
)
where RDEAD is in kΩ. Greatest accuracy is obtained for values
of RDEAD between 6 and 60 kΩ, which are shown in figure 3.
The IDEAD current can be estimated by:
1.2
IDEAD
.
=
(2)
RDEAD
If the dead time is to be generated externally, for example by
the PWM output of a microcontroller, then connect the RDEAD
pin to the AGND pin to set the internally-generated dead time to
zero. Note that this configuration can allow cross-conduction, and
appropriate care should be taken, as described in the Cross-Con-
duction section. The maximum internally-generated dead time,
6 μs typical, can be set by connecting the RDEAD and VDD pins.
0
10
20
30
40
50
60
70
RDEAD (kΩ)
Figure 3. Dead time versus RDEAD
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A4935
Automotive 3-Phase MOSFET Driver
is set to zero by connecting RDEAD to AGND, the fault blank
The voltage drop across the bootstrap capacitor as the FET is
time typically defaults to 2 μs.
being turned on, ∆V, can be approximated by:
QGATE
.
∆V
≈
(4)
CBOOT
Cross-Conduction
So, for a factor of 20, ∆V would be approximately 5% of VBOOT
.
In some circumstances it is desirable to allow activation of both
high-side and low-side FETs in a single phase of the power
bridge. This can be used, with care, to reduce diode conduction
during synchronous rectification, which improves overall effi-
ciency and reduces electromagnetic emissions.
The maximum voltage across the bootstrap capacitor under
normal operating conditions is VREG(max). However, in some
circumstances the voltage may transiently reach 18 V, the clamp
voltage of the Zener diodes between the Cx and Sx pins. In most
applications, with a good ceramic capacitor the working voltage
can be limited to 16 V.
This mode of operation is also useful for independently control-
ling each phase of a VR motor or to turn on all high-side and low-
side FETs together, to cause a supply short circuit for blowing a
safety fuse.
Bootstrap Charging
Cross-conduction can occur only when the internally-generated
dead time is set to zero by connecting RDEAD to AGND and, at
the same time, CCEN is high. If zero internally-generated dead
time is required, but cross-conduction is to be prevented, then
CCEN can be tied to AGND. When the dead time is set to zero
it is still possible for some overlap to be present at the switching
instants due to the relative switching time of the FETs.
It is good practice to ensure the high-side bootstrap capacitor is
completely charged before a high-side PWM cycle is requested.
The time required to charge the capacitor, tCHARGE (μs), is
approximated by:
CBOOT × ∆V
tCHARGE
,
=
(5)
100
where CBOOT is the value of the bootstrap capacitor, in nF, and
∆V is the required voltage of the bootstrap capacitor.
Bootstrap Capacitor Selection
At power-up and when the drives have been disabled for a long
time, the bootstrap capacitor can be completely discharged. In
this case ∆V can be considered to be the full high-side drive
voltage, 12 V. Otherwise, ∆V is the amount of voltage dropped
during the charge transfer, which should be 400 mV or less.
The capacitor is charged whenever the Sx pin is pulled low and
current flows from VREG through the internal bootstrap diode
The bootstrap capacitors, CBOOTx, must be correctly selected to
ensure proper operation of the A4935. If the capacitances are too
high, time will be wasted charging the capacitor, resulting in a
limit on the maximum duty cycle and the PWM frequency. If the
capacitances are too low, there can be a large voltage drop at the
time the charge is transferred from CBOOTx to the FET gate, due
to charge sharing.
circuit to CBOOT
.
To keep this voltage drop small, the charge in the bootstrap
capacitor, QBOOT, should be much larger than the charge required
by the gate of the FET, QGATE. A factor of 20 is a reasonable
value, and the following formula can be used to calculate the
Bootstrap Charge Management
The A4935 provides automatic bootstrap capacitor charge
management. The bootstrap capacitor voltage for each phase
is continuously checked to ensure that it is above the bootstrap
under-voltage threshold, VBOOTUV. If the bootstrap capacitor volt-
age drops below this threshold, the A4935 will turn on the neces-
sary low-side FET, and continue charging until the bootstrap
capacitor exceeds the undervoltage threshold plus the hysteresis,
value for CBOOT
:
QBOOT CBOOT × VBOOT = QGATE × 20 ,
=
therefore:
QGATE × 20
,
CBOOT
=
(3)
VBOOT
V
BOOTUV + VBOOTUVhys. The minimum charge time is typically
where VBOOT is the voltage across the bootstrap capacitor.
7 μs, but may be longer for very large values of bootstrap capaci-
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A4935
Automotive 3-Phase MOSFET Driver
tor (>1000 nF). If the bootstrap capacitor voltage does not reach
the threshold within approximately 200 μs, an undervoltage fault
will be flagged.
attention should be paid to ensure the operating conditions allow
the A4935 to remain in a safe range of junction temperature.
The power consumed by the A4935, PD, can be estimated by:
PD PBIAS + PCPUMP + PSWITCHING
,
=
(6)
(7)
VREG Capacitor Selection
given:
The internal reference, VREG, supplies current for the low-side
gate drive circuits and the charging current for the bootstrap
capacitors. When a low-side FET is turned on, the gate-drive
circuit will provide the high transient current to the gate that is
necessary to turn on the FET quickly. This current, which can be
several hundred milliamperes, cannot be provided directly by the
limited output of the VREG regulator, and must be supplied by an
external capacitor connected to VREG.
PBIAS VBB × IBB
;
=
PCPUMP [( 2 VBB) – VREG
]
IAV
, for VBB < 15 V,
=
or
[VBB – VREG
]
IAV
, for VBB ≥ 15 V,
=
(8)
(9)
PSWITCHING QGATE × VREG × N × fPWM × Ratio ;
=
where:
The turn-on current for the high-side FET is similar in value to
that for the low-side FET, but is mainly supplied by the boot-
strap capacitor. However the bootstrap capacitor must then be
recharged from the VREG regulator output. Unfortunately the
bootstrap recharge can occur a very short time after the low-
side turn-on occurs. This requires that the value of the capacitor
connected between VREG and AGND should be high enough to
minimize the transient voltage drop on VREG for the combina-
tion of a low-side FET turn-on and a bootstrap capacitor recharge.
A value of 20 × CBOOT is a reasonable value. The maximum
working voltage will never exceed VREG , so the capacitor can be
rated as low as 15 V. This capacitor should be placed as close as
possible to the VREG pin.
IAV QGATE × N × fPWM
,
=
N is the number of FETs switching during a PWM cycle, and
10
.
Ratio
=
RGATE + 10
Braking
The A4935 can be used to perform dynamic braking by either
forcing all low-side FETs on and all high-side FETs off or, con-
versely, by forcing all low-side FETs off and all high-side FETs
on. This will effectively short-circuit the back EMF of the motor,
creating a breaking torque.
Supply Decoupling
During braking, the load current can be approximated by:
Because this is a switching circuit, there are current spikes from all
supplies at the switching points. As with all such circuits, the power
supply connections should be decoupled with a ceramic capacitor,
typically 100 nF, between the supply pin and ground. These capaci-
tors should be connected as close as possible to the device supply
pins VBB and VDD, and the power ground pin, PGND.
VBEMF
,
IBRAKE
=
(10)
RL
where VBEMF is the voltage generated by the motor and RL is the
resistance of the phase winding.
Care must be taken during braking to ensure that the maximum
ratings of the power FETs are not exceeded. Dynamic braking is
equivalent to slow decay with synchronous rectification and all
phases enabled.
Power Dissipation
In applications where a high ambient temperature is expected, the
on-chip power dissipation may become a critical factor. Careful
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A4935
Automotive 3-Phase MOSFET Driver
The A4935 can also be used to perform regenerative braking.
tor, RG, between CSP and AGND also have matched values. The
This is equivalent to reversing the motor commutation sequence
or using fast decay with synchronous rectification. Note that
phase commutation must continue for regenerative braking to
operate and the supply must be capable of managing the reverse
current, such as by connecting a resistive load or dumping the
current to a battery or capacitor.
gain of the sense amplfier, G, is determined by the relative values
of RF and RN, and is approximately:
RF
,
G
=
(11)
RN
If an output offset is required, for example to allow reverse cur-
rent measurement, then this can be generated by adding offset to
the CSP input through the RG resistor. Because the amplifier is
operating in a closed loop, any offset added to CSP will be mir-
rored at the output.
Current Sense Amplifier
The gain of the current sense amplifier is set using external input
and feedback resistors. Output offset can also be added using
external resistors. Care must be taken to ensure that the input
impedances seen from either end of the sense resistor match.
Figure 4B shows suitable resistor values for a gain, G, of 20 and
an output offset, VOS , of 250 mV.
For the basic configuration shown in figure 4A, the two input
resistors, RN and RP, have matched values. The feedback resistor,
RF, between CSN and CSOUT, and the ground reference resis-
Layout Recommendations
Careful consideration must be given to PCB layout when design-
ing high frequency, fast switching, high current circuits. The
following are recommendations regarding some of these consid-
erations:
RF
A4935
RP
• The A4935 analog ground, AGND, and power ground, PGND,
should be connected together at the package pins. This common
point, and the high-current return of the external FETs, should
return separately to the negative side of the motor supply
filtering capacitor. This will minimize the effect of switching
noise on the device logic and analog reference.
CSP
CSN
RS
CSOUT
RN
RG
G = RF / RN
RG = RF
• The exposed thermal pad and all NC pins of the package should
be connected to the common point of AGND and PGND.
• Minimize stray inductance by using short, wide copper traces at
the drain and source terminals of all power FETs. This includes
motor lead connections, the input power bus, and the common
source of the low-side power FETs. This will minimize voltages
induced by fast switching of large load currents.
R
P = RN
(A) Basic configuration
80 kΩ
A4935
4 kΩ
• Consider the use of small (100 nF) ceramic decoupling
capacitors across the sources and drains of the power FETs to
limit fast transient voltage spikes caused by the inductance of
the circuit trace.
• Keep the gate discharge return connections Sx and LSS as short
as possible. Any inductance on these traces will cause negative
transitions on the corresponding A4935 pins, which may exceed
the absolute maximum ratings. If this is likely, consider the use
of clamping diodes to limit the negative excursion on these pins
with respect to AGND.
CSP
CSN
RS
CSOUT
4 kΩ
76 kΩ
5 V
G = 20
OS= 250 mV
V
76 kΩ
4 kΩ
(B) Typical Configuration
Figure 4. Current sense amplifier configurations
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A4935
Automotive 3-Phase MOSFET Driver
• Sensitive connections such as RDEAD and VDSTH, which
GHx, GLx, Sx, and LSS should be as short as possible to reduce
have very little ground current, should be connected to the Quiet
ground (refer to figure 5), which is connected independently,
closest to the AGND pin. These sensitive components should
never be connected directly to the supply common or to a
common ground plane. They must be referenced directly to the
AGND pin.
• The supply decoupling for VBB, VREG, and VDD should be
connected to the Controller Supply ground, which is connected
independently, close to the PGND pin. The decoupling
capacitors should also be connected as close as practicable to
the relevant supply pin.
• If layout space is limited, then the Quiet ground and the
Controller Supply ground may be combined. In this case, ensure
that the ground return of the dead time resistor is close to the
AGND pin.
• Check the peak voltage excursion of the transients on the LSS
pin with reference to the AGND pin, using a close grounded (tip
and barrel) probe. If the voltage at LSS exceeds the absolute
maximum shown in this datasheet, add additional clamping and
capacitance between the LSS pin and the AGND pin as shown
in figure 5.
the circuit trace inductance.
• Provide an independent connection from LSS to the common
point of the power bridge. It is not recommended to connect
LSS directly to an xGND pin, as this may inject noise into
sensitive functions such as the timer for dead time. The LSS
connection should not be used for the CSP connection.
• The inputs to the sense amplifier, CSP and CSN, should have
independent circuit traces. For best results, they should be
matched in length and route.
• A low-cost diode can be placed in the connection to VBB to
provide reverse battery protection. In reverse battery conditions,
it is possible to use the body diodes of the power FETs to clamp
the reverse voltage to approximately 4 V. In this case, the
additional diode in the VBB connection will prevent damage
to the A4935 and the VDRAIN input will survive the
reverse voltage.
Note that the above are only recommendations. Each application
is different and may encounter different sensitivities. A driver
running a few amps will be less susceptible than one running with
150 A, and each design should be tested at the maximum current
to ensure any parasitic effects are eliminated.
• Gate charge drive paths and gate discharge return paths may
carry a large transient current pulse. Therefore, the traces from
Optional reverse
battery protection
VBB
VDRAIN
+ Supply
GHC
GHB
VREG
GHA
A4935
VDD
SA
SB
SC
Motor
GLA
GLB
GLC
VDSTH
RDEAD
LSS
AGND PGND
RS
Optional components
to limit LSS transients
Supply
Common
Quiet Ground
Power Ground
Controller Supply Ground
Figure 5. Supply routing suggestions
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A4935
Automotive 3-Phase MOSFET Driver
Input and Output Structures
CP1
CP2
VREG
VDRAIN
VBB
VDD
Cx
20 V
19 V
18 V
18 V
18 V
18 V
19 V
20 V
18 V
18 V
GHx
6 V
18 V
18 V
(B) Supply protection structures
Sx
VREG
18 V
GLx
LSS
10 Ω
3 kΩ
10 Ω
18 V
FF2
FF1
8.5 V
(A) Gate drive outputs
(C) Fault output
(D) Fault input/output
COAST
3 kΩ
3 kΩ
ESF
PWMx
xHI
3 kΩ
CCEN
RESET
xLO
50 kΩ
50 kΩ
8.5 V
8.5 V
6 V
6 V
(E) Logic inputs, no pulldown
(F) Logic input, with pulldown
(G) RESET input
VREG
1.2 V
100 Ω
22 V 22 V
4 kΩ
4 kΩ
1 kΩ
RDEAD
CSOUT
VDSTH
CSN
CSP
8.5 V
8.5 V
(J) VDS monitor threshold input
(I) RDEAD
(H) Current sense amplifier
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A4935
Automotive 3-Phase MOSFET Driver
Pin-out Diagram
NC
NC
SA
GLA
CB
GHB
SB
GLB
CC
GHC
SC
NC
NC
RESET
ESF
FF2
FF1
ALO
AHI
BHI
1
2
36
35
34
33
32
31
30
29
28
27
26
25
3
4
5
6
PAD
7
8
BLO
CLO
CHI
9
10
11
12
PWML
Terminal List
Pin
Pin Name
Pin Description
Pin
22
23
26
27
28
29
30
31
32
33
34
38
39
40
41
43
44
45
46
47
48
–
Pin Name
GLC
Pin Description
Low-side gate drive phase C
Analog ground
1, 24, 25, 35,
36, 37, 42
NC
No internal connection; connect to AGND
AGND
SC
Motor connection phase C
High-side gate drive phase C
Bootstrap capacitor phase C
Low-side gate drive phase B
Motor connection phase B
High-side gate drive phase B
Bootstrap capacitor phase B
Low-side gate drive phase A
Motor connection phase A
High-side gate drive phase A
Bootstrap capacitor phase A
Gate drive supply output
High-side drain voltage sense
Pump capacitor
2
3
RESET
ESF
Standby mode control
GHC
CC
Enable Stop on Fault input
Fault Flag 2 and serial clock input
Fault Flag 1 and serial data output
Control input phase A low-side
Control input phase A high-side
Control input phase B high-side
Control input phase B low-side
Control input phase C low-side
Control input phase C high-side
Low-side PWM input
4
FF2
GLB
5
FF1
SB
6
ALO
GHB
CB
7
AHI
8
BHI
GLA
9
BLO
SA
10
11
12
13
14
15
16
17
18
19
20
21
CLO
GHA
CA
CHI
PWML
CCEN
PWMH
COAST
CSOUT
VDD
VREG
VDRAIN
CP2
Cross-conduction enable
High-side PWM input
Coast input
CP1
Pump capacitor
Current sense output
PGND
VBB
Power ground
Logic supply
Main power supply
CSN
Current sense negative input
Current sense positive input
Fault threshold voltage
AGND
RDEAD
PAD
Analog ground
CSP
Dead time setting
VDSTH
LSS
Exposed thermal pad; connect to AGND
Low-side source
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A4935
Automotive 3-Phase MOSFET Driver
Package JP 48-Pin LQFP with Exposed Thermal Pad
0.30
9.00 ±0.20
7.00 ±0.20
0.50
1.70
4° ±4
+0.05
–0.06
0.15
C
B
9.00 ±0.20 7.00 ±0.20
5.00
8.60
5.00±0.04
0.60 ±0.15
(1.00)
48
A
1
48
2
1
2
0.25
5.00±0.04
SEATING PLANE
GAGE PLANE
5.00
8.60
C
PCB Layout Reference View
48X
C
SEATING
PLANE
0.08
C
For Reference Only
(reference JEDEC MS-026 BBCHD)
0.22 ±0.05
0.50
1.60 MAX
1.40 ±0.05
0.10 ±0.05
Dimensions in millimeters
Dimensions exclusive of mold flash, gate burrs, and dambar protrusions
Exact case and lead configuration at supplier discretion within limits shown
Terminal #1 mark area
A
B
Exposed thermal pad (bottom surface)
Reference land pattern layout (reference IPC7351
QFP50P900X900X160-48M); adjust as necessary to meet
application process requirements and PCB layout
tolerances; when mounting on a multilayer PCB, thermal
vias at the exposed thermal pad land can improve thermal
dissipation (reference EIA/JEDEC Standard JESD51-5)
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A4935
Automotive 3-Phase MOSFET Driver
Revision History
Revision
Revision Date
May 30, 2012
Description of Revision
Change in Absolute Maximum Ratings,
IBIAS and IOS
Rev. 4
Copyright ©2007-2012, Allegro MicroSystems, Inc.
Allegro MicroSystems, Inc. reserves the right to make, from time to time, such departures from the detail specifications as may be required to per-
mit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that the
information being relied upon is current.
Allegro’s products are not to be used in life support devices or systems, if a failure of an Allegro product can reasonably be expected to cause the
failure of that life support device or system, or to affect the safety or effectiveness of that device or system.
The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, Inc. assumes no responsibility for its use;
nor for any infringement of patents or other rights of third parties which may result from its use.
Allegro MicroSystems, Inc.
115 Northeast Cutoff
25
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
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