AP6502 [DIODES]
340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER; 340kHz 23V 2A同步DC / DC降压转换器
| 型号： | AP6502 |
| 厂家： | 
DIODES INCORPORATED |
| 描述： | 340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER |
| 文件： | 总12页 (文件大小：405K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
AP6502
340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER
Description
Pin Assignments
The AP6502 is a 340kHz switching frequency external
compensated synchronous DCDC buck converter. It has
integrated low RDSON high and low side MOSFETs.
( Top View )
The AP6502 enables continues load current of up to 2A with
efficiency as high as 95%.
BS
IN
1
2
8
7
SS
EN
The AP6502 features current mode control operation, which
enables fast transient response times and easy loop
stabilization.
3
4
6
5
COMP
FB
SW
The AP6502 simplifies board layout and reduces space
requirements with its high level of integration and minimal
need for external components, making it ideal for distributed
power architectures.
GND
SO-8EP
The AP6502 is available in a standard Green SO-8EP
package with exposed PAD for improved thermal
performance and is RoHS compliant.
Figure 1. Package Pin Out
Features
Applications
•
•
•
•
•
•
•
VIN 4.75V to 23V
•
•
•
•
•
•
•
•
•
Gaming Consoles
Flat Screen TV sets and Monitors
Set Top Boxes
2A continuous Output Current, 3A Peak
VOUT adjustable to 0.925 to 20V
Distributed power systems
Home Audio
340kHz switching frequency
Programmable Soft-Start
Enable pin
Consumer electronics
Network Systems
Protection
FPGA, DSP and ASIC Supplies
Green Electronics
o
o
OCP
Thermal Shutdown
•
Lead Free Finish/ RoHS Compliant (Note 1)
Note:
1. EU Directive 2002/95/EC (RoHS). All applicable RoHS exemptions applied. Please visit our website at
http://www.diodes.com/products/lead_free.html.
Typical Application Circuit
100
90
V
= 5V
IN
V
= 12V
IN
80
70
60
50
V
= 3.3V
OUT
L = 10µH
40
0
0.4
0.8
1.2
1.6
2
LOAD CURRENT (A)
Efficiency vs. Load Current
Figure 2. Typical Application Circuit
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© Diodes Incorporated
AP6502
Document Number: DS35423 Rev. 2 - 2
AP6502
340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER
Pin Descriptions
Pin #
Name
Description
High-Side Gate Drive Boost Input. BS supplies the drive for the high-side N-Channel MOSFET
switch. Connect a 0.01µF or greater capacitor from SW to BS to power the high side switch.
1
BS
Power Input. IN supplies the power to the IC, as well as the step-down converter switches.
Drive IN with a 4.75V to 23V power source. Bypass IN to GND with a suitably large capacitor
to eliminate noise on the input to the IC. See Input Capacitor.
2
IN
Power Switching Output. SW is the switching node that supplies power to the output. Connect
the output LC filter from SW to the output load. Note that a capacitor is required from SW to
BS to power the high-side switch.
3
4
5
SW
GND
FB
Ground (Connect the exposed pad to Pin 4).
Feedback Input. FB senses the output voltage and regulates it. Drive FB with a resistive
voltage divider connected to it from the output voltage. The feedback threshold is 0.925V. See
Setting the Output Voltage.
Compensation Node. COMP is used to compensate the regulation control loop. Connect a
series RC network from COMP to GND. In some cases, an additional capacitor from COMP to
GND is required. See Compensation Components.
6
7
8
COMP
EN
Enable Input. EN is a digital input that turns the regulator on or off. Drive EN high to turn on
the regulator; low to turn it off. Attach to IN with a 100kΩ pull up resistor for automatic startup.
Soft-Start Control Input. SS controls the soft-start period. Connect a capacitor from SS to GND
to set the soft-start period. A 0.1µF capacitor sets the soft-start period to 15ms. To disable the
soft-start feature, leave SS floating.
SS
Functional Block Diagram
Figure 3. Functional Block Diagram
+
-
IN
2
OVP
CURRENT
SENSE
AMPLIFIER
+
RAMP
1.1V
OSCILLATOR
E
-
+
-
5
FB
100/340 KHz
CLK
BS
1
Logic
0.3 V
100mꢀ
+
-
SW
3
SS
-
8
6
+
+
CURRENT
COMPARATOR
100mꢀ
ERROR
AMPLIFIER
6uA
0.923 V
GND
4
COMP
+
-
2.5V
EN OK
disable
LOCKOUT
COMPARATOR
IN < 4.10V
IN
EN
+
-
7
INTERNAL
REGULATORS
5V
SHUTDOWN
COMPARATOR
0.9V
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© Diodes Incorporated
AP6502
Document Number: DS35423 Rev. 2 - 2
AP6502
340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER
Absolute Maximum Ratings (Note 2)
Symbol
VIN
Parameter
Rating
-0.3 to 26
Unit
V
Supply Voltage
VSW
VBS
Switch Node Voltage
Bootstrap Voltage
Feedback Voltage
Enable/UVLO Voltage
Comp Voltage
-1.0 to VIN+0.3
VSW-0.3 to VSW + 6
–0.3V to +6
–0.3V to +6
–0.3V to +6
-65 to +150
+150
V
V
VFB
V
VEN
VCOMP
TST
V
V
Storage Temperature
Junction Temperature
Lead Temperature
°C
°C
°C
TJ
TL
+260
ESD Susceptibility (Note 3)
HBM
MM
Human Body Model
4
400
1
kV
V
Machine Model
CDM
Charged Device Model
kV
Notes:
only;
2. Stresses greater than the 'Absolute Maximum Ratings' specified above, may cause permanent damage to the device. These are stress ratings
functional operation of the device at these or any other conditions exceeding those indicated in this specification is not implied. Device
reliability may
be affected by exposure to absolute maximum rating conditions for extended periods of time.
3. Semiconductor devices are ESD sensitive and may be damaged by exposure to ESD events. Suitable ESD precautions should be taken when
handling and transporting these devices.
Thermal Resistance (Note 4)
Symbol
θJA
Parameter
Junction to Ambient
Junction to Case
Rating
74
Unit
°C/W
°C/W
θJC
16
Note: 4. Test condition for SO-8EP: Measured on approximately 1” square of 1 oz copper
Recommended Operating Conditions (Note 5)
Symbol
VIN
Parameter
Min
4.75
-40
Max
23
Unit
V
Supply Voltage
Operating Ambient Temperature Range
TA
+85
°C
Note: 5. The device function is not guaranteed outside of the recommended operating conditions.
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© Diodes Incorporated
AP6502
Document Number: DS35423 Rev. 2 - 2
AP6502
340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER
Electrical Characteristics (VIN = 12V, TA = +25°C, unless otherwise noted)
Symbol
Parameter
Shutdown Supply Current
Supply Current (Quiescent)
Test Conditions
VEN = 0V
Min
Typ.
0.3
Max
3.0
Unit
µA
IIN
IIN
VEN = 2.0V, VFB = 1.0V
0.6
1.5
mA
High-Side Switch On-Resistance
(Note 6)
RDS(ON)1
RDS(ON)2
100
100
mꢀ
mꢀ
Low-Side Switch On-Resistance
(Note 6)
ILimit
ILimit
HS Current Limit
LS Current Limit
Minimum duty cycle
From Drain to Source
4.4
0.9
A
A
VEN = 0V, VSW = 0V,
Vsw=12V
High-Side Switch Leakage Current
0
10
μA
AVEA
GEA
Error Amplifier Voltage Gain (Note 5)
Error Amplifier Transconductance
800
V/V
ΔIC = ±10μA
1000
uA/V
COMP to Current Sense
Transconductance
GCS
2.8
A/V
FSW
FFB
Oscillator Frequency
VFB = 0.75V
VFB = 0V
300
900
340
0.30
90
380
950
kHz
fSW
%
Fold-back Frequency
DMAX
TON
VFB
Maximum Duty Cycle
VFB = 800mV
Minimum On Time
200
925
1.1
ns
mV
V
Feedback Voltage
TA = -40°C to +85°C
Feedback Overvoltage Threshold
EN Rising Threshold
VEN_Rising
0.7
2.2
0.8
0.9
2.7
V
EN Lockout Threshold Voltage
EN Lockout Hysteresis
VIN Under Voltage Threshold Rising
2.5
V
220
4.05
mV
V
INUVVth
3.80
4.40
VIN Under Voltage Threshold
Hysteresis
INUVHYS
250
mV
Soft-Start Current
Soft-Start Period
Thermal Shutdown
VSS = 0V
6
μA
ms
°C
CSS = 0.1µF
15
TSD
150
Note: 6. Guaranteed by design
4 of 12
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© Diodes Incorporated
AP6502
Document Number: DS35423 Rev. 2 - 2
AP6502
340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER
Typical Performance Graphs (VIN = 12V, VOUT=3.3V ,TA = +25°C, unless otherwise noted)
0.6
0.074
6.2
6
0.064
0.054
0.044
0.034
0.58
0.56
5.8
5.6
0.54
0.52
5.4
5.2
0.024
0.014
0.5
5
4.8
0.004
0.48
0
5
10
INPUT VOLTAGE (V)
Shutdown Supply Current vs. Input Voltage
15
20
25
-60 -40 -20
0
20
TEMPERATURE (C)
Current Limit vs. Temperature
40
60
80 100
0
5
10
INPUT VOLTAGE (V)
Quiescent Supply Current vs. Input Voltage
15
20
25
3.33
0.92
0.918
0.916
0.914
0.912
375
370
365
3.329
3.328
3.327
3.326
3.325
3.324
0.91
360
355
350
0.908
3.323
3.322
0.906
0.904
3.321
3.32
0.902
0.9
-60 -40
-20
0
20
TEMPERATURE (°C)
Oscillator Frequency vs. Temperature
40
60
80 100
4.75
9.75
14.75
INPUT VOLTAGE (V)
Line Regulation
19.75
24.75
-60 -40
-20
0
20
TEMPERATURE (°C)
Feedback Voltage vs. Temperature
40
60
80 100
90
85
80
90
85
80
100
90
80
70
75
70
65
60
V
= 5V
75
70
65
IN
V = 12V
IN
V
= 5V
IN
V
= 12V
IN
60
60
50
55
50
45
55
50
45
V
= 1.2V
V
V
= 12V
OUT
L = 3.3µH
IN
V
= 1.8V
OUT
L = 3.3µH
= 5V
OUT
L = 10µH
40
40
40
0
0.4
0.8
1.2
1.6
2
0
0.4
0.8
1.2
1.6
2
0
0.4
0.8
LOAD CURRENT (A)
Efficiency vs. Load Current
1.2
1.6
2
LOAD CURRENT (A)
Efficiency vs. Load Current
LOAD CURRENT (A)
Efficiency vs. Load Current
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© Diodes Incorporated
AP6502
Document Number: DS35423 Rev. 2 - 2
AP6502
340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER
Typical Performance Characteristics
(VIN = 12V, VOUT=3.3V ,L=3.3µH, C1=22uF, C2=47uF, TA = +25°C, unless otherwise noted)
Steady State Test no load
Steady State Test 2A
Startup Through Enable_no load
Time-10ms/div
Time-2us/div
Time-2us/div
Shutdown Through Enable_no load
Startup Through Enable 2A
Shutdown Through Enable 2A
Time-10ms/div
Time-5ms/div
Time-2ms/div
Load Transient Test 1.0A to 2.0A
Short Circuit Test
Short Circuit Recovery
Time-20us/div
Time-100us/div
Time-20us/div
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AP6502
Document Number: DS35423 Rev. 2 - 2
AP6502
340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER
Applications Information
Theory of Operation
External Soft Start
The AP6502 is a 2A current mode control, synchronous
buck regulator with built in power MOSFETs. Current
mode control assures excellent line and load regulation
and a wide loop bandwidth for fast response to load
transients.
Soft start is traditionally implemented to prevent the
excess inrush current. This in turn prevents the
converter output voltage from overshooting when it
reaches regulation. The AP6502 has an internal current
source with a soft start capacitor to ramp the reference
voltage from 0V to 0.925V. The soft start current is 6uA.
The soft start sequence is reset when there is a Thermal
Shutdown, Under Voltage Lockout (UVLO) or when the
part is disabled using the EN pin.
Figure 3 depicts the functional block diagram of AP6502.
The operation of one switching cycle can be explained as
follows. At the beginning of each cycle, HS (high-side)
MOSFET is off. The EA output voltage is higher than the
current sense amplifier output, and the current
comparator’s output is low. The rising edge of the 340kHz
oscillator clock signal sets the RS Flip-Flop. Its output
turns on HS MOSFET. The current sense amplifier is reset
for every switching cycle.
External Soft Start can be calculated from the formula
below:
DV
I
= C *
SS
DT
When the HS MOSFET is on, inductor current starts to
increase. The Current Sense Amplifier senses and
amplifies the inductor current. Since the current mode
control is subject to sub-harmonic oscillations that peak at
half the switching frequency, Ramp slope compensation is
utilized. This will help to stabilize the power supply. This
Ramp compensation is summed to the Current Sense
Amplifier output and compared to the Error Amplifier
output by the PWM Comparator. When the sum of the
Current Sense Amplifier output and the Slope
Compensation signal exceeds the EA output voltage, the
RS Flip-Flop is reset and HS MOSFET is turned off.
Where;
Iss = Soft Start Current
C = External Capacitor
DV=change in feedback voltage from 0V to maximum
voltage
DT = Soft Start Time
Current Limit Protection
In order to reduce the total power dissipation and to
protect the application, AP6502 has cycle-by-cycle
current limiting implementation. The voltage drop across
the internal high-side MOSFET is sensed and compared
with the internally set current limit threshold. This voltage
drop is sensed at about 30ns after the HS turns on.
When the peak inductor current exceeds the set current
limit threshold, current limit protection is activated.
During this time the feedback voltage (VFB) drops down.
When the voltage at the FB pin reaches 0.3V, the internal
oscillator shifts the frequency from the normal operating
frequency of 340Khz to a fold-back frequency of 102Khz.
The current limit is reduced to 70% of nominal current
limit when the part is operating at 102Khz. This low Fold-
back frequency prevents runaway current.
For one whole cycle, if the sum of the Current Sense
Amplifier output and the Slope Compensation signal does
not exceed the EA output, then the falling edge of the
oscillator clock resets the Flip-Flop. The output of the Error
Amplifier increases when feedback voltage (VFB) is lower
than the reference voltage of 0.925V. This also increases
the inductor current as it is proportional to the EA voltage.
If in one cycle the current in the power MOSFET does not
reach the COMP set current value, the power MOSFET
will be forced to turn off. When the HS MOSFET turns off,
the synchronous LS MOSFET turns on until the next clock
cycle begins. There is a “dead time” between the HS turn
off and LS turn on that prevents the switches from
“shooting through” from the input supply to ground.
Under Voltage Lockout (UVLO)
Under Voltage Lockout is implemented to prevent the IC
from insufficient input voltages. The AP6502 has a
UVLO comparator that monitors the input voltage and the
internal bandgap reference. If the input voltage falls
below 4.0V, the AP6502 will latch an under voltage fault.
In this event the output will be pulled low and power has
to be re-cycled to reset the UVLO fault.
The voltage loop is compensated through an internal
transconductance amplifier and can be adjusted through
the external compensation components.
Enable
The enable (EN) input allows the user to control turning on
or off the regulator. To enable the regulator EN must be
pulled above the ‘EN Rising Threshold’ and to disable the
regulator EN must be pulled below ‘EN falling Threshold’
(EN rising threshold – En threshold Hysteresis).
Over Voltage Protection
When the AP6502 FB pin exceeds 20% of the nominal
regulation voltage of 0.925V, the over voltage comparator
is tripped and the COMP pin and the SS pin are
discharged to GND, forcing the high-side switch off.
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AP6502
Document Number: DS35423 Rev. 2 - 2
AP6502
340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER
Applications Information (cont.)
Thermal Shutdown
Compensation Components
The AP6502 has on-chip thermal protection that prevents
damage to the IC when the die temperature exceeds safe
margins. It implements a thermal sensing to monitor the
operating junction temperature of the IC. Once the die
temperature rises to approximately 150°C, the thermal
protection feature gets activated .The internal thermal
sense circuitry turns the IC off thus preventing the power
switch from damage.
The AP6502 has an external COMP pin through which
system stability and transient response can be controlled.
COMP pin is the output of the internal trans-conductance
error amplifier. A series capacitor-resistor combination
sets
a
pole-zero combination to control the
characteristics of the control system. The DC gain of the
voltage feedback loop is given by:
A hysteresis in the thermal sense circuit allows the device
to cool down to approximately 120°C before the IC is
enabled again through soft start. This thermal hysteresis
feature prevents undesirable oscillations of the thermal
protection circuit.
V
FB
A
= R
LOAD
×G
CS
× A
×
VEA
VDC
V
OUT
Where VFB is the feedback voltage (0.925V), RLOAD is the
load resistor value, GCS is the current sense trans-
conductance and AVEA is the error amplifier voltage gain.
The control loop transfer function incorporates two poles
one is due to the compensation capacitor (C3) and the
output resistor of error amplifier, and the other is due to
the output capacitor and the load resistor. These poles
are located at:
Setting the Output Voltage
The output voltage can be adjusted from 0.925V to 18V
using an external resistor divider. Table 1 shows a list of
resistor selection for common output voltages. Resistor
R1 is selected based on a design tradeoff between
efficiency and output voltage accuracy. For high values of
R1 there is less current consumption in the feedback
network. However the trade off is output voltage accuracy
due to the bias current in the error amplifier. R2 can be
determined by the following equation:
G
EA
f
=
P1
2π×C3× A
VEA
1
V
⎛
⎞
f
=
OUT
P2
R
1
= R ⋅ ⎜
− 1⎟
2
2π × C2×R
⎜
⎟
LOAD
0.925
⎝
⎠
Where GEA is the error amplifier trans-conductance.
One zero is present due to the compensation capacitor
(C3) and the compensation resistor (R3). This zero is
located at:
1
f
=
Z1
2π×C3×R3
Figure 4. Feedback Divider Network
The goal of compensation design is to shape the
converter transfer function to get a desired loop gain. The
system crossover frequency where the feedback loop
has the unity gain is crucial.
When output voltage is low, network as shown in Figure 4
is recommended.
A rule of thumb is to set the crossover frequency to below
one-tenth of the switching frequency. Use the following
procedure to optimize the compensation components:
Vout(V)
5
3.3
2.5
1.8
R1(KΩ)
45.3
26.1
16.9
9.53
3
R2(KΩ)
10
10
10
10
10
1. Choose the compensation resistor (R3) to set the
desired crossover frequency. Determine the R3 value by
the following equation:
1.2
Table 1—Resistor Selection for Common Output
Voltages
V
V
OUT
2π × C2× fc
2π × C2× 0.1× fs
OUT
R3 =
×
<
×
×G
G
× G
V
G
V
CS
EA
CS
FB
FB
EA
Where fC is the crossover frequency, which is typically
less than one tenth of the switching frequency.
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AP6502
Document Number: DS35423 Rev. 2 - 2
AP6502
340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER
Applications Information (cont.)
Input Capacitor
Compensation Components (cont.)
The input capacitor reduces the surge current drawn from
the input supply and the switching noise from the device.
The input capacitor has to sustain the ripple current
produced during the on time on the upper MOSFET. It
must hence have a low ESR to minimize the losses.
2. Choose the compensation capacitor (C3) to achieve the
desired phase margin set the compensation zero, fZ1, to
below one fourth of the crossover frequency to provide
sufficient phase margin. Determine the C3 value by the
following equation:
The RMS current rating of the input capacitor is a critical
parameter that must be higher than the RMS input
current. As a rule of thumb, select an input capacitor
which has RMs rating that is greater than half of the
maximum load current.
2
C3 >
π×R3× fc
Where R3 is the compensation resistor value.
Cin/C1
(µF)
Cout/C2
(µF)
Rc/R3
(kΩ)
Cc/C3
(nF)
L1
(µH)
VOUT
(V)
Due to large dI/dt through the input capacitors,
electrolytic or ceramics should be used. If a tantalum
must be used, it must be surge protected. Otherwise,
capacitor failure could occur. For most applications, a
4.7µF ceramic capacitor is sufficient.
1.2
1.8
2.5
3.3
5
22
22
22
22
22
22
47
47
47
47
47
47
3.24
6.8
6.8
6.8
6.8
6.8
6.8
6.8
6.8
6.8
6.8
6.8
3.3
3.3
10
10
10
15
Output Capacitor
12
The output capacitor keeps the output voltage ripple
small, ensures feedback loop stability and reduces the
overshoot of the output voltage. The output capacitor is a
basic component for the fast response of the power
supply. In fact, during load transient, for the first few
microseconds it supplies the current to the load. The
converter recognizes the load transient and sets the duty
cycle to maximum, but the current slope is limited by the
inductor value.
Table 2—Resistor
Component Selection
Inductor
Calculating the inductor value is a critical factor in
designing a buck converter. For most designs, the
following equation can be used to calculate the inductor
value;
V
⋅(V − V )
OUT
V
IN OUT
L =
⋅ ΔI ⋅ f
L SW
IN
Maximum capacitance required can be calculated from
the following equation:
ESR of the output capacitor dominates the output voltage
ripple. The amount of ripple can be calculated from the
equation below:
Where ΔIL
is the inductor ripple current.
And fSW is the buck converter switching frequency.
Choose the inductor ripple current to be 30% of the
maximum load current. The maximum inductor peak
current is calculated from:
Vout
= ΔI * ESR
inductor
capacitor
An output capacitor with ample capacitance and low ESR
is the best option. For most applications, a 22µF ceramic
capacitor will be sufficient.
ΔI
L
I
= I +
L(MAX) LOAD
2
ΔIinductor
2
L(Iout
+
)
2
Co =
Peak current determines the required saturation current
rating, which influences the size of the inductor. Saturating
the inductor decreases the converter efficiency while
increasing the temperatures of the inductor and the
internal MOSFETs. Hence choosing an inductor with
appropriate saturation current rating is important.
2
(Δ V + Vout )2 − Vout
Where ΔV is the maximum output voltage overshoot.
PC Board Layout
This is a high switching frequency converter. Hence
attention must be paid to the switching currents
interference in the layout. Switching current from one
power device to another can generate voltage transients
across the impedances of the interconnecting bond wires
and circuit traces. These interconnecting impedances
should be minimized by using wide, short printed circuit
traces.
A 1µH to 10µH inductor with a DC current rating of at least
25% percent higher than the maximum load current is
recommended for most applications.
For highest efficiency, the inductor’s DC resistance
should be less than 200mꢀ. Use a larger inductance
for improved efficiency under light load conditions.
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AP6502
Document Number: DS35423 Rev. 2 - 2
AP6502
340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER
Applications Information (cont.)
efficiency of the regulator. This solution is also applicable
for D > 65%. The bootstrap diode can be a low cost one
such as BAT54 or a schottky that has a low Vf.
External
feedback
resistor dividers
must be placed
close to the FB
34mm
Input capacitor C1
must be placed as
close as possible
to the IC and to L1.
52mm
Figure 7—External Bootstrap
Compensation Components
AP6502 is exposed at the bottom of the package and must
be soldered directly to a well designed thermal pad on the
PCB. This will help to increase the power dissipation.
Recommended Diodes:
Voltage/Current
Part Number
Vendor
Rating
30V, 1A
30V, 1A
External Bootstrap Diode
B130
SK13
Diodes Inc
Diodes Inc
It is recommended that an external bootstrap diode be
added when the input voltage is no greater than 5V or the
5V rail is available in the system. This helps to improve the
Ordering Information
13” Tape and Reel
Packaging
(Note 7)
Package
Code
Device
Quantity
Part Number Suffix
AP6502SP-13
SP
SO-8EP
2500/Tape & Reel
-13
Note: 7. Pad layout as shown on Diodes Inc. suggested pad layout document AP02001, which can be found on our website at
http://www.diodes.com/datasheets/ap02001.pdf.
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© Diodes Incorporated
AP6502
Document Number: DS35423 Rev. 2 - 2
AP6502
340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER
Marking Information
Package Outline Dimensions (All Dimensions in mm)
Detail "A"
7°~9°
Exposed pad
45°
1
1
7°~9°
0.15/0.25
3.3Ref.
Bottom View
0.3/0.5
1.27typ
Gauge Plane
Seating Plane
4.85/4.95
0.62/0.82
1
Detail "A"
8x-0.60
Exposed pad
6x-1.27
Land Pattem Recommendation
(Unit:mm)
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© Diodes Incorporated
AP6502
Document Number: DS35423 Rev. 2 - 2
AP6502
340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER
IMPORTANT NOTICE
DIODES INCORPORATED MAKES NO WARRANTY OF ANY KIND, EXPRESS OR IMPLIED, WITH REGARDS TO THIS
DOCUMENT, INCLUDING, BUT NOT LIMITED TO, THE IMPLIED WARRANTIES OF MERCHANTABILITY AND FITNESS FOR A
PARTICULAR PURPOSE (AND THEIR EQUIVALENTS UNDER THE LAWS OF ANY JURISDICTION).
Diodes Incorporated and its subsidiaries reserve the right to make modifications, enhancements, improvements, corrections or other
changes without further notice to this document and any product described herein. Diodes Incorporated does not assume any liability
arising out of the application or use of this document or any product described herein; neither does Diodes Incorporated convey any
license under its patent or trademark rights, nor the rights of others. Any Customer or user of this document or products described
herein in such applications shall assume all risks of such use and will agree to hold Diodes Incorporated and all the companies
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Diodes Incorporated does not warrant or accept any liability whatsoever in respect of any products purchased through unauthorized
sales channel.
Should Customers purchase or use Diodes Incorporated products for any unintended or unauthorized application, Customers shall
indemnify and hold Diodes Incorporated and its representatives harmless against all claims, damages, expenses, and attorney fees
arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized application.
Products described herein may be covered by one or more United States, international or foreign patents pending. Product names
and markings noted herein may also be covered by one or more United States, international or foreign trademarks.
LIFE SUPPORT
Diodes Incorporated products are specifically not authorized for use as critical components in life support devices or systems without
the express written approval of the Chief Executive Officer of Diodes Incorporated. As used herein:
A. Life support devices or systems are devices or systems which:
1. are intended to implant into the body, or
2. support or sustain life and whose failure to perform when properly used in accordance with instructions for use provided
in the labeling can be reasonably expected to result in significant injury to the user.
B. A critical component is any component in a life support device or system whose failure to perform can be reasonably expected
to cause the failure of the life support device or to affect its safety or effectiveness.
Customers represent that they have all necessary expertise in the safety and regulatory ramifications of their life support devices or
systems, and acknowledge and agree that they are solely responsible for all legal, regulatory and safety-related requirements
concerning their products and any use of Diodes Incorporated products in such safety-critical, life support devices or systems,
notwithstanding any devices- or systems-related information or support that may be provided by Diodes Incorporated. Further,
Customers must fully indemnify Diodes Incorporated and its representatives against any damages arising out of the use of Diodes
Incorporated products in such safety-critical, life support devices or systems.
Copyright © 2011, Diodes Incorporated
www.diodes.com
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© Diodes Incorporated
AP6502
Document Number: DS35423 Rev. 2 - 2
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