EL7571C [ELANTEC]
Programmable PWM Controller; 可编程PWM控制器
| 型号： | EL7571C |
| 厂家： | 
ELANTEC SEMICONDUCTOR |
| 描述： | Programmable PWM Controller |
| 文件： | 总23页 (文件大小：478K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
EL7571C
Programmable PWM Controller
Features
General Description
• Pentium® II Compatible
The EL7571C is a flexible, high efficiency, current mode, PWM step
down controller. It incorporates five bit DAC adjustable output voltage
control which conforms to the Intel Voltage Regulation Module (VRM)
Specification for Pentium® II and Pentium® Pro class processors. The
controller employs synchronous rectification to deliver efficiencies
greater than 90% over a wide range of supply voltages and load condi-
tions. The on-board oscillator frequency is externally adjustable, or may
be slaved to a system clock, allowing optimization of RFI performance in
critical applications. In single supply operation, the high side FET driver
supports boot-strapped operation. For maximum flexibility, system oper-
ation is possible from either a 5V rail, a single 12V rail, or dual supply
rails with the controller operating from 12V and the power FETs from
5V.
• 5 bit DAC Controlled Output Voltage
• Greater than 90% Efficiency
• 4.5V to 12.6V Input Range
• Dual NMOS Power FET Drivers
• Fixed frequency, Current Mode
Control
• Adjustable Oscillator with External
Sync. Capability
• Synchronous Switching
• Internal Soft-Start
• User Adjustable Slope
Compensation
• Pulse by Pulse Current Limiting
• 1% Typical Output Accuracy
• Power Good Signal
• Output Power Down
• Over Voltage Protection
Connection Diagram
R2
5W
D1
Applications
C6 0.1µF
ENABLE
1.4V
1
2
3
4
5
6
7
8
9
OTEN
CSLOPE
COSC
REF
VH1 20
HSD 19
LX 18
L2
C3 240pF
C3 240pF
4.5V
to
12.6V
• Pentium® II Voltage Regulation
Modules (VRMs)
• PC Motherboards
• DC/DC Converters
• GTL Bus Termination
• Secondary Regulation
1.5µH
C1
C8
Q1
1µF 1000µF
x3
V
OUT
C3
1.3V to
3.5V
VIN 17
VINP 16
LSD 15
GNDP 14
GND 13
CS 12
L1
R2
5W
0.1µF
5.1µH
C2
C7
PWRGD
VIDO
POWER
GOOD
1µF
1000µF
x6
Q2
D2
Ordering Information
VID1
Voltage
I.D.
(VID
Part No
Temp. Range
Package
Outline #
EL7571C
0°C to +70°C
20-Pin SO
MDP0027
VID2
(0:4))
VID3
10 VID4
FB 11
Q1, Q2: Siliconix, Si4410, x2
C1: Sanyo, 16MV 1000GX, 1000µF x3
C2: Sanyo, 6MV 1000GX, 1000µF x6
L1: Pulse Engineering, PE-53700, 5.1µH
L2: Micrometals, T30-26, 7T AWG #20, 1.5µH
R1: Dale, WSL-25-12, 15mW, x2
D1: BAV99
D2: IR, 32CTQ030
Note: All information contained in this data sheet has been carefully checked and is believed to be accurate as of the date of publication; however, this data sheet cannot be a “controlled document”. Current revisions, if any, to these
specifications are maintained at the factory and are available upon your request. We recommend checking the revision level before finalization of your design documentation.
© 2001 Elantec Semiconductor, Inc.
EL7571C
Programmable PWM Controller
Absolute Maximum Ratings (T = 25°C)
A
Supply Voltage:
Input Pin Voltage:
VHI
-0.5V to 14V
Operating Temperature Range:
Operating Junction Temperature:
Peak Output Current:
0°C to +70°C
125°C
-.03 below Ground, +0.3 above Supply
-0.5V to 27V
3A
Storage Temperature Range:
65°C to +150°C
Power Dissipation:
SO20 500mW
Important Note:
All parameters having Min/Max specifications are guaranteed. Typ values are for information purposes only. Unless otherwise noted, all tests are at the
specified temperature and are pulsed tests, therefore: TJ = TC = TA.
DC Electrical Characteristics
TA = 25°C, VIN = 5V, COSC = 330pF, CSLOPE = 390pF, RSENSE = 7.5mW unless otherwise specified.
Parameter
VIN
VUVLO HI
VUVLO LO
Description
Input Voltage Range
Condition
Min
4.5
Typ
Max
12.6
4.4
Unit
V
Input Under Voltage Lock out Upper Limit
Input Under Voltage Lock out Lower Limit
Positive going input voltage
Negative going input voltage
See VID table
3.6
4
V
3.15
1.3
3.5
3.85
3.5
V
VOUT RANGE Output Voltage Range
V
VOUT 1
Steady State Output Voltage Accuracy, VID = IL = 6.5A, VOUT = 2.8V
10111
2.74
2.82
1.81
2.90
V
VOUT 2
Steady State Output Voltage Accuracy, VID = IL = 6.5A, VOUT =1.8V
00101
1.74
1.9
V
VREF
Reference Voltage
1.396
125
-40
-18
8
1.41
154
-5
1.424
185
20
V
mV
mV
%
VILIM
Current Limit Voltage
VILIM = (VCS-VFB)
VIREV
VOUT PG
Current Reversal Threshold
VIREV = (VCS-VFB
VOUT = 2.05V
)
Output Voltage Power Good Lower Level
Output Voltage Power Good Upper Level
Over-Voltage Protection Threshold
Power Down Input Low Level
Power Down Input High Level
Voltage I.D. Input Low Level
Voltage I.D. Input High Level
Oscillator Voltage Swing
-14
12
-10
16
%
VOVP
+9
+13
+17
1.5
%
VOTEN LO
VOTEN HI
VID LO
VIN = -10uA
V
(VIN-1.5)
(VIN-1.5)
V
1.5
V
VID HI
V
VOSC
0.85
4.8
VP-P
V
VPWRGD LO
RDS ON
Power Good Output Low Level
HSD, LSD Switch On-Resistance
IOUT = 1mA
0.5
6
VIN, VINP = 12V, IOUT = 100mA, (VHI-
LX) = 12V
W
RFB
FB Input Impedance
9.5
115
1.2
kW
kW
mA
mA
A
RCS
CS Input Impedance
IVIN
Quiescent Supply Current
Supply Current in Output Disable Mode
VOTEN>(VIN-0.5)V
VOTEN<1.5V
2
1
IVIN DIS
0.76
2.5
ISOURCE/SINK Peak Driver Output Current
VIN,VINP = 12V, Measured at HSD, LSD,
(VHI-LX) = 12V
IRAMP
CSLOPE Ramp Current
High Side Switch Active
1.2>VOSC>0.35V
8.5
14
50
2
20
µA
µA
mA
IOSC CHARGE Oscillator Charge Current
IOSC
Oscillator Discharge Current
1.2>VOSC>0.35V
DISCHARGE
IREFMAX
IVID
VREF Output Current
25
7
µA
µA
µA
VID Input Pull up Current
OTEN Input Pull up Current
3
3
5
5
IOTEN
7
2
EL7571C
Programmable PWM Controller
AC Electrical Characteristics
TA = 25°C, VIN = 5V, COSC = 330pF, CSLOPE = 390pF unless otherwise specified.
Parameter
fOSC
Description
Nominal Oscillator Frequency
Clock Frequency
Conditions
Min
140
50
Typ
190
500
100
Max
240
Unit
kHz
kHz
ns
COSC = 330pF
VOTEN>1.5V
fCLK
1000
tOTEN
tSYNC
Shutdown Delay
Oscillator Sync. Pulse Width
Oscillator i/p (COSC) driven with HCMOS
gate
20
800
ns
TSTART
DMAX
Soft-start Period
VOUT = 3.5V
100/fCLK
97
us
%
Maximum Duty Cycle
Pin Descriptions
Pin
Name
Pin
Pin No.
Type [1]
Function
1
2
OTEN
I
I
Chip enable input, internal pull up (5mA typical). Active high.
CSLOPE
With a capacitor attached from CSLOPE to GND, generates the voltage ramp compensation for the PWM current mode con-
troller. Slope rate is determined by an internal 14uA pull up and the CSLOPE capacitor value. VCSLOPE is reset to ground at
the termination of the high side cycle.
3
COSC
I
Multi-function pin: with a timing capacitor attached, sets the internal oscillator rate fS (kHz) = 57/COSC (µF); when pulsed
low for a duration tSYNC synchronizes device to an external clock.
4
5
REF
PWRGD
VID0
VID1
VID2
VID3
VID4
FB
O
O
I
Band gap reference output. Decouple to GND with 0.1uF.
Power good, open drain output. Set low whenever the output voltage is not within ±13% of the programmed value.
Bit 0 of the output voltage select DAC. Internal pull up sets input high when not driven.
Bit 1 of the output voltage select DAC. Internal pull up sets input high when not driven.
Bit 2 of the output voltage select DAC. Internal pull up sets input high when not driven.
Bit 3 of the output voltage select DAC. Internal pull up sets input high when not driven.
Bit 4 of the output voltage select DAC. Internal pull up sets input high when not driven.
Voltage regulation feedback input. Tie to VOUT for normal operation.
6
7
I
8
I
9
I
10
11
12
I
I
CS
I
Current sense. Current feedback input of PWM controller and over current capacitor input. Current limit threshold set at
+154mV with respect to FB. Connect sense resistor between CS and FB for normal operation.
13
14
15
16
17
18
19
GND
GNDP
LSD
VINP
VIN
S
S
Ground
Power ground for low side FET driver. Tie to GND for normal operation.
Low side gate drive output.
O
S
Input supply voltage for low side FET driver. Tie to VIN for normal operation.
Input supply voltage for control unit.
S
LX
S
Negative supply input for high side FET driver.
HSD
O
High side gate drive output. Driver ground referenced to LX. Driver supply may be bootstrapped to enhance low controller
input voltage operation.
20
VH1
S
Positive supply input for high side FET driver.
1. Pin designators: I = Input, O = Output, S = Supply
3
EL7571C
Programmable PWM Controller
Typical Performance Curves
+12V Supply Sync Line Regulation
5V Supply Line Regulation
0.004
0.30
0.20
0.003
0.002
0.001
0
0.10
0.00
-0.10
-0.20
-0.30
-0.40
-0.001
-0.002
-0.003
13.5
13.0
12.5
12.0
11.5
(V)
11.0
10.5
10.0
5.50
5.25
5.00
(V)
4.75
4.50
V
V
IN
IN
+12V Supply Sync Load Regulation
VRM +5V Supply +12V Controller Sync w/o
Schottky Load Regulation
0.04
0.03
0.02
0.01
0
6.00
5.00
4.00
3.00
2.00
1.00
0
V
= 1.8V
OUT
V
= 2.1V
OUT
V
= 2.8V
OUT
V
OUT
= 2.8V
V
OUT
= 3.5V
V
= 1.3V
3
OUT
-0.01
-0.02
-1.00
-2.00
V
OUT
= 1.8V
0
1
5
I
7
9
11
13
0
1
3
5
7
9
11
13
I
(A)
OUT
(A)
OUT
+5V Supply Non-Sync Load Regulation
+12V Supply Sync Efficiency
5.00
4.00
3.00
2.00
1.00
0
1.0
0.9
0.8
0.7
0.6
0.5
V
= 1.3V
OUT
V
= 1.8V
OUT
V
= 3.5V
OUT
V
= 2.8V
OUT
V
OUT
= 2.8V
V
= 3.5V
OUT
V
= 1.8V
5
OUT
-1.00
-2.00
0
1
3
5
7
9
11
13
0
1
3
7
9
11
13
I
(A)
OUT
I (A)
OUT
4
EL7571C
Programmable PWM Controller
Typical Performance Curves
+5V Supply Sync with Schottky Load
+5V Supply +12V Controller Sync w/o Schottky
VRM Efficiency
2.5
1.0
0.9
0.8
0.7
0.6
0.5
V
OUT
= 3.5V
1.5
0.5
0
V
OUT
= 2.8V
V
V
= 3.5V
= 1.8V
OUT
OUT
V
= 1.8V
OUT
-0.5
-1.5
-2.5
V
V
= 2.8V
= 1.3V
OUT
OUT
V
OUT
= 1.3V
0
1
3
5
7
9
11
13
0.02
1.02
3.04
5.04
I
7.04
(A)
9.04 11.04 13.04
I (A)
OUT
OUT
+5V Supply Non-Sync VRM Efficiency
+5V Supply Sync with Schottky VRM Efficiency
1.0
0.9
0.8
0.7
0.6
0.5
1.0
0.9
0.8
0.7
0.6
0.5
V
V
= 3.5V
V
V
= 3.5V
= 2.8V
= 1.8V
= 1.3V
OUT
OUT
OUT
= 2.8V
= 1.8V
OUT
V
V
OUT
OUT
OUT
V
OUT
= 1.3V
V
0
1
3
5
7
9
11
13
0
1
3
5
7
9
11
13
I (A)
OUT
I (A)
OUT
12V Transient Response
5V Non-sync Transient Response
1
1
5
EL7571C
Programmable PWM Controller
Typical Performance Curves
5V Sync Transient Response
5V Input 12V Controller Transient Response
1
1
Efficiency vs Temperature
V
vs Temperature
REF
92.6
1.425
1.420
1.415
1.410
1.405
1.400
1.395
1.390
92.5
92.4
92.2
92.0
91.8
91.6
-45
-30
-15
0
15
30
45
60
-45
-30
-15
0
15
30
45
60
Temperature (°C)
Temperature (°C)
Frequency vs Temperature
280
270
260
250
240
230
220
210
200
-45
-30
-15
0
15
30
45
60
Temperature (°C)
6
EL7571C
Programmable PWM Controller
Applications Information
Circuit Description
General
sating ramp signals together. The relative gains of the
comparator input stages are weighed. The ratio of volt-
age feedback to current feedback to compensating ramp
defines the load regulation and open loop voltage gain
for the system, respectively. The compensating ramp is
required to maintain large system signal system stability
for PWM duty cycles greater than 50%. Compensation
ramp amplitude is user adjustable and is set with a single
external capacitor (CSLOPE). The ramp voltage is
ground referenced and is reset to ground whenever the
high side drive signal is low. In operation, the DAC out-
put voltage is compared to the regulator output, which
has been internally attenuated. The resulting error volt-
age is compared with the compensating ramp and
current feedback voltage. PWM duty cycle is adjusted
by the comparator output such that the combined com-
parator input sums to zero. A weighted comparator
scheme enhances system operation over traditional volt-
age error amplifier loops by providing cycle-by-cycle
adjustment of the PWM output voltage, eliminating the
need for error amplifier compensation. The dominant
pole in the loop is defined by the output capacitance and
equivalent load resistance, the effect of the output induc-
tor having been canceled due to the current feedback. An
output enable (OUTEN) input allows the regulator out-
put to be disabled by an external logic control signal.
The EL7571C is a fixed frequency, current mode, pulse
width modulated (PWM) controller with an integrated
high precision reference and a 5 bit Digital-to-Analog
Converter (DAC). The device incorporates all the active
circuitry required to implement a synchronous step
down (buck) converter which conforms to the Intel Pen-
tium® II VRM specification. Complementary switching
outputs are provided to drive dual NMOS power FET’s
in either synchronous or non-synchronous configura-
tions, enabling the user to realize a variety of high
efficiency and low cost converters.
Reference
A precision, temperature compensated band gap refer-
ence forms the basis of the EL7571C. The reference is
trimmed during manufacturing and provides 1% set
point accuracy for the overall regulator. AC rejection of
the reference is optimized using an external bypass
capacitor CREF
.
Main Loop
A current mode PWM control loop is implemented in
the EL7571C (see block diagram). This configuration
employs dual feedback loops which provide both output
voltage and current feedback to the controller. The
resulting system offers several advantages over traditi-
tional voltage control systems, including simpler loop
design, pulse by pulse current limiting, rapid response to
line variaion and good load step response. Current feed-
back is performed by sensing voltage across an external
shunt resistor. Selection of the shunt resistance value
sets the level of current feedback and thereby the load
regulation and current limit levels. Consequently, opera-
tion over a wide range of output currents is possible. The
reference output is fed to a 5 bit DAC with step weigh-
ing conforming to the Intel VRM Specification. Each
DAC input includes an internal current pull up which
directly interfaces to the VID output of a Pentium® II
class microprocessor. The heart of the controller is a tri-
ple-input direct summing differential comparator, which
sums voltage feedback, current feedback and compen-
Auxiliary Comparators
The current feedback signal is monitored by two addi-
tional comparators which set the operating limits for the
main inductor current. An over current comparator ter-
minates the PWM cycle independently of the main
summing comparator output whenever the voltage
across the sense resistor exceeds 154mV. For a 7.5mW
resistor this corresponds to a nominal 20A current limit.
Since output current is continuously monitored, cycle-
by-cycle current limiting results. A second comparator
senses inductor current reverse flow. The low side drive
signal is terminated when the sense resistor voltage is
less than -5mV, corresponding to a nominal reverse cur-
rent of -0.67A, for a 7.5mW sense resistor. Additionally,
under fault conditions, with the regulator output over-
7
EL7571C
Programmable PWM Controller
voltage, inductor current is prevented from ramping to a
high level in the reverse direction. This prevents the par-
asitic boost action of the local power supply when the
fault is removed and potential damage to circuitry con-
nected to the local supply.
voltage differs from it’s selected value by more than
±13%. PWRGD is an open drain output. A third watch-
dog function disables PWM output switching during
over-voltage fault conditions, displaying both external
FET drives, whenever the output voltage is greater than
13% of its selected value, thereby anticipating reverse
inductor current ramping and conforming to the VRM
over-voltage specification, which requires the regulator
output to be disabled during fault conditions. Switching
is enabled after the fault condition is removed.
Oscillator
A system clock is generated by an internal relaxation
oscillator. Operating frequency is simple to adjust using
a single external capacitor COSC. The ratio of charge to
discharge current in the oscillator is well defined and
sets the maximum duty cycle for the system at around
96%.
Output Drivers
Complementary control signals developed by the PWM
control loop are fed to dual NMOS power FET drivers
via a level shift circuit. Each driver is capable of deliver-
ing nominal peak output currents of 2A at 12V. To
prevent shoot-through in the external FET’s, each driver
is disabled until the gate voltage of the complementary
power FET has fallen to less than 1V. Supply connec-
tions for both drivers are independent, allowing the
controller to be configured with a boot-strapped high
side drive. Employing this technique a single supply
voltage may be used for both power FET’s and control-
ler. Alternatively, the application may be simplified
using dual supply rails with the power FET’s connected
to a secondary supply voltage below the controller’s,
typically 12V and 5V. For applications where efficiency
is less important than cost, applications can be further
simplified by replacing the low side power FET with a
Schottky diode, resulting in non-synchronous operation.
Soft-start
During start-up, potentially large currents can flow into
the regulator output capacitors due to the fast rate of
change of output voltage caused during start-up,
although peak inrush current will be limited by the over
current comparator. However an additionally internal
switch capacitor soft-start circuit controls the rate of
change of output voltage during start-up by overriding
the voltage feedback input of the main summing com-
parator, limiting the start-up ramp to around 1ms under
typical operating conditions. The soft-start ramp is reset
whenever the output enable (OUTEN) is reset or when-
ever the controller supply falls below 3.5V.
Watchdog
A system watchdog monitors the condition of the con-
troller supply and the integrity of the generated output
voltage. Modern logic level power FET’s rapidly
increase in resistivity (Rdson) as their gate drive is
reduced below 5V. To prevent thermal damage to the
power FET’s under load, with a reduced supply voltage,
the system watchdog monitors the controller supply
(VIN) and disables both PWM outputs (HSD, LSD)
when the supply voltage drops below 3.5V. When the
supply voltage is increased above 4V the watchdog ini-
tiates a soft-start ramp and enables PWM operation. The
difference between enable and disable thresholds intro-
duces hysteresis into the circuit operation, preventing
start-up oscillation. In addition, output voltage is also
monitored by the watchdog. As called out by the Intel
Pentium® II VRM specification, the watchdog power
good output (PWRGD) is set low whenever the output
Applications Information
The EL7571C is designed to meet the Intel 5 bit VRM
specification. Refer to the VID decode table for the con-
troller output voltage range.
The EL7571C may be used in a number converter topol-
ogies. The trade-off between efficiency, cost, circuit
complexity, line input noise, transient response and
availability of input supply voltages will determine
which converter topology is suitable for a given applica-
8
EL7571C
Programmable PWM Controller
tion. The following table lists some of the differences
between the various configurations:
Converter Topologies
Transient
Topology
5V only Non-synchronous
5V only Synchronous
Diagram
figure 1
figure 2
figure 3
figure 4
Efficiency
92%
Cost
low
Complexity
low
Input Noise
high
Response
good
95%
higher
lowest
high
higher
lowest
high
high
good
5V &12V Non-synchronous
5V & 12V Synchronous
12V only Synchronous
92%
high
good
95%
high
good
Connection
Diagram
92%
highest
highest
high
best
Circuit schematics and Bills of Material (BOMs) for the
various topologies are provided at the end of this data
sheet. If your application requirements differ from the
included samples, the following design guide lines
should be used to select the key component values.
Refer to the front page connection diagram for compo-
nent locations.
where:
IPEAK = peak ripple current
TON = top switch on time
VIN = input voltage
FSW = switching frequency
VOUT = output voltage
IMIN = minimum load
Output Inductor, L1
Two key converter requirements are used to determine
inductor value:
Since inductance value tends to decrease with current,
ripple current will generally be greater than 21MIN at
higher output current.
• IMIN- minimum output current; the current level at
which the converter enters the discontinuous mode of
operation (refer to Elantec application note #18 for a
detailed discussion of discontinuous mode)
Once the minimum output inductance is determined, an
off the shelf inductor with current rating greater than the
maximum DC output required can be selected. Pulse
Engineering and Coil Craft are two manufactures of
high current inductors. For converter designers who
want to design their own high current inductors, for
experimental purposes or to further reduce costs, we rec-
ommend the Micrometals Powered Iron Cores data
sheet and applications note as a good reference and start-
ing point.
• IMAX- maximum output current
Although many factors influence the choice of the
inductor value, including efficiency, transient response
and ripple current, one practical way of sizing the induc-
tor is to select a value which maintains continuous mode
operation, i.e. inductor current positive for all condi-
tions. This is desirable to optimize load regulation and
light load transient response. When the minimum induc-
tor ripple current just reaches zero and with the mean
ripple current set to IMIN, peak inductor ripple current is
twice IMAX, independent of duty cycle. The minimum
inductor value is given by:
Current Sense Resistor, R1
Inductor current is monitored indirectly via a low value
resistor R1. The voltage developed across the current
sense resistor is used to set the maximum operating cur-
rent, the current reversal threshold and the system load
regulation. To ensure reliable system operation it is
important to sense the actual voltage drop across the
resistor. Accordingly a four wire Kelvin connection
should be made to the controller current sense inputs.
(V – V
) ´ T
(V – V
) ´ V
OUT OUT
IN
OUT
ON
IN
L
= ---------------------------------------------------- = --------------------------------------------------------
1MIN
1
V
´ F ´ 2 ´ I
IN SW MIN
PEAK
9
EL7571C
Programmable PWM Controller
There are two criteria for selecting the resistor value and
type. Firstly, the minimum value is limited by the maxi-
mum output current. The EL7571C current limit
capacitor has a typical threshold of 154mV, 125mV
minimum. When the voltage across the sense resistor
exceeds this threshold, the conduction cycle of the top
switch terminates immediately, providing pulse by pulse
current limiting. A resistor value must be selected which
guarantees operation under maximum load. That is:
where:
PD = power dissipated in current sense resistor
PD must be less than the power rating of the current
sense resistor. High current applications may require
parallel sense resistors to dissipate sufficient power.
Current Sense Resistor Table below lists some popular
current sense resistors: the WLS-2512 series of Power
Metal Strip Resistors from Dale Electronics, OARS
series Iron Alloy resistor from IRC, and Copper Magna-
nin (CuNi) wire resistor from Mills Resistors. Mother
board copper trace is not recommended because of its
high temperature coefficient and low power dissipation.
The trade-off between the different types of resistors are
cost, space, packaging and performance. Although
Power Metal Strip Resistors are relatively expensive,
they are available in surface mount packaging with
tighter tolerances. Consequently, less board space is
used to achieve a more accurate current sense. Alterna-
tively, Magnanin copper wire has looser tolerance and
higher parasitic inductance. This results in a less current
sense but at a much lower cost. Metal track on the PCB
can also be used as current sense resistor. The trade-offs
are ±30% tolerance and ±4000 ppm temperature coeffi-
cient. Ultimately, the selection of the type of current
sense element must be made on an application by appli-
cation basis.
V
OCMIN
R
= ---------------------
1
1
MAX
where:
VOCMIN = minimum over current voltage threshold
IMAX = maximum output current
Secondly, since the load current passes directly through
the sense resistor, its power rating must be sufficient to
handle the power dissipated during maximum load (cur-
rent limit) conditions. Thus:
2
P
= 1
´ R
1
D
OUTMAX
Bill of Materials
Temperature
Manufacturer
Dale
Part No.
WSL 2512
Tolerance
±1%
Coefficient
Power Rating
1 W
Phone No.
402-563-6506
800-472-6467
916-422-5461
Fax No.
±75ppm
402-563-6418
800-472-3282
906-422-1409
IRC
OARS Series
±5%
±20ppm
1W - 5W
1.2W
Mills Resistor
PCB Trace Resistor
MRS1367-TBA
±10%
±30%
±20ppm
±4000ppm
50A/in (1oz Cu)
cause premature failure. Maximum input ripple current
occurs when the duty cycle is 50%, a current of Iout/2
RMS.
Input Capacitor, C1
In a buck converter, where the output current is greater
than 10A, significant demand is placed on the input
capacitor. Under steady state operation, the high side
FET conducts only when it is switched “on” and con-
ducts zero current when it is turned “off”. The result is a
current square wave drawn from the input supply. Most
of this input ripple current is supplied from the input
capacitor C1. The current flow through C1’s equivalent
series resistance (ESR) can heat up the capacitor and
Worst case power dissipation is:
2
I
OUT
æ
ö
P
=
· ESR
------------
D
IN
è
ø
2
where:
ERSIN = input capacitor ESR
10
EL7571C
Programmable PWM Controller
For safe and reliable operation, PD must be less than the
capacitor’s data sheet rating.
ESL = output capacitor ESL
DIOUT = output current step
Input Inductor, L2
di/dt = rate of change of output current
The input inductor (L2) isolates switching noise from
the input supply line by diverting buck converter input
ripple current into the input capacitor. Buck regulators
generate high levels of input ripple current because the
load is connected directly to the supply through the top
switch every cycle, chopping the input current between
the load current and zero, in proportion to the duty cycle.
The input inductor is critical in high current applications
where the ripple current is similarly high. An exclu-
sively large input inductor degrades the converter’s load
transient response by limiting the maximum rate of
change of current at the converter input. A 1.5µH input
inductor is sufficient in most applications.
Power MOSFET, Q1 and Q2
The EL7571C incorporates a boot-strap gate drive
scheme to allow the usage of N-channel MOSFETs. N-
channel MOSFETs are preferred because of their rela-
tive low cost and low on resistance. The largest amount
of the power loss occurs in the power MOSFETs, thus
low on resistance should be the primary characteristic
when selecting power MOSFETs. In the boot-strap gate
drive scheme, the gate drive voltage can only go as high
as the supply voltage, therefore in a 5V system, the
MOSFETs must be logic level type, Vgs<4.5V. In addi-
tion to on resistance and gate to source threshold, the
gate to source capacitance is also very important. In the
region when the output current is low (below5A),
switching loss is the dominant factor. Switching loss is
determined by:
Output Capacitor, C2
During steady state operation, output ripple current is
much less than the input ripple current since current flow
is continuous, either via the top switch or the bottom
switch. Consequently, output capacitor power dissipa-
tion is less of a concern than the input capacitor’s.
However, low ESR is still required for applications with
very low output ripple voltage or transient response
requirements. Output ripple voltage is given by:
2
P = C ´ V ´ F
where:
C is the gate to source capacitance of the MOSFET
V is the supply voltage
V
= I
´ ESR
RIP OUT
RIP
F is the switching frequency
Another undesirable reason for a large MOSFET gate to
source capacitance is that the on resistance of the MOS-
FET driver can not supply the peak current required to
turn the MOSFET on and off fast. This results in addi-
tional MOSFET conduction loss. As frequency
increases, this loss also increases which leads to more
power loss and lower efficiency.
where:
IRIP = output ripple current
ESROUT = output capacitor ESR
During a transient response, the output voltage spike is
determined by the ESR and the equivalent series induc-
tance (ESL) of the output capacitor in addition to the rate
of change and magnitude of the load current step. The
output voltage transient is given by:
Finally, the MOSFET must be able to conduct the maxi-
mum current and handle the power dissipation.
The EL7571C is designed to boot-strap to 12V for 12V
only input converters. In this application, logic level
MOSFETs are not required.
d
æ
ö
÷
ø
i
DV
=
ESR
´ DI
+ ESL ´
OUT
ç
----
OUT
OUT
d
t
è
Table below lists a few popular MOSFETs and their crit-
ical specifications.
where:
ESROUT = output capacitor ESR
11
EL7571C
Programmable PWM Controller
Manufacturer
MegaMos
MegaMos
Siliconix
Fuji
Model
Mi4410
Vgs
4.5V
4.5V
4.5V
4V
Ron (max)
20mW
22mW
20mW
37mW
8mW
Cgs
ID
VDS
30V
30V
30V
Package
SO-8
6.4nF
6.3nF
4.3nF
±10A
±15A
±10A
±17.5A
±98A
±75A
Mip30N03A
Si4410
TO-220
SO-8
2SK1388
TO-220
D2Pak
TO-220
IR
IRF3205S
MTB75N05HD
4
17nF (max)
7.1nF
55V
50V
Motorola
4
7mW
forward voltage drop. The product of forward voltage
drop and condition current is a primary source of power
dissipation in the convertor. The Schottky diode selected
is the International Rectifier 32CTQ030 which has 0.4V
of forward voltage drop at 15A.
Skottky Diode, D2
In the non-synchronous scheme a flyback diode is
required to provide a current path to the output when the
high side power MOSFET, Q1, is switched off. The crit-
ical criteria for selecting D2 is that it must have low
12
EL7571C
Programmable PWM Controller
Block Diagram
In
Regulation
ENABLE
0.1µF
REF
1.5µH
L
C
V
IN
OTEN
FB
C
PWRGD
V
INP
2
1
S
4.5V to
12.6V
3mF
+
-
Reference
+
-
V
HI
4V
UVLO HI
+
-
HSD
Current Reversal
+
-
UVLO LOW
0.1µF
5.1µH
+
-
LX
3.5V
+
-
L
V
OUT
1
VID
(0:4)
DAC
+
-
PWM
Control Logic
å
LSD
7.5mW
C
2
6mF
+
-
C
SLOPE
Soft
Start
Ramp Control
240pF
220pF
ENABLE
C
OSC
Oscillator
GND GNDP
13
EL7571C
Programmable PWM Controller
Voltage ID Code Output Voltage Settings
VID4
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
VID3
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
VID2
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
VID1
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
VID0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
VOUT
1.3
1.35
1.4
1.45
1.5
1.55
1.6
1.65
1.7
1.75
1.8
1.85
1.9
1.95
2.0
2.05
0, No CPU
2.1
2.2
2.3
2.4
2.5
2.6
2.7
2.8
2.9
3.0
3.1
3.2
3.3
3.4
3.5
Application Circuits
To assist the evaluation of EL7571C, several VRM
applications have been developed. These are described
in the converter topologies table earlier in the data sheet.
The demo board can be configured to operate with either
a 5V or 12V controller supply, using a 5V FET supply.
14
EL7571C
Programmable PWM Controller
5V Input, Boot-Strapped Non-Synchronous DC:DC Converter
5W
R2
D1
C6
ENABLE
1
2
3
4
5
6
7
8
9
OTEN
CSLOPE
COSC
REF
VH1 20
HSD
0.1µF
Q1
240pF
1µH
L2
19
C3
C4
C8
C1
5V
1µF 1000µF
x3
LX 18
220pF
VOUT
1.4V
V1H 17
L1
R1
7.5mW
C5
0.1µF
POWER
C7
0.1µF
C2
5.1µH
VINP
16
15
14
PWRGD
VIDO
1000µ
F
D2
GOOD
LSD
VID1
GNDP
VID2
GND 13
CS
Voltage
LD.
(VID(0:4))
VID3
12
FB 11
10 VID4
EL7571C 5V VRM Bill of Materials - 5V Non Sync Solution
Component
Manufacturer
Part Number
Value
Unit
3
C1
Sanyo
Sanyo
6MV1000GX
1000µF
1000µF
240pf
220pf
0.1µF
1µF
C2
6MV1000GX
6
C3
Chip Capacitors
Chip Capacitors
Chip Capacitors
Chip Capacitors
1
C4
1
C5, C6
C7, C8
D1
2
2
GI
Schotty diode SS12GICT-ND
EL7571CM
1
IC1
Elantec
1
L1
Pulse Engineering
Micrometals
DALE
PE-53700
5.1µH
1µH
1
L2
T30-26,7T AWG #20
WSL-2512
1
R1
15mW
5W
2
R2
Chip Resistor
IR32CTQ030
Si4410
1
D2
IR
1
Q1
Siliconix
2
15
EL7571C
Programmable PWM Controller
5V Input Boot-Strapped Synchronous DC:DC Converter
R2
D1
5W
C6
ENABLE
1
2
3
4
5
6
7
8
9
OTEN
CSLOPE
COSC
REF
VH1 20
HSD
0.1µF
Q1
240pF
1.5µH
L2
19
C3
C4
C8
C1
5V
1µF 1000µF
x3
LX 18
220pF
1.4V
V1H 17
VOUT
L1
R1
7.5mW
C5
0.1µF
POWER
C7
C2
5.1µH
D2
VINP
16
15
14
PWRGD
VIDO
1000µ
F
0.1µF
GOOD
LSD
Q2
VID1
GNDP
VID2
GND 13
CS
Voltage
LD.
(VID(0:4))
VID3
12
FB 11
10 VID4
EL7571C 5V VRM Bill of Materials - 5V Non Sync Solution
Component
Manufacturer
Part Number
Value
Unit
C1
Sanyo
Sanyo
6MV1000GX
1000µF
1000µF
240pf
220pf
0.1µF
1µF
3
C2
6MV1000GX
6
C3
Chip Capacitors
Chip Capacitors
Chip Capacitors
Chip Capacitors
1
C4
1
C5, C6
C7, C8
D1
2
2
GI
Schotty diode SS12GICT-ND
EL7571CM
1
IC1
Elantec
1
L1
Pulse Engineering
Micrometals
DALE
PE-53700
5.1µH
1µH
1
L2
T30-26,7T AWG #20
WSL-2512
1
R1
15mW
5W
2
R2
Chip Resistor
IR32CTQ030
Si4410
1
1
D2
IR
Q1, Q2
Siliconix
2 each
16
EL7571C
Programmable PWM Controller
5V Input, 12V Controller, Non-Sync Solution
12V
5W
ENABLE
1
2
3
4
5
6
7
8
9
OTEN
CSLOPE
COSC
REF
VH1 20
HSD
R2
220pF
1µH
19
Q1
C3
C4
C8
C1
L2
5V
1000µF
x3
1µF
LX 18
220pF
1.4V
V1H 17
L1
5.1µH
R1
7.5mW
VOUT
C2
C5
0.1µF
POWER
C7
VINP
16
15
14
PWRGD
VIDO
1000µ
F
0.1µF
GOOD
LSD
Q2
VID1
GNDP
VID2
GND 13
CS
Voltage
LD.
(VID(0:4))
VID3
12
FB 11
10 VID4
EL7571C 5V VRM Bill of Materials - 5V Non Sync Solution
Component
Manufacturer
Part Number
Value
1000µF
1000µF
240pF
220pF
0.1µF
1µF
Unit
3
C1
C2
Sanyo
Sanyo
6MV1000GX
6MV1000GX
Chip Capacitors
Chip Capacitors
Chip Capacitors
Chip Capacitors
EL7571CM
6
C3
1
C4
1
C5
1
C7, C8
IC1
L1
2
Elantec
1
Pulse Engineering
Micrometals
DALE
PE-53700
5.1µH
1µH
1
L2
T30-26,7T AWG #20
WSL-2512
1
R1
15mW
5W
2
R2
Chip Resistor
IR32CTQ030
Si4410
1
D2
IR
1
Q1
Siliconix
2
17
EL7571C
Programmable PWM Controller
5V Input, 12V Controller, Synchronous DC:DC Converter
12V
C6
0.1µF
ENABLE
1
2
3
4
5
6
7
8
9
OTEN
CSLOPE
COSC
REF
VH1 20
HSD
330pF
1.5µH
L2
19
Q1
C3
C4
C8
C1
5V
1µF 1000µF
x3
LX 18
330pF
1.4V
V1H 17
L1
R1
7.5mW
VOUT
C2
C5
0.1µF
POWER
C7
0.1µF
5.1µH
VINP
LSD
16
15
14
PWRGD
VIDO
1000µ
F
GOOD
D2
VID1
GNDP
VID2
GND 13
CS
Voltage
LD.
(VID(0:4))
VID3
12
FB 11
10 VID4
EL7571C 5V VRM Bill of Materials - 5V Input, 12V Controller Sync Solution
Component
Manufacturer
Part Number
Value
1000µF
1000µF
330pf
330pf
0.1µF
1µF
Unit
C1
Sanyo
Sanyo
6MV1000GX
6MV1000GX
Chip Capacitors
Chip Capacitors
Chip Capacitors
Chip Capacitors
EL7571CM
3
C2
6
C3
1
C4
1
C5, C6
C7, C8
IC1
2
2
Elantec
1
L1
Pulse Engineering
Micrometals
DALE
PE-53700
5.1µH
1µH
1
L2
T30-26,7T AWG #20
WSL-2512
1
R1
15mW
2
1
D2
IR
IR32CTQ030
Si4410
Q1, Q2
Siliconix
2 each
18
EL7571C
Programmable PWM Controller
PCB Layout Considerations
1. Place the power MOSFET’s as close to the con-
troller as possible. Failure to do so will cause
large amounts of ringing due to the parasitic
inductance of the copper trace. Additionally, the
parasitic capacitance of the trace will weaken the
effective gate drive. High frequency switching
noise may also couple to other control lines.
4. Connect the power and signal grounds at the out-
put capacitors. Output capacitor ground is the
quietest point in the converter and should be
used as the reference ground.
5. The power MOSFET’s output inductor and
Schottky diode should be grouped together to
contain high switching noise in the smallest area.
2. Always place the by-pass capacitors (0.1µF and
1µF) as close to the EL7571C as possible. Long
lead lengths will lessen the effectiveness.
6. Current sense traces running from pin 11 and pin
12 to the current sense resistor should run paral-
lel and close to each other and be Kelvin
connected (no high current flow). In high current
applications performance can be improved by
connecting low Pass filter (typical values 4.7W,
0.1µF) between the sense resistor and the IC
inputs.
3. Separate the power ground (input capacitor
ground and ground connections of the Schottky
diode and the power MOSFET’s) and signal
grounds (ground pins of the by-pass capacitors
and ground terminals of the EL7571C). This will
isolate the highly noisy switching ground from
the very sensitive signal ground.
19
EL7571C
Programmable PWM Controller
Layout Example
To demonstrate the points discussed above, below
shows two reference layouts - a synchronous 5V only
VRM layout and a synchronous 5V only PC board lay-
out. Both layouts can be modified to any application
circuit configuration shown on this data sheet. Gerber
files of the layouts are available from the factory.
Top Layer Silkscreen
Bottom Layer Silkscreen
20
EL7571C
Programmable PWM Controller
Top Layer Metal
Bottom Layer Metal
Top Layer Silkscreen
21
EL7571C
Programmable PWM Controller
Top Layer Metal
Bottom Layer Metal
22
EL7571C
Programmable PWM Controller
General Disclaimer
Specifications contained in this data sheet are in effect as of the publication date shown. Elantec, Inc. reserves the right to make changes in the cir-
cuitry or specifications contained herein at any time without notice. Elantec, Inc. assumes no responsibility for the use of any circuits described
herein and makes no representations that they are free from patent infringement.
WARNING - Life Support Policy
Elantec, Inc. products are not authorized for and should not be used
within Life Support Systems without the specific written consent of
Elantec, Inc. Life Support systems are equipment intended to sup-
port or sustain life and whose failure to perform when properly used
in accordance with instructions provided can be reasonably
Elantec Semiconductor, Inc.
675 Trade Zone Blvd.
Milpitas, CA 95035
Telephone: (408) 945-1323
(888) ELANTEC
expected to result in significant personal injury or death. Users con-
templating application of Elantec, Inc. Products in Life Support
Systems are requested to contact Elantec, Inc. factory headquarters
to establish suitable terms & conditions for these applications. Elan-
tec, Inc.’s warranty is limited to replacement of defective
components and does not cover injury to persons or property or
other consequential damages.
Fax:
(408) 945-9305
European Office: 44-118-977-6020
Japan Technical Center: 81-45-682-5820
Printed in U.S.A.
23
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