RM4391D [FAIRCHILD]
Inverting and Step-Down Switching Regulator; 反相和降压型开关稳压器
| 型号： | RM4391D |
| 厂家： | 
FAIRCHILD SEMICONDUCTOR |
| 描述： | Inverting and Step-Down Switching Regulator |
| 文件： | 总22页 (文件大小：154K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
PRODUCT SPECIFICATION
RC4391
Pin Descriptions
Pin Assignments
Pin
Number
LBR
LBD
1
2
3
4
8
7
6
V
V
FB
Pin Function Description
1
2
3
4
5
6
7
8
Low Battery Resistor (LBR)
Low Battery Detector (LBD)
REF
C
+V
S
X
Timing Capacitor (C )
X
GND
5
L
X
Ground
65-3471-02
External Inductor (L )
X
+Supply Voltage (+V )
S
+1.25V Reference Voltage (V
)
REF
Feedback Voltage (V
)
FB
Absolute Maximum Ratings
Parameter
Conditions
Min
Typ
Max
500
+30
70
Unit
mW
V
Internal Power Dissipation
Supply Voltage1
(Pin 6 to Pin 4 or Pin 6 to Pin 5)
Operating Temperature
RC4391
RV4391
RM4391
0
°C
-25
-55
-65
85
°C
125
150
125
175
375
468
833
300
300
°C
Storage Temperature
Junction Temperature
°C
PDIP, SOIC
CerDIP
Peak
°C
°C
Switch Current (I
)
mA
mW
mW
mW
°C
MAX
P
D
T <50˚C
A
PDIP
CerDIP
SOIC
Lead Soldering Temperature
Note:
(10 seconds)
1. The maximum allowable supply voltage (+V ) in inverting applications will be reduced by the value of the negative output
S
voltage, unless an external power transistor is used in place of Q1.
Thermal Characteristics
8-Lead Plastic DIP 8-Lead Ceramic DIP Small Outline SO-8
Therm. Res q
—
45°C/W
150°C/W
—
JC
Therm. Res. q
160°C/W
6.25 mW/°C
240°C/W
4.17 mW/°C
JA
For T >50˚C Derate at
8.33 mW/°C
A
2
RC4391
PRODUCT SPECIFICATION
Electrical Characteristics
(V = +6.0V, over the full operating temperature range unless otherwise noted)
S
Symbol
Parameters
Condition
Min
Typ
Max
30
Units
V
+V
Supply Voltage
Supply Current
Reference Voltage
Output Voltage
(Note 1)
4.0
S
I
V = +25V
S
300
1.25
-5.0
500
1.36
-4.5
-13.5
mA
V
SY
V
REF
V
OUT
1.13
-5.5
V
V
V
= -5.0V
= -15V
= -5.0V,
V
OUT nom
OUT nom
OUT nom
-16.5
-15.0
LI
1
Line Regulation
%V
OUT
C = 150pF
X
V = +5.8V to +15V
2.0
1.5
0.2
4.0
3.0
0.5
S
V
= -15V,
OUT nom
C = 150pF
X
V = +5.8V to +15
S
L0
Load Regulation
V
= -5.0V,
%V
OUT
1
OUT nom
C = 350pF, V = +4.5V,
X
S
P
V
= 0mW to 75mW
LOAD
OUT nom
= -15V,
C = 350pF, V = +4.5V,
X
S
P
= 0mW to 75mW
0.2
0.1
0.3
30
LOAD
I
Switch Leakage Current
Pin 5 = -20V
mA
CO
Note:
1. The maximum allowable supply voltage (+V ) in inverting applications will be reduced by the value of the negative output
S
voltage, unless an external power transistor is used.
3
PRODUCT SPECIFICATION
RC4391
Electrical Characteristics
(V = +6.0V, T = +25°C unless otherwise noted)
S
A
Symbol Parameters
Condition
Min
Typ
Max
Units
I
Supply Voltage
V = +4.0V,
S
170
250
mA
SY
No External Loads
V = +25V
S
300
500
No External Loads
V
Output Voltage
Line Regulation
V
V
V
= -5.0V
= -15V
= -5.0V
-5.35
-5.0
-4.65
V
OUT
OUT nom
OUT nom
OUT nom
-15.85
-15.0
-14.15
LI
%V
OUT
1
C = 150pF,
1.5
1.0
3.0
2.0
0.4
X
V = +5.8V to +15V
S
V
= -15V,
OUT nom
C = 150pF
X
V = +5.8V to +15V
S
L0
Load Regulation
V
= -5.0V,
%V
OUT
1
OUT nom
C = 350pF, V = +4.5V,
X
0.2
S
P
V
= 0mW to 75mW
LOAD
= -15V,
OUT nom
C = 350pF, V = +4.5V,
0.07
0.14
1.32
X
S
P
LOAD
= 0mW to 75mW
V
Reference Voltage
Switch Current
1.18
75
1.25
100
0.01
10
V
REF
I
I
I
I
I
I
Pin 5 = 5.5V
mA
mA
mA
mA
mA
mA
SW
Switch Leakage Current
Cap. Charging Current
LBD Leakage Current
LBD On Current
Pin 5 = -24V
5.0
14
CO
Pin 3 = 0V
6.0
CX
Pin 1 = 1.5V, Pin 2 = 6.0V
Pin 1 = 1.1V, Pin 2 = 0.4V
Pin 1 = 1.5V
0.01
600
0.7
5.0
LBDL
LBD0
LBRB
210
LBR Bias Current
4
RC4391
PRODUCT SPECIFICATION
Typical Performance Characteristics
6.5
6.0
5.5
5.0
4.5
4.0
3.5
3.0
8
7
6
5
4
3
2
1
0
0
5
10
15
+VS (V)
20
25
-55
0
25
70
125
TA (¡C)
Figure 1. Oscillator Frequency vs. Supply Voltage
Figure 2. Oscillator Frequency vs. Temperature
1.260
1.255
1.250
1.245
1.240
1.260
1.255
1.250
1.245
1.240
4
6
10
20
30
-55
0
25
70
125
TA (¡C)
+VS (V)
Figure 3. Reference Voltage vs. Temperature
Figure 4. Reference Voltage vs. Supply Voltage
4
3
2
1
0
600
500
400
300
200
100
20
10
0
8
1
2
3
5
7
-55
0
25
70
125
4
6
VCE (SAT) (V)
Figure 5. Collector Current vs. Q1 Saturation Voltage
TA (¡C)
Figure 6. Minimum Supply Voltage vs. Temperature
5
PRODUCT SPECIFICATION
RC4391
reference. Because C is initially discharged a positive
F
Principles of Operation
The basic switching inverter circuit is the building block on
which the complete inverting application is based.
voltage is applied to the comparator, and the output of the
comparator gates the squarewave oscillator. This gated
squarewave signal turns on, then off, the PNP output transis-
tor. This turning on and off of the output transistor performs
the same function as opening and closing the ideal switch in
the simplified diagram; i.e., it stores energy in the inductor
during the on time and releases it into the capacitor during
the off time.
A simplified diagram of the voltage inverter circuit with
ideal components and no feedback circuitry is shown in
Figure 7. When the switch S is closed, charging current from
the battery flows through the inductor L, which builds up a
magnetic field, increasing as the switch is held closed. When
the switch is opened, the magnetic field collapses, and the
energy stored in the magnetic field is converted into a current
which flows through the inductor in the same direction as the
changing current. Because there is no path for this current to
flow through the switch, the current must flow through the
diode to charge the capacitor C. The key to the inversion is
the ability of the inductor to become a source when the
charging current is removed.
The comparator will continue to allow the oscillator to turn
the switch transistor on and off until enough energy has been
stored in the output capacitor to make the comparator input
voltage decrease to less than 0V. The voltage applied to the
comparator is set by the output voltage, the reference volt-
age, and the ratio of R1 to R2.
The equation V = L (di/dt) gives the maximum possible
voltage across the inductor; in the actual application, feed-
back circuitry and the output capacitor will decrease the
output voltage to a regulated fixed value.
S
(–)
D
+V
S
V
L
OUT
C
A complete schematic for the standard inverting application
is shown in Figure 8. The ideal switch in the simplified
diagram is replaced by the PNP transistor switch between
(+)
pins 5 and 6. C functions as the output filter capacitor, and
F
65-1601
D1 and L replace D and L.
X
Figure 7. Simple Inverting Regulator
When power is first applied, the ground sensing comparator
(pin 8) compares the output voltage to the +1.25V voltage
C F*
33µF
V
-
OUT
To
+V
R3
260K
s
LBR
-15V
Output
Parts
List
-5.0V
Output
R1
C2
Q2
R4
590K
VFB
C1
W
R1 =
R2 =
900 k
300 k W
75 k W
150 pF
75 k W
R2
R6
100K
VREF
C
L
=
150 pF
x
+1.25V
REF/Bias
=
1.0 mH Dale TE3 Q4 TA
x
C1
0.1µF
VREF
LBD
CX
F
D1
LBD
Output
= Optional
R1
1N914
+VS
B
+V
s
OSC
-VOUT = (1.25V) (
)
R2
A
C
x
D
Q1
LX
GND
L
x
E
RC4391
65-1602
*Caution: Use current limiting protection circuit for high values of C (Figure 13)
F
Figure 8. Inverting Regulator – Standard Circuit
6
RC4391
PRODUCT SPECIFICATION
1.78V
CX
A
0.62V
OSC
(Internal)
B
I L
I LOAD
0 mA
C
D
+VS
(Internal)
+VS - 0.7V
Max
VBEQ1
VOUT
LX
VBAT
L X
I LX
0 mA
I MAX
E
I D
0 mA
F
+VS - VSW
Ground
VLX
G
-VOUT - VD
65-2472
Figure 9. Inverting Regulator Waveforms
This feedback system will vary the duration of the on time in
response to changes in load current or battery voltage (see
Figure 9). If the load current increases (waveform C), then
the transistor will remain on (waveform D) for a longer por-
tion of the oscillator cycle, (waveform B) to build up to a
higher peak value. The duty cycle of the switch transistor
varies in response to changes in load and line.
L
S
(+)
+VS
RL
C
VOUT
D
(-)
Figure 10. Simple Step-Down Regulator
65-2473
Step-Down Regulator
The step-down circuit function is similar to inversion; it uses
the same components (switch, inductor, diode, filter capaci-
tor), and charges and discharges the inductor by closing and
opening the switch. The great difference is that the inductor
is in series with the load; therefore, both the charging current
and the discharge current flow into the load. In the inverting
circuit only the discharge current flows into the load. Refer
to Figure 10.
rent will be greater than in an inverting circuit. The signifi-
cance of that is that for equal load currents the step-down
circuit will require less peak inductor current than an invert-
ing circuit. Therefore, the inductor will not require as large
of a core, and the switch transistor will not be stressed as
heavily for equal load currents.
Figure 11 depicts a complete schematic for a step-down cir-
cuit using the RC4391. Observe that the ground lead of the
4391 is not connected to circuit ground; instead, it is tied to
the output voltage. It is by this rearrangement that the feed-
back system, which senses voltages more negative than the
ground lead, can be used to regulate a non-negative output
voltage.
When the switch S is closed, current flows from the battery,
through the inductor, and through the load resistor to ground.
After the switch is opened, stored energy in the inductor
causes current to keep flowing through the load, the circuit
being completed by the catch diode D. Since current flows to
the load during charge and discharge, the average load cur-
7
PRODUCT SPECIFICATION
RC4391
When power is first applied, the output filter capacitor is dis-
charged so the ground lead potential starts at 0V. The refer-
ence voltage is forced to +1.25V above the ground lead and
pulls the feedback input (pin 8) more positive than the
ground lead. This positive voltage forces the control network
to begin pulsing the switch transistor. As the switching
action pumps up the output voltage, the ground lead rises
with the output until the voltage on the ground lead is equal
to the feedback voltage. At that point, the control network
reduces the time on time of the switch to maintain a constant
output.
inductor current (waveform E) to build up to a higher peak
value. The duty cycle of the switch transistor varies in
response to changes in load and line.
Design Equations
The inductor value and timing capacitor (C ) value must be
X
carefully tailored to the input voltage, input voltage range,
output voltage, and load current requirements of the applica-
tion. The key to the problem is to select the correct inductor
value for a given oscillator frequency, such that the inductor
current rises to a high enough peak value (I ) to meet the
MAX
average load current drain. The selection of this inductor
value must take into account the variation of oscillator
frequency from unit to unit and the drift of frequency over
temperature. Use ±30% as a maximum variation of oscillator
frequency.
This control network will vary the on time of the switch in
response to changes in load current or battery voltage (see
Figure 12). If the load current increases (waveform C), then
the transistor will remain on (waveformD) for a longer por-
tion of the oscillator cycle, (waveform B), thus allowing the
R1
LBR
VFB
C2
Q2
C1
R2
VREF
LBD
+1.25V
VREF
REF/Bias
+VS
B
CX
+V
s
OSC
A
C
x
D
Q1
LX
GND
+VOUT
RC4391
E
F
L
x
D1
1N914
CF
R1
+VOUT = (1.25V) (
)
R2
65-2475
Important Note: This circuit must have a minimum load ³ 1 mA always connected.
Figure 11. Step-Down Regulator – Standard Circuit
8
RC4391
PRODUCT SPECIFICATION
1.78V
CX
A
0.62V
B
(Internal)
OSC
I L
ILOAD
C
D
0 mA
+VS
VBEQ1
(Internal)
+V - 0.7V
VOUT - VBAT
L X
VBAT
LX
I MAX
ILX
E
F
0 mA
+VS - VSW
VOUT
VLX
VS ( -0.7V)
65-2474
Figure 12. Step-Down Regulator Waveforms
The oscillator creates a squarewave using a method similar
to the 555 timer IC, with a current steering flip-flop con-
trolled by two voltage sensing comparators. The oscillator
frequency is set by the timing capacitor (C ) according to
X
the following equation.
2. Find the maximum on time T (add 3mS for the turn off
ON
base recombination delay of Q1):
1
TON = --------- + 3mS
2FO
3. Calculate the peak inductor current IMAX (if this value
is greater than 375mA then an external power transistor
must be used in place of Q1):
–6
4.1x10
F
(Hz) = -----------------------
O
C (pF)
x
(VOUT + VD)2IL
The squarewave output of the oscillator is internal and
cannot be directly measured, but is equal in frequency to the
triangle waveform measurable at pin 3. The switch transistor
is normally on when the triangle waveform is ramping up
and off when ramping down. Capacitor selection depends on
the application; higher operating frequencies will reduce the
output voltage ripple and will allow the use of an inductor
with a physically smaller inductor core, but excessively
high frequencies will reduce load driving capability and
efficiency.
IMAX = ---------------------------------------------------------
(FO)(TON)(VS – VSW
)
Where:
V = Supply Voltage
= Saturation Voltage of Q1 (typically 0.5V)
SW
V = Diode Forward Voltage (typically 0.7V)
S
V
D
I = DC Load Current
L
4. Find an inductance value for LX:
VS – V
Inverting Design Procedure
æ
SWö
------------------------
LX(Henries) =
(TON)
è
ø
IMAX
1. Select an operating frequency and timing capacitor value
as shown above (frequencies from 10kHz to 50kHz are
typical).
The inductor chosen must exhibit this value of inductance
and have a current rating equal to I
.
MAX
9
PRODUCT SPECIFICATION
RC4391
Step-Down Design Procedure
Compensation
1. Select an operating frequency.
When large values (> 50 kW) are used for the voltage setting
resistors (R1 and R2 of Figure 8) stray capacitance at the
2. Determine the maximum on time T
design procedure.
as in the inverting
ON
V
FB
input can add lag to the feedback response, destabiliz-
ing the regulator, increasing low frequency ripple, and lower-
ing efficiency. This can often be avoided by minimizing the
stray capacitance at the V node. It can also be remedied by
FB
3. Calculate I
:
MAX
2IL
IMAX = ----------------------------------------------------------------------------
(VS – VOUT
(FO)(TON) --------------------------------- + 1
(VOUT – VD)
adding a lead compensation capacitor of 100 pF to 10 nF. In
inverting applications, the capacitor connects between
)
-V
OUT
and V ; for step-down circuits it connects between
FB
ground and V . Most applications do not require this
FB
capacitor.
4. Calculate L :
X
VS – V
æ
SWö
------------------------
Inductors
LX(Henries) =
(TON)
è
ø
IMAX
Efficiency and load regulation will improve if a quality high
Q inductor is used. A ferrite pot core is recommended; the
wind-yourself type with an air gap adjustable by washers or
spacers is very useful for bread-boarding prototypes. Care
must be taken to choose a core with enough permeability to
Alternate Design Procedure
The design equations above will not work for certain input/
output voltage ratios, and for these circuits another method
of defining component values must be used. If the slope of
the current discharge waveform is much less than the slope
of the current charging waveform, then the inductor current
will become continuous (never discharging completely), and
the equations will become extremely complex. So, if the
voltage applied across the inductor during the charge time is
greater than during the discharge time, use the design proce-
dure below. For example, a step-down circuit with 20V input
and 5V output will have approximately 15V across the
inductor when charging, and approximately 5V when dis-
charging. So in this example the inductor current will be con-
tinuous and the alternate procedure will be necessary. The
alternate procedure may also be used for discontinuous cir-
cuits.
handle the magnetic flux produced at I . If the core satu-
MAX
rates, then efficiency and output current capability are
severely degraded and excessive current will flow through
the switch transistor. A pot core inductor design section is
provided later in this datasheet.
An isolated AC current probe for an oscilloscope (example:
Tektronix P6042) is an excellent tool for saturation prob-
lems; with it the inductor current can be monitored for non-
linearity at the peaks (a sign of saturation).
Low Battery Detector
An open collector signal transistor Q2 with comparator C2
provides the designer with a method of signaling a display or
computer whenever the battery voltage falls below a pro-
grammed level (see Figure 13). This level is determined by
the +1.25V reference level and by the selection of two exter-
nal resistors according to the equation:
1. Select an operating frequency based on efficiency and
component size requirements (a value between 10kHz
and 50kHz is typical).
2. Build the circuit and apply the worst case conditions to
it, i.e., the lowest battery voltage and the highest load
current at the desired output voltage.
R4
R5
æ
ö
------ + 1
VTH = V
REFè
ø
When the battery drops below this threshold Q2 will turn on
and sink typically 600mA. The low battery detection circuit
can also be used for other less conventional applications such
as the voltage dependent oscillator circuit of Figure 18.
3. Adjust the inductor value down until the desired output
voltage is achieved, then decrease its value by 30% to
cover manufacturing tolerances.
4. Check the output voltage with an oscilloscope for ripple,
at high supply voltages, at voltages as high as are
expected. Also check for efficiency by monitoring supply
and output voltages and currents:
+V
s
R4
LBD
Q2
2
LBR
1
(VOUT)(IOUT
)
C2
æ
ö
ILBD
eff = -----------------------------------------
è
ø
(+VS)(ISY)x100
R5
5. If the efficiency is poor, go back to Step 1 and start over.
If the ripple is excessive, then increase the output filter
capacitor value or start over.
VREF
1.25V
65-1651A
Figure 13. Low Battery Detector
10
RC4391
PRODUCT SPECIFICATION
The following external power transistor circuits may demand
some adjustment to resistor values to satisfy various power
Device Shutdown
The entire device may be shut down to an extremely low cur-
rent non-operating condition by disconnecting the ground
(pin 4). This can be easily done by putting an NPN transistor
in series with ground pin and switching it with an external
signal. This switch will not affect the efficiency of operation,
but will add to and increase the reference voltage by an
amount equal to the saturation voltage of the transistor used.
A mechanical switch can also be used in series between
circuit ground and pin 4, without introducing any reference
offset.
levels and input/output voltages. C and L values must be
X
X
selected according to the design equations (pages 2-213 and
2-214).
Inverting Medium Power Application
Figure 8 is a schematic of an inverting medium power supply
(250mW to 1W) using an external PNP switch transistor.
Supply voltage is applied to the IC via R3: when the internal
switch transistor is turned on current through R4 is also
drawn through R3; creating a voltage drop from base to
emitter of the external switch transistor. This drop turns on
the external transistor.
Power Transistor Interfaces
The most important consideration in selecting an external
power transistor is the saturation voltage at I = I
.
C
MAX
Voltage pulses on the supply lead (pin 6) do not affect circuit
operation because the internal reference and bias circuitry
have good supply rejection capabilities. A power Schottky
diode is used for higher efficiency.
The lower the saturation voltage is, the better the efficiency
will be. Also, a higher beta transistor requires less base drive
and therefore less power will be.
Also, a higher beta transistor requires less base drive and
therefore less power will be consumed in driving it, improv-
ing efficiency losses in the interface. The part numbers given
in the following applications are recommended, but other
types may be more appropriate depending on voltage and
power levels.
Inverting High Power Application
For higher power applications (500mW to 5W), refer to
Figure 9. This circuit uses an extra external transistor to pro-
vide well controlled drive current in the correct phase to the
power switch transistor. The value of R3 sets the drive
current to the switch by making the interface transistor act as
a current source. R4 and R5 must be selected such that the
RC time constant of R4 and the base capacitance of Q2 do
not slow the response time (and affect duty cycle), but not so
low in value that excess power is consumed and efficiency
suffers. The resistor values chosen should be proportional to
the supply voltage (values shown are for +5V).
When troubleshooting external power transistor circuits,
ensure that clean, sharp-edged waveforms are driving the
interface and power transistors. Monitor these waveforms
with an oscilloscop—disconnect the inductor, and tie the
V
FB
input (pin 8) high through a 10K resistor. This will
cause the regulator to pulse at maximum duty cycle without
drawing excessive inductor currents. Check for expected on
time and off time, and look for slow rise times that might
cause the power transistor to enter its linear operating region.
Step-Down Power Applications
Figures 16 and 17 show medium and high power interfaces
modified to perform step-down functioning. The design
+5V
R3
1k½
C1
Q1
2N3635
R2
62 k½
0.1µF
-24V
Motorola
MBR030
7
6
5
x
5
VFB
CF
100µF
VREF
L
+V
s
R4
50½
4391
220µH
R1
1.2 M½
C
x
3
GND
4
150 pF
C
x
65-2476
Figure 14. Inverting Medium Power Application
11
PRODUCT SPECIFICATION
RC4391
+V
s
R6
1K
Q1
R5
2K
TIP116
C1
R2
Q2
2N33904
0.1µF
MBR140P
-VOUT
7
6
5
x
5
VFB
VREF
L
+V
s
R4
4.7K
CF
R3
750½
L
4391
x
R1
C
GND
4
x
3
C
x
65-2478
Figure 9. Inverting High Power Application
equations and suggestions for the circuits of Figures 14 and
15 also apply to these circuits. For a certain range of load
power, the RC4193 can be used for step-sown applications.
A load range from 400mW to 2W can be sustained with
fewer components (especially when stepping down greater
than 30V) than the comparable RC4391 circuit. Refer to
Fairchild Semiconductor's RC4191/4192/4193 data sheet for
a schematic of this medium power step-down application.
The threshold is programmed exactly as the normal low bat-
tery detector connection:
R4
R5
æ
ö
------ + 1
VTH = V
REFè
ø
When the battery voltage reaches this threshold the compara-
tor will turn on the open collector transistor at pin 2, effec-
tively pulling C in parallel with C . This added
Y
X
capacitance will reduce the oscillator frequency, according to
the following equation:
Voltage Dependent Oscillator
The RC4391's ability to supply load current at low battery
voltages depends on the inductor value and the oscillator fre-
quency. Low values of inductance or a low oscillator fre-
quency will cause a higher peak inductor current and
therefore increase the load current capability. A large induc-
tor current is not necessarily best , however, because the
large amount of energy delivered with each cycle will cause
a large voltage ripple at the output, especially at high input
voltages. This trade-off between load current capability and
output ripple can be improved with the circuit connection
shown in Figure 18. This circuit uses the low battery detector
to sense for a low battery voltage condition and will
decrease the oscillator frequency after a pre- programmed
threshold is reached.
–6
4.1x10
F
(Hz) = ------------------------------------------------
O
C
(pF) + C (pF)
Y
X
Current Limiting
The oscillator (C ) pin can be used to add short circuit pro-
X
tection and to protect against over current at start-up (when
using large values for the output filter capacitor —greater
than 100 mF). A transistor V is used as a current sensing
BE
comparator which resets the oscillator upon sensing an over
current condition, thus providing cycle-by-cycle current lim-
iting. Figure 19 shows how this is applied.
12
RC4391
PRODUCT SPECIFICATION
+V
s
C1
0.1 µF
R3
1K
R2
2N3635
8
7
6
5
VFB
VREF
L
x
+V
s
MBR030
R1
R4
30 - 100½
L
4391
x
C
GND
4
x
3
C
x
+VOUT
CF
Note: A minimum load ³1mA must be connected.
65-2479
Figure 16. Step-Down Medium Power Application
MBR140P
TIP116
*
500½
6
+1.3V
+VS
7
8
VREF
5
R2
5K
250µH
LX
4391
2N3904
VBAT
VFB
CX
GND
4
R3
1K
R4
20K
3
R1
5K
CX
470 pF
470 pF
(+)
VOUT
470µF
CF
(+5V at 1A as shown)
(-)
Note: A minimum load ³1mA must be connected.
65-2077
*Optional — Extends supply voltage range.
Figure 17. Step-Down High Power Application
+VS
1
W
CX
3
2
To
+VS
OSC
+1.25V
LBR
CX
CV
LBD
+VS
R4
R5
2N3906
or Equivalent
C2
Q2
3
1
C
4391
X
C
X
65-2053
65-2159
Figure 18. Voltage Dependent Oscillator
Figure 18. Current Limiting
13
PRODUCT SPECIFICATION
RC4391
Simplified Schematic Diagram
14
RC4391
PRODUCT SPECIFICATION
Troubleshooting Chart
Symptom
Possible Problems
Inductance value too low.
Draws excessive supply current on star-up.
Output frequency (F ) too low.
O
Combination of low resistance inductor and high
value filter capacitor — needs current limiting circuit
(Figure 13).
Output voltage is low.
Inductance value too high for F or core saturating.
O
Inductor "sings" with audible hum.
Not potted well or bolted loosely.
Normal operating condition.
Inductor is saturating:
L pin appears noisy — scope will not synchronize.
X
-IMAX
1. Core too small.
ILX
2. Core too hot.
3. Operating frequency too low.
Time
Inductor current shows nonlinear waveform.
Waveform has resistive component:
1. Wire size too small.
-IMAX
ILX
2. Power transistor lacks base drive.
3. Components not rated high enough.
4. Battery has high series resistance.
Time
Inductor current shows nonlinear waveform.
External transistor lacks base drive or beta is too
low.
-IMAX
ILX
Time
Inductor current is linear until high current is reached.
Poor efficiency.
Core saturating.
Diode or transistor:
1. Not fast enough.
2. Not rated for current level (high V SAT).
CE
High series resistance.
Operating frequency too high.
Motorboating (erratic current pulses).
Loop stability problem — needs feedback from
V
OUT
to V (pin 8), 100pF to 1000pF
FB
15
PRODUCT SPECIFICATION
RC4391
Pot Core Inductor Design
Electrical Circuit
Magentic Circuit
I
E = I * R
1
H =B •
R
E
U
North
South
Flux
65-3464-07
Figure 20. Electricity vs. Magnetism
the point where the permeability decreases, the magnetic
field has realigned all of the magnetic domains in the core
material. Once all of the domains have been aligned the core
will then carry no more flux than just air, it becomes as if
there were no core at all. This phenomenon is called satura-
tion. Because the inductance value, L, is dependent on the
amount of flux, core saturation will cause the value of L to
decrease dramatically, in turn causing excessive and possibly
destructive inductor current.
Electricity Versus Magnetism
Electrically the inductor must meet just one requirement, but
that requirement can be hard to satisfy. The inductor must
exhibit the correct value of inductance (L, in Henrys) as the
inductor current rises to its highest operating value (I
).
MAX
This requirement can be met most simply by choosing a very
large core and winding it until it reaches the correct induc-
tance value, but that brute force technique wastes size,
weight and money. A more efficient design technique must
be used.
6000
Question: What happens if too small a core is used?
+25¡C
5000
+85¡C
First, one must understand how the inductor's magnetic field
works. The magnetic circuit in the inductor is very similar to
a simple resistive electrical circuit. There is a magnetizing
force (H, in oersteds), a flow of magnetism, or flux density
(B, in Gauss), and a resistance to the flux, called permeability
(U, in Gauss per oersted). H is equivalent to voltage in the
electrical model, flux density is like current flow, and perme-
ability is like resistance (except for two important differences
discussed to the right).
4000
+125¡C
3000
2000
1000
0
Stackpole Ceramag 24B
Hysteresis Loop vs. Temperature
-0.5 0 0.5 1
2 2.5 3
H Oersteds
5
7
9
Figure 21. Typical Manufacturer’s Curve Showing
Saturation Effects
First Difference: Permeability instead of being analogous to
resistance, is actually more like conductance (1/R). As per-
meability increases, flux increases.
Pot Cores for RC4391
Pot core inductors are best suited for the RC4391 switching
regulator for several reasons:
Second Difference: Resistance is a linear function. As volt-
age increases, current increases proportionally, and the resis-
tance value stays the same. In a magnetic circuit the value of
permeability varies as the applied magnetic force varies. This
nonlinear characteristic is usually shown in graph form in
ferrite core manufacturer's data sheet.
1. They are available in a wide range of sizes. RC4391
applications are usually low power with relatively low
peak currents (less than 500mA). A small inexpensive pot
core can be chosen to meet the circuit requirements.
As the applied magnetizing force increases, at some point the
permeability will start decreasing, and therefore the amount
of magnetic flux will not increase any further, even as the
magnetizing force increases. The physical reality is that, at
2. Pot cores are easily mounted. They can be bolted
directly to the PC card adjacent to the regulator IC.
16
RC4391
PRODUCT SPECIFICATION
3. Pot cores can be easily air-gapped. The length of the
gap is simply adjusted using different washer
thicknesses. cores are also available with predetermined
air gaps.
Use of the Design Aid Graph
1. From the application requirement, determine the
inductor value (L) and the required peak current (I
).
MAX
2. Observe the curves of the design aid graph and determine
the smallest core that meets both the L and I
requirements.
4. Electromagnetic interference (EMI) is kept to a
minimum. the completely enclosed design of a pot core
reduces stray electromagnetic radiation—an important
consideration if the regulator circuit is built on a PC card
with other circuitry.
3. Note the approximate air gap at IMAX for the selected
core, and order the core with the gap. (If the gapping is
done by the user, remember that a washer lspacer results
in an air gap of twice the washer thickness, because two
gaps will be created, one at the center post and one at the
rim, like taking two bites from a doughnut.)
Not quite. Core size is dependent on the amount of energy
stored, not on load power. Raising the operating frequency
allows smaller cores and windings. Reduction of the size of
the magnetics is the main reason switching regulator design
tends toward higher operating frequency. Designs with the
RC4391 should use 75 kHz as a maximum running fre-
quency, because the turn off delay of the power transistor
and stray capacitive coupling begin to interfere. Most appli-
cations are in the 10 to 50 kHz range, for efficiency and EMI
reasons.
4. If the required inductance is equal to the indicated value
on the graph, then wind the core with the number of turns
shown in the table of sizes. The turns given are the
maximum number for that gauge of wire that can be
easily wound in cores winding area.
5. If the required inductance is less than the value indicated
on the graph, a simple calculation must be done to find
The peak inductor current (I ) must reach a high enough
MAX
value to meet the load current and simultaneously the induc-
tor value is decreased, then the core can be made smaller.
For a given core size and winding, an increase in air gap
spacing (an air gap is a break in the material in the magnetic
path, like a section broken off a doughnut) will cause the
the adjusted number of turns. Find A (inductance index)
L
for a specific air gap.
inHenries
-------------------------
L(indicated)
æ
ö
--------------------------------- = A
Lè
ø
Turn2
Turns2
inductance to decrease and I
MAX
before saturation )to increase.
(the usable peak current
Then divide the required inductance value by A to give the
L
actual turns squared, and take the square root to find the
actual turns needed.
The curves shown are typical of the ferrite manufacturer's
power HF material, such as Siemens N27 or Stackpole 24B,
which are usually offered in standard millimeter sizes
including the sizes shown.
L(required)
ActualTurns = ------------------------------
AL
22X
13 mm
24 Gauge
70 Turns
DCW = 0.5W
Air Gap = 0.02"
Air Gap = 0.012"
#1
#2
3A
18X
11 mm
26 Gauge
70 Turns
DCW = 0.7W
2A
1A
0
Air Gap = 0.006"
#1
14X
28 Gauge
60 Turns
8 mm
No Air Gap
#2
#3
#4
DCW = 0.6W
#3
#4
1 mH
Inductor Value (Henries)
*Includes safety margin (25%) to ensure nonsaturation
11X
7 mm
2 mH
3 mH
30 Gauge
50 Turns
DCW = 1W
Figure 22. Inductor Design Aid
17
PRODUCT SPECIFICATION
RC4391
If the actual number of turns is significantly less than the
number from the table then the wire size can be increased to
use up the leftover winding area and reduce resistive losses.
Where:
N = number of turns
Ae = core area from data sheet (in cm2)
le = magnetic path length from data sheet (in cm)
ue =permeability of core from manufacturer's graph
g = center post air gap (in cm)
6. Wind and gap the core as per calculations, and measure
the value with an inductance meter. Some adjustment of
the number of turns may be necessary.
Manufacturers
The saturation characteristics may be checked with the
inductor wired into the switching regulator application
circuit. To do so, build and power up the circuit. Then clamp
an oscilloscope current probe (recommend Tektronix P6042
or equivalent) around the inductor lead and monitor the cur-
rent in the inductor. Draw the maximum load current from
the application circuit so that the regulator is running at close
to full duty cycle. Compare the waveform you see to those
pictured.
Below is a list of several pot core manufacturers:
Ferroxcube Company
5083 Kings Highway
Saugerties, NY 12477
Indiana General Electronics
Keasley, NJ 08832
Check for saturation at the highest expected ambient
temperature.
Siemens Company
186 Wood Avenue South
Iselin, NJ 08830
7. After the operation in circuit has been checked,
reassemble and pot the core using a potting compound
recommended by the manufacturer.
Stackpole Company
201 Stackpole Street
St. Mary, PA 15857
If the core material differs greatly in magnetic
characteristics from the standard power material shown
in Figure 16, then the following general equation can be
used to help in winding and gapping. This equation can
be used for any core geometry, such as an E-E core.
TDK Electronics
13-1, 1-Chrome
Nihonbaski, Chuo-ku, Tokyo
(1.26)(N2)(Ae)(108)
LX = -----------------------------------------------------
g= (le/ue)
Improper Operation
(Waveform is Nonlinear, Inductor
Is Saturating)
Proper Operation
(Waveform is Fairly Linear)
I
MAX
I
MAX
0
0
65-3464-08
Figure 23. Inductor Current Waveforms
18
RC4391
PRODUCT SPECIFICATION
Mechanical Dimensions
8-Lead Ceramic DIP Package
Notes:
Inches
Millimeters
Min. Max.
Symbol
Notes
1. Index area: a notch or a pin one identification mark shall be located
adjacent to pin one. The manufacturer's identification shall not be
used as pin one identification mark.
Min.
Max.
A
—
.200
.023
.065
.015
.405
.310
—
.36
1.14
.20
—
5.08
.58
2. The minimum limit for dimension "b2" may be .023 (.58mm) for leads
number 1, 4, 5 and 8 only.
b1
b2
c1
D
.014
.045
.008
—
8
2, 8
1.65
.38
3. Dimension "Q" shall be measured from the seating plane to the base
plane.
8
4
10.29
7.87
4. This dimension allows for off-center lid, meniscus and glass overrun.
E
.220
5.59
4
5. The basic pin spacing is .100 (2.54mm) between centerlines. Each
pin centerline shall be located within ±.010 (.25mm) of its exact
longitudinal position relative to pins 1 and 8.
5, 9
7
e
.100 BSC
.300 BSC
2.54 BSC
7.62 BSC
eA
L
.125
.200
.060
—
3.18
5.08
1.52
—
6. Applies to all four corners (leads number 1, 4, 5, and 8).
Q
s1
a
.015
.005
90¡
.38
.13
90¡
3
6
7. "eA" shall be measured at the center of the lead bends or at the
centerline of the leads when "a" is 90¡.
105¡
105¡
8. All leads – Increase maximum limit by .003 (.08mm) measured at the
center of the flat, when lead finish applied.
9. Six spaces.
D
4
1
8
Note 1
E
5
s1
eA
e
A
Q
c1
a
L
b2
b1
19
PRODUCT SPECIFICATION
RC4391
Mechanical Dimensions (continued)
8-Lead Plastic DIP Package
Notes:
Inches
Millimeters
Min. Max.
Symbol
Notes
1. Dimensioning and tolerancing per ANSI Y14.5M-1982.
Min.
Max.
2. "D" and "E1" do not include mold flashing. Mold flash or protrusions
shall not exceed .010 inch (0.25mm).
A
—
.210
—
—
.38
5.33
—
A1
A2
B
.015
.115
.014
.045
.008
3. Terminal numbers are for reference only.
.195
.022
.070
.015
2.93
.36
4.95
.56
4. "C" dimension does not include solder finish thickness.
5. Symbol "N" is the maximum number of terminals.
B1
C
1.14
.20
1.78
.38
4
2
D
.348
.005
.300
.240
.430
—
.325
.280
8.84
.13
10.92
—
D1
E
7.62
6.10
8.26
7.11
2
5
E1
e
.100 BSC
2.54 BSC
eB
L
—
.430
.160
—
10.92
4.06
.115
2.92
N
8¡
8¡
D
1
4
E1
D1
5
8
e
E
A2
A
A1
C
L
eB
B1
B
20
RC4391
PRODUCT SPECIFICATION
Mechanical Dimensions (continued)
8-Lead SOIC Package
Notes:
Inches
Millimeters
Symbol
Notes
1. Dimensioning and tolerancing per ANSI Y14.5M-1982.
Min.
Max.
Min.
Max.
2. "D" and "E" do not include mold flash. Mold flash or
protrusions shall not exceed .010 inch (0.25mm).
A
.053
.004
.013
.008
.189
.150
.069
.010
.020
.010
.197
.158
1.35
0.10
0.33
0.20
4.80
3.81
1.75
0.25
0.51
0.25
5.00
4.01
A1
B
3. "L" is the length of terminal for soldering to a substrate.
4. Terminal numbers are shown for reference only.
5. "C" dimension does not include solder finish thickness.
6. Symbol "N" is the maximum number of terminals.
C
D
E
5
2
2
e
.050 BSC
1.27 BSC
H
h
.228
.010
.016
.244
.020
.050
5.79
0.25
0.40
6.20
0.50
1.27
L
3
6
N
a
8
8
0¡
8¡
0¡
8¡
ccc
—
.004
—
0.10
8
5
E
H
1
4
h x 45¡
D
C
A1
A
a
SEATING
PLANE
– C –
L
e
LEAD COPLANARITY
ccc C
B
21
PRODUCT SPECIFICATION
RC4391
Ordering Information
Part Number
RC4391N
RC4391M
RV4391N
RM4391D
Package
Operating Temperature Range
0˚C to +70°C
8 Lead Plastic DIP
8 Lead Plastic SOIC
8 Lead Plastic DIP
8 Lead Ceramic DIP
0˚C to +70°C
-25°C to +85°C
-55˚C to +125°C
LIFE SUPPORT POLICY
FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES
OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR
CORPORATION. As used herein:
1. Life support devices or systems are devices or systems
which, (a) are intended for surgical implant into the body,
or (b) support or sustain life, and (c) whose failure to
perform when properly used in accordance with
instructions for use provided in the labeling, can be
reasonably expected to result in a significant injury of the
user.
2. A critical component in any component of a life support
device or system whose failure to perform can be
reasonably expected to cause the failure of the life support
device or system, or to affect its safety or effectiveness.
www.fairchildsemi.com
5/20/98 0.0m 001
Stock#DS30004391
Ó 1998 Fairchild Semiconductor Corporation
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