FP6711MSPTR [FITIPOWER]
High-Efficiency, 1-Cell and 2-Cell Boost Converter;
| 型号： | FP6711MSPTR |
| 厂家： | 
Fitipower |
| 描述： | High-Efficiency, 1-Cell and 2-Cell Boost Converter |
| 文件： | 总13页 (文件大小：678K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
fitipower integrated technology lnc.
High-Efficiency, 1-Cell and
2-Cell Boost Converter
Description
Features
The FP6711 is a high efficiency, fixed frequency
500KHz, current mode PWM boost DC/DC converter
which could operate from single/dual-cell NiCd,
NiMH or alkaline battery such as input voltage below
1V. The converter output voltage can be adjusted
from 1.8V to maximum 4V by an external resistor
divider. Besides the converter includes a 0.35Ω
N-channel MOSFET switch and 0.45Ω P-channel
synchronous rectifier. So no external Schottky
diode is required, and it could get better efficiency
near 94%.
● Synchronous Rectification: 94% Efficiency
● Very Low Start-up Voltage at 0.85V
● Automatically Switch to PFM Mode for Improving
Efficiency at Light Load
● Built-in True Shutdown: Isolation of Load from
Battery during Shutdown
● Internal Anti-Ringing Switch across Inductor
● Low Battery Warning Display
● Fixed Frequency Operation at 500kHz
● Very Low Shutdown Current at 1μA
● Small 10-Pin MSOP Package
● RoHS Compliant
The converter is based on a fixed frequency, current
mode, pulse-width-modulation PWM controller that
goes automatically into PFM mode at light load
which quiescent current is only 25μA in this mode
operation.
Applications
● Handheld Instrument
● Cordless Phone
● Wireless Handset
● GPS Receiver
● MP3
The converter features a special function that the
load is completely isolated from the battery during
shutdown.
Besides it also has auto-discharge
function which could discharge the output capacitor
immediately during shutdown.
When converter operates into discontinuous mode,
the internal anti-ringing switch will reduce
interference and radiated electromagnetic energy.
The FP6711 is available in a space-saving 10-lead
MSOP package for portable application.
Pin Assignments
Ordering Information
FP6711□□□
MS Package (MSOP-10)
TR: Tape/Reel
1
2
3
4
5
10
9
EN
COMP
FB
LBO
LBI
P: Green
G: Green
8
ADEN
SW
Package Type
MS: MSOP-10
7
GND
VOUT
6
VIN
Figure 1. Pin Assignment of FP6711
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Typical Application Circuit
L1
10µH
SW
VIN
VOUT
LBO
VOUT
COUT
VIN
R5
R1
R2
22µF
R3
CIN
10µF
LBI
Low Battery
Warning
FP6711
GND
ADEN
EN
FB
ON
R4
COMP
OFF
RC
CC
680K
100P
Figure 2. Typical Application Circuit of FP6711
Functional Pin Description
Pin Name
Pin Function
EN
COMP
FB
Chip-enable input. Pull the pin high to enable IC. Pull the pin low to shutdown IC.
The gm error amplifier output. A frequency compensation network is connected from this pin to ground to
compensate the loop.
The feedback input for adjusting output voltage. This pin connects resistor divider that output voltage could be
adjusted from 1.8V to 4V. The feedback voltage is typical at 0.5V.
GND
VOUT
VIN
Ground pin
Output voltage pin
Input voltage pin
SW
Switch input pin which is connected to inductor
Auto-discharge enable input pin. The auto-discharge function will be enabled when this pin is connected to logic
high. It will be disabled when this pin is connected to logic low.
ADEN
Low battery detector input. A low battery warning signal is generated at LBO when the voltage on LBI drops
below the threshold voltage of 500mV. Connect LBI to GND or VIN when low battery detector function is not
used. Don’t leave this pin floating.
LBI
Open drain low battery detector output. This Pin will be pulled low when the voltage on LBI drops below the
threshold voltage of 500mV. An external pull-up resistor has to be connected between LBO and VOUT.
LBO
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Block Diagram
ADEN
VIN
SW
PMOS
VOUT
ANTIRING
I2
On/Off
Control
EN
NMOS
PFM
Control
Body-Diode
Switch
Anti-Reverse
Comparator
Isense/current limit
Ramp generator
OSC
PWM
Control
Logic
LBI
COMP
FB
Error Amp
Bandgap
Reference
UVLO
LBO
VREF
VIN
GND
Figure 3. Block Diagram of FP6711
COMP
Absolute Maximum Ratings
● Supply Input Voltage (VIN ,VOUT, EN, LBI, COMP, FB, ADEN, LBO) ------------------------------ -0.3V to +4V
● SW Voltage (SW) ------------------------------------------------------------------------------------------------ -0.3V to +7V
+630mW
● Power Dissipation @TA=25C, MSOP-10 (PD) -----------------------------------------------------------
● Package Thermal Resistance, MSOP-10 (θJA) ----------------------------------------------------------- +160C/W
● Maximum Junction Temperature (TJ) -----------------------------------------------------------------------
+150C
● Storage Temperature Range (T ) ----------------------------------------------------------------------------
-65C to +150C
S
● Lead Temperature (Soldering, 10 sec.) (TLEAD) -----------------------------------------------------------
+260C
Note 1：Stresses beyond those listed under “Absolute Maximum Ratings" may cause permanent damage to the device.
Recommended Operating Conditions
● Input Voltage (VIN) ----------------------------------------------------------------------------------------------- +0.85V to VOUT
● Operating Temperature Range (TOPR) ----------------------------------------------------------------------
-40C to +85C
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Electrical Characteristics
(VIN=1.2V, EN=VIN, TA=25C, unless otherwise specified)
Parameter
Symbol
Conditions
Min
Typ
Max
Unit
V
Start-up Voltage
VST
IOUT =1mA
IOUT =1mA
VFB>0.7V
0.85
Output Voltage Range
Quiescent Current (No Switching)
Switch Current Limit (Note2)
Feedback Voltage
VOUT
IQ
1.8
4
V
25
1
40
µA
A
ILIM
VOUT =3.3V
VFB
490
420
500
500
85
510
780
mV
kHz
%
Oscillation Frequency
fOSC
DMAX
Maximum Duty Cycle
NMOS Switch ON Resistance
(Note2)
PMOS Switch ON Resistance
(Note2)
RDS(ON) VOUT =3.3V
RDS(ON) VOUT =3.3V
0.35
0.45
0.3
0.1
300
Ω
Ω
Line Regulation
Load Regulation
VIN =2V to 2.4V Io =100mA
%
VLINE
VIN =2V IOUT =50 to 100mA
%
VLOAD
Auto-Discharge Switch Resistance
(Note2)
Residual Output Voltage after
Discharge
400
0.4
Ω
ADEB =VIN EN =GND
VLBI voltage decreasing
V
LBI Voltage Threshold
LBI Input Hysteresis
VLBI
480
500
10
520
mV
mV
µA
V
LBI Input Current
0.1
1
LBO Output Low Voltage
LBO Output Leakage Current
FB Input Bias Current
VLBO
VLBI =0V, VOUT =3.3V
0.2
VLBI =650mV, VLBO =VO
0.1
0.1
1
1
µA
µA
V
I(FB)
VIL
EN/ADEN Input Low Voltage
EN/ADEN Input High Voltage
EN/ADEN Input Current
0.8V<VIN<5V
VIN0.1
VIH
0.8V<VIN<5V
V
VIN0.9
EN/ADEN =GND or VIN
EN =0V, ADEN= VIN
0.1
1
1
5
µA
µA
C
C
Shutdown Current from Power
Source
IOFF
TSD
150
20
Over-Temperature Protection
(Note2)
Δ TSD
Hysteresis
Note 2：The specification is guaranteed by design, not production tested.
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Typical Performance Curves
700
600
500
400
300
200
100
0
600
500
400
300
200
100
VOUT=3.3V
VOUT=2.5V
0.9
1.2
1.5
1.8
2.1
2.4
1.0
1.5
2.0
2.5
3.0
Input Voltage(V)
Input Voltage(V)
Figure 4. Maximum Output Current vs. Input Voltage
Figure 5. Maximum Output Current vs. Input Voltage
100
100
VBAT=1.2V
VBAT=1.2V
90
90
80
70
60
50
40
30
20
10
VOUT=2.5V
VOUT=3.3V
80
70
60
50
40
30
20
10
0.1
1
10
100
1000
0.1
1
10
100
1000
IOUT Output Current(mA)
IOUT Output Current(mA)
Figure 6. Efficiency vs. Output Current
Figure 7. Efficiency vs. Output Current
100
90
80
70
60
50
40
30
20
10
100
90
80
70
60
50
40
30
20
10
VBAT=2.5V
VOUT=3.3V
VBAT=1.5V
VOUT=3.3V
0.1
1
10
100
1000
0.1
1
10
100
1000
IOUT Output Current(mA)
IOUT Output Current(mA)
Figure 8. Efficiency vs. Output Current
FP6711-1.4-DEC-2011
Figure 9. Efficiency vs. Output Current
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Typical Performance Curves (Continued)
3.50
100
VIN=1.2V
90
IO=10mA
80
IO=100mA
3.25
70
IO=150mA
60
50
3.00
0.5
1.0
1.5
2.0
2.5
3.0
3.5
0.1
1
10
100
1000
Output Current (mA)
Input Voltage (V)
Figure 10. Efficiency vs. Input Voltage (VOUT=3.3V)
Figure 11. Output voltage vs. Output Current
50
45
40
35
30
25
20
15
10
2.75
2.50
2.25
VIN=1.2V
TA=850C
TA=250C
TA=-400C
1.0
1.5
2.0
2.5
3.0
0.1
1
10
100
1000
Output Current (mA)
Input Voltage(V)
Figure 12. Output voltage vs. Output Current
Figure 13. Quiescent Current vs. Input Voltage
0.520
0.515
0.510
0.505
0.500
0.495
0.490
0.485
0.480
750
700
650
600
550
500
450
400
350
-40
-20
0
20
40
60
80
-40
-20
0
20
40
60
80
Temperature (0C)
Temperature (0C)
Figure 14. Feedback Voltage vs. Temperature
FP6711-1.4-DEC-2011
Figure 15. Oscillator Frequency vs. Temperature
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Typical Performance Curves (Continued)
3.30
3.25
3.20
3.15
3.10
3.05
3.00
2.95
2.90
2.85
2.80
5
4
3
2
1
0
V
IN=1.5V
VOUT=3.0V
IO=10mA
TA=850C
TA=250C
TA=-400C
1.2
1.5
1.8
2.1
2.4
2.7
3.0
-40
-20
0
20
40
60
80
Temperature (0C)
Input Voltage (V)
Figure 16. Output Voltage vs. Temperature
Figure 17.Shutdown Supply Current vs. Input Voltage
CH1: VSW ,CH2: VOUT ,CH4: IL (VIN=1.5V, VOUT=3.3V, IOUT=20mA)
CH1: VSW ,CH2: VOUT ,CH4: IL (VIN=1.2V, VOUT=3.3V, IOUT=100mA)
Figure 18. Dynamic Test
Figure 19. Dynamic Test
CH2: VOUT ,CH4: IOUT (VIN=2V, VOUT=3V, IOUT=50mA 100mA)
CH1: VIN ,CH2: VOUT (VIN=1.2V~1.8V, VOUT=3.3V, IOUT=50mA)
Figure 20. Load Transient Response
Figure 21. Line Transient Response
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Typical Performance Curves (Continued)
VOUT
VSW
SHDN
IIN
VOUT
VIN=1.5V, VOUT=3.3V
IOUT=10mA100mA (PFMPWM)
IL
SW
Figure 22. Load Transient Response
Figure 23. Converter Start-up Time after Enable
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Application Information
Controller Circuit
PFM Mode
The device is based on a current-mode control
The FP6711 is designed for high efficiency over a
wide output current range. Even at light load, the
efficiency stays high because the switching losses
of the converter are minimized by effectively
reducing the switching frequency. The controller
will enter a power saving mode if certain conditions
are met. In this mode, the controller only switches
on the transistor if the output voltage trips below a
set threshold voltage. It ramps up the output
voltage with one or several pulses, and goes again
into PFM mode once the output voltage exceeds a
set threshold voltage.
topology and uses
a
constant frequency
pulse-width modulator to regulate the output
voltage. The controller limits the current through
the power switch on a pulse by pulse basis. The
current sensing circuit is integrated in the device;
therefore, no additional components are required.
Due to the nature of the boost converter topology
used here, the peak switch current is the same as
the peak inductor current, which will be limited by
the integrated current limiting circuits under normal
operating conditions.
The control loop must be externally compensated
with an R-C network connected to the COMP pin.
Device Enable
The device will be shut down when EN is set to
GND. In this mode, the regulator stops switching,
all internal control circuitry including the low-battery
comparator will be switched off, and the load is
disconnected from the input (as described in above
synchronous rectifier section). This also means
that the output voltage may drop below the input
voltage during shutdown.
Synchronous Rectifier
The device integrates an N-channel and a P-
channel MOSFET transistor to realize
a
synchronous rectifier. There is no additional
Schottky diode required. Because the device
uses a integrated low RDS(ON) PMOS switch for
rectification, the power conversion efficiency
reaches 94%.
The device is put into operation when EN is set
high. During start-up of the converter, the duty
cycle is limited in order to avoid high peak currents
drawn from the battery. The limit is set internally
by the current limit circuit and is proportional to the
voltage on the COMP pin.
A special circuit is applied to disconnect the load
from the input during shutdown of the converter.
In conventional synchronous rectifier circuits, the
backgate diode of the high-side PMOS is forward
biased in shutdown and allows current flowing from
the battery to the output. This device, however,
uses a special circuit to disconnect the backgate
diode of the high-side PMOS and so, disconnects
the output circuitry from the source when the
regulator is not enabled (EN = low).
Under-Voltage Lockout
Under-voltage lockout function prevents the device
from starting up if the supply voltage on VBAT is
lower than approximately 0.7V. This under-voltage
lockout function is implemented in order to prevent
the malfunctioning of the converter. When the
battery is being discharged, the device will
automatically enter the shutdown mode if the
voltage on VBAT drops below approximately 0.7V.
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Application Information (Continued)
Auto-Discharge
Adjustable Output Voltage
The auto-discharge function is useful for
applications where the supply voltage of a µC, µP,
or memory has to be removed during shutdown in
order to ensure a defined state of the system.
The accuracy of the output voltage is determined by
the accuracy of the internal voltage reference, the
controller topology, and the accuracy of the external
resistor. The reference voltage has an accuracy of
± 4%.
The controller switches between fixed
frequency and PFM mode, depending on load
current. The tolerance of the resistors in the
feedback divider determines the total system
accuracy.
The auto-discharge function will be enabled when
the ADEN is set high; and it will be disabled when
the ADEN is set to GND.
When the
auto-discharge function is enabled, the output
capacitor will be discharged after the device is shut
down by setting EN to GND. The capacitors
connected to the output are discharged by an
integrated switch of 300Ω, hence the discharge
time depends on the total output capacitance. The
residual voltage on VOUT is less than 0.4V after
auto-discharge.
Design Procedure
The FP6711 boost converter family is intended for
systems that are powered by a single-cell NiCd or
NiMH battery with a typical terminal voltage
between 0.9V to 1.6V. It can also be used in
systems that are powered by two-cell NiCd or NiMH
batteries with a typical stack voltage between 1.8V
to 3.2V. Additionally, single or dual-cell, primary
and secondary alkaline battery cells can be the
power source in systems where the FP6711 is used.
The resistive divider scales down the battery
voltage to a voltage level of 500mV, which is then
compared to the LBI threshold voltage. The LBI
pin has a built-in hysteresis of 10mV. See the
application section for more details about the
programming of the LBI threshold.
(1) Programming the Output Voltage
If the low-battery detection circuit is not used, the
LBI pin should be connected to GND (or to VBAT)
and the LBO pin can be left unconnected. Do not
let the LBI pin float.
The output voltage of the FP6711 can be
adjusted with an external resistor divider. The
typical value of the voltage on the FB pin is
500mV in fixed frequency operation.
The
maximum allowed value for the output voltage is
3.3V. The current through the resistive divider
should be about 100 times greater than the
current into the FB pin. The typical current into
the FB pin is 0.01µA, and the voltage across R4
is typically 500mV. Based on those two values,
the recommended value for R4 is in the range of
500kΩ in order to set the divider current at 1µA.
From that, the value of resistor R3, depending on
the needed output voltage (VO), can be
calculated using Equation 1.
Low-Battery Detector Circuit (LBI and LBO)
The low-battery detector circuit is typically used to
supervise the battery voltage and generate an error
flag when the battery voltage drops below user-set
threshold voltage. The function is active only
when the device is enabled. When the device is
disabled, the LBO pin will be high impedance. The
LBO pin goes active low when the voltage on the
LBI pin decreases below the set threshold voltage
of 500 mV ±15 mV, which is equal to the internal
reference voltage. The battery voltage, at the
detection circuit switches, can be programmed with
a resistive divider connected to the LBI pin.
VO
VO
R3 R4(
-1) 500k(
.....(1)
-1)
VFB
500 mV
Anti-Ringing Switch
The device integrates a circuit which removes the
ringing that typically appears on the SW node when
the converter enters the discontinuous current
mode. In this case, the current through the
inductor ramps to zero and the integrated PMOS
switch turns off to prevent a reverse current from
the output capacitors back to the battery. Due to
remaining energy that is stored in parasitic
components of the semiconductors and the
inductor, a ringing on the SW pin is induced. The
integrated anti-ringing switch clamps this voltage
internally to VBAT; therefore, dampens this ringing.
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Application Information (Continued)
In this example, the desired inductor has the value
of 12µH. With this calculated value and currents,
it is possible to choose a suitable inductor. Care
must be taken that load transients and losses in the
circuit can lead to higher currents. Also, the
losses in the inductor caused by magnetic
hysteresis losses and copper losses are a major
parameter for total circuit efficiency.
(2) Programming the Low Battery Comparator
Threshold Voltage
The current through the resistive divider should
be about 100 times greater than the current into
the LBI pin. The typical current into the LBI pin
is 0.01µA; the voltage across R2 is equal to the
reference voltage that is generated on-chip,
which has a value of 500mV±15mV.
The
recommended value for R2 is therefore in the
range of 500 kΩ. From that, the value of
resistor R1, depending on the desired minimum
battery voltage VBAT, can be calculated using
Equation 2.
(4) Capacitor Selection
The major parameter necessary to define the
output capacitor is the maximum allowed output
voltage ripple of the converter. This ripple is
determined by two parameters of the capacitor, the
capacitance and the ESR.
It is possible to
VBAT
VREF
VBAT
…..(2)
-1)
R1 R2(
-1) 500k(
calculate the minimum capacitance needed for the
defined ripple, supposing that the ESR is zero, by
using Equation 4.
500mV
For example, if the low-battery detection circuit
should flag an error condition on the LBO output
pin at a battery voltage of 1V, a resistor in the
range of 500kΩ should be chosen for R1. The
output of the low battery comparator is a simple
open-drain output that goes active low if the
battery voltage drops below the programmed
threshold voltage on LBI. The output requires a
pull-up resistor with a recommended value of
1MΩ, and should only be pulled up to the VO. If
not used, the LBO pin can be left floating or tied
to GND.
IOUT × (VOUT - VBAT
)
…..(4)
CMIN
=
f × ΔV× VOUT
Parameter f is the switching frequency and △V is
the maximum allowed ripple.
With a chosen ripple voltage of 15mV, a minimum
capacitance of 10 µF is needed. The total ripple is
larger due to the ESR of the output capacitor.
This additional component of the ripple can be
calculated using Equation 5.
ΔVESR =IOUT × RESR …..(5)
(3) Inductor Selection
A boost converter normally requires two main
passive components for storing energy during
the conversion. A boost inductor is required and
a storage capacitor at the output. To select the
boost inductor, it is recommended to keep the
possible peak inductor current below the current
limit threshold of the power switch in the chosen
configuration.
An additional ripple of 30mV is the result of using a
tantalum capacitor with a low ESR of 300mΩ. The
total ripple is the sum of the ripple caused by the
capacitance and the ripple caused by the ESR of
the capacitor. In this example, the total ripple is
45mV. It is possible to improve the design by
enlarging the capacitor or using smaller capacitors
in parallel to reduce the ESR or by using better
capacitors with lower ESR, like ceramics. For
example, a 10µF ceramic capacitor with an ESR of
50mΩ is used on the evaluation module (EVM).
Tradeoffs must be made between performance and
costs of the converter circuit.
The second parameter for choosing the inductor
is the desired current ripple in the inductor.
Normally, it is advisable to work with a ripple of
less than 20% of the average inductor current.
A smaller ripple reduces the magnetic hysteresis
losses in the inductor, as well as output voltage
ripple and EMI. But in the same way, regulation
time at load changes rises. In addition, a larger
inductor increases the total system cost. With
those parameters, it is possible to calculate the
value for the inductor by using Equation 3.
A 10µF input capacitor is recommended to improve
transient behavior of the regulator. A ceramic or
tantalum capacitor with a 100nF in parallel placed
close to the IC is recommended.
VBAT × (VOUT - VBAT
)
…..(3)
L =
ΔIL × f × VOUT
Parameter f is the switching frequency and ΔIL is
the ripple current in the inductor, i.e, 20% x IL.
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Application Information (Continued)
(5) Compensation of the Control Loop
Thermal Information
An R/C network must be connected to the COMP
pin in order to stabilize the control loop of the
converter. Both the pole generated by the
inductor L1 and the zero caused by the ESR and
capacitance of the output capacitor must be
compensated. The network shown in Figure 24
satisfies these requirements.
The maximum junction temperature (TJ) of the
FP6711 devices is recommended to 125°C. The
thermal resistance of the 10-pin MSOP package is
JA=160°C/W. Specified regulator operations are
assured to a maximum ambient temperature (TA) of
70°C. Therefore, the maximum power dissipation
is about 340mW. More power can be dissipated if
the maximum ambient temperature of the
application is lower.
VOUT
0.5V
TJ(MAX) - TA
125C - 70C
160C/W
R3
ERROR
AMP
PD(MAX)
340mW
JA
8
R4
Layout Considerations
9
CC
Rc
As for all switching power supplies, the layout is an
important step in the design, especially at high peak
currents and high switching frequencies. If the
layout is not carefully done, the regulator could
show stability problems as well as EMI problems.
Therefore, use wide and short traces for the main
current path as indicated in bold in Figure 25. The
input capacitor, output capacitor and the inductor
should be placed as close to the IC as possible.
Use a common ground node as shown in Figure 25
to minimize the effects of ground noise. The
compensation circuit and the feedback divider
should be placed as close to the IC as possible. To
layout the control ground, it is recommended to use
short traces as well, separated from the power
ground traces. Connect both grounds close to the
ground pin of the IC as indicated in the layout
diagram in Figure25. This avoids ground shift
problems, which can occur due to superimposition
of power ground current and control ground current.
Figure 24. Compensation of Control Loop
Resistor RC and capacitor CC depend on the
chosen inductance. The equation for the loop
dynamics is shown as below :
1
HZ
fZER01
=
2 x
π
x Rc x CC
The FP6711 uses current mode control with
internal adaptive slope compensation. Current
mode control eliminates the 2nd order filter due to
the inductor and output capacitor exhibited in
voltage mode controllers and simplifies it to a
single-pole filter response.
L1
VOUT
SW
R5
OUTPUT
VBAT
LBI
LBO
FB
LBO
Battery
C4
R3
R1
R2
C1
FP6711
RC
COMP
GND
ADEN
EN
R4
CC
Figure 25. Layout Diagram
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Outline Information
MSOP-10 Package (Unit: mm)
DIMENSION IN MILLIMETER
SYMBOLS
UNIT
MIN
0.75
0.00
0.75
0.17
2.90
4.80
2.90
0.40
0.40
MAX
1.10
0.15
0.95
0.33
3.10
5.00
3.10
0.60
0.80
A
A1
A2
B
D
E
E1
e
L
Carrier dimensions
Life Support Policy
Fitipower’s products are not authorized for use as critical components in life support devices or other medical systems.
FP6711-1.4-DEC-2011
13
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