MC33215_技术文档

Freescale Semiconductor, Inc._{Order this document by MC33215/D}

52

1

FB SUFFIX

PLASTIC PACKAGE

CASE 848B

The MC33215 is developed for use in fully electronic telephone sets with

speakerphone functions. The circuit performs the ac and dc line termination,

2–4 wire conversion, line length AGC and DTMF transmission. The

speakerphone part includes a half duplex controller with signal and noise

monitoring, base microphone and loudspeaker amplifiers and an efficient

supply. The circuit is designed to operate at low line currents down to 4.0 mA

enabling parallel operation with a classical telephone set.

(TQFP–52)

• Highly Integrated Cost Effective Solution

• Straightforward AC and DC Parameter Adjustments

• Efficient Supply for Loudspeaker Amplifier and Peripherals

• Stabilized Supply Point for Handset Microphone

• Stabilized Supply Point for Base Microphone

• Loudspeaker Amplifier can be Powered and Used Separately

• Smooth Switch–Over from Handset to Speakerphone Operation

• Adjustable Switching Depth for Handsfree Operation

42

1

B SUFFIX

PLASTIC PACKAGE

CASE 858

(SDIP–42)

ORDERING INFORMATION

Operating

Temperature Range

Device

Package

MC33215FB

MC33215B

TQFP–52

SDIP–42

T = –20° to +70°C

A

Simplified Application

AC

Impedance

DC Offset

Line Current

Telephone

Line

Current

Splitter

1:10

DTMF

MF

HM

BM

V

Supply

CC

Handset

Microphone

Line

Driver

DC Slope

Base

Microphone

Attenuator

Duplex

Controller

Receive Signal

V

or

CC

External Supply

R

x

Handset Earpiece

Attenuator

LS

Base Loudspeaker

Auxiliary Input

This device contains 2782 active transistors.

This document contains information on a new product. Specifications and information herein

are subject to change without notice.

Motorola, Inc. 1997

Rev 0

1

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MC33215

FEATURES

Line Driver and Supply

• Separate Input for DTMF and Auxiliary Signals

• Parallel Operation Down to 4.0 mA of Line Current

• AC and DC Termination of Telephone Line

• Adjustable Set Impedance for Real and Complex

Termination

• Efficient Supply Point for Loudspeaker Amplifier and

Peripherals

• Two Stabilized Supply Points for Handset and Base

Microphones

• Separate Supply Arrangement for Handset and

Speakerphone Operation

• Handsfree Operation via Loudspeaker and Base

Microphone

• Integrated Microphone and Loudspeaker Amplifiers

• Differential Microphone Inputs

• Loudspeaker Amplifier can be Powered and Used

Separately from the Rest of the Circuit

• Integrated Switches for Smooth Switch–Over from

Handset to Speakerphone Operation

• Signal and Background Noise Monitoring in Both

Channels

Handset Operation

• Transmit and Receive Amplifiers

• Differential Microphone Inputs

• Sidetone Cancellation Network

• Line Length AGC

• Adjustable Switching Depth for Handsfree Operation

•

Switch–Over

• Dial Tone Detector in the Receive Channel

• Microphone and Earpiece Mute

Figure 1. Pin Connections

1

2

V

42

41

40

39

38

37

36

PGD

LSO

VLS

LSB

CC

VLN

3

VHF

VMC

SLB

52

51

50

49

48

47

46

45

44

43

42

41

40

4

5

1

2

3

4

5

6

7

8

9

SLB

REG

SLP

MFI

LSF

BVO

PPL

39

38

REG

6

BVO

7

SLP

MFI

PPL 37

LSI 36

8

LSI 35

9

HM1

HM2

34

33

32

31

30

29

28

27

26

25

24

23

22

VOL

HM1

HM2

VOL 35

SWD 34

10

11

12

13

14

15

16

17

18

19

20

21

SWD

REF

AGC

Gnd

RLS

RSA

BM2

BM1

REF

AGC

33

32

BM2

BM1

SDIP–42

TQFP–52

V

DD

V

Gnd 31

RLS 30

RSA 29

RSE 28

DD

TSA

TSE

TBN

10 TSA

11 TSE

12 TBN

13 MUT

RSE

RBN

RXI

MUT

SPS

PRS

RBN

27

GRX

14

15

16

17

18

19

20

21

22

23

24

25

26

SWT

LSM

RXO

RXS

(Top View)

2

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MC33215

MAXIMUM RATINGS

Rating

Min

–0.5

–

Max

12

Unit

V

Peak Voltage at VLN

Maximum Loop Current

160

12

mA

V

Voltage at VLS (if Powered Separately)

Voltage at VHF (if Externally Applied)

Voltage at SPS, MUT, PRS, LSM

Maximum Junction Temperature

Storage Temperature Range

–0.5

–

5.5

7.5

150

V

°C

–65

NOTE: ESD data available upon request.

RECOMMENDED OPERATING CONDITIONS

Characteristic

Min

2.4

4.0

2.4

0

Max

Unit

V

Biasing Voltage at VLN

10

130

8.0

5.0

70

Loop Current

mA

V

Voltage at VLS

Voltage at VHF (if Externally Applied)

Voltage at SPS, MUT, PRS, LSM

Operating Ambient Temperature Range

V

–20

°C

ELECTRICAL CHARACTERISTICS (All parameters are specified at T = 25°C, I

= 18 mA, VLS = 2.9 V, f = 1000 Hz,

line

PRS = high, MUT = high, SPS = low, LSM = high, test figure in Figure 17 with S1 in position 1, unless otherwise stated.)

Characteristic

Min

Typ

Max

Unit

DC LINE VOLTAGE

Line Voltage V

V

line

Parallel Operation, I = 4.0 mA

–

3.9

4.8

2.4

4.2

5.2

–

4.5

5.6

line

I

= 20 mA

= 70 mA

line

SUPPLY POINT V

DD

Internal Current Consumption from V

–

1.2

1.5

mA

DD

V

DD

= 2.5 V

SUPPLY POINT VMC

DC Voltage at VMC (= VMC0)

1.6

1.0

1.75

–

1.9

–

V

Current Available from VMC

VMC = VMC0 – 200 mV

mA

SUPPLY POINT VHF

DC Voltage at VHF (= VHF0)

2.6

–

2.8

1.4

3.0

2.0

V

Internal Current Consumption from VHF

VHF = VHF0 + 100 mV

mA

Current Available from VHF

VHF = VHF0 – 300 mV

2.0

–

mA

SUPPLY POINT V

CC

Current Available from V

13

–

15

–

mA

V

CC

V

CC

= 2.4 V, I

= 20 mA

line

DC Voltage Drop Between VLN and V

1.0

1.5

CC

I

= 20 mA

line

SUPPLY INPUT VLS

Internal Current Consumption from VLS

–

1.0

1.5

mA

3

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MC33215

ELECTRICAL CHARACTERISTICS (continued) (All parameters are specified at T = 25°C, I

= 18 mA, VLS = 2.9 V, f = 1000 Hz,

line

PRS = high, MUT = high, SPS = low, LSM = high, test figure in Figure 17 with S1 in position 1, unless otherwise stated.)

Characteristic

Min

Typ

Max

Unit

LOGIC INPUTS

Logic Low Level Pins PRS, MUT, SPS, LSM

Logic High Level Pins PRS, MUT, SPS, LSM

Internal Pull Up Pins PRS, MUT, LSM

Internal Pull Down Pin SPS

–

2.0

–

0.4

5.0

–

V

–

100

kΩ

–

T CHANNEL, HANDSET MICROPHONE AMPLIFIER

x

Voltage Gain from V

to V

46

60

47

–

48

–

dB

HM

line

V

HM

= 1.5 mVrms

Gain Reduction in Mute Condition

MUT = Low or PRS = Low or SPS = High

Input Impedance at HM1 and HM2

Common Mode Rejection Ratio

Total Harmonic Distortion at VLN

14

–

18

50

–

22

–

kΩ

dB

%

–

2.0

V

HM

= 4.5 mVrms

Psophometrically Weighted Noise Level at V

HM1 and HM2 Shorted with 200 Ω

–

–72

–

dBmp

dB

line

T CHANNEL, BASE MICROPHONE AMPLIFIER (SPS = HIGH, T MODE FORCED)

x

Voltage Gain from V

to V

53

55.5

58

BM

line

V

BM

= 0.5 mVrms

Input Impedance at BM1 and BM2

Common Mode Rejection Ratio

Total Harmonic Distortion at VLN

14

–

18

50

–

22

–

kΩ

dB

%

–

2.0

V

BM

= 1.5 mV

Psophometrically Weighted Noise Level at V

BM1 and BM2 Shorted with 200 Ω

–

–62

–

dBmp

dB

line

Gain Reduction in Mute Condition

60

MUT = Low or PRS = Low or SPS = Low

T CHANNEL, DTMF AMPLIFIER (MUT = LOW OR PRS = LOW)

x

Voltage Gain from V to V

34

35

36

dB

MF

line

V

MF

= 7.5 mVrms

Input Impedance at MFI

14

60

18

–

22

–

kΩ

Gain Reduction in Mute Condition

MUT = High or PRS = Low

dB

R CHANNEL, EARPIECE AMPLIFIER

x

Voltage Gain from V

to V

(Note 1)

EAR

23

60

24

–

25

–

dB

RXI

V

line

= 20 mVrms

Gain Reduction in Mute Condition

MUT = Low or SPS = Low

Input Impedance at RXI

24

–

30

36

–

kΩ

Psophometrically Weighted Noise Level at V

130

µVrms

EAR

RXI Shorted to Gnd via 10 µF

Confidence Level During DTMF Dialing

10

15

–

20

–

mVrms

mVpp

V

MF

= 7.5 mVrms, MUT = Low

Output Swing Capability into 150 Ω

THD ≤2%

680

Output Swing Capability into 450 Ω

1800

–

THD ≤2%, R

= 360 kΩ

RXO

NOTE: 1. Corresponding to –0.6 dB gain from the line to output RXO in the typical application.

4

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MC33215

ELECTRICAL CHARACTERISTICS (continued) (All parameters are specified at T = 25°C, I

= 18 mA, VLS = 2.9 V, f = 1000 Hz,

line

PRS = high, MUT = high, SPS = low, LSM = high, test figure in Figure 17 with S1 in position 1, unless otherwise stated.)

Characteristic

Min

Typ

Max

Unit

R CHANNEL, LOUDSPEAKER PRE–AMPLIFIER (SPS = HIGH, R MODE FORCED)

x

Voltage Gain from V

to V

(Note 2)

RLS

21

60

24

–

27

–

dB

RXI

V

line

= 20 mVrms

Gain Reduction in Mute Condition

SPS = Low or MUT = Low

R CHANNEL, LOUDSPEAKER AMPLIFIER

x

Voltage Gain from V to V

25

26

27

dB

LSI

LSP

V

LSI

= 10 mVrms

Attenuation at Delta R

= 47 kΩ

–

32

–

dB

VOL

Psophometrically Weighted Noise Level at V

1.2

mVrms

LSP

RXI Shorted to Gnd via 10 µF

Confidence Level During DTMF Dialing

150

200

250

mVrms

V

MF

= 7.5 mVrms MUT = Low

Available Peak Current from LSO

110

1.8

–

mApeak

Vpp

Output Capability into 25 Ω

THD ≤2%, V = 55 mVrms

LSI

Output Capability into 25 Ω

2.7

60

–

Vpp

dB

THD ≤2%, V = 5.0 V, V = 90 mVrms

LS

LSI

Gain Reduction in Mute Condition

LSM = Low

R CHANNEL PEAK–TO–PEAK LIMITER

x

Peak–to–Peak Limiter Attack Time

–

300

–

5.0

–

ms

%

V

LSI

Jumps from 40 mVrms to 120 mVrms

Peak–to–Peak Limiter Release Time

Jumps from 120 mVrms to 40 mVrms

V

LSI

THD at 10 dB Overdrive

= 120 mVrms

7.0

0.1

V

LSI

Peak–to–Peak Limiter Disable Threshold at PPL

–

V

AUTOMATIC GAIN CONTROL

Gain Reduction in Transmit and Receive Channel with Respect to I

= 18 mA

4.5

–

6.0

–

7.5

1.5

dB

line

I

= 70 mA

line

Gain Variation in Transmit and Receive Channel with Respect to I

=18 mA with

line

AGC Disabled (AGC to V

)

DD

Highest Line Current for Maximum Gain

Lowest Line Current for Minimum Gain

BALANCE RETURN LOSS

–

20

50

–

mA

Balance Return Loss with Respect to 600 Ω

SIDETONE

20

–

dB

Voltage Gain from V

S1 in Position 2

to V

28

HM

EAR

LOGARITHMIC AMPLIFIERS AND ENVELOPE DETECTORS

Voltage Gain from RXI to RSA

18

17.5

40

20

18.5

–

22

19.5

–

dB

V

RXI

= 15 mVrms

Voltage Gain from BMI to TSA

= 0.5 mVrms

V

BM

Dynamic Range of Logarithmic Compression from TSA to TSE and RSA to RSE

and I from 2.5 µA to 250 µA

I

TSA

RSA

Envelope Tracking Between TSE and RSE and Between TBN and RBN

Maximum Source Current from TSE or RSE

–

±3.0

–

dB

0.3

0.4

0.5

µA

NOTE: 2. Corresponding to –0.6 dB gain from the line to output RLS in the typical application.

5

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MC33215

ELECTRICAL CHARACTERISTICS (continued) (All parameters are specified at T = 25°C, I

= 18 mA, VLS = 2.9 V, f = 1000 Hz,

line

PRS = high, MUT = high, SPS = low, LSM = high, test figure in Figure 17 with S1 in position 1, unless otherwise stated.)

Characteristic

Min

Typ

Max

Unit

LOGARITHMIC AMPLIFIERS AND ENVELOPE DETECTORS

Maximum Sink Current into TSE or RSE

100

0.7

100

–

1.3

–

µA

Maximum Sink Current into TBN and RBN

1.0

–

Maximum Source Current from TBN or RBN

µA

Dial Tone Detector Threshold at V

20

4.5

–

mVrms

dB

line

Speech Noise Threshold Both Channels

ATTENUATOR CONTROL

–

Switching Depth

46

24

–

50

–

54

60

–

dB

µA

Adjustable Range for Switching Depth

Gain Variation in Idle Mode for Both Channels

Current Sourced from SWT

25

10

7.0

13

T Mode

x

Current Sunk into SWT

7.0

10

13

µA

R Mode

x

PIN FUNCTION DESCRIPTION

SDIP–42 TQFP–52

Name

Description

Supply Output for Loudspeaker Amplifier and Peripherals

Line Connection Input

1

2

47

48

49

50

51

52

1

V

CC

VLN

VHF

VMC

N/C

3

Supply Output for Speakerphone Section and Base Microphone

Supply Output for Handset Microphone

Not Connected

4

–

N/C

Not Connected

5

SLB

REG

SLP

MFI

SLP Buffered Output

6

2

Regulation of Line Voltage Adjustment

DC Slope Adjustment

7

3

8

4

DTMF Input

9

5

HM1

HM2

BM2

BM1

Handset Microphone Input 1

Handset Microphone Input 2

Base Microphone Input 2

10

11

12

13

14

15

16

17

–

6

7

8

Base Microphone Input 1

9

V

DD

Supply Input for Speech Part

Transmit Sensitivity Adjustment

Transmit Signal Envelope Timing Adjustment

Transmit Background Noise Envelope Timing Adjustment

Transmit and Receive Mute Input

Not Connected

10

11

12

13

14

15

16

17

18

19

TSA

TSE

TBN

MUT

N/C

–

N/C

Not Connected

18

19

20

21

SPS

PRS

SWT

LSM

Speakerphone Select Input

Privacy Switch Input

Switch–Over Timing Adjustment

Loudspeaker Mute Input

6

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MC33215

PIN FUNCTION DESCRIPTION (continued)

SDIP–42 TQFP–52

Name

N/C

Description

–

20

21

22

23

24

25

26

27

28

29

30

31

32

33

34

35

36

37

38

39

40

41

42

43

44

45

46

Not Connected

22

23

24

25

–

RXS

RXO

GRX

RXI

Receive Amplifier Stability

Receive Amplifier Output

Earpiece Amplifier Feedback Input

Receive Amplifier Input

N/C

Not Connected

–

N/C

Not Connected

26

27

28

29

30

31

32

33

34

35

36

37

38

–

RBN

RSE

RSA

RLS

Gnd

AGC

REF

SWD

VOL

LSI

Receive Background Noise Envelope Timing Adjustment

Receive Signal Envelope Timing Adjustment

Receive Sensitivity Adjustment

Receive Output for Loudspeaker Amplifier

Small Signal Ground

Line Length AGC Adjustment

Reference Current Set

Switching Depth Adjustment for Handsfree

Volume Control Adjustment

Loudspeaker Amplifier Input

Peak–to–Peak Limiter Timing Adjustment

Bias Voltage for Loudspeaker Amplifier Output

Loudspeaker Amplifier Feedback Input

Not Connected

PPL

BVO

LSF

N/C

–

N/C

Not Connected

39

40

41

42

–

LSB

VLS

LSO

PGD

N/C

Loudspeaker Amplifier Bootstrap Output

Supply Input for Loudspeaker Amplifier

Loudspeaker Amplifier Output

Power Ground

Not Connected

7

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MC33215

DESCRIPTION OF THE CIRCUIT

Based on the typical application circuit as given in

into a small part for biasing the internal line drive transistor

and into a large part for supplying the speakerphone. The

ratio between these two currents is fixed to 1:10. The dc set

impedance or dc setting of the telephone as created by the

line driver and its external components can be approximated

with the equivalent of a zener voltage plus a series resistor

according to:

Figure 18, the MC33215 will be described in three parts: line

driver and supplies, handset operation, and handsfree

operation. The data used refer to typical data of the

characteristics.

LINE DRIVER AND SUPPLIES

The line driver and supply part performs the ac and dc

telephone line termination and provides the necessary

supply points.

VLN

V

ILN x R

zener

slope

AC Set Impedance

With:

V

The ac set impedance of the telephone as created by the

line driver and its external components can be approximated

with the equivalent circuit shown in Figure 2.

R

REG1

0.2 x

1

10 µA x R

zener

REG1

R

REG2

ILN

R

I

– I

line

VDD

Figure 2. Equivalent of the AC impedance

R

SLP

11

REG1

x

1

Inductor

slope

R

REG2

Z

620

R

REG1

360 k

VDD

Z

bal

If R_REG2is not mounted, the term between the brackets

becomes equal to 1.

With the values shown in the typical application and under

the assumption that I_VDD= 1.0 mA, the above formulas can

be simplified to:

C

10 n

C

100

R

2.2 k

C

220 n

R

REG

∞

VLN

VDD

µ

SLB

REG

Slope

R

SLP

11

VLN

3.8 V

I

– 1.0 mA x 20

Inductor

Slope

R

x C

x

line

REG1

REG

R

I

x 20

line

R

SLP

11

REG1

REG2

x

1

In the typical application this leads to a line voltage of 4.2 V

at 20 mA of line current with a slope of 20 Ω. Adding a 1.5 V

voltage drop for the diode bridge and the interruptor, the dc

voltage at tip–ring will equal 5.7 V.

If the dc mask is to be adapted to a country specific

requirement, this can be done by adjusting the resistors

R_REG1and R_REG2, as a result, the zener voltage and the

slope are varied. It is not advised to change the resistor R_SLP

since this changes many parameters. The influence of R_REG1

and R_REG2is shown in Figure 4.

R

With the component values of the typical application, the

inductor calculates as 1.6 H. Therefore, in the audio range of

300 Hz to 3400 Hz, the set impedance is mainly determined

by Z_VDD. As a demonstration, the impedance matching or

Balance Return Loss BRL is shown in Figure 3.

Figure 3. Balance Return Loss

40

35

30

25

20

15

10

5.0

0

Figure 4. Influence of R_REG1and R_REG2

on the DC Mask

.

12

R

= 470 k

= 220 k

REG1

REG2

10

8.0

6.0

4.0

2.0

0

R

= 365 k

= 220 k

REG1

REG2

R

= 365 k

= Infinite

REG1

REG2

100

1000

10000

R

= 470 k

= Infinite

REG1

REG2

f, FREQUENCY (Hz)

The influence of the frequency dependent parasitic

components is seen for the lower frequencies (Inductor) and

the higher frequencies (C_VLN) by a decreasing BRL value.

0

20

40

60

80

100

I

(mA)

line

DC Set Impedance

As can be seen in Figure 4, for low line currents below

10 mA, the given dc mask relations are no longer valid. This

is the result of an automatic decrease of the current drawn

The line current flowing towards the MC33215 application

is partly consumed by the circuitry connected to V_DDwhile

the rest flows into Pin VLN. At Pin VLN, the current is split up

8

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MC33215

from Pin REG by the internal circuit (the 10 µA term in the

formulas). This built–in feature drops the line voltage and

therefore enables parallel operation.

The voltage over the line driver has to be limited to 12 V to

protect the device. A zener of 11 V at VLN is therefore the

maximum advised.

If, during parallel operation, a high current is required from

VMC, a 2.7 k resistor between VMC and VHF can be applied.

In Figure 5, the VMC voltage under different microphone

currents, is shown.

VHF Supply

VHF is a stabilized supply which powers the internal

duplex controller part of the MC33215, and which is also

meant to power the base microphone or other peripherals.

The base microphone however, can also be connected to

VMC, which is preferred in case of microphones with a poor

supply rejection. Another possibility is to create an additional

filter at VHF, like is shown in the typical application. The

supply capability of VHF is guaranteed as 2.0 mA for line

currents of 20 mA and greater.

Since in parallel operation not enough line current is

available to power a loudspeaker and thus having a

speakerphone working, the current internally supplied to VHF

is cut around 10 mA of line current to save current for the

handset operated part. A small hysteresis is built in to avoid

system oscillations.

When the current to VHF is cut, the voltage at VHF will

drop rapidly due to the internal consumption of 1.4 mA and

the consumption of the peripherals. When VHF drops below

2.0 V, the device internally switches to the handset mode,

neglecting the state of the speakerphone select Pin SPS.

In case an application contains a battery pack or if it is

mains supplied, speakerphone operation becomes possible

under all line current conditions. In order to avoid switch–over

to handset operation below the 10 mA, VHF has to be

supplied by this additional power source and preferably kept

above 2.4 V.

V

_DDSupply

The internal circuitry for the line driver and handset

interface is powered via V_DD. This pin may also be used to

power peripherals like a dialer or microcontroller. The voltage

at V_DDis not internally regulated and is a direct result of the

line voltage setting and the current consumption at V_DD

internally (I_VDD) and externally (I_PER). It follows that:

V

VLN – I

I

x R

set

DD

VDD

PER

For correct operation, it must be ensured that V_DDis

biased at 1.8 V higher than SLP. This translates to a

maximum allowable voltage drop across Z_VDDof

V_zener– 1.8 V. In the typical application, this results in a

maximum allowable current consumption by the peripherals

of 2.0 mA.

VMC Supply

At VMC, a stabilized voltage of 1.75 V is available for

powering the handset microphone. Due to this stabilized

supply, microphones with a low supply rejection can be used

which reduces system costs. In order to support the parallel

operation of the telephone set, the voltage at VMC will be

maintained even at very low line currents down to 4.0 mA.

Under normal supply conditions of line currents of 20 mA

and above, the supply VMC is able to deliver a guaranteed

minimum of 1.0 mA. However, for lower line currents, the

supply capability of VMC will decrease.

V

_CCSupply

At V_CCthe major part of the line current is available for

Figure 5. VMC Under Different Microphone Loads

powering the loudspeaker amplifier and peripheral circuitry.

This supply pin should be looked at as a current source since

the voltage on V_CCis not stabilized and depends on the

instantaneous line voltage and the current to and consumed

1.8

1.7

I

= 20 mA

line

from V_CC

.

I

= 4.0 mA

line

1.6

1.5

1.4

1.3

1.2

1.1

1.0

2.7 k VMC–VHF

The maximum portion of the line current which is available

at V_CCis given by the following relation:

10

11

I

x

I

– I

I

= 4.0 mA

VCC

line

VDD

VMC

VHF

line

This formula is valid when the voltage drop from VLN to

V_CCis sufficient for the current splitter to conduct all this

current to V_CC. When the drop is not sufficient, the current

source saturates and the surplus of current is conducted to

the power ground PGD to avoid distortion in the line driver. In

fact, when no current is drawn from V_CC, the voltage at V_CC

will increase until the current splitter is in balance. In Figure 6

this behavior is depicted.

0

0.2

0.4

0.6

0.8

(mA)

1.0

1.2

1.4

1.6

I

VMC

9

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Figure 6. Available Current at V_CC

100

90

80

70

60

50

40

30

20

10

0

3.5

3.0

2.5

2.0

1.5

1.0

I

at 98% of

VCC

VCC(max)

I

at 50% of

VCC

VCC(max)

I

/l

(%)

VCC line

V

to VLS

Open

CC

V

CC

I

VCC(max) (mA)

0.5

0

20

40

60

80

100

0

20

40

60

80

100

I

(mA)

I

(mA)

line

A. Maximum Available Current at V

B. Voltage Drop to V

CC

For instance, at a line current of 20 mA a maximum of

15 mA of current is available at V_CC. If all this current is

taken, V_CCwill be 1.7 V below VLN. When not all this current

is drawn from V_CC, but for instance only 1.0 mA for biasing of

the loudspeaker amplifier, the voltage at V_CCwill be 1.2 V

below VLN. Although the measurements for Figure 6 are

done with R_REG1= 365 k, the results are also globally valid

for other dc settings.

by adjusting the sensitivity of the handset microphone by

adjusting the resistors R_HM1and R_HM2. It is not advised to

adjust the gain by including series resistors towards the Pins

HM1 and HM2.

A high pass filter is introduced by the coupling capacitors

C

_HM1and C_HM2in combination with the input impedance. A

low pass filter can be created by putting capacitors in parallel

with the resistors R_HM1and R_HM2

.

As can be seen from Figure 6, the voltage at V_CCis limited

by the voltage at VLN minus 1.0 V. This means that the

voltage at V_CCis limited by the external zener at VLN. If it is

necessary to limit the voltage at V_CCin order to protect

peripheral circuits, a zener from V_CCto Gnd can be added. If

The transmit noise is measured as –72 dBmp with the

handset microphone inputs loaded with a capacitively

coupled 200 Ω. In a real life application, the inputs will be

loaded with a microphone powered by VMC. Although VMC

is a stablized supply voltage, it will contain some noise which

can be coupled to the handset microphone inputs, especially

when a microphone with a poor supply rejection is used. An

additional RC filter on VMC can improve the noise figure, see

also the base microphone section.

the supply of the loudspeaker VLS is also connected to V_CC

it is advisable that V_CCdoes not exceed 8.0 V.

,

The high efficiency of the V_CCpower supply contributes

to a high loudspeaker output power at moderate line

currents. More details on this can be found in the handsfree

operation paragraph.

Handset Earpiece Amplifier

The handset earpiece is to be capacitively connected to

the RXO output. Here, the receive signal is available which is

amplified from the line via the sidetone network and the R_x

and EAR amplifiers. The sidetone network attenuates the

receive signal from the line via the resistor divider composed

of R_SLBand Z_bal, see also the sidetone section. The

attenuation in the typical application by this network equals

24.6 dB. Then the signal from the sidetone network is

pre–amplified by the amplifier R_xwith a typical gain of 6.0 dB.

This amplifier also performs the AGC and MUTE functions,

see the related paragraphs. Finally, the signal is amplified by

the noninverting voltage amplifier EAR. The overall receive

gain A_RXfrom the line to the earpiece output then follows as:

HANDSET OPERATION

During handset operation, the MC33215 performs the

basic telephone functions for the handset microphone and

earpiece. It also enables DTMF transmission.

Handset Microphone Amplifier

The handset microphone is to be capacitively connected

to the circuit via the differential input HM1 and HM2. The

microphone signal is amplified by the HMIC amplifier and

modulates the line current by the injection of the signal into

the line driver. This transfer from the microphone inputs to the

line current is given as 15/(R_SLP/11), which makes a total

transmit voltage gain A_HMfrom the handset microphone

inputs to the line of:

V

R

RXO

GRX

A

x A

x

1

RX

ST

RXI

V

Z

x Z

set

line

HM

15

line

A

x

HM

V

Z

R

11

set

line

With: A_ST= Attenuation of the Sidetone Network

_RXI= Gain of the Pre–Amplifier R_x

SLP

With the typical application and Z_line= 600 Ω the transmit

gain calculates as 47 dB.

A

For the typical application an overall gain from the line to

the earpiece is close to 0 dB.

In case an electret microphone is used, it can be supplied

from the stabilized microphone supply point VMC of 1.75 V

properly biased with resistors R_HM1and R_HM2. This allows

the setmaker to use an electret microphone with poor supply

rejection to reduce total system costs. Since the transmit gain

The receive gain can be adjusted by adjusting the resistor

ratio R_RXOover R_GRX. However, R_RXOalso sets the

confidence tone level during dialing which leaves R_GRXto be

chosen freely. A high pass filter is introduced by the coupling

capacitor C_RXItogether with the input impedance of the input

A

_HMis fixed by the advised R_SLP= 220 Ω and the constraints

of set impedance and line impedance, the transmit gain is set

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RXI. A second high pass filtering is introduced by the

Automatic Gain Control

combination of C_GRXand R_GRX. A low pass filter is created

by C_RXOand R_RXO. The coupling capacitor at the output

RXO is not used for setting a high pass filter but merely for dc

decoupling.

In combination with dynamic ear capsules, the EAR

amplifier can become unstable due to the highly inductive

characteristic of some of the capsules. To regain stability, a

100 nF capacitor can be connected from RXS to Gnd in

those cases. An additional 10 nF at the RXI input, as shown

in the typical application, improves the noise figure of the

receiver stage.

To obtain more or less constant signal levels for transmit

and receive regardless of the telephone line length, both the

transmit and receive gain can be varied as a function of line

current when the AGC feature is used. The gain reduction as

a function of line current, and thus line length, is depicted in

Figure 8.

Figure 8. Automatic Gain Control

0

–1.0

R

= 20 k

AGC

–2.0

Sidetone Cancellation

R

= 30 k

AGC

The line driver and the receiver amplifier of the MC33215

are tied up in a bridge configuration as depicted in Figure 7.

This bridge configuration performs the so–called hybrid

function which, in the ideal case, prevents transmitted signals

from entering the receive channel.

–3.0

–4.0

–5.0

–6.0

Figure 7. Sidetone Bridge

0

10

20

30

40

(mA)

50

60

70

VLN

I

line

For small line currents, and thus long lines, no gain

reduction is applied and thus the transmit and receive gains

are at their maximum. For line currents higher than I_start, the

gain is gradually reduced until a line current I_stopis reached.

This should be the equivalent of a very short line, and the

gain reduction equals 6.0 dB. For higher line currents the

gain is not reduced further. For the start and stop currents the

following relations are valid:

Z

//Z

Z

bal

line set

V

x 15

11

HM

RXI

Receive

Transmit

R

Gnd

SLP

Gnd

R

/11

R

SLB

SLP

1

I

stop

As can be seen from Figure 7 by inspection, the receiver

will not pick up any transmit signal when the bridge is in

balance, that is to say when:

R

11

SLP

20 µ x R

AGC

11

1

Z

I

–

set

11

start

bal

line

R

11

R

SLP

R

SLB

SLP

For the typical application, where R_AGC= 30 kΩ, the gain

will start to be reduced at I_start= 20 mA while reaching 6.0 dB

of gain reduction at I_stop= 50 mA. When AGC is connected to

The sidetone suppression is normally measured in an

acoustic way. The signal at the earpiece when applying a

signal on the microphone is compared with the signal at the

earpiece when applying a signal on the line. The suppression

takes into account the transmit and receive gains set. In fact

the sidetone suppression can be given as a purely electrical

parameter given by the properties of the sidetone bridge

itself. For the MC33215, this so–called electrical sidetone

suppression A_STEcan be given as:

V

_DD, the AGC function is disabled leading to no gain

reduction for any line current. This is also sometimes called

PABX mode.

The automatic gain control takes effect in the HMIC and R_x

amplifiers as well as in the BMIC amplifier. In this way the

AGC is also active in speakerphone mode, see the handsfree

operation paragraph.

Z

R

11

bal

SLP

Privacy and DTMF Mode

During handset operation a privacy and a DTMF mode can

be entered according the logic Table 1.

A

1 –

x

STE

R

Z

set

SLB

line

Values of –12 dB or better, thus A_STE< 0.25, can easily be

reached in this way.

Table 1. Logic Table for Handset Mode

Logic Inputs

Amplifiers

DTMF

SPS

MUT

PRS

HMIC

On

BMIC

Off

R

RX

EAR

On

Mode

x

att

0

1

0

1

0

Handset Normal

Handset Privacy

Handset DTMF

Off

On

Off

On

X

Off

On

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Table 2. Logic Table for Handsfree Mode

Logic Inputs

Amplifiers

DTMF

SPS

MUT

PRS

HMIC

Off

BMIC

On

R

RX

EAR

Off

Mode

x

att

1

0

1

0

Handsfree Normal

Handsfree Privacy

Handsfree DTMF

Off

On

Off

On

Off

On

Off

X

Off

By applying a logic 0 to Pin MUT, the DTMF mode is

entered where the DTMF amplifier is enabled and where the

R_xamplifier is muted. A DTMF signal can be sent to the line

via the MFI input for which the gain A_DTMFis given as:

With the typical application and Z_line= 600 Ω the transmit

gain calculates as 55 dB.

The electret base microphone can be supplied directly

from VHF but it is advised to use an additional RC filter to

obtain a stable supply point, as shown in the typical

application. The microphone can also be supplied by VMC.

The transmit gain is set by adjusting the sensitivity of the

V

Z

x Z

set

line

MFI

3.75

SLP

line

A

x

DTMF

V

Z

R

11

set

line

In the typical application, the gain equals 35 dB. The

DTMF gain can be controlled by a resistor divider at the input

MFI as shown in the typical application. The signal has to be

capacitively coupled to the input via C_MFIwhich creates a

high pass filter with the input impedance. The line length

AGC has no effect on the DTMF gains.

base microphone by adjusting the resistors R_BM1and R_BM2

It is not advised to adjust the gain by including series

resistors towards the Pins BM1 and BM2.

.

A high pass filter is introduced by the coupling capacitors

C

_BM1and C_BM2in combination with the input impedance. A

low pass filter can be created by putting capacitors in parallel

The signal applied to the MFI input is made audible at the

earpiece output for confidence tone. The signal is internally

applied to the GRX pin where it is amplified via the EAR

amplifier which is used as a current to voltage amplifier. The

with the resistors R_BM1and R_BM2

.

Loudspeaker Amplifier

The loudspeaker amplifier of the MC33215 has three major

benefits over most of the existing speakerphone loudspeaker

amplifiers: it can be supplied and used in a telephone line

powered application but also stand alone, it has an all NPN

bootstrap output stage which provides maximum output

swing under any supply condition, and it includes a

peak–to–peak limiter to limit the distortion at the output.

The loudspeaker amplifier is powered at Pin VLS. In

telephone line powered applications, this pin should be

connected to V_CCwhere most of the line current is available,

see the V_CCsupply paragraph. In an application where an

external power supply is used, VLS and thus the loudspeaker

amplifier can be powered separately from the rest of the

circuit. The amplifier is grounded to PGD, which is the circuits

power ground shared by both the loudspeaker amplifier and

the current splitter of the V_CCsupply. Half the supply voltage

of VLS is at BVO, filtered with a capacitor to VLS. This

voltage is used as the reference for the output amplifier.

The receive signal present at RLS can be capacitively

coupled to LSI via the resistor R_LSI. The overall gain from

RLS to LSO follows as:

gain is therefore proportional to the feedback resistor R_RXO

.

For R_RXO= 180 kΩ the gain equals 6.0 dB. The confidence

tone is also audible at the loudspeaker output when the

loudspeaker amplifier is activated, see speakerphone

operation.

By applying a logic 0 to Pin PRS, the MC33215 enters

privacy mode. In this mode, both handset and handsfree

microphone amplifiers are muted while the DTMF amplifier is

enabled. Through the MFI input, a signal, for example music

on hold, can be sent to the line. In the same way, the MFI

input can also be used to couple in signals from, for instance,

an answering machine.

HANDSFREE OPERATION

Handsfree operation, including DTMF and Privacy modes,

can be performed by making Pin SPS high according Table 2.

The handset amplifiers will be switched off while the base

amplifiers will be activated. The MC33215 performs all the

necessary functions, such as signal monitoring and

switch–over, under supervision of the duplex controller.

With the MC33215 also a group listening–in application

can be built. For more information on this subject please refer

to application note AN1574.

V

R

LSO

RLS

LSF

A

–

x 4.0

LS

V

R

LSI

In the typical application this leads to a loudspeaker gain

_LSof 26 dB. The above formula follows from the fact that the

overall amplifier architecture from RLS to LSO can be looked

at as an inverting voltage amplifier with an internal current

gain from LSI to LSF of 4. The input LSI is a signal current

summing node which allows other signals to be applied here.

Base Microphone Amplifier

A

The base microphone can be capacitively connected to

the circuit via the differential input BM1 and BM2. The setup

is identical to the one for the handset microphone amplifier.

The total transmit voltage gain A_BMfrom the base

microphone inputs to the line is:

V

Z

x Z

set

line

BM

37.5

SLP

line

A

x

BM

V

Z

R

11

set

line

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Figure 9. Loudspeaker Output Stage

0.5 VLS

0

1.5 VLS

VLS

–0.5 VLS

0.5 VLS

Loudspeaker

LSB

VLS

C

LSO

T2

T1

LSO

PGD

VLS

0.5 VLS

0

Figure 10. Loudspeaker Amplifier Output Power with External Supply

140

120

100

80

300

R

= 25

= 50

Ω

LSP

250

R

= 25 Ω

LSP

200

150

100

60

40

20

0

LSP

R

= 50 Ω

LSP

50

0

2.0

4.0

6.0

2.0

3.0

5.0

7.0

3.0

5.0

7.0

8.0

4.0

6.0

8.0

VLS (V)

A. Peak–to–Peak Limiter Active

B. Peak–to–Peak Limiter Disabled

The total gain from the telephone line to the loudspeaker

output includes, besides the loudspeaker amplifier gain, also

the attenuation of the sidetone network and the internal gain

from RXI to RLS. When in receive mode, see under duplex

controller, the gain from RXI to RLS is maximum and equals

24 dB at maximum volume setting. The attenuation of the

sidetone network in the typical application equals 24.6 dB

which makes an overall gain from line to loudspeaker of

25.4 dB. Due to the influence of the line length AGC on the R_x

amplifier, the gain will be reduced for higher line currents.

The output stage of the MC33215 is a modified all NPN

bootstrap stage which ensures maximum output swing under

all supply conditions. The major advantage of this type of

output stage over a standard rail–to–rail output is the higher

stability. The principle of the bootstrap output stage is

explained with the aid of Figure 9.

T2 to be supplied for output signals with positive excursions

up to VLS without distorting the output signal. The resulting

ac signal over the loudspeaker will equal the signal at LSO.

As an indication of the high performance of this type of

amplifier, in Figure 10, the output power of the loudspeaker

amplifier as a function of supply voltage is depicted for 25 Ω

and 50 Ω loads with both the peak–to–peak limiter active and

disabled. As can be seen, in case the peak–to–peak limiter is

disabled, the output power is roughly increased with 6.0 dB,

this at the cost of increased distortion levels up to 30%.

In a telephone line powered application, the loudspeaker

amplifier output power is limited not only by the supply

voltage but also by the telephone line current. This means

that in telephones the use of 25 Ω or 50 Ω speakers is

preferred over the use of the cheaper 8.0 Ω types. Figure 11

gives the output power into the loudspeaker for a line

powered application and two different dc settings with the

peak–to–peak limiter active. In case the peak–to–peak limiter

is disabled the output power will be increased for the higher

line currents up to 6.0 dB.

The output LSO is biased at half the supply VLS while the

filtering of the loudspeaker with the big capacitor C_LSO

requires that LSB is biased at VLS. In fact, because of the

filtering, LSB is kept at VLS/2 above the LSO output even if

LSO contains an ac signal. This allows the output transistor

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Figure 11. Loudspeaker Amplifier Output

Power when Line Powered

loudspeaker amplifier is muted which is needed for correct

handset operation.

100

90

The volume of the loudspeaker signal can be varied via a

potentiometer at VOL. A fixed current of 10 µA is running

through the potentiometer and the resulting voltage at VOL

is a measure for the gain reduction. The relation between

the voltage at VOL and the obtained gain reduction is given

in Figure 13.

R

= 365 k

= 220 k

REG1

REG2

LSP

= 25

Ω

80

R

= 365 k

= 220 k

REG1

REG2

LSP

70

R

= 365 k

= Infinite

= 50

Ω

REG1

REG2

LSP

60

50

40

30

20

10

0

= 25

Ω

R

= 365 k

= Infinite

REG1

REG2

LSP

Figure 13. Volume Reduction

0

= 50

Ω

–5.0

–10

0

20

40

60

80

100

–15

–20

I

(mA)

line

The quality of the audio output of the loudspeaker amplifier

is mainly determined by the distortion level. To keep high

quality under difficult supply conditions, the MC33215

incorporates a peak–to–peak limiter. The peak–to–peak

limiter will detect when the output stage gets close to its

maximum output swing and will then reduce the gain from LSI

to LSF. The attack and release of the limiter is regulated by

the C_PPLcapacitor. Figure 12 depicts the limiter’s attack

behavior with C_PPL= 100 nF. The release time is given as

3 x C_PPLx R_PPL. In the typical application this leads to a

release time of 300 ms.

–25

–30

–35

–40

0

100

200

(mV), dA

300

(dB)

400

500

V

VOL

LSP

It can be seen from Figure 13 that a linear variation of

R_VOLwill give a logarithmic gain reduction which adapts

better to the human ear than a linear gain reduction.

During DTMF dialing, see Table 2, a confidence tone is

audible at the loudspeaker of which the level is proportional

to the feedback resistor R_LSFonly. At R_LSF= 180 kΩ the gain

from MFI to LSO equals 28.5 dB.

Figure 12. Peak–to–Peak Limiter Dynamic Behavior

Half Duplex Controller

V

LSO

To avoid howling during speakerphone operation, a half

duplex controller is incorporated. By monitoring the signals in

both the transmit and receive channel the duplex controller

will reduce the gain in the channel containing the smallest

signal. A typical gain reduction will be between 40 dB and

52 dB depending on the setting, see below. In case of equal

signal levels or by detection of noise only, the circuit goes into

idle mode. In this mode the gain reduction in both channels is

halfway, leading to 20 dB to 26 dB of reduction.

V

PPL

V

in

0

2.0

5.0

In a speakerphone built around the MC33215, following

the signal path from base microphone to the line and via

sidetone, loudspeaker and acoustic coupling back to the

microphone, the loop gain can be expressed as a sum of the

gains of the different stages. However, since the transmit and

receive gains are dependent on AGC and the sidetone

suppression is dependent on matching with the different lines

we are mostly interested by the maximum possible loop gain

1.0

3.0

4.0

6.0

t, TIME (ms)

Figure 12 clearly shows that due to the action of the

peak–to–peak limiter, the output swing and thus the output

power is reduced with respect to the maximum possible as

already indicated in Figure 10. The peak–to–peak limiter can

be disabled by connecting the PPL pin to ground.

On top of the peak–to–peak limiter, the MC33215

incorporates a supply limiter, which reduces the gain rapidly

when the supply voltage VLS drops too much. This will

avoid malfunctioning of the amplifier and unwanted

oscillations. The voltage drop is detected via the BVO input

and for that reason the C_BVOhas to be connected to VLS

and not to Gnd.

The amplifier can be activated by making Pin LSM high. In

the typical application this pin is connected to SPS, which

activates the loudspeaker amplifier automatically when the

speakerphone mode is entered. When LSM is made low, the

A

_LOOP(max). It follows:

A

_LOOP(max)= A_BMRX(max)+ A_RXBM(max)– A_SWD(dB)

With: A_BMRX(max)= Maximum gain from BM1 and BM2 to

RXI as a function of line length AGC and line

impedance matching

A_RXBM(max)= Maximum gain from RXI to BM1 and

BM2 as a function of line length AGC and acoustic

coupling

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A_SWD= Switching depth as performed in the

mode, due to the coupling of the high loudspeaker signal, is

attenuators

automatically taken into account.

In the table, two particulars can be found. At first, the set

will go to idle mode if the signals are not at least 4.5 dB

greater then the noise floor, which calculates as a 13 mV

voltage difference in envelopes. This avoids continuous

switching over between the modes under slight variations of

the background noise due to, for instance, typing on a

keyboard. Second, a dial tone detector threshold is

implemented to avoid that the set goes to idle mode in

presence of a continuous strong receive signal like a dial

tone. The dial tone detector threshold is proportional to the

R_RSAresistor. In the typical application with R_RSA= 3.3 kΩ,

the threshold is at 1.26 mVrms at the input RXI or 20 mVrms

at the line. Line length AGC is of influence on the dial tone

detector threshold, increasing the level depending on the line

current with a maximum of 6.0 dB.

To avoid howling, the maximum possible loop gain should

be below 0 dB and preferably below –10 dB for comfort. In a

practical telephone design, both the A_BMRX(max)and the

A

_RXBM(max)will be less than 20 dB thus a switching depth of

50 dB will give a loop gain of less than –10 dB. An optimized

sidetone network is of high importance for handsfree

operation. The better the network matches with the

telephone line the less local feedback and the smaller the

switching range can be.

The amount of gain reduction A_SWDobtained by the

duplex controller is set via resistor R_SWDaccording:

2

3.6 x R

SWD

A

20 log

(dB)

SWD

R

REF

In order to perform a correct comparison between the

signal strengths, the sensitivity of the envelope detectors can

be adjusted via the resistors connected to TSA and RSA.

Based on the above, and on the fact that there is an effective

gain of 20 dB in the transmit monitor, it can be derived that for

stable operation the following two relations are valid:

In the typical application the gain reduction will be 50 dB.

To compare the transmit and receive signals with each

other, they have to be monitored. This is done by making a

signal envelope and a background noise envelope via the

_TSE, C_TBNcapacitors for the transmit channel and via the

_RSE, C_RBNcapacitors for the receive channel. In Figure 14,

C

20 log R

– A

20 (dB)

a schematic behavior of the envelopes is depicted which is

equal for both transmit and receive.

TSA

20 log R

RSA

20 log R

BMRX(max)

– A

The voltage signal at the input is first transferred to a

current via the sensitivity adjust network. Then this current is

led through a diode which gives a logarithmic compression in

voltage. It is this voltage from which the signal envelope is

created by means of asymmetric charge and discharge of the

signal envelope capacitor. The noise envelope voltage then

follows in a similar way. Based on the envelope levels, the

MC33215 will switch to transmit, receive or idle mode

following Table 3. The fact that in receive mode the signal on

the base microphone is greater than it is in case of transmit

TSA

RSA

RXBM(max)

– A

20 (dB)

SW

By measuring the gains and choosing the R_RSA, the limits

for R_TSAfollow. The choice for the sensitivity resistors is not

completely free. The logarithmic compressors and the

amplifier stages have a certain range of operation and, on the

receive side, the choice for R_RSAis given by the desired dial

tone detector threshold. Figure 15 indicates the available

dynamic range for the selected value of the sensitivity

resistors.

Figure 14. Signal and Noise Envelopes

1.8 V

Internal

VHF

C

TSE

C

TSE

TBN

Microphone

Input Signal

TSA

R

TSA

C

TSA

15

MOTOROLA ANALOG IC DEVICE DATA

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MC33215

Table 3. Logic Table for Switch–Over

TSE > RSE

TSE > TBN + 13 mV

RSE > V

RSE > RBN + 13 mV

Mode

DDT

1

0

1

0

X

1

0

X

1

Transmit

Idle

X

Receive

Idle

0

The resistors for the sensitivity setting have to be coupled

capacitively to the pins for dc decoupling, and also to create

a high pass filter to suppress low frequent background noises

like footsteps and 50 Hz.

The switch–over timing is performed by charging and

discharging the C_SWTcapacitor. The switch–over from

transmit to receive or vice versa is fast, on the order of

milliseconds, and is proportional to the value of C_SWT. The

switch–over to idle mode is slow, in the order of a few

seconds, and is proportional to the product of the values of

R_SWTand C_SWT. Figure 16 depicts a typical switch–over

behavior when applying transmit and receive stimuli.

The electrical characteristics and the behavior of the

MC33215 are not the only factor in designing a handsfree

speakerphone. During the design the acoustics have to be

taken into account from the beginning. The choice of the

transducers and the design of the cabinet are of great

influence on the speakerphone performance. Also, to

achieve a proper handsfree operation, the fine tuning of the

components around the duplex controller have to be done

with the final choice of the cabinet and the transducers.

Figure 15. Compression Range of the Signal Monitors

100.0E–3

10.0E–3

1.0E–3

100.0E–3

Upper Limit of

Compression

Upper Limit of

Compression

Dial Tone

Threshold

10.0E–3

Lower Limit of

Compression

Lower Limit of

Compression

1.0E–3

100.0E–6

10.0E–6

100.0E–6

1000

10000

100000

1000

10000

100000

100

R

(Ω)

R

(Ω)

RSA

TSA

A. Receive Monitor

B. Transmit Monitor

Figure 16. Switch–Over Behavior

Receive

Transmit

VMC + 0.5

VMC – 0.5

SWT

16

MOTOROLA ANALOG IC DEVICE DATA

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MC33215

Figure 17. Test Circuit

Z

VDD

620

Z

33 k

R

360 k

bal

REG

600

Vac

C

100

C

REG

220 n

VDD

µ

V

line

DD

I

line

R

2.2 k

SLB

Gnd

V

VLN

REG

DD

VMC

VHF

C

VMC

10

µ

Supply

1:10

Supply

MC33215

VHF

V

47

µ

CC

C

HM1

MHM

VCC

µ

33 n

HM1

HM2

470

Driver

SLB

SLP

1x

HMIC

BMIC

V

HM

R

SLP

0.2 V

MDF

220

C

HM2

AGC

MBM

33 n

C

BM1

33 n

BM1

BM2

C

MF1

47 n

MFI

C

BM2

V

T Attenuator

BM

x

DTMF

33 n

V

MF

SPS

MUT

PRS

LSM

AGC

C

470 n

R

TSA

2.2 k

TSA

V

SPS

TSA

TSE

MHM

MBM

MDF

MRX

MRA

T

Log–Amp

and

Envelope

Detectors

x

V

MUT

Logic

Control

Block

C

TSE

VHF

330 n

V

PRS

TBN

C

TBN

4.7

AGC

30 k

µ

V

M

LSM

EAR

Attenuator

Control

R

SWT

RBN

AGC

REF

C

4.7

RBN

µ

R

V

REF

SWT

20 k

AGC

R

Log–Amp

x

C

R

RSE

VHF

SWD

100 k

Analog

Control

Block

and Envelope

Detectors

330 n

SWD

VOL

RSE

RSA

C

RSA

470 n

R

3.3 k

RSA

R

47 k

MRX

VOL

RLS

2

RXI

R Attenuator

x

R

x

S1

C

47 n

1

RXI

V

RLS

BVO

VLS

V

RXI

AGC

C

BVO

MRA

220 n

M

EAR

VLS

V

LSP

25

C

EAR

V

LSB

LSO

10

µ

RXO

GRX

C

47

EAR

LSO

µ

LSP

Peak

Limiter

R

V

RXO

180 k

V

EAR

PGD

R

LSO

180 k

R

24 k

GRX

LSF

PPL

LSI

RXS

C

R

LSI

C

100 n

C

47 n

PPL

1.0 M

PPL

RXS

GRX

36 k

100 n

C

LSI

47 n

V

LSI

17

MOTOROLA ANALOG IC DEVICE DATA

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MC33215

Figure 18. Typical Application

T1

R

REG2

Z

R

365 k

VDD

620

REG1

Z1

10 V

Z

bal

33 k

0.01

Hook

Switch

C

VDD

100

C

V

REG

220 n

DD

µ

R

2.2 k

SLB

Gnd

V

REG

VLN

DD

VMC

VHF

T2

C

10

VMC

µ

Supply

1:10

Supply

V

DD

MC33215

C

VHF

VMC

47

µ

V

CC

C

R

HM1

33 n

C

470

HM1

1.0 k

MHM

VCC

µ

HM1

HM2

Driver

SLB

SLP

1x

C

HM2

HMIC

33 n

R

220

SLP

0.2 V

R

HM2

1.0 k

VHF

AGC

MBM

Tip

C

BM1

33 n

R

1.0 k

BM1

1.0 k

MDF

BM1

BM2

C

MF1

47 n

MFI

C

BM2

33 n

T

Attenuator

10 µF

BMIC

x

DTMF

Ring

Dialer or

Microcontroller

R

1

4

7

*

2

5

8

0

3

6

9

#

BM2

1.0 k

SPS

MUT

AGC

R

TSA

470

TSA

TSE

MHM

MBM

C

TSA

1.0

Privacy

Button

Speakerphone

Button

T

Log–Amp

µ

F

x

and Envelope

Detectors

MDF

MRX

MRA

PRS

LSM

Logic Control

Block

C

TSE

VHF

330 n

TBN

C

TBN

4.7

C

VMC

SWT

M

µ

EAR

100 n

SWT

Attenuator

Control

R

AGC

30 k

AGC

REF

R

C

4.7

SWT

REF

20 k

RBN

µ

2.2 M

RBN

AGC

C

330 n

R

100 k

RSE

SWD

Analog

Control

Block

R

Log–Amp

x

VHF

SWD

VOL

RSE

RSA

and Envelope

Detectors

C

RSA

470 n

R

50 k

R

3.3 k

VOL

RSA

MRX

RLS

RXI

R

Attenuator

MRA

R

x

C

RXI

33 n

10 n

BVO

VLS

AGC

C

BVO

220 n

V

CC

M

EAR

C

EAR

10

LSB

LSO

µ

RXO

GRX

25

Ω

C

R

LSO

LSP

EAR

47

µ

R

Peak Limiter

RXO

180 k

150 Ω

PGD

LSF

LSO

180 k

R

24 k

GRX

PPL

LSI

RXS

C

47 n

R

C

100 n

RXS

GRX

PPL

1.0 M

PPL

100 n

C

R

RLS

33 n

LSI

36 k

18

MOTOROLA ANALOG IC DEVICE DATA

For More Information On This Product,

Go to: www.freescale.com

Freescale Semiconductor, Inc.

MC33215

OUTLINE DIMENSIONS

FB SUFFIX

PLASTIC PACKAGE

CASE 848B–04

(TQFP–52)

ISSUE C

B

L

39

27

26

40

–A–, –B–, –D–

DETAIL A

–B–

–A–

L

F

J

N

14

13

52

1

BASE METAL

D

–D–

M

S

B

0.02 (0.008)

C

A–B

D

M

S

0.20 (0.008)

H

A–B

D

SECTION B–B

0.05 (0.002) A–B

NOTES:

1. DIMENSIONING AND TOLERANCING PER ANSI

Y14.5M, 1982.

V

M

S

0.20 (0.008)

C

A–B

2. CONTROLLING DIMENSION: MILLIMETER.

3. DATUM PLANE –H– IS LOCATED AT BOTTOM OF

LEAD AND IS COINCIDENT WITH THE LEAD WHERE

THE LEAD EXITS THE PLASTIC BODY AT THE

BOTTOM OF THE PARTING LINE.

4. DATUMS –A–, –B– AND –D– TO BE DETERMINED AT

DATUM PLANE –H–.

5. DIMENSIONS S AND V TO BE DETERMINED AT

SEATING PLANE –C–.

DETAIL C

M

C

E

6. DIMENSIONS A AND B DO NOT INCLUDE MOLD

PROTRUSION. ALLOWABLE PROTRUSION IS 0.25

(0.010) PER SIDE. DIMENSIONS A AND B DO

INCLUDE MOLD MISMATCH AND ARE DETERMINED

AT DATUM PLANE –H–.

7. DIMENSION D DOES NOT INCLUDE DAMBAR

PROTRUSION. ALLOWABLE DAMBAR PROTRUSION

SHALL BE 0.08 (0.003) TOTAL IN EXCESS OF THE D

DIMENSION AT MAXIMUM MATERIAL CONDITION.

DAMBAR CANNOT BE LOCATED ON THE LOWER

RADIUS OR THE FOOT.

DATUM

–H–

PLANE

0.10 (0.004)

H

SEATING

PLANE

–C–

M

G

MILLIMETERS

INCHES

DIM

A

B

C

D

E

MIN

9.90

2.10

0.22

2.00

0.22

MAX

10.10

2.45

0.38

2.10

MIN

MAX

0.398

0.096

0.015

0.083

0.013

0.390

0.083

0.009

0.079

0.009

U

F

0.33

G

H

J

K

L

0.65 BSC

0.026 BSC

–––

0.13

0.65

0.25

0.23

0.95

–––

0.005

0.026

0.010

0.009

0.037

R

Q

7.80 REF

0.307 REF

M

N

Q

R

S

T

U

V

5

0.13

0

0.13

12.95

0.13

0

12.95

0.35

1.6 REF

10

0.17

7

0.30

13.45

–––

13.45

0.45

5

0.005

0

0.005

0.510

0.005

0

0.510

0.014

0.063 REF

10

0.007

7

0.012

0.530

–––

0.530

0.018

K

T

W

X

DETAIL C

W

X

19

MOTOROLA ANALOG IC DEVICE DATA

For More Information On This Product,

Go to: www.freescale.com

Freescale Semiconductor, Inc.

MC33215

OUTLINE DIMENSIONS

B SUFFIX

PLASTIC PACKAGE

CASE 858–01

(SDIP–42)

ISSUE O

NOTES:

–A–

1. DIMENSIONING AND TOLERANCING PER ANSI

Y14.5M, 1982.

2. CONTROLLING DIMENSION: INCH.

3. DIMENSION L TO CENTER OF LEAD WHEN

FORMED PARALLEL.

4. DIMENSIONS A AND B DO NOT INCLUDE MOLD

FLASH. MAXIMUM MOLD FLASH 0.25 (0.010).

42

22

21

–B–

INCHES

MILLIMETERS

1

DIM

A

B

C

D

F

MIN

MAX

1.465

0.560

0.200

0.022

0.046

MIN

36.45

13.72

3.94

0.36

0.81

MAX

37.21

14.22

5.08

0.56

1.17

L

1.435

0.540

0.155

0.014

0.032

H

C

K

G

H

J

K

L

0.070 BSC

0.300 BSC

1.778 BSC

7.62 BSC

0.008

0.115

0.015

0.135

0.20

2.92

0.38

3.43

–T–

SEATING

PLANE

0.600 BSC

15.24 BSC

N

G

M

N

0

15

0.040

0

0.51

15

1.02

M

F

0.020

J 42 PL

0.25 (0.010)

D 42 PL

M

S

T

B

M

S

0.25 (0.010)

T A

Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regarding

the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and

specificallydisclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters which may be provided in Motorola

datasheetsand/orspecificationscananddovaryindifferentapplicationsandactualperformancemayvaryovertime. Alloperatingparameters,including“Typicals”

must be validated for each customer application by customer’s technical experts. Motorola does not convey any license under its patent rights nor the rights of

others. Motorola products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other

applicationsintended to support or sustain life, or for any other application in which the failure of the Motorola product could create a situation where personal injury

ordeathmayoccur. ShouldBuyerpurchaseoruseMotorolaproductsforanysuchunintendedorunauthorizedapplication,BuyershallindemnifyandholdMotorola

and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees

arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that

Motorola was negligent regarding the design or manufacture of the part. Motorola and

Opportunity/Affirmative Action Employer.

are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal

Mfax is a trademark of Motorola, Inc.

How to reach us:

USA/EUROPE/Locations Not Listed: Motorola Literature Distribution;

P.O. Box 5405, Denver, Colorado 80217. 303–675–2140 or 1–800–441–2447

JAPAN: Nippon Motorola Ltd.; Tatsumi–SPD–JLDC, 6F Seibu–Butsuryu–Center,

3–14–2 Tatsumi Koto–Ku, Tokyo 135, Japan. 81–3–3521–8315

Mfax : RMFAX0@email.sps.mot.com – TOUCHTONE 602–244–6609

ASIA/PACIFIC: Motorola Semiconductors H.K. Ltd.; 8B Tai Ping Industrial Park,

– US & Canada ONLY 1–800–774–1848 51 Ting Kok Road, Tai Po, N.T., Hong Kong. 852–26629298

INTERNET: http://www.mot.com/SPS/

MC33215/D

◊

20

MOTOROLA ANALOG IC DEVICE DATA

For More Information On This Product,

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型号：	MC33215
厂家：	Freescale
描述：	Telephone Line Interface and Speakerphone Circuit 电话线接口和扬声器电路电话
文件：	总20页 (文件大小：490K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

MC33215 [FREESCALE]

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