IR3832WMTR1PBF [INFINEON]
HIGHLY EFFICIENT INTEGRATED SYNCHRONOUS BUCK REGULATOR FOR DDR APPLICATIONS; 高效集成同步降压稳压器，用于DDR应用
| 型号： | IR3832WMTR1PBF |
| 厂家： | 
Infineon |
| 描述： | HIGHLY EFFICIENT INTEGRATED SYNCHRONOUS BUCK REGULATOR FOR DDR APPLICATIONS |
| 文件： | 总32页 (文件大小：716K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
PD-97508
IR3832WMPbF
TM
SupIRBuck
HIGHLY EFFICIENT INTEGRATED
SYNCHRONOUS BUCK REGULATOR FOR DDR APPLICATIONS
Features
Description
•
•
•
•
•
Wide Input Voltage Range 1.5V to 16V
The IR3832W SupIRBuckTM is an easy-to-use,
fully integrated and highly efficient DC/DC
regulator. The MOSFETS co-packaged with the
on-chip PWM controller make IR3832W a space-
efficient solution, providing accurate power
delivery for DDR memory applications.
Wide Output Voltage Range 0.6V to 0.9*Vin
Continuous 4A Load Capability
Integrated Bootstrap-diode
High Bandwidth E/A for excellent transient
performance
•
•
•
•
•
•
•
•
•
•
•
•
Programmable Switching Frequency up to 1.5 MHz
Programmable Over Current Protection
PGood output
IR3832W is configured to generate termination
voltage (VTT) for DDR memory applications.
Hiccup Current Limit
IR3832W offers programmability of start up time,
switching frequency and current limit while
operating in wide input and output voltage range.
Programmable Soft-Start
Enable Input with Voltage Monitoring Capability
Enhanced Pre-Bias Start-up
The switching frequency is programmable from
250kHz to 1.5MHz for an optimum solution.
Vp input for DDR Tracking applications
-40oC to 125oC operating junction temperature
Thermal Protection
It also features important protection functions,
such as Pre-Bias startup, hiccup current limit and
thermal shutdown to give required system level
security in the event of fault conditions.
5mm x 6mm Power QFN Package, 0.9 mm height
Halogen Free, Lead Free and RoHS Compliant
Applications
•
•
•
Server Applications
•
•
Distributed Point of Load Power Architectures
Netcom Applications
Storage Applications
Embedded Telecom Systems
1.5V <Vin<16V
4.5V <Vcc<5.5V
Enable
Vin
Boot
Vo
Vcc
VDDQ
SW
PGood
PGood
OCSet
Fb
Vp
Rt
Comp
PGnd
SS/ SD
Gnd
Fig. 1. Typical application diagram
1
Rev 3.0
PD-97508
IR3832WMPbF
ABSOLUTE MAXIMUM RATINGS
(Voltages referenced to GND unless otherwise specified)
•
•
•
•
•
•
•
•
•
•
•
•
Vin ……………………………………………………. -0.3V to 25V
Vcc ……………….….…………….……..……….…… -0.3V to 8V (Note2)
Boot
SW
……………………………………..……….…. -0.3V to 33V
…………………………………………..……… -0.3V to 25V(DC), -4V to 25V(AC, 100ns)
Boot to SW ……..…………………………….…..….. -0.3V to Vcc+0.3V (Note1)
OCSet ………………………………………….……. -0.3V to 30V
Input / output Pins ……………………………….. ... -0.3V to Vcc+0.3V (Note1)
PGND to GND ……………...………………………….. -0.3V to +0.3V
Storage Temperature Range ................................... -55°C To 150°C
Junction Temperature Range ................................... -40°C To 150°C (Note2)
ESD Classification …………………………… ……… JEDEC Class 1C
Moisture sensitivity level………………...………………JEDEC Level 3@260 °C
Note1: Must not exceed 8V
Note2: Vcc must not exceed 7.5V for Junction Temperature between -10oC and -40oC
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the
device. These are stress ratings only and functional operation of the device at these or any other
conditions beyond those indicated in the operational sections of the specifications are not implied.
PACKAGE INFORMATION
5mm x 6mm POWER QFN
SW
VIN
11
PGnd
12
10
θJA = 35oC / W
θJ-PCB = 2oC / W
15
Gnd
13
VCC
Boot
9
8
14
PGood
Enable
1
4
7
2
3
5
6
Vp FB COMP Gnd Rt
SS OCSet
ORDERING INFORMATION
PACKAGE
DESIGNATOR
PACKAGE
PIN COUNT
PARTS PER
REEL
DESCRIPTION
M
M
IR3832WMTRPbF
IR3832WMTR1PbF
15
15
4000
750
2
Rev 3.0
PD-97508
IR3832WMPbF
Block Diagram
Fig. 2. Simplified block diagram of the IR3832W
3
Rev 3.0
PD-97508
IR3832WMPbF
Pin Description
Pin Name
Description
Track pin. Use External resistors from VDDQ rail. The Vp voltage can
be set to 0.9V for DDR2 application and 0.75 or 0.6V for DDR3
application.
Inverting input to the error amplifier. This pin is connected directly to the
output of the regulator via resistor divider to set the output voltage and
provide feedback to the error amplifier.
Output of error amplifier. An external resistor and capacitor network is
typically connected from this pin to Fb pin to provide loop
compensation.
1
2
3
Vp
Fb
Comp
4
5
Gnd
Rt
Signal ground for internal reference and control circuitry.
Set the switching frequency. Connect an external resistor from this pin
to Gnd to set the switching frequency.
Soft start / shutdown. This pin provides user programmable soft-start
function. Connect an external capacitor from this pin to Gnd to set the
start up time of the output voltage. The converter can be shutdown by
pulling this pin below 0.3V.
6
SS/S¯D¯
Current limit set point. A resistor from this pin to SW pin will set the
current limit threshold.
7
8
OCSet
PGood
VCC
Power Good status pin. Output is open drain. Connect a pull up resistor
from this pin to Vcc. If unused, it can be left open.
This pin powers the internal IC and drivers. A minimum of 1uF high
frequency capacitor must be connected from this pin to the power
ground (PGnd).
9
Power Ground. This pin serves as a separated ground for the MOSFET
drivers and should be connected to the system’s power ground plane.
10
11
PGnd
SW
Switch node. This pin is connected to the output inductor.
Input voltage connection pin.
VIN
12
Supply voltage for high side driver. Connect a 0.1uF capacitor from this
pin to SW.
13
14
Boot
Enable
Enable pin to turn on and off the device.
15
Gnd
Signal ground for internal reference and control circuitry.
4
Rev 3.0
PD-97508
IR3832WMPbF
Recommended Operating Conditions
Symbol
Definition
Min
Max
Units
Vin
Vcc
Boot to SW
Vo
Io
Input Voltage
Supply Voltage
Supply Voltage
Output Voltage
1.5
4.5
4.5
0.6
0
16
5.5
5.5
V
0.90*Vin
4
Output Current
A
Fs
Tj
Switching Frequency
Junction Temperature
225
-40
1650
125
kHz
oC
Electrical Specifications
Unless otherwise specified, these specification apply over 4.5V< Vcc<5.5V, Vp=0.6V, Vin=12V,
0oC<Tj< 125oC. Typical values are specified at Ta = 25oC.
Parameter
Power Loss
Symbol
Test Condition
Min
TYP
MAX
Units
Power Loss
Ploss
Vcc=5V, Vin=12V, Vo=0.75V, Io=4A,
0.51
W
Fs=400kHz, L=1.5uH, Note4
MOSFET Rds(on)
Top Switch
Rds(on)_Top
Rds(on)_Bot
VBoot -Vsw =5V, ID=10A, Tj=25oC
Vcc=5V, ID=10A, Tj=25oC
22.6
15.1
29
20
mΩ
Bottom Switch
Supply Current
VCC Supply Current (Standby)
ICC(Standby)
ICC(Dyn)
SS=0V, No Switching, Enable low
500
μA
Vcc Supply Current (Dyn)
SS=3V, Vcc=5V, Fs=500kHz
Enable high
10
mA
Under Voltage Lockout
VCC-Start-Threshold
VCC_UVLO_Start
VCC_UVLO_Stop
Vcc Rising Trip Level
Vcc Falling Trip Level
3.95
3.65
4.15
3.85
4.35
4.05
VCC-Stop-Threshold
Enable-Start-Threshold
Enable_UVLO_Start
Supply ramping up
1.14
0.9
1.2
1.0
1.36
V
Enable-Stop-Threshold
Enable leakage current
Enable_UVLO_Stop
Ien
Supply ramping down
Enable=3.3V
1.06
15
μA
5
Rev 3.0
PD-97508
IR3832WMPbF
Electrical Specifications (continued)
Unless otherwise specified, these specification apply over 4.5V< Vcc<5.5V, Vp=0.6V, Vin=12V,
0oC<Tj< 125oC. Typical values are specified at Ta = 25oC.
Parameter
Oscillator
Symbol
Test Condition
Min
TYP
MAX
Units
Rt Voltage
0.625
225
0.7
0.775
275
V
Frequency
FS
Rt=59K
250
500
Rt=28.7K
450
550
kHz
Rt=9.31K, Note4
1350
1500
1.8
1650
Ramp Amplitude
Ramp Offset
Vramp
Note4
Vp-p
V
Ramp (os)
Dmin(ctrl)
Note4
0.6
50
Min Pulse Width
Fixed Off Time
Max Duty Cycle
Note4
ns
%
Note4
130
200
+5
Dmax
Fs=250kHz
92
-5
Error Amplifier
Input Offset Voltage
Vos
Vfb-Vp
0
mV
Vp=0.6V
Input Bias Current
Input Bias Current
Sink Current
IFb(E/A)
IVp(E/A)
Isink(E/A)
Isource(E/A)
SR
-1
-1
+1
+1
μA
0.40
8
0.85
10
12
mA
Source Current
Slew Rate
13
Note4
7
12
20
V/μs
MHz
dB
Gain-Bandwidth Product
DC Gain
GBWP
Note4
20
100
3.4
30
40
Gain
Note4
110
3.5
120
120
3.75
220
1
Maximum Voltage
Minimum Voltage
Common Mode Voltage
Vmax(E/A)
Vmin(E/A)
Vcc=4.5V
V
mV
V
Note4
0
Soft Start/SD
Soft Start Current
ISS
Vss(clamp)
SD
Source
14
20
26
3.3
0.3
μA
Soft Start Clamp Voltage
2.7
3.0
V
Shutdown Output Threshold
Over Current Protection
OCSET Current
IOCSET
Fs=250kHz
Fs=500kHz
20.8
43
23.6
48.8
26.4
54.6
μA
Fs=1500kHz
136
-10
154
0
172
+10
OC Comp Offset Voltage
SS off time
VOFFSET
Note4
mV
SS_Hiccup
4096
Cycles
Bootstrap Diode
Forward Voltage
I(Boot)=30mA
180
5
260
470
30
mV
ns
Deadband
Deadband time
Note4
10
6
Rev 3.0
PD-97508
IR3832WMPbF
Electrical Specifications (continued)
Unless otherwise specified, these specification apply over 4.5V< Vcc<5.5V, Vp=0.6V, Vin=12V,
0oC<Tj< 125oC. Typical values are specified at Ta = 25oC.
Parameter
SYM
Test Condition
Min
TYP
MAX
Units
Thermal Shutdown
Thermal Shutdown
Note4
Note4
140
20
oC
Hysteresis
Power Good
Power Good upper
Threshold
Upper Threshold
Delay
Power Good lower
Threshold
Lower Threshold
Delay
Delay Comparator
Threshold
VPG(upper)
VPG(upper)_Dly
VPG(lower)
Fb Rising
Fb Rising
Fb Falling
Fb Falling
0.660
0.480
0.690
256/Fs
0.510
256/Fs
2.1
0.720
0.540
V
S
V
VPG(lower)_Dly
PG(Delay)
s
Relative to charge voltage, SS rising
2
2.3
V
Delay Comparator
Hysteresis
Delay(hys)
Note4
260
300
340
mV
PGood Voltage Low
PG(voltage)
Ileakage
IPGood=-5mA
0.5
10
V
Leakage Current
0
μA
Switch Node
SW Bias Current
SW=0V, Enable=0V
Isw
μA
6
SW=0V,Enable=high,SS=3V,Vp=0V,
Note4
Note3: Cold temperature performance is guaranteed via correlation using statistical quality control. Not tested in production.
Note4: Guaranteed by Design but not tested in production.
7
Rev 3.0
PD-97508
IR3832WMPbF
TYPICAL OPERATING CHARACTERISTICS (-40oC - 125oC) Fs=500 kHz
Icc(Standby)
Icc(Dyn)
10.5
10.4
10.3
10.2
10.1
10.0
9.9
290
270
250
230
210
190
170
150
9.8
9.7
9.6
9.5
-40
-20
-20
-20
0
0
0
20
40
60
80
80
80
100
100
100
120
120
120
-40
-20
0
20
40
60
80
100
120
Temp[ oC]
Temp[ oC]
IOCSET(500kHz)
FREQUENCY
550
540
530
520
510
500
490
480
470
460
450
54.0
53.0
52.0
51.0
50.0
49.0
48.0
47.0
46.0
45.0
44.0
43.0
-40
-20
0
20
40
60
80
100
120
-40
20
40
60
Temp[ oC]
Temp[ oC]
Vcc(UVLO) Stop
Vcc(UVLO) Start
4.46
4.41
4.36
4.31
4.26
4.21
4.16
4.11
4.06
4.16
4.11
4.06
4.01
3.96
3.91
3.86
3.81
3.76
-40
20
40
60
-40
-20
0
20
40
60
80
100
120
Temp[ oC]
Temp[ oC]
Enable(UVLO) Start
Enable(UVLO) Stop
1.36
1.34
1.32
1.30
1.28
1.26
1.24
1.22
1.20
1.18
1.16
1.14
1.06
1.04
1.02
1.00
0.98
0.96
0.94
0.92
0.90
-40
-20
0
20
40
60
80
100
120
-40
-20
0
20
40
60
80
100
120
Temp[ οC]
Temp[ oC]
ISS
26.0
24.0
22.0
20.0
18.0
16.0
14.0
-40
-20
0
20
40
60
80
100
120
Temp[ oC]
8
Rev 3.0
PD-97508
IR3832WMPbF
Rdson of MOSFETs Over Temperature at Vcc=5V
30
28
26
24
22
20
18
16
14
12
-40
-20
0
20
40
60
80
100
120
140
Temperature [°C]
Sync-FET
Ctrl-FET
9
Rev 3.0
PD-97508
IR3832WMPbF
Typical Efficiency and Power Loss Curves
Vin=12V, Vcc=5V, Vo=0.75V, Io=0.5A-4A, Fs=400kHz, L=1.5uH (MPO104-1R5 from Delta), Room
Temperature, No Air Flow
89
88
87
86
85
84
83
82
81
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
Load Current (A)
0.55
0.50
0.45
0.40
0.35
0.30
0.25
0.20
0.15
0.10
0.05
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
Load Current (A)
10
Rev 3.0
PD-97508
IR3832WMPbF
Typical Efficiency and Power Loss Curves
Vin=5V, Vcc=5V, Vo=0.75V, Io=0.5A-4A, Fs=400kHz, L=1.5uH (MPO104-1R5 from Delta), Room
Temperature, No Air Flow
90
89
88
87
86
85
84
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
Load Current (A)
0.50
0.45
0.40
0.35
0.30
0.25
0.20
0.15
0.10
0.05
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
Load Current (A)
11
Rev 3.0
PD-97508
IR3832WMPbF
Circuit Description
THEORY OF OPERATION
If the input to the Enable pin is derived from the
bus voltage by a suitably programmed resistive
divider, it can be ensured that the IR3832W does
not turn on until the bus voltage reaches the
desired level. Only after the bus voltage reaches
or exceeds this level will the voltage at Enable
pin exceed its threshold, thus enabling the
IR3832W. Therefore, in addition to being a logic
input pin to enable the IR3832W, the Enable
feature, with its precise threshold, also allows the
user to implement an Under-Voltage Lockout for
the bus voltage Vin. This is desirable particularly
for high output voltage applications, where we
might want the IR3832W to be disabled at least
until Vin exceeds the desired output voltage level.
Introduction
The IR3832W uses a PWM voltage mode control
scheme with external compensation to provide
good noise immunity and maximum flexibility in
selecting inductor values and capacitor types.
The switching frequency is programmable from
250kHz to 1.5MHz and provides the capability of
optimizing the design in terms of size and
performance.
IR3832W provides precisely regulated output
voltage programmed via two external resistors
from 0.7V to 0.9*Vin.
The IR3832W operates with an external bias
supply from 4.5V to 5.5V, allowing an extended
operating input voltage range from 1.5V to 16V.
The device utilizes the on-resistance of the low
side MOSFET as current sense element, this
method enhances the converter’s efficiency and
reduces cost by eliminating the need for external
current sense resistor.
IR3832W includes two low Rds(on) MOSFETs
using IR’s HEXFET technology. These are
specifically designed for high efficiency
applications.
Fig. 3a. Normal Start up, Device turns on
when the Bus voltage reaches 10.2V
Figure 3b. shows the recommended start-up
sequence for the non-tracking operation of
IR3832W, when Enable is used as a logic input.
Under-Voltage Lockout and POR
The under-voltage lockout circuit monitors the
input supply Vcc and the Enable input. It assures
that the MOSFET driver outputs remain in the off
state whenever either of these two signals drop
below the set thresholds. Normal operation
resumes once Vcc and Enable rise above their
thresholds.
The POR (Power On Ready) signal is generated
when all these signals reach the valid logic level
(see system block diagram). When the POR is
asserted the soft start sequence starts (see soft
start section).
Enable
The Enable features another level of flexibility for
start up. The Enable has precise threshold which
is internally monitored by Under-Voltage Lockout
(UVLO) circuit. Therefore, the IR3832W will turn
on only when the voltage at the Enable pin
exceeds this threshold, typically, 1.2V.
Fig. 3b. Recommended startup sequence,
Non-Tracking operation
12
Rev 3.0
PD-97508
IR3832WMPbF
Figure 3c. shows the recommended startup
sequence for tracking operation of IR3832W with
Enable used as logic input.
Fig. 5. Pre-Bias startup pulses
Soft-Start
The IR3832W has a programmable soft-start to
control the output voltage rise and limit the
current surge at the start-up. To ensure correct
start-up, the soft-start sequence initiates when
the Enable and Vcc rise above their UVLO
thresholds and generate the Power On Ready
(POR) signal. The internal current source
(typically 20uA) charges the external capacitor
Css linearly from 0V to 3V. Figure 6 shows the
waveforms during the soft start.
Fig. 3c. Recommended startup sequence,
Sequenced operation
The start up time can be estimated by:
Pre-Bias Startup
Vp *CSS
IR3832W is able to start up into pre-charged
Tstart
=
- - - - - - - - - - - - - - - - - - - -(1)
output,
which
prevents
oscillation
and
20μA
disturbances of the output voltage.
During the soft start the OCP is enabled to
protect the device for any short circuit and over
current condition.
The output starts in asynchronous fashion and
keeps the synchronous MOSFET off until the first
gate signal for control MOSFET is generated.
Figure 4 shows a typical Pre-Bias condition at
start up.
The synchronous MOSFET always starts with a
narrow pulse width and gradually increases its
duty cycle with a step of 25%, 50%, 75% and
100% until it reaches the steady state value. The
number of these startup pulses for the
synchronous MOSFET is internally programmed.
Figure 5 shows a series of 32, 16, 8 startup
pulses.
Fig. 6. Theoretical peration waveforms
during soft-start
Fig. 4. Pre-Bias startup
13
Rev 3.0
PD-97508
IR3832WMPbF
Operating Frequency
1400
IOCSet (μA) =
...................................(2)
The switching frequency can be programmed
between 250kHz – 1500kHz by connecting an
external resistor from Rt pin to Gnd. Table 1
tabulates the oscillator frequency versus Rt.
Rt (kΩ)
Table 1. shows IOCSet at different switching
frequencies. The internal current source
develops a voltage across ROCSet. When the low
side MOSFET is turned on, the inductor current
flows through the Q2 and results in a voltage at
OCSet which is given by:
Table 1. Switching Frequency and IOCSet vs.
External Resistor (Rt)
Rt (kΩ)
47.5
35.7
28.7
23.7
20.5
17.8
15.8
14.3
12.7
11.5
10.7
9.76
9.31
Fs (kHz)
300
Iocset (μA)
29.4
VOCSet = (IOCSet ∗ROCSet )−(RDS(on) ∗IL )..........(3)
400
39.2
500
48.7
600
59.07
68.2
700
800
78.6
900
88.6
1000
1100
1200
1300
1400
1500
97.9
110.2
121.7
130.8
143.4
150.3
Fig. 7. Connection of over current sensing resistor
An over current is detected if the OCSet pin goes
below ground. Hence, at the current limit
threshold, VOCset=0. Then, for a current limit
setting ILimit, ROCSet is calculated as follows:
Shutdown
The IR3832W can be shutdown by pulling the
Enable pin below its 1 V threshold. This will tri-
state both, the high side driver as well as the low
side driver. Alternatively, the output can be
shutdown by pulling the soft-start pin below 0.3V.
Normal operation is resumed by cycling the
voltage at the Soft Start pin.
R
DS(on) * ILimit
ROCSet
=
......................(4)
IOCSet
An overcurrent detection trips the OCP
comparator, latches OCP signal and cycles the
soft start function in hiccup mode.
Over-Current Protection
The hiccup is performed by shorting the soft-start
capacitor to ground and counting the number of
switching cycles. The Soft Start pin is held low
until 4096 cycles have been completed. The
OCP signal resets and the converter recovers.
After every soft start cycle, the converter stays in
this mode until the overload or short circuit is
removed.
The over current protection is performed by
sensing current through the RDS(on) of low side
MOSFET. This method enhances the converter’s
efficiency and reduces cost by eliminating a
current sense resistor. As shown in figure 7, an
external resistor (ROCSet) is connected between
OCSet pin and the switch node (SW) which sets
the current limit set point.
The OCP circuit starts sampling current typically
160 ns after the low gate drive rises to about 3V.
This delay functions to filter out switching noise.
An internal current source sources current (IOCSet
) out of the OCSet pin. This current is a function
of the switching frequency and hence, of Rt.
14
Rev 3.0
PD-97508
IR3832WMPbF
Thermal Shutdown
Power Good Output
Temperature sensing is provided inside
IR3832W. The trip threshold is typically set to
140oC. When trip threshold is exceeded, thermal
shutdown turns off both MOSFETs and
discharges the soft start capacitor.
The IC continually monitors the output voltage via
Feedback (Fb pin). The Power Good signal is
flagged when the Fb pin voltage is above 0.5V
and between 85% to 115 % of Vp. This pin is
open drain and it needs to be externally pulled
high. High state indicates that output is in
regulation. Fig. 8a shows the PGood timing
diagram for non-tracking operation. In this case,
during startup, PGood goes high after the SS
voltage reaches 2.1V if the Fb voltage is within
the PGood comparator window. Fig. 8a. and Fig
8b. also show a 256 cycle delay between the Fb
voltage entering within the thresholds defined by
the PGood window and PGood going high.
Automatic restart is initiated when the sensed
temperature drops within the operating range.
There is a 20oC hysteresis in the thermal
shutdown threshold.
TIMING DIAGRAM OF PGOOD FUNCTION
Fig.4A IR3832W Non-Tracking Operation
Fig.8b IR3832W Tracking Operation
15
Rev 3.0
PD-97508
IR3832WMPbF
Minimum on time Considerations
Maximum Duty Ratio Considerations
The minimum ON time is the shortest amount of
time for which the Control FET may be reliably
turned on, and this depends on the internal
timing delays. For the IR3832W, the typical
minimum on-time is specified as 50 ns.
Any design or application using the IR3832W
must ensure operation with a pulse width that is
higher than this minimum on-time and preferably
higher than 100 ns. This is necessary for the
circuit to operate without jitter and pulse-
skipping, which can cause high inductor current
ripple and high output voltage ripple.
A fixed off-time of 200 ns maximum is specified
for the IR3832W. This provides an upper limit on
the operating duty ratio at any given switching
frequency. It is clear, that higher the switching
frequency, the lower is the maximum duty ratio at
which the IR3832W can operate. To allow a
margin of 50ns, the maximum operating duty
ratio in any application using the IR3832W
should still accommodate about 250 ns off-time.
Fig 9. shows a plot of the maximum duty ratio v/s
the switching frequency, with 250 ns off-time.
Max Duty Cycle
D
ton
=
Fs
Vout
Vin × Fs
95
90
85
80
75
70
65
60
55
=
In any application that uses the IR3832W, the
following condition must be satisfied:
250
450
650
850
1050
1250
1450
1650
ton(min) ≤ ton
Switching Frequency (kHz)
Vout
∴ ton(min)
≤
Vin × Fs
Vout
Fig. 9. Maximum duty cycle v/s switching
frequency.
∴Vin × Fs ≤
ton(min)
The minimum output voltage for the IR3832W is
limited to Vout(min) = 0.6 V.
Vout(min)
∴ Vin × Fs
≤
ton(min)
0.6 V
∴ Vin × Fs ≤
= 6 ×106 V/s
100 ns
Furthermore, for the IR3832W, especially for
active bus termination applications, it is strongly
recommended to use a switching frequency of
400 kHz to obtain clean and jitter free operation
in sourcing as well as sinking modes. Therefore,
the maximum input voltage that may be stepped
down to 0.6V at 400 kHz without jitter or pulse
skipping is 15 V.
16
Rev 3.0
PD-97508
IR3832WMPbF
When an external resistor divider is connected to
the output as shown in figure 10.
Equation (5) can be rewritten as:
Application Information
Design Example:
The following example is a typical application for
IR3832W. The application circuit is shown on
page 23.
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
Vp
R9 = R8 ∗
...............................(8)
Vo−Vp
For low voltage applications, such as this
design, it is often advisable to eliminate the bias
resistor R9 from Fb to ground. For the calculated
value of R8 see feedback compensation section.
V =12V (13.2Vmax)
in
V = 0.75V
o
Io = 4 A
ΔV ≤ 22.5mV
o
F = 400 kHz
s
Enabling the IR3832W
As explained earlier, the precise threshold of
the Enable lends itself well to implementation of
a UVLO for the Bus Voltage.
Vin
IR3832W
Fig. 10. Typical application of the IR3832W for
programming the output voltage
R1
Enable
R2
Further, the tracking reference Vp may be itself
derived from some master reference by means
of a resistive divider as shown in Fig. 9. This is
common in active bus termination circuits such
as Voltage Tracking Termination (VTT) where
the tracking reference Vp may be obtained as
half of the master reference VDDQ which forms
the input to one or more memory banks.
In this design,
For a typical Enable threshold of VEN = 1.2 V
R2
Vin(min)
*
= VEN = 1.2.......... (5)
R1 + R2
VEN
Vin( min ) −VEN
R2 = R1
..........(6)
VDDQ=1.5V
Rp1=Rp2=1.5 kΩ
Vp=0.75V
For a Vin (min)=10.2V, R1=49.9K and R2=7.50K is
a good choice.
Soft-Start Programming
The soft-start timing can be programmed by
selecting the soft-start capacitance value. From
(1), for a desired start-up time of the converter,
the soft start capacitor can be calculated by
using:
Programming the frequency
For Fs = 400 kHz, select Rt = 35.7 kΩ, using
Table. 1.
Output Voltage Programming
CSS (μF) = Tstart ( ms )× 0.02857 .......... (9)
Output voltage is programmed by the tracking
reference voltage at Vp and external voltage
divider. The divider is ratioed such that the
voltage at the Fb pin is equal to the voltage at the
Vp pin pin when the output is at its desired value.
The output voltage is defined by using the
following equation:
Where Tstart is the desired start-up time (ms).
For tracking applications the output is generally
required to track Vp even at start-up. Hence, it is
necessary to ensure that the SS pin is already
up to 3 V before the tracking reference signal is
applied to the Vp pin. This can be done by
choosing
a small value for the soft-start
⎛
⎜
⎝
⎞
⎟
⎟
⎠
R8
R9
capacitor to ensure that the voltage at the SS pin
rises to 3 V quickly. A 0.022 uF capacitor is
chosen for this purpose.
⎜
Vo =Vp ∗ 1+
................................(7)
17
Rev 3.0
PD-97508
IR3832WMPbF
Bootstrap Capacitor Selection
Vo
D =
.............................(13)
To drive the Control FET, it is necessary to
supply a gate voltage at least 4V greater than the
voltage at the SW pin, which is connected the
source of the Control FET . This is achieved by
using a bootstrap configuration, which comprises
the internal bootstrap diode and an external
bootstrap capacitor (C6), as shown in Fig. 11..
The operation of the circuit is as follows: When
the lower MOSFET is turned on, the capacitor
node connected to SW is pulled down to ground.
The capacitor charges towards Vcc through the
internal bootstrap diode, which has a forward
voltage drop VD. The voltage Vc across the
bootstrap capacitor C6 is approximately given as
Vin
Where:
D is the Duty Cycle
IRMS is the RMS value of the input capacitor
current.
Io is the output current.
For Io=4 A and D = 0.0625, the IRMS = 0.97 A
Ceramic capacitors are recommended due to
their peak current capabilities. They also feature
low ESR and ESL at higher frequency which
enables better efficiency. For this application, it is
advisable to have 2x10uF 16V ceramic capacitors
ECJ-3YB1C106M from Panasonic. In addition to
these, although not mandatory, a 330uF 25V
SMD capacitor EEV-FK1E332P from Panasonic
may be used as a bulk capacitor, and is
recommended if the input power supply is not
located close to the converter.
Vc ≅ Vcc −VD ........................................(10)
When the upper MOSFET turns on in the next
cycle, the capacitor node connected to SW rises
to the bus voltage Vin. However, if the value of C6
is appropriately chosen, the voltage Vc across C6
remains approximately unchanged and the
voltage at the Boot pin becomes
Inductor Selection
The inductor is selected based on output power,
operating frequency and efficiency requirements.
A low inductor value results in a smaller size and
faster response to a load transient but poor
efficiency and high output noise due to large
ripple current. Generally, the selection of the
inductor value can be reduced to the desired
VBoot ≅ Vin +Vcc −VD ........................................(11)
(Δi)
maximum ripple current in the inductor
optimum point is usually found between 20% and
50% ripple of the output current.
. The
For the buck converter, the inductor value for the
desired operating ripple current can be
determined using the following relation:
Δi
Δt
1
Vin −Vo = L ∗ ; Δt = D ∗
Fs
....................... (14)
Vo
L =
(
Vin −Vo ∗
)
Vin ∗ Δi * Fs
Fig. 11. Bootstrap circuit to generate
Vc voltage
Where:
Vin = Maximum input voltage
Vo = Output Voltage
Δi = Inductorripple current
Fs= Switching frequency
Δt = Turn on time
A bootstrap capacitor of value 0.1uF is suitable
for most applications.
Input Capacitor Selection
The ripple current generated during the on time of
the upper MOSFET should be provided by the
input capacitor. The RMS value of this ripple is
expressed by:
D = Duty cycle
If Δi ≈ 30%(Io), then the output inductor is
calculated to be 1.46μH. Select L=1.50 μH.
The MPO104-1R5 from Delta provides
compact, low profile inductor suitable for this
application.
a
IRMS =Io ∗ D∗(1−D) ........................(12)
18
Rev 3.0
PD-97508
IR3832WMPbF
Output Capacitor Selection
The output LC filter introduces a double pole,
–40dB/decade gain slope above its corner
resonant frequency, and a total phase lag of 180o
(see figure 12). The resonant frequency of the LC
filter is expressed as follows:
The voltage ripple and transient requirements
determine the output capacitors type and values.
The criteria is normally based on the value of the
Effective Series Resistance (ESR). However the :
actual capacitance value and the Equivalent Series
Inductance (ESL)
components. These components can be described
as
are other contributing
1
FLC
=
..............................(16)
2∗π Lo ∗Co
ΔVo = ΔVo(ESR ) + ΔVo(ESL) + ΔVo(C )
Figure 12 shows gain and phase of the LC filter.
Since we already have 180o phase shift from the
output filter alone, the system runs the risk of
being unstable.
ΔVo( ESR ) = ΔIL * ESR
V −V
⎛
⎜
⎞
⎟
in
o
ΔVo( ESL)
=
* ESL
L
⎝
⎠
Gain
0 dB
Phase
00
.........................(15)
ΔIL
8* Co * F
ΔVo(C )
=
s
-40dB/decade
Frequency
ΔVo = Output voltage ripple
ΔIL = Inductor ripple current
-1800
Frequency
FLC
FLC
Since the output capacitor has a major role in the
overall performance of the converter and
determines the result of transient response,
selection of the capacitor is critical. The
IR3832W can perform well with all types of
capacitors.
Fig. 12. Gain and Phase of LC filter
The IR3832W uses a voltage-type error amplifier
with high-gain (110dB) and wide-bandwidth. The
output of the error amplifier is available for DC
gain control and AC phase compensation.
As a rule, the capacitor must have low enough
ESR to meet output ripple and load transient
requirements.
The error amplifier can be compensated either in
type II or type III compensation.
The goal for this design is to meet the voltage
ripple requirement in the smallest possible
capacitor size. Therefore it is advisable to select
ceramic capacitors due to their low ESR and ESL
and small size. Six of the Panasonic ECJ-
2FB0J226ML (22uF, 6.3V, 3mOhm) capacitors is
a good choice.
Local feedback with Type II compensation is
shown in Fig. 13.
This method requires that the output capacitor
should have enough ESR to satisfy stability
requirements. In general the output capacitor’s
ESR generates a zero typically at 5kHz to 50kHz
which is essential for an acceptable phase
margin.
Feedback Compensation
The IR3832W is a voltage mode controller. The
control loop is a single voltage feedback path
including error amplifier and error comparator. To
achieve fast transient response and accurate
output regulation, a compensation circuit is
necessary. The goal of the compensation
network is to provide a closed-loop transfer
function with the highest 0 dB crossing frequency
and adequate phase margin (greater than 45o).
The ESR zero of the output capacitor is
expressed as follows:
1
FESR
=
...........................(17)
2 ∗π*ESR*Co
19
Rev 3.0
PD-97508
IR3832WMPbF
Where:
Vin = Maximum Input Voltage
Vosc = Oscillator Ramp Voltage
Fo = Crossover Frequency
VOUT
ZIN
CPOLE
FESR = Zero Frequency of the Output Capacitor
FLC = Resonant Frequency of the Output Filter
R8 = Feedback Resistor
R3
C4
R8
Zf
To cancel one of the LC filter poles, place the
zero before the LC filter resonant frequency pole:
Fb
Ve
E/A
R9
Comp
Fz = 75%F
LC
VREF
Gain(dB)
1
Fz = 0.75*
.....................................(22)
H(s) dB
2π Lo *Co
Use equations (20), (21) and (22) to calculate
C4.
One more capacitor is sometimes added in
parallel with C4 and R3. This introduces one
more pole which is mainly used to suppress the
switching noise.
FPOLE Frequency
FZ
Fig. 13. Type II compensation network
and its asymptotic gain plot
The additional pole is given by:
1
FP =
.................................(23)
The transfer function (Ve/Vo) is given by:
C4 *CPOLE
C4 +CPOLE
2π *R3 *
Ve
Vo
Zf
1+sRC4
3
= H(s) = −
= −
.....(18)
ZIN
sRC4
8
The pole sets to one half of the switching
frequency which results in the capacitor CPOLE
:
The (s) indicates that the transfer function varies
as a function of frequency. This configuration
introduces a gain and zero, expressed by:
1
1
CPOLE
=
≅
....................(24)
1
π*R*F
3
s
π*R*F −
3
s
C4
R3
H
(
s
)
=
.....................................(19)
............................(20)
R8
For a general solution for unconditional stability
for any type of output capacitors, and a wide
range of ESR values, we should implement local
feedback with a type III compensation network.
The typically used compensation network for
voltage-mode controller is shown in figure 14.
1
Fz =
2π *R3 *C4
First select the desired zero-crossover frequency
(Fo):
Again, the transfer function is given by:
F > FESR andF ≤
(
1/5~1/10
)
*F
s
o
o
Ve
Vo
Zf
= H(s) = −
Use the following equation to calculate R3:
ZIN
By replacing Zin and Zf according to figure 14,
the transfer function can be expressed as:
Vosc *F *FESR*R8
o
R3 =
...........................(21)
V *F2
in
LC
(1+ sR3C4 )
1+ sC7
(
R8 + R10
)
]
H(s) = −
....(25)
⎡
⎤
⎞
⎟
⎟
⎠
⎛
⎜
⎜
⎝
C4 *C3
sR (C +C ) 1+ sR
(1+ sR C )
⎢
⎥
8
4
3
3
10 7
C4 +C3
⎢
⎥
⎦
⎣
20
Rev 3.0
PD-97508
IR3832WMPbF
VOUT
Compensator
Type
Output
Capacitor
ZIN
F
ESR vs Fo
C3
C7
R3
C4
Electrolytic
Tantalum
Type II
Type III
FLC<FESR<Fo<Fs/2
FLC<Fo<FESR
R
10
R8
Zf
Tantalum
Ceramic
Fb
Ve
E/A
R9
Comp
VREF
Gain(dB)
The higher the crossover frequency, the
potentially faster the load transient response.
However, the crossover frequency should be low
enough to allow attenuation of switching noise.
Typically, the control loop bandwidth or crossover
frequency is selected such that
H(s) dB
Frequency
FZ1
FZ2
FP2
FP3
Fo ≤
1/5 ~ 1/10 * Fs
)
The DC gain should be large enough to provide
high DC-regulation accuracy. The phase margin
should be greater than 45o for overall stability.
Fig.14. Type III Compensation network and
its asymptotic gain plot
The compensation network has three poles and
two zeros and they are expressed as follows:
For this design we have:
Vin=12V
Vo=0.75V
Vosc=1.8V
Vp=0.75V
Lo=1.5 uH
FP1 = 0............................................................(26)
1
FP2
FP3
=
=
...........................................(27)
2π *R10 *C7
Co=6x22uF, ESR=3mOhm each
1
1
≅
..............(28)
2π *R3 *C3
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
C4 *C3
C4 +C3
It must be noted here that the value of the
capacitance used in the compensator design
must be the small signal value. For instance, the
small signal capacitance of the 22uF capacitor
used in this design is 12uF at 0.75 VDC bias and
400 kHz frequency. It is this value that must be
used for all computations related to the
compensation. The small signal value may be
obtained from the manufacturer’s datasheets,
design tools or SPICE models. Alternatively, they
may also be inferred from measuring the power
stage transfer function of the converter and
measuring the double pole frequency FLC and
using equation (16) to compute the small signal
Co.
2π * R3
1
FZ1
FZ2
=
=
.........................................(29)
2π *R3 *C4
1
1
≅
..........(30)
2π *C7 *(R8 + R10) 2π *C7 *R8
Cross over frequency is expressed as:
V
1
in
F = R3 *C7 *
*
................................(31)
o
Vosc 2π *Lo *Co
Based on the frequency of the zero generated by
the output capacitor and its ESR, relative to
crossover frequency, the compensation type can
be different. The table below shows the
compensation types and location of the
crossover frequency.
These result to:
FLC=15.31 kHz
F
F
ESR=4.4 MHz
s/2=200 kHz
Select crossover frequency: Fo= 60 kHz
Since FLC<Fo<Fs/2<FESR, TypeIII is selected to
place the pole and zeros.
21
Rev 3.0
PD-97508
IR3832WMPbF
Programming the Current-Limit
Detailed calculation of compensation TypeIII
The Current-Limit threshold can be set by
connecting a resistor (ROCSET) from the SW pin
to the OCSet pin. The resistor can be calculated
by using equation (4). This resistor ROCSET must
be placed close to the IC.
o
Desired Phase Margin Θ=70
1−sin Θ
o
F =F
=10.58 kHz
Z2
1+sin Θ
The RDS(on) has
coefficient and it should be considered for the
worst case operation.
a
positive temperature
1+sin Θ
1−sin Θ
F =F
=340.28kHz
P2
o
ROCSet∗IOCSet
Select: F =0.5*F = 5.29 kHz and
Z1
Z2
ISET =IL(critica)l
=
.......................(32)
RDS(on)
F =0.5*F =200 kHz
P3
s
RDS(on) =14.3mΩ*1.25=17.87 mΩ
ISET ≅Io(LIM) =4 A*1.5=6 A
Select:C =2.2nF
7
(50% over nominal output current)
IOCSet =39.22μA (at F =400kHz)
s
Calculate R , C and C :
3
3
4
ROCSet =2.73kΩ Select R7 =2.74 kΩ
2π*F *L *C *V
o
o
o
R =
osc ;R =2.78 kΩ
3
3
C *V
7
in
Setting the Power Good Threshold
A window comparator internally sets a lower
Power Good threshold at 85% of Vp and an
upper Power Good threshold at 115% of Vp.
When the voltage at the FB pin is within the
window set by these thresholds, PGood is
asserted.
Select: R =3.48kΩ
3
1
C =
;C =10.75nF, Select: C =10 nF
4
4
4
2π*F *R
Z1
3
1
The PGood is an open drain output. Hence, it is
necessary to use a pull up resistor RPG from
PGood pin to Vcc. The value of the pull-up
resistor must be chosen such as to limit the
current flowing into the PGood pin, when the
output voltage is not in regulation, to less than 5
mA. A typical value used is 10kΩ.
C =
;C =228 pF, Select:C =220 pF
3
3
3
2π*F *R
P3
3
Calculate R , R and R :
10
8
9
1
R =
; R =215 Ω, Select:R =210 Ω
10
10
10
2π*C *F
7
P2
1
R =
-R ; R =6.63kΩ,
8
10
8
2π*C *F
7
Z2
Select:R =6.65kΩ
8
22
Rev 3.0
PD-97508
IR3832WMPbF
Application Diagram:
Fig. 15. Application circuit diagram for a 12V to 0.75 V, 4 A Point Of Load Converter
Suggested Bill of Materials for the application circuit:
Part Reference Quantity Value
Description
SMD Elecrolytic, Fsize, 25V, 20%
1206, 16V, X5R, 20%
0603, 25V, X7R, 10%
11.5x10x4mm, 20%, 1.7mOhm
0805, 6.3V, X5R, 20%
Thick Film, 0603,1/10 W,1%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
Manufacturer
Panasonic
TDK
Panasonic
Delta
Panasonic
Rohm
Rohm
Part Number
1
2
1
1
6
1
1
1
330uF
10uF
0.1uF
1.5uH
22uF
49.9k
7.5k
EEV-FK1E331P
C3216X5R1E106M
ECJ-1VB1E104K
MPO104-1R5
ECJ-2FB0J226ML
MCR03EZPFX4992
MCR03EZPFX7501
MCR03EZPFX3572
Cin
Lo
Co
R1
R2
Rt
35.7k
Rohm
Rocset
RPG
Css
1
1
2.74k
10k
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
Rohm
Rohm
MCR03EZPFX2741
MCR03EZPFX1002
1
1
1
1
1
1
1
1
1
0.022uF
3.48k
220pF
0.1uF
2200pF
6.65k
210
2200pF
10nF
0603, 25V, X7R, 10%
Thick Film, 0603,1/10W,1%
50V, 0603, NPO, 5%
0603, 25V, X7R, 10%
0603, 50V, X7R, 10%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
0603, 50V, X7R, 10%
0603, 50V, X7R, 10%
Panasonic
Rohm
Panasonic
Panasonic
Panasonic
Rohm
Rohm
Panasonic
Panasonic
ECJ-1VB1E223K
MCR03EZPFX3481
ECJ-1VC1H221J
ECJ-1VB1E104K
ECJ-1VB1H223K
MCR03EZPFX6651
ERJ-3EKF2100V
ECJ-1VB1H222K
ECJ-1VB1H103K
R3
C3
C6
C4
R8
R10
C7
Cp2
CVcc
U1
1
1
1.0uF
IR3832W
0603, 16V, X5R, 20%
SupIRBuck, 4A, PQFN 5x6mm
Panasonic
International Rectifier
ECJ-BVB1C105M
IR3832WMPbF
23
Rev 3.0
PD-97508
IR3832WMPbF
TYPICAL OPERATING WAVEFORMS
Vin=12.0V, Vcc=5V, Vo=0.75V, Io=0- ±4A, Room Temperature, No Air Flow
Fig. 16: Start up at 4A, sourcing current
Ch1:PGood, Ch2:Vout, Ch3:VDDQ, Ch4:SS
Fig. 17: Start up with Prebias, 0A Load
Ch1:PGood, Ch2:Vout,Ch3:VDDQ, Ch4:SS
Fig. 19: Inductor node at -3A, sinking
current, Ch3:SW , Ch4:Iout
Fig. 18: Inductor node at 4A, sourcing
current, Ch3:SW, Ch4:Iout
Fig. 20: Output Voltage Ripple, 4A,
sourcing current, Ch2: Vout
Fig. 21: Short (Hiccup) Recovery
Ch2:Vout, Ch3:VSS , Ch4:PGood
24
Rev 3.0
PD-97508
IR3832WMPbF
TYPICAL OPERATING WAVEFORMS
Vin=12V, Vcc=5V, Vo=0.75V, Room Temperature, No Air Flow
Fig. 23: Tracking -3A load, sinking current,
Ch2:Vout, Ch3:VDDQ, Ch4:PGood
Fig. 22: Tracking 4A, sourcing current,
Ch2:Vout, Ch3:VDDQ, Ch4:PGood
Fig. 24: Transient Response, 1A/us
-0.5A to +0.5A load , Ch2:Vout, Ch4:Io
25
Rev 3.0
PD-97508
IR3832WMPbF
TYPICAL OPERATING WAVEFORMS
Vin=12V, Vcc=5V, Vo=0.75V, Io=+4A, Room Temperature, No Air Flow
Fig.25: Bode Plot at 4A load (sourcing current) shows a bandwidth of 65kHz and phase margin of
60 degrees
26
Rev 3.0
PD-97508
IR3832WMPbF
Layout Considerations
The connection between the OCSet resistor and
the Sw pin should not share any trace with the
connection between the bootstrap capacitor and
the Sw pin. Instead, it is recommended to use a
Kelvin connection of the trace from the OCSet
The layout is very important when designing high
frequency switching converters. Layout will affect
noise pickup and can cause a good design to
perform with less than expected results.
Make all the connections for the power
components in the top layer with wide, copper
filled areas or polygons. In general, it is desirable
to make proper use of power planes and
polygons for power distribution and heat
dissipation.
The inductor, output capacitors and the IR3832W
should be as close to each other as possible.
This helps to reduce the EMI radiated by the
power traces due to the high switching currents
through them. Place the input capacitor directly at
the Vin pin of IR3832W.
The feedback part of the system should be kept
away from the inductor and other noise sources.
The critical bypass components such as
capacitors for Vcc should be close to their
respective pins. It is important to place the
feedback components including feedback
resistors and compensation components close to
Fb and Comp pins.
Vin
PGnd
resistor and the trace from the bootstrap
Vin
PGnd
capacitor at the Sw pin.
In a multilayer PCB use one layer as a power
ground plane and have a control circuit ground
Vout
to which all signals are
AGnd ground),
(analog
referenced. The goal is to localize the high
current path to a separate loop that does not
Vout
iAnGtenrdfere with the more sensitive analog control
function. These two grounds must be connected
together on the PC board layout at a single point.
The Power QFN is a thermally enhanced
package. Based on thermal performance it is
recommended to use at least a 4-layers PCB. To
effectively remove heat from the device the
exposed pad should be connected to the ground
plane using vias. Figure 26 illustrates the
implementation of the layout guidelines outlined
above, on a 4 layer board.
Enough copper &
minimum length
ground path between
Input and Output
PGnd
Compensation parts
should be placed as close
as possible to
All bypass caps
should be placed as
close as possible to
their connecting
the Comp pin.
Vin
pins.
AGnd
Resistors Rt and
Rocset should be
placed as close as
possible to their pins.
Vout
Fig. 26a. IR3832W layout
considerations – Top Layer
27
Rev 3.0
PD-97508
IR3832WMPbF
PGnd
Feedback
trace should
be kept
away form
noise
sources
Fig. 26b. IR3832W layout considerations –
Bottom Layer
Analog
Ground
plane
Power
Vin
Ground
Plane
Single point
connection between
AGND & PGND,
should be close to
the SupIRBuck, kept
away from noise
sources.
AGnd
Fig. 26c. IR3832W layout considerations –
Mid Layer 1
Use separate traces
for connecting Boot
cap and Rocset to the
switch node and with
the minimum length
traces. Avoid big
loops.
Fig. 26d. IR3832W layout considerations –
Mid Layer 2
28
Rev 3.0
PD-97508
IR3832WMPbF
PCB Metal and Components Placement
Lead lands (the 11 IC pins) width should be equal to nominal part lead width. The
minimum lead to lead spacing should be ≥ 0.2mm to minimize shorting.
Lead land length should be equal to maximum part lead length + 0.3 mm outboard
extension. The outboard extension ensures a large and inspectable toe fillet.
Pad lands (the 4 big pads other than the 11 IC pins) length and width should be equal to
maximum part pad length and width. However, the minimum metal to metal spacing
should be no less than 0.17mm for 2 oz. Copper; no less than 0.1mm for 1 oz. Copper
and no less than 0.23mm for 3 oz. Copper.
29
Rev 3.0
PD-97508
IR3832WMPbF
Solder Resist
It is recommended that the lead lands are Non Solder Mask Defined (NSMD). The solder
resist should be pulled away from the metal lead lands by a minimum of 0.025mm to ensure
NSMD pads.
The land pad should be Solder Mask Defined (SMD), with a minimum overlap of the solder
resist onto the copper of 0.05mm to accommodate solder resist mis-alignment.
Ensure that the solder resist in-between the lead lands and the pad land is ≥ 0.15mm due to
the high aspect ratio of the solder resist strip separating the lead lands from the pad land.
30
Rev 3.0
PD-97508
IR3832WMPbF
Stencil Design
•
The Stencil apertures for the lead lands should be approximately 80% of the area of
the lead lads. Reducing the amount of solder deposited will minimize the
occurrences of lead shorts. If too much solder is deposited on the center pad the part
will float and the lead lands will be open.
•
The maximum length and width of the land pad stencil aperture should be equal to
the solder resist opening minus an annular 0.2mm pull back to decrease the
incidence of shorting the center land to the lead lands when the part is pushed into
the solder paste.
31
Rev 3.0
PD-97508
IR3832WMPbF
BOTTOM VIEW
IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (320) 252-7105
TAC Fax: (320) 252-7903
This product has been designed and qualified for the Consumer market
Visit us at www.irf.com for sales contact information
Data and specifications subject to change without notice. 01/09
32
Rev 3.0
相关型号：
IR38363MTRPBFAUMA1
Switching Regulator, Voltage-mode, 25A, 1500kHz Switching Freq-Max, PXSS26, QFN-26
INFINEON
IR38365M
OptiMOS™ IPOL DC-DC converter single-input voltage, 15 A buck regulators with SVID
INFINEON
IR3837MTRPBF
Switching Regulator, Voltage-mode, 14A, 1650kHz Switching Freq-Max, 5 X 6 MM, 0.90 MM HEIGHT, HALOGEN FREE AND ROHS COMPLIANT, PQFN-17
INFINEON
IR3838MTR1PBF
Switching Regulator, Voltage-mode, 10A, 1650kHz Switching Freq-Max, 5 X 6 MM, 0.90 MM HEIGHT, HALOGEN FREE AND ROHS COMPLIANT, PQFN-17
INFINEON
©2020 ICPDF网 联系我们和版权申明