IR3856WMTRPBF [INFINEON]
HIGHLY EFFICIENT INTEGRATED 6A, SYNCHRONOUS BUCK REGULATOR; 高效的综合6A同步降压稳压器
| 型号： | IR3856WMTRPBF |
| 厂家： | 
Infineon |
| 描述： | HIGHLY EFFICIENT INTEGRATED 6A, SYNCHRONOUS BUCK REGULATOR |
| 文件： | 总34页 (文件大小：1022K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
PD-97506
IR3856WMPbF
TM
HIGHLY EFFICIENT
SupIRBuck
INTEGRATED 6A, SYNCHRONOUS BUCK REGULATOR
Features
Description
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Greater than 95% Maximum Efficiency
Wide Input Voltage Range 1.5V to 16V
Wide Output Voltage Range 0.7V to 0.9*Vin
Continuous 6A Load Capability
The IR3856W SupIRBuckTM is an easy-to-use,
fully integrated and highly efficient DC/DC
synchronous Buck regulator. The MOSFETs co-
packaged with the on-chip PWM controller make
IR3856W a space-efficient solution, providing
accurate power delivery for low output voltage
applications.
Integrated Bootstrap-diode
High Bandwidth E/A for excellent transient
performance
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•
•
•
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•
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Programmable Switching Frequency up to 1.5MHz
Programmable Over Current Protection
PGood output
IR3856W is a versatile regulator which offers
programmability of start up time, switching
frequency and current limit while operating in
wide input and output voltage range.
Hiccup Current Limit
Precision Reference Voltage (0.7V, +/-1%)
Programmable Soft-Start
The switching frequency is programmable from
250kHz to 1.5MHz for an optimum solution.
Enable Input with Voltage Monitoring Capability
Enhanced Pre-Bias Start-up
Seq input for Tracking applications
-40oC to 125oC operating junction temperature
Thermal Protection
It also features important protection functions,
such as Pre-Bias startup, hiccup current limit and
thermal shutdown to give required system level
security in the event of fault conditions.
4mm x 5mm Power QFN Package
Halogen Free, Lead Free and RoHS compliant
Applications
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Netcom Applications
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Server Applications
Computing Peripheral Voltage Regulators
General DC-DC Converters
Storage Applications
Embedded Telecom Systems
Distributed Point of Load Power Architectures
Fig. 1. Typical application diagram
1
Rev 9.0
PD-97506
IR3856WMPbF
ABSOLUTE MAXIMUM RATINGS
(Voltages referenced to GND unless otherwise specified)
•
•
•
•
•
•
•
•
•
•
•
•
Vin ……………………………………….……………. -0.3V to 25V
PVcc, Vcc ……………….……..….……..……….…… -0.3V to 8V (Note2)
Boot
SW
……………………………………..……….…. -0.3V to 33V
…………………………………………..……… -0.3V to 25V(DC), -4V to 25V(AC, 100ns)
Boot to SW ……..…………………………….…..….. -0.3V to Vcc+0.3V (Note1)
OCSet ………………………………………….……. -0.3V to 30V (Max 30mA)
Input / output Pins ……………………………….. ... -0.3V to Vcc+0.3V (Note1)
PGND to GND ……………...………………………….. -0.3V to +0.3V
Storage Temperature Range ................................... -55°C To 150°C
Junction Temperature Range ................................... -40°C To 150°C (Note2)
ESD Classification …………………………… ……… JEDEC Class 1C
Moisture sensitivity level………………...………………JEDEC Level 3@260 °C (Note5)
Note1: Must not exceed 8V
Note2: Vcc must not exceed 7.5V for Junction Temperature between -10oC and -40oC
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to
the device. These are stress ratings only and functional operation of the device at these or any other
conditions beyond those indicated in the operational sections of the specifications are not implied.
PACKAGE INFORMATION
4mm x 5mm POWER QFN
θJA
θJA
= 45o C / W *
= 45o C / W *
( Sync _ FET )
( Ctrl _ FET )
11
12
SW
13
VIN
θJ-PCB = 2o C / W
PGnd
* Exposed pads on underside
are connected to copper
pads of 4-layer (2 oz.) PCB
board design
14
PVcc
VCC
Boot
10
17
Gnd
Enable
9
8
15
16
PGood
Seq
1
4
7
2
3
5
6
Fb
NC COMP Gnd Rt SS/SD OCSet
ORDERING INFORMATION
PACKAGE
PACKAGE
PIN COUNT
PARTS PER
REEL
DESIGNATOR
DESCRIPTION
M
M
IR3856WMTRPbF
IR3856WMTR1PbF
17
17
4000
750
2
Rev 9.0
PD-97506
IR3856WMPbF
Block Diagram
Fig. 2. Simplified block diagram of the IR3856W
3
Rev 9.0
PD-97506
IR3856WMPbF
Pin Description
Pin Name
Description
1
Fb
Inverting input to the error amplifier. This pin is connected directly to the
output of the regulator via resistor divider to set the output voltage and
provide feedback to the error amplifier.
2
3
NC
No connect
Comp
Output of error amplifier. An external resistor and capacitor network is
typically connected from this pin to Fb pin to provide loop
compensation.
4
5
6
Gnd
Rt
Signal ground for internal reference and control circuitry.
Set the switching frequency. Connect an external resistor from this pin
to Gnd to set the switching frequency.
SS/SD
Soft start / shutdown. This pin provides user programmable soft-start
function. Connect an external capacitor from this pin to Gnd to set the
start up time of the output voltage. The converter can be shutdown by
pulling this pin below 0.3V.
7
8
OCSet
PGood
VCC
Current limit set point. A resistor from this pin to SW pin will set the
current limit threshold.
Power Good status pin. Output is open drain. Connect a pull up resistor
from this pin to Vcc.
9
This pin powers the internal IC. A minimum of 1uF high frequency
capacitor must be connected from this pin to Analog ground (AGnd).
10
11
12
13
PVcc
PGnd
SW
This pin powers the drivers. A minimum of 1uF high frequency capacitor
must be connected from this pin to the power ground (PGnd).
Power Ground. This pin serves as a separated ground for the MOSFET
drivers and should be connected to the system’s power ground plane.
Switch node. This pin is connected to the output inductor.
VIN
Input voltage connection pin.
14
15
Boot
Supply voltage for high side driver. A 0.1uF capacitor must be
connected from this pin to SW.
Enable pin to turn on and off the device. Use two external resistors to
set the turn on threshold (see Enable section). Connect this pin to Vcc if
it is not used.
Enable
16
17
Seq
Gnd
Sequence pin. Use two external resistors to set Simultaneous Power up
sequencing. If this pin is not used connect to Vcc.
Signal ground for internal reference and control circuitry.
4
Rev 9.0
PD-97506
IR3856WMPbF
Recommended Operating Conditions
Symbol
Definition
Min
Max
Units
Vin
P Vcc, Vcc
Boot to SW
Vo
Io
Input Voltage
Supply Voltage
Supply Voltage
Output Voltage
1.5
4.5
4.5
0.7
0
16
5.5
5.5
0.9*Vin
6
V
Output Current
A
Fs
Tj
Switching Frequency
Junction Temperature
225
-40
1650
125
kHz
oC
Electrical Specifications
Unless otherwise specified, these specification apply over 4.5V< PVcc=Vcc<5.5V, Vin=12V
0oC<Tj< 125oC. Typical values are specified at Ta = 25oC.
Parameter
Power Loss
Symbol
Test Condition
Min
TYP
MAX
Units
Power Loss
Plo ss
Vcc=5V, Vin=12V, Vo=1.8V, Io=6A,
1.3
W
Fs=600kHz, L=0.82uH, Note4
MOSFET Rds(on)
Top Switch
Rds(on)_Top
Rds(on)_Bot
VBoot -Vsw =5V, ID=6A, Tj=25oC
Vcc=5V, ID=6A, Tj=25oC
22.6
14.3
29.0
19
mΩ
Bottom Switch
Reference Voltage
Feedback Voltage
Accuracy
VFB
0.7
V
0oC<Tj<125oC
-1.0
+1.0
+2.0
%
-40oC<Tj<125oC, Note3
+2.0
Supply Current
VCC Supply Current (Standby)
Vcc Supply Current (Dyn)
PVCC Supply Current (Standby)
PVcc Supply Current (Dyn)
ICC(Standby)
ICC(Dyn)
IPCC(Standby)
IPCC(Dyn)
SS=0V, No Switching, Enable low
500
6
μA
SS=3V, Vcc=5V, Fs=500kHz
Enable high
3
7
mA
SS=0V, No Switching, Enable low
μA
SS=3V, PVcc=5V, Fs=500kHz
Enable high
mA
Under Voltage Lockout
VCC-Start-Threshold
VCC_UVLO_Start
VCC_UVLO_Stop
Enable_UVLO_Start
Enable_UVLO_Stop
Ien
Vcc Rising Trip Level
Vcc Falling Trip Level
Supply ramping up
Supply ramping down
Enable=3.3V
3.95
3.65
1.14
0.9
4.15
3.85
1.2
4.35
4.05
1.36
1.06
15
VCC-Stop-Threshold
Enable-Start-Threshold
Enable-Stop-Threshold
Enable leakage current
V
1.0
μA
5
Rev 9.0
PD-97506
IR3856WMPbF
Electrical Specifications (continued)
Unless otherwise specified, these specifications apply over 4.5V< PVcc=Vcc<5.5V, Vin=12V
0oC<Tj< 125oC. Typical values are specified at Ta = 25oC.
Parameter
Oscillator
Symbol
Test Condition
Min
TYP
MAX
Units
Rt Voltage
0.665
225
0.7
0.735
275
V
Frequency
FS
Rt=59K
250
500
Rt=28.7K
450
550
kHz
Rt=9.31K, Note4
1350
1500
1.8
1650
Ramp Amplitude
Ramp Offset
Vramp
Note4
Vp-p
V
Ramp (os)
Dmin(ctrl)
Note4
0.6
50
Min Pulse Width
Fixed Off Time
Max Duty Cycle
Note4
ns
%
Note4
130
200
+10
Dmax
Fs=250kHz
92
Error Amplifier
Input Offset Voltage
Vos
Vfb-Vseq,
-10
0
mV
Vseq=0.8V
Input Bias Current
Input Bias Current
Sink Current
IFb(E/A)
IVp(E/A)
Isink(E/A)
Isource(E/A)
SR
-1
-1
+1
+1
μA
0.40
8
0.85
10
1.2
13
mA
Source Current
Slew Rate
Note4
7
12
20
V/μs
MHz
dB
Gain-Bandwidth Product
DC Gain
GBWP
Note4
20
100
3.4
30
40
Gain
Note4
110
3.5
120
120
3.75
220
1
Maximum Voltage
Minimum Voltage
Common Mode Voltage
Vmax(E/A)
Vmin(E/A)
Vcc=4.5V
V
mV
V
Note4
0
Soft Start/SD
Soft Start Current
ISS
Vss(clamp)
SD
Source
14
20
26
3.3
0.3
μA
Soft Start Clamp Voltage
2.7
3.0
V
Shutdown Output
Threshold
Over Current Protection
OCSET Current
IOCSET
Fs=250kHz
Fs=500kHz
Fs=1500kHz
Note4
20.8
43
23.6
48.8
154
0
26.4
54.6
172
+10
μA
136
-10
OC Comp Offset Voltage
SS off time
VOFFSET
mV
SS_Hiccup
4096
Cycles
Bootstrap Diode
Forward Voltage
I(Boot)=30mA
180
5
260
10
470
30
mV
ns
Deadband time
Note4
6
Rev 9.0
PD-97506
IR3856WMPbF
Electrical Specifications
Unless otherwise specified, these specifications apply over 4.5V< PVcc=Vcc<5.5V, Vin=12V
0oC<Tj< 125oC. Typical values are specified at Ta = 25oC.
Parameter
SYM
Test Condition
Min
TYP
MAX
Units
Thermal Shutdown
Thermal Shutdown
Note4
Note4
140
20
oC
Hysteresis
Power Good
Power Good upper
Threshold
Upper Threshold
Delay
Power Good lower
Threshold
Lower Threshold
Delay
Delay Comparator
Threshold
VPG(upper)
VPG(upper)_Dly
VPG(lower)
Fb Rising
Fb Rising
Fb Falling
Fb Falling
0.770
0.560
0.810
256/Fs
0.6
0.840
0.630
V
s
V
VPG(lower)_Dly
PG(Delay)
256/Fs
2.1
s
Relative to charge voltage, SS rising
2
2.3
V
Delay Comparator
Hysteresis
Delay(hys)
Note4
260
300
340
mV
PGood Voltage Low
PG(voltage)
Ileakage
IPGood=-5mA
0.5
10
V
Leakage Current
0
μA
Switch Node
SW Bias Current
SW=0V, Enable=0V
Isw
μA
6
SW=0V,Enable=high,SS=3V,Vseq=0V
, Note4
Note3: Cold temperature performance is guaranteed via correlation using statistical quality control. Not tested in production.
Note4: Guaranteed by Design but not tested in production.
Note5: Upgrade to industrial/MSL2 level applies from date codes 1227 (marking explained on application note AN1132 page 2).
Products with prior date code of 1227 are qualified with MSL3 for Consumer market.
7
Rev 9.0
PD-97506
IR3856WMPbF
TYPICAL OPERATING CHARACTERISTICS (-40oC - 125oC) Fs=500 kHz
Icc(Dyn)
Icc(Standby)
5
4
3
2
1
290
270
250
230
210
190
170
150
-40
-20
0
20
40
60
80
100
120
Temp[oC]
-40
-20
0
20
40
60
80
100
120
Temp[oC]
FREQUENCY
IOCSET(500kHz)
550
540
530
520
510
500
490
480
470
460
450
54
53
52
51
50
49
48
47
46
45
44
43
-40
-20
0
20
40
60
80
100
120
-40
-20
0
20
40
Temp[oC]
Vcc(UVLO) Stop
60
80
100
120
Temp[oC]
Vcc(UVLO) Start
4.05
4.00
3.95
3.90
3.85
3.80
3.75
3.70
3.65
4.35
4.30
4.25
4.20
4.15
4.10
4.05
4.00
3.95
-40
-20
0
20
40
60
80
100
120
-40
-20
0
20
40
60
80
100
120
Temp[oC]
Temp[oC]
Enable(UVLO) Stop
Enable(UVLO) Start
1.36
1.34
1.32
1.30
1.28
1.26
1.24
1.22
1.20
1.18
1.16
1.14
1.06
1.04
1.02
1.00
0.98
0.96
0.94
0.92
0.90
-40
-20
0
20
40
60
80
100
120
-40
-20
0
20
40
60
80
100 120
Temp[oC]
Vfb
Temp[oC]
ISS
26.0
24.0
22.0
20.0
18.0
16.0
14.0
711
706
701
696
691
686
-40
-20
0
20
40
60
80
100
120
-40
-20
0
20
40
60
80
100
120
Temp[oC]
Temp [oC]
8
Rev 9.0
PD-97506
IR3856WMPbF
TYPICAL OPERATING CHARACTERISTICS (-40oC - 125oC) Fs=500 kHz
Ipcc(Standby)
Ipcc(Dyn)
2.0
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0.0
8.0
7.5
7.0
6.5
6.0
5.5
5.0
-40
-20
0
20
40
60
80
100
120
-40
-20
0
20
40
60
80
100
120
Temp[oC]
Temp[oC]
Rdson of MOSFETs Over Temperature at Vcc=5V
30
28
26
24
22
20
18
16
14
12
-40
-20
0
20
40
60
80
100
120
140
Temperature [°C]
Sync-FET
Ctrl-FET
9
Rev 9.0
PD-97506
IR3856WMPbF
Typical Efficiency and Power Loss Curves
Vin=12V, Vcc=5V, Io=1A-6A, Fs=600kHz, Room Temperature, No Air Flow
The table below shows the inductors used for each of the output voltages
in the efficiency measurement.
Vo (V)
0.9
1.0
1.1
1.2
1.5
1.8
2.5
3.3
L (uH) P/N
DCR (mOhm)
0.68
0.68
0.82
0.82
1.0
PCMB065T-R68MS
3.9
3.9
4.6
4.6
4.7
4.7
6.7
6.7
11.2
PCMB065T-R68MS
IHLP2525EZERR82M
IHLP2525EZERR82M
SPM6550T-1R0M100A
SPM6550T-1R0M100A
PCMB065T-1R5MS
PCMB065T-1R5MS
PCMB065T-2R2MS
1.0
1.5
1.5
2.2
5.0
96
94
92
90
88
86
84
82
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
Load Current (A)
0.9V
1.0V
1.1V
1.2V
1.5V
1.8V
2.5V
3.3V
5.0V
1.9
1.7
1.5
1.3
1.1
0.9
0.7
0.5
0.3
0.1
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
Load Current (A)
0.9V
1.0V
1.1V
1.2V
1.5V
1.8V
2.5V
3.3V
5.0V
10
Rev 9.0
PD-97506
IR3856WMPbF
Typical Efficiency and Power Loss Curves
Vin=5V, Vcc=5V, Io=1A-6A, Fs=600kHz, Room Temperature, No Air Flow
The table below shows the inductors used for each of the output voltages
in the efficiency measurement.
Vo (V)
0.7
0.75
0.9
1.0
1.1
1.2
1.5
1.8
2.5
L (uH) P/N
DCR (mOhm)
0.56
0.56
0.56
0.56
0.68
0.68
0.68
0.82
0.82
0.82
IHLP2525EZERR56M
3.4
3.4
3.4
3.4
3.9
3.9
3.9
4.6
4.6
4.6
IHLP2525EZERR56M
IHLP2525EZERR56M
IHLP2525EZERR56M
PCMB065T-R68MS
PCMB065T-R68MS
PCMB065T-R68MS
IHLP2525EZERR82M
IHLP2525EZERR82M
IHLP2525EZERR82M
3.3
96
94
92
90
88
86
84
82
80
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
Load Current (A)
0.7V
0.75V
0.9V
1.0V
1.1V
1.2V
1.5V
1.8V
2.5V
3.3V
1.3
1.1
0.9
0.7
0.5
0.3
0.1
1
1.5
2
2.5
3
3.5
4
4.5
5
5.5
6
Load Current (A)
0.7Vout
1.2Vout
0.75Vout
1.5Vout
0.9Vout
1.8Vout
1.0Vout
2.5Vout
1.1Vout
3.3Vout
11
Rev 9.0
PD-97506
IR3856WMPbF
Circuit Description
THEORY OF OPERATION
Introduction
If the input to the Enable pin is derived from the
bus voltage by a suitably programmed resistive
divider, it can be ensured that the IR3856W does
not turn on until the bus voltage reaches the
desired level. Only after the bus voltage reaches
or exceeds this level will the voltage at Enable
pin exceed its threshold, thus enabling the
IR3856W. Therefore, in addition to being a logic
input pin to enable the IR3856W, the Enable
feature, with its precise threshold, also allows the
user to implement an Under-Voltage Lockout for
the bus voltage Vin. This is desirable particularly
for high output voltage applications, where we
might want the IR3856W to be disabled at least
until Vin exceeds the desired output voltage level.
The IR3856W uses a PWM voltage mode control
scheme with external compensation to provide
good noise immunity and maximum flexibility in
selecting inductor values and capacitor types.
The switching frequency is programmable from
250kHz to 1.2MHz and provides the capability of
optimizing the design in terms of size and
performance.
IR3856W provides precisely regulated output
voltage programmed via two external resistors
from 0.7V to 0.9*Vin.
The IR3856W operates with an external bias
supply from 4.5V to 5.5V, allowing an extended
operating input voltage range from 1.5V to 21V.
The device utilizes the on-resistance of the low
side MOSFET as current sense element, this
method enhances the converter’s efficiency and
reduces cost by eliminating the need for external
current sense resistor.
IR3856W includes two low Rds(on) MOSFETs
using IR’s HEXFET technology. These are
specifically designed for high efficiency
applications.
Fig. 3a. Normal Start up, Device turns on
when the Bus voltage reaches 10.2V
Figure 3b. shows the recommended start-up
sequence for the non-sequenced operation of
IR3856W.
Under-Voltage Lockout and POR
The under-voltage lockout circuit monitors the
input supply Vcc and the Enable input. It assures
that the MOSFET driver outputs remain in the off
state whenever either of these two signals drop
below the set thresholds. Normal operation
resumes once Vcc and Enable rise above their
thresholds.
The POR (Power On Ready) signal is generated
when all these signals reach the valid logic level
(see system block diagram). When the POR is
asserted the soft start sequence starts (see soft
start section).
Enable
The Enable features another level of flexibility for
start up. The Enable has precise threshold which
is internally monitored by Under-Voltage Lockout
(UVLO) circuit. Therefore, the IR3856W will turn
on only when the voltage at the Enable pin
exceeds this threshold, typically, 1.2V.
Fig. 3b. Recommended startup sequence,
Non-Sequenced operation
Figure 3c. shows the recommended startup
sequence for sequenced operation of IR3856W
12
Rev 9.0
PD-97506
IR3856WMPbF
Fig. 5. Pre-Bias startup pulses
Soft-Start
The IR3856W has a programmable soft-start to
control the output voltage rise and to limit the
current surge at the start-up. To ensure correct
start-up, the soft-start sequence initiates when
the Enable and Vcc rise above their UVLO
thresholds and generate the Power On Ready
(POR) signal. The internal current source
(typically 20uA) charges the external capacitor
Css linearly from 0V to 3V. Figure 6 shows the
waveforms during the soft start.
Fig. 3c. Recommended startup sequence,
Sequenced operation
Pre-Bias Startup
IR3856W is able to start up into pre-charged
output, which prevents oscillation and
disturbances of the output voltage.
The output starts in asynchronous fashion and
keeps the synchronous MOSFET off until the first
gate signal for control MOSFET is generated.
Figure 4 shows a typical Pre-Bias condition at
start up.
The start up time can be estimated by:
1.4- 0.7
)
*CSS
Tstart
=
- - - - - - - - - - - - - - - - - - - - (1)
20μA
The synchronous MOSFET always starts with a
narrow pulse width and gradually increases its
duty cycle with a step of 25%, 50%, 75% and
100% until it reaches the steady state value. The
number of these startup pulses for the
synchronous MOSFET is internally programmed.
Figure 5 shows a series of 32, 16, 8 startup
pulses.
During the soft start the OCP is enabled to
protect the device for any short circuit and over
current condition.
Fig. 4. Pre-Bias startup
Fig. 6. Theoetical operation waveforms
uring soft-start
13
Rev 9.0
PD-97506
IR3856WMPbF
Operating Frequency
1400
IOCSet (μA) =
...................................(2)
The switching frequency can be programmed
between 250kHz – 1200kHz by connecting an
external resistor from Rt pin to Gnd. Table 1
tabulates the oscillator frequency versus Rt.
Rt (kΩ)
Table 1. shows IOCSet at different switching
frequencies. The internal current source
develops a voltage across ROCSet. When the low
side MOSFET is turned on, the inductor current
flows through the Q2 and results in a voltage at
OCSet which is given by:
Table 1. Switching Frequency and IOCSet vs.
External Resistor (Rt)
VOCSet = (IOCSet ∗ROCSet )−(RDS(on) ∗IL )..........(3)
Rt (kΩ)
47.5
35.7
28.7
23.7
20.5
17.8
15.8
14.3
12.7
11.5
Fs (kHz)
300
Iocset (μA)
29.4
39.2
48.7
59.07
68.2
78.6
88.6
97.9
110.2
121.7
400
500
600
700
800
900
1000
1100
1200
Fig. 7. Connection of over current sensing resistor
Shutdown
The IR3856W can be shutdown by pulling the
Enable pin below its 1 V threshold. This will tri-
state both, the high side driver as well as the low
side driver. Alternatively, the output can be
shutdown by pulling the soft-start pin below 0.3V.
Normal operation is resumed by cycling the
voltage at the Soft Start pin.
An over current is detected if the OCSet pin goes
below ground. Hence, at the current limit
threshold, VOCset=0. Then, for a current limit
setting ILimit, ROCSet is calculated as follows:
R
DS(on) * ILimit
ROCSet
=
......................(4)
IOCSet
An overcurrent detection trips the OCP
comparator, latches OCP signal and cycles the
soft start function in hiccup mode.
Over-Current Protection
The over current protection is performed by
sensing current through the RDS(on) of low side
MOSFET. This method enhances the converter’s
efficiency and reduces cost by eliminating a
current sense resistor. As shown in figure 7, an
external resistor (ROCSet) is connected between
OCSet pin and the switch node (SW) which sets
the current limit set point.
The hiccup is performed by shorting the soft-start
capacitor to ground and counting the number of
switching cycles. The Soft Start pin is held low
until 4096 cycles have been completed. The
OCP signal resets and the converter recovers.
After every soft start cycle, the converter stays in
this mode until the overload or short circuit is
removed.
An internal current source sources current (IOCSet
) out of the OCSet pin. This current is a function
of the switching frequency and hence, of Rt.
The OCP circuit starts sampling current typically
160 ns after the low gate drive rises to about 3V.
This delay functions to filter out switching noise.
14
Rev 9.0
PD-97506
IR3856WMPbF
1.5V <Vin<16V
4.5V <Vcc<5.5V
Thermal Shutdown
Temperature sensing is provided inside
IR3856W. The trip threshold is typically set to
140oC. When trip threshold is exceeded, thermal
shutdown turns off both MOSFETs and
discharges the soft start capacitor.
Enable
Vin
Boot
SW
PVcc
Vcc
Vo(master)
PGood
PGood
OCSet
Fb
RA
Automatic restart is initiated when the sensed
temperature drops within the operating range.
There is a 20oC hysteresis in the thermal
shutdown threshold.
Seq
Rt
RB
Comp
PGnd
SS/ SD
Gnd
Output Voltage Sequencing
1.5V <Vin<16V
4.5V <Vcc<5.5V
The
IR3856W
can
accommodate
user
programmable sequencing using Seq, Enable
and Power Good pins.
Enable
Vin
Boot
PVcc
Vcc
Vo(slave)
Vo(master)
PGood
RE
SW
PGood
OCSet
Fb
RC
Vo1
Vo2
Seq
Rt
RF
RD
Comp
PGnd
SS/ SD
Gnd
Simultaneous Powerup
Fig. 8b. Application Circuit for Simultaneous
Sequencing
Fig. 8a. Simultaneous Power-up of the slave
with respect to the master.
Power Good Output
The IC continually monitors the output voltage via
Feedback (Fb pin). The feedback voltage forms
an input to a window comparator whose upper
and lower thresholds are 0.805V and 0.595V
respectively. Hence, the Power Good signal is
flagged when the Fb pin voltage is within the
PGood window, i. e. between 0.595V to 0.805V,
as shown in Fig .9. The PGood pin is open drain
and it needs to be externally pulled high. High
state indicates that output is in regulation. Fig. 9a
shows the PGood timing diagram for non-
tracking operation. In this case, during startup,
PGood goes high after the SS voltage reaches
2.1V if the Fb voltage is within the PGood
comparator window. Fig. 9a. and Fig 9.b. also
show a 256 cycle delay between the Fb voltage
entering within the thresholds defined by the
PGood window and PGood going high.
Through these pins, voltage sequencing such as
simultaneous
and
sequential
can
be
implemented. Figure 8. shows simultaneous
sequencing configurations. In simultaneous
power-up, the voltage at the Seq pin of the slave
reaches 0.7V before the Fb pin of the master. For
RE/RF =RC/RD, therefore, the output voltage of
the slave follows that of the master until the
voltage at the Seq pin of the slave reaches 0.7 V.
After the voltage at the Seq pin of the slave
exceeds 0.85V, the internal 0.7V reference of
the slave dictates its output voltage.
It is recommended that irrespective of the
sequencing configuration used, the input voltage
should be allowed to come up to its nominal
value first, followed by Vcc and Enable, before the
sequencing signal is applied.
For non-sequenced operation, the Seq pin
should be tied to a voltage greater than 0.85V,
such as 3.3V or Vcc. Again, the input voltage
should be allowed to come up before Vcc and
Enable.
15
Rev 9.0
PD-97506
IR3856WMPbF
TIMING DIAGRAM OF PGOOD FUNCTION
Fig.9a IR3856W Non-Tracking Operation (Seq=Vcc)
Fig.9b IR3856W Tracking Operation
16
Rev 9.0
PD-97506
IR3856WMPbF
Minimum on time Considerations
Maximum Duty Ratio Considerations
The minimum ON time is the shortest amount of
time for which the Control FET may be reliably
turned on, and this depends on the internal
timing delays. For the IR3856W, the typical
minimum on-time is specified as 50 ns.
Any design or application using the IR3856W
must ensure operation with a pulse width that is
higher than this minimum on-time and preferably
higher than 100 ns. This is necessary for the
circuit to operate without jitter and pulse-
skipping, which can cause high inductor current
ripple and high output voltage ripple.
A fixed off-time of 200 ns maximum is specified
for the IR3856W. This provides an upper limit on
the operating duty ratio at any given switching
frequency. It is clear that, higher the switching
frequency, the lower is the maximum duty ratio at
which the IR3856W can operate. To allow a
margin of 50 ns, the maximum operating duty
ratio in any application using the IR3856W
should still accommodate about 250 ns off-time.
Fig 10. shows a plot of the maximum duty ratio
v/s the switching frequency, with 250 ns off-time.
95
90
85
80
75
70
D
ton
=
Fs
Vout
Vin × Fs
=
In any application that uses the IR3856W, the
following condition must be satisfied:
ton(min) ≤ ton
Vout
∴ ton(min)
≤
Vin × Fs
Vout
300
400
500
600
700
800
900
1000
1100
1200
Switching Frequency (kHz)
∴Vin × Fs ≤
ton(min)
The minimum output voltage is limited by the
reference voltage and hence Vout(min) = 0.7 V.
Therefore, for Vout(min) = 0.7 V,
Fig. 10. Maximum duty cycle v/s switching
frequency.
Vout(min)
∴ Vin × Fs
≤
ton(min)
0.7 V
∴ Vin × Fs ≤
= 7 ×106 V/s
100 ns
Therefore, at the maximum recommended input
voltage 21V and minimum output voltage, the
converter should be designed at a switching
frequency that does not exceed 330 kHz.
Conversely, for operation at the maximum
recommended operating frequency 1.2 MHz and
minimum output voltage, any voltage above
5.83V may not be stepped down reliably without
pulse-skipping.
17
Rev 9.0
PD-97506
IR3856WMPbF
when an external resistor divider is connected to
the output as shown in figure 11.
Equation (7) can be rewritten as:
Application Information
Design Example:
The following example is a typical application for
IR3856W. The application circuit is shown on
page 24.
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
Vref
R9 = R8 ∗
...............................(8)
Vo−Vref
Vin =12 V (13.2Vmax)
Vo =1.8 V
For the calculated values of R8 and R9 see
feedback compensation section.
Io =6 A
VOUT
ΔVo ≤ 54mV
Fs = 600kHz
IR3856W
R8
Fb
R9
Enabling the IR3856W
As explained earlier, the precise threshold of
the Enable lends itself well to implementation of
a UVLO for the Bus Voltage.
Fig. 11. Typical application of the IR3856W for
programming the output voltage
Soft-Start Programming
Vin
The soft-start timing can be programmed by
selecting the soft-start capacitance value. From
(1), for a desired start-up time of the converter,
the soft start capacitor can be calculated by
using:
IR3856W
R1
Enable
R2
CSS (μF) = Tstart ( ms )× 0.02857 .......... (9)
Where Tstart is the desired start-up time (ms).
For a start-up time of 3.5ms, the soft-start
capacitor will be 0.099μF. Choose a 0.1μF
ceramic capacitor.
For a typical Enable threshold of VEN = 1.2 V
R2
Vin(min)
*
= VEN = 1.2.......... (5)
R1 + R2
Bootstrap Capacitor Selection
VEN
R2 = R1
..........(6)
Vin( min ) −VEN
To drive the Control FET, it is necessary to
supply a gate voltage at least 4V greater than
the voltage at the SW pin, which is connected
the source of the Control FET . This is achieved
by using a bootstrap configuration, which
comprises the internal bootstrap diode and an
external bootstrap capacitor (C6), as shown in
Fig. 12. The operation of the circuit is as follows:
When the lower MOSFET is turned on, the
capacitor node connected to SW is pulled down
to ground. The capacitor charges towards Vcc
through the internal bootstrap diode, which has a
forward voltage drop VD. The voltage Vc across
the bootstrap capacitor C6 is approximately
given as
For a Vin (min)=10.2V, R1=49.9K and R2=7.5K is a
good choice.
Programming the frequency
For Fs = 600 kHz, select Rt = 23.7 kΩ, using
Table. 1.
Output Voltage Programming
Output voltage is programmed by reference
voltage and external voltage divider. The Fb pin
is the inverting input of the error amplifier, which
is internally referenced to 0.7V. The divider is
ratioed to provide 0.7V at the Fb pin when the
output is at its desired value. The output voltage
is defined by using the following equation:
Vc ≅ Vcc −VD ..........................(10)
When the upper MOSFET turns on in the next
cycle, the capacitor node connected to SW rises
to the bus voltage Vin. However, if the value of
C6 is appropriately chosen,
⎛
⎜
⎝
⎞
⎟
⎟
⎠
R8
R9
⎜
Vo =Vref ∗ 1+
................................(7)
18
Rev 9.0
PD-97506
IR3856WMPbF
the voltage Vc across C6 remains approximately
unchanged and the voltage at the Boot pin
becomes
advisable to have 2x10uF 25V ceramic capacitors
C3216X5R1E106M from TDK. In addition to
these, although not mandatory, a 1X330uF, 25V
SMD capacitor EEV-FK1E331P may also be used
as a bulk capacitor and is recommended if the
input power supply is not located close to the
converter.
VBoot ≅ Vin +Vcc −VD ........................................(11)
Inductor Selection
The inductor is selected based on output power,
operating frequency and efficiency requirements.
A low inductor value causes large ripple current,
resulting in the smaller size, faster response to a
load transient but poor efficiency and high output
noise. Generally, the selection of the inductor
value can be reduced to the desired maximum
ripple current in the inductor
point is usually found between 20% and 50%
ripple of the output current.
. The optimum
(Δi)
Fig. 12. Bootstrap circuit to generate
Vc voltage
For the buck converter, the inductor value for the
desired operating ripple current can be
determined using the following relation:
A bootstrap capacitor of value 0.1uF is suitable
for most applications.
For applications with 21V input voltage, the switch
node may ring above the 25V absolute maximum
voltage rating. To prevent this, in addition to
using best layout practices, it may be necessary
to provide a 10ohm resistor in series with the boot
capacitor.
Δi
Δt
1
Vin −Vo = L ∗ ; Δt = D ∗
Fs
............................... (14)
Vo
L =
(
Vin −Vo ∗
)
Vin ∗ Δi * Fs
Where:
Vin = Maximum input voltage
Vo = Output Voltage
Δi = Inductorripple current
Fs= Switching frequency
Δt = Turn on time
Input Capacitor Selection
The ripple current generated during the on time of
the upper MOSFET should be provided by the
input capacitor. The RMS value of this ripple is
expressed by:
D = Duty cycle
If Δi ≈ 42%(Io), then the output inductor is
calculated to be 1.01μH. Select L=1 μH.
IRMS =Io ∗ D∗(1−D) ........................(12)
Vo
The SPM6550T-1R0M from TDK provides a
compact inductor suitable for this application
D =
.............................(13)
Vin
Where:
D is the Duty Cycle
IRMS is the RMS value of the input capacitor
current.
Io is the output current.
For Io=6A and D = 0.15, the IRMS = 2.14 A.
Ceramic capacitors are recommended due to
their peak current capabilities. They also feature
low ESR and ESL at higher frequency which
enables better efficiency. For this application, it is
19
Rev 9.0
PD-97506
IR3856WMPbF
Output Capacitor Selection
The output LC filter introduces a double pole,
–40dB/decade gain slope above its corner
resonant frequency, and a total phase lag of 180o
(see figure 13). The resonant frequency of the LC
filter is expressed as follows:
The voltage ripple and transient requirements
determine the output capacitors type and values.
The criteria is normally based on the value of the
Effective Series Resistance (ESR). However the
actual capacitance value and the Equivalent Series
Inductance (ESL)
components. These components can be described
as
are other contributing
1
FLC
=
................................(16)
2∗π Lo ∗Co
ΔVo = ΔVo(ESR) + ΔVo(ESL) + ΔVo(C)
Figure 13 shows gain and phase of the LC filter.
Since we already have 180o phase shift from the
output filter alone, the system runs the risk of
being unstable.
ΔVo(ESR) = ΔIL * ESR
V
−Vo
⎛
⎞
in
ΔVo(ESL) = ⎜
⎟ * ESL
⎜
⎟
L
⎝
⎠
Gain
0 dB
Phase
00
.........................(15)
ΔIL
8 * Co * Fs
ΔVo(C)
=
-40dB/decade
Frequency
ΔVo = Output voltage ripple
ΔIL = Inductor ripple current
-1800
Frequency
FLC
FLC
Since the output capacitor has a major role in the
overall performance of the converter and
determines the result of transient response,
selection of the capacitor is critical. The
IR3856W can perform well with all types of
capacitors.
Fig. 13. Gain and Phase of LC filter
The IR3856W uses a voltage-type error amplifier
with high-gain (110dB) and wide-bandwidth. The
output of the amplifier is available for DC gain
control and AC phase compensation.
As a rule, the capacitor must have low enough
ESR to meet output ripple and load transient
requirements.
The error amplifier can be compensated either in
type II or type III compensation.
The goal for this design is to meet the voltage
ripple requirement in the smallest possible
capacitor size. Therefore it is advisable to select
ceramic capacitors due to their low ESR and ESL
and small size. Four of the Panasonic ECJ-
2FB0J226ML (22uF, 6.3V, 3mOhm) capacitors is
a good choice.
Local feedback with Type II compensation is
shown in Fig. 14.
This method requires that the output capacitor
should have enough ESR to satisfy stability
requirements. In general the output capacitor’s
ESR generates a zero typically at 5kHz to 50kHz
which is essential for an acceptable phase
margin.
Feedback Compensation
The IR3856W is a voltage mode controller. The
control loop is a single voltage feedback path
including error amplifier and error comparator. To
achieve fast transient response and accurate
output regulation, a compensation circuit is
necessary. The goal of the compensation
network is to provide a closed-loop transfer
function with the highest 0 dB crossing frequency
and adequate phase margin (greater than 45o).
The ESR zero of the output capacitor is
expressed as follows:
1
FESR
=
...........................(17)
2 ∗π*ESR*Co
20
Rev 9.0
PD-97506
IR3856WMPbF
Where:
Vin = Maximum Input Voltage
Vosc = Oscillator Ramp Voltage
Fo = Crossover Frequency
VOUT
ZIN
CPOLE
FESR = Zero Frequency of the Output Capacitor
FLC = Resonant Frequency of the Output Filter
R8 = Feedback Resistor
R3
C4
R8
Zf
To cancel one of the LC filter poles, place the
zero before the LC filter resonant frequency pole:
Fb
Ve
E/A
R9
Comp
Fz = 75%F
LC
VREF
Gain(dB)
1
Fz = 0.75*
.....................................(22)
H(s) dB
2π Lo *Co
Use equations (20), (21) and (22) to calculate
C4.
One more capacitor is sometimes added in
parallel with C4 and R3. This introduces one
more pole which is mainly used to suppress the
switching noise.
FPOLE Frequency
FZ
Fig. 14. Type II compensation network
and its asymptotic gain plot
The additional pole is given by:
1
FP =
.................................(23)
The transfer function (Ve/Vo) is given by:
C4 *CPOLE
C4 +CPOLE
2π *R3 *
Ve
Vo
Zf
1+sRC4
3
= H(s) = −
= −
.....(18)
ZIN
sRC4
8
The pole sets to one half of the switching
frequency which results in the capacitor CPOLE
:
The (s) indicates that the transfer function varies
as a function of frequency. This configuration
introduces a gain and zero, expressed by:
1
1
CPOLE
=
≅
....................(24)
1
π*R3*F
s
π*R3*F −
s
C4
R3
H
(
s
)
=
.....................................(19)
............................(20)
R8
For a general solution for unconditional stability
for any type of output capacitors, and a wide
range of ESR values, we should implement local
feedback with a type III compensation network.
The typically used compensation network for
voltage-mode controller is shown in figure 15.
1
Fz =
2π *R3 *C4
First select the desired zero-crossover frequency
(Fo):
Again, the transfer function is given by:
F > FESR andF ≤
(
1/5~1/10
)
*F
s
o
o
Ve
Vo
Zf
= H(s) = −
Use the following equation to calculate R3:
ZIN
By replacing Zin and Zf according to figure 15,
the transfer function can be expressed as:
Vosc *F *FESR*R8
o
R3 =
...........................(21)
V *F2
in
LC
(1+ sR3C4 )
1+ sC7
(
R8 + R10
)
]
H(s) = −
....(25)
⎡
⎤
⎞
⎟
⎟
⎠
⎛
⎜
⎜
⎝
C4 *C3
sR (C +C ) 1+ sR
(1+ sR C )
⎢
⎥
8
4
3
3
10 7
C4 +C3
⎢
⎥
⎦
⎣
21
Rev 9.0
PD-97506
IR3856WMPbF
VOUT
Compensator
Type
Output
Capacitor
ZIN
F
ESR vs Fo
C3
C7
R3
C4
Electrolytic
Tantalum
Type II
Type III
FLC<FESR<Fo<Fs/2
FLC<Fo<FESR
R
10
R8
Zf
Tantalum
Ceramic
Fb
Ve
E/A
R9
Comp
VREF
Gain(dB)
The higher the crossover frequency, the
potentially faster the load transient response.
However, the crossover frequency should be low
enough to allow attenuation of switching noise.
Typically, the control loop bandwidth or crossover
frequency is selected such that
H(s) dB
Frequency
FZ1
FZ2
FP2
FP3
F ≤
1/5~1/10
)
*F
o
s
The DC gain should be large enough to provide
high DC-regulation accuracy. The phase margin
should be greater than 45o for overall stability.
Fig.15. Type III Compensation network and
its asymptotic gain plot
The compensation network has three poles and
two zeros and they are expressed as follows:
For this design we have:
Vin=12V
Vo=1.8V
FP1 = 0............................................................(26)
Vosc=1.8V
1
Vref=0.7V
Lo=1 uH
Co=4x22uF, ESR=3mOhm each
FP2
FP3
=
=
...........................................(27)
2π *R10 *C7
1
1
≅
..............(28)
2π *R3 *C3
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
C4 *C3
C4 +C3
It must be noted here that the value of the
capacitance used in the compensator design
must be the small signal value. For instance, the
small signal capacitance of the 22uF capacitor
used in this design is 12uF at 1.8 V DC bias and
600 kHz frequency. It is this value that must be
used for all computations related to the
compensation. The small signal value may be
obtained from the manufacturer’s datasheets,
design tools or SPICE models. Alternatively, they
may also be inferred from measuring the power
stage transfer function of the converter and
measuring the double pole frequency FLC and
using equation (16) to compute the small signal
Co.
2π * R3
1
FZ1
FZ2
=
=
.........................................(29)
2π *R3 *C4
1
1
≅
..........(30)
2π *C7 *(R8 + R10) 2π *C7 *R8
Cross over frequency is expressed as:
V
1
in
F = R3 *C7 *
*
................................(31)
o
Vosc 2π *Lo *Co
Based on the frequency of the zero generated by
the output capacitor and its ESR, relative to
crossover frequency, the compensation type can
be different. The table below shows the
compensation types and location of the
crossover frequency.
These result to:
FLC=22.97 kHz
FESR=4.4 MHz
Fs/2=300 kHz
Select crossover frequency: Fo=100 kHz
Since FLC<Fo<Fs/2<FESR, TypeIII is selected to
place the pole and zeros.
22
Rev 9.0
PD-97506
IR3856WMPbF
Programming the Current-Limit
Detailed calculation of compensation Type-III
The Current-Limit threshold can be set by
connecting a resistor (ROCSET) from the SW pin
to the OCSet pin. The resistor can be calculated
by using equation (4). This resistor ROCSET must
be placed close to the IC.
o
Desired Phase Margin Θ=70
1−sin Θ
1+sin Θ
F =F
=17.63 kHz
Z2
o
1+sin Θ
1−sin Θ
The RDS(on) has
a
positive temperature
F
P2 =F
=567.1kHz
o
coefficient and it should be considered for the
worst case operation.
Select: F =0.5*F = 8.82 kHz and
Z1
Z2
ROCSet∗IOCSet
F =0.5*F =300 kHz
P3
s
ISET =IL(critica)l
=
.......................(32)
RDS(on)
Selec:tC7 =2.2nF
Calculate R3, C3 and C4 :
RDS(on) =14.3 mΩ*1.25=17.875mΩ
ISET ≅ Io(LIM) = 6 A*1.5= 9 A
2π*F *L *Co *V
o
o
R3 =
osc ;R =2.56 kΩ
3
(50% over nominal output current )
IOCSet = 59.07μA (at Fs = 600kHz)
ROCSet = 2.694kΩ Select R7 = 2.67 kΩ
C7*V
in
Select: R3 =2.55kΩ
1
C4 =
C3 =
;C4 =8.84 nF, Selec:t C4 =10 nF
;C3 =258.79pF, Selec:tC3 =220pF
Setting the Power Good Threshold
2π *F *R
Z1
3
Power Good threshold is internally set at 88% of
Vref. When the voltage at the FB pin exceeds the
threshold, PGood is asserted.
1
2π*F *R3
P3
Calculate R , R8 and R9 :
The PGood is an open drain output. Hence, it is
necessary to use a pull up resistor RPG from
PGood pin to Vcc. The value of the pull-up
resistor must be chosen such as to limit the
current flowing into the PGood pin, when the
output voltage is not in regulation, to less than
5mA. A typical value used is 10kΩ.
10
1
R =
; R =128Ω, Selec:tR =130Ω
10 10
10
2π*C7*F
P2
1
R8 =
-R ; R8 =3.97kΩ,
10
2π*C7*F
Z2
Selec:tR8 =4.02kΩ
V
ref
R9 =
*R ;R9 =2.57kΩ Selec:tR =2.55kΩ
8 9
V -V
o
ref
23
Rev 9.0
PD-97506
IR3856WMPbF
Application Diagram:
Fig. 16. Application circuit diagram for a 12V to 1.8 V, 6 A Point Of Load Converter
Suggested Bill of Materials for the application circuit:
Part Reference Quantity Value
Description
SMD Elecrolytic, Fsize, 25V, 20%
1206, 16V, X5R, 20%
0603, 25V, X7R, 10%
6.9x6.5x5mm, 20%, 4.7mOhm
0805, 6.3V, X5R, 20%
Thick Film, 0603,1/10 W,1%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
Manufacturer
Panasonic
TDK
Panasonic
TDK
Panasonic
Rohm
Rohm
Part Number
EEV-FK1E331P
1
2
1
1
4
1
1
1
330uF
10uF
0.1uF
1.0uH
22uF
49.9k
7.5k
Cin
C3216X5R1E106M
ECJ-1VB1E104K
SPM6550T-1R0M100A
ECJ-2FB0J226ML
MCR03EZPFX4992
MCR03EZPFX7501
MCR03EZPFX2372
Lo
Co
R1
R2
Rt
23.7k
Rohm
Rocset
RPG
1
1
2.67k
10k
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
Rohm
Rohm
MCR03EZPFX2671
MCR03EZPFX1002
C
ss C6
2
1
1
1
1
1
1
1
2
1
0.1uF
2.05k
220pF
10nF
4.02k
2.55k
130
2200pF
1.0uF
IR3856W
0603, 25V, X7R, 10%
Thick Film, 0603,1/10W,1%
50V, 0603, NPO, 5%
Panasonic
Rohm
Panasonic
Panasonic
Rohm
ECJ-1VB1E104K
MCR03EZPFX2051
ECJ-1VC1H221J
ECJ-1VB1H103K
MCR03EZPFX4021
MCR03EZPFX2551
ERJ-3EKF1300V
ECJ-1VB1H222K
ECJ-BVB1C105M
R3
C3
C4
R8
R9
R10
C7
0603, 50V, X7R, 10%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
0603, 50V, X7R, 10%
0603, 16V, X5R, 20%
SupIRBuck, 6A, PQFN 4x5mm
Rohm
Rohm
Panasonic
Panasonic
International Rectifier IR3856WMPbF
C
U1
Vcc CPVcc
24
Rev 9.0
PD-97506
IR3856WMPbF
TYPICAL OPERATING WAVEFORMS
Vin=12.0V, Vcc=5V, Vo=1.8V, Io=0-6A, Room Temperature, No Air Flow
Fig. 17. Start up at 6A Load
Fig. 18. Start up at 6A Load,
Ch1:Vin, Ch2:Vo, Ch3:Vss, Ch4:Enable
Ch1:Vin, Ch2:Vo, Ch3:Vss, Ch4:VPGood
Fig. 20. Output Voltage Ripple, 6A
load Ch3: Vo
Fig. 19. Start up with 1.62V Pre
Bias, 0A Load, Ch2:Vo, Ch3:VSS
Fig. 21. Inductor node at 6A load
Ch3:LX
Fig. 22. Short (Hiccup) Recovery
Ch2:Vo , Ch3:VSS
25
Rev 9.0
PD-97506
IR3856WMPbF
TYPICAL OPERATING WAVEFORMS
Vin=12V, Vcc=5V, Vo=1.8V, Io=3A-6A, Room Temperature, No Air Flow
Fig. 23. Transient Response, 3A to 6A step
2.5A/μs
Ch3:Vo, Ch4:Io
26
Rev 9.0
PD-97506
IR3856WMPbF
TYPICAL OPERATING WAVEFORMS
Vin=12V, Vcc=5V, Vo=1.8V, Io=6A, Room Temperature, No Air Flow
Fig. 24. Bode Plot at 6A load shows a bandwidth of 104kHz and phase margin of 54 degrees
27
Rev 9.0
PD-97506
IR3856WMPbF
Simultaneous Tracking at Power Up and Power Down
Vin=12V, Vo=1.8V, Io=6A, Room Temperature, No Air Flow
3.3V
VOUT
IR3856W
4.02K
2.55K
R
8
Rs1
4.02K
Seq
Fb
2..55K
Rs2
R9
Fig. 25: Simultaneous Tracking a 3.3V input at power-up and shut-down with 6A load.
Ch2: Vo2 (1.8Vout) Ch3:SS(1.8V) Ch4: Vo1 (3.3V)
28
Rev 9.0
PD-97506
IR3856WMPbF
Layout Considerations
The connection between the OCSet resistor and
the Sw pin should not share any trace with the
connection between the bootstrap capacitor and
the Sw pin. Instead, it is recommended to use a
Kelvin connection of the trace from the OCSet
The layout is very important when designing high
frequency switching converters. Layout will affect
noise pickup and can cause a good design to
perform with less than expected results.
Make all the connections for the power
components in the top layer with wide, copper
filled areas or polygons. In general, it is desirable
to make proper use of power planes and
polygons for power distribution and heat
dissipation.
The inductor, output capacitors and the IR3856W
should be as close to each other as possible.
This helps to reduce the EMI radiated by the
power traces due to the high switching currents
through them. Place the input capacitor directly at
the Vin pin of IR3856W.
The feedback part of the system should be kept
away from the inductor and other noise sources.
The critical bypass components such as
capacitors for Vcc should be close to their
respective pins. It is important to place the
feedback components including feedback
resistors and compensation components close to
Fb and Comp pins.
Vin
PGnd
resistor and the trace from the bootstrap
Vin
PGnd
capacitor at the Sw pin.
In a multilayer PCB use one layer as a power
ground plane and have a control circuit ground
Vout
AGnd ground), to which all signals are
(analog
referenced. The goal is to localize the high
current path to a separate loop that does not
iAnGtenrdfere with the more sensitive analog control
Vout
function. These two grounds must be connected
together on the PC board layout at a single point.
The Power QFN is a thermally enhanced
package. Based on thermal performance it is
recommended to use at least a 4-layers PCB. To
effectively remove heat from the device the
exposed pad should be connected to the ground
plane using vias. Figure 26 illustrates the
implementation of the layout guidelines outlined
above, on the IRDC3856W 4 layer demoboard.
Compensation parts
should be placed as close
as possible to
the Comp pin.
Vout
Vin
All bypass caps
should be placed as
close as possible to
their connecting
pins.
AGnd
Vin
Resistors Rt, Rocset,
and SS capacitor
should be placed as
close as possible to
their pins.
PGnd
Fig. 26a. IRDC3856W demo-board layout
considerations – Top Layer
29
Rev 9.0
PD-97506
IR3856WMPbF
PGnd
Fig. 26b. IRDC3856 demoboard layout
considerations – Bottom Layer
Analog
Ground
plane
Vin
Single point
connection between
AGND & PGND,
should be close to
the SupIRBuck, kept
away from noise
sources.
Power
Ground
Plane
AGnd
Fig. 26c. IRDC3856 demoboard layout
considerations – Mid Layer 1
Feedback trace
should be kept
away form
noise sources
Fig. 26d. IRDC3856 demoboard layout
considerations – Mid Layer 2
30
Rev 9.0
PD-97506
IR3856WMPbF
PCB Metal and Components Placement
31
Rev 9.0
PD-97506
IR3856WMPbF
Solder Resist
32
Rev 9.0
PD-97506
IR3856WMPbF
Stencil Design
33
Rev 9.0
PD-97506
IR3856WMPbF
IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105
TAC Fax: (310) 252-7903
This product has been designed and qualified for the Industrial market (Note5)
Visit us at www.irf.com for sales contact information
Data and specifications subject to change without notice. 08/12
34
Rev 9.0
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