ISL6401CB-T [INTERSIL]
Synchronizing Current Mode PWM for Subscriber Line Interface Circuits (SLICs); 同步电流模式PWM用于用户线路接口电路（ SLIC组件）
| 型号： | ISL6401CB-T |
| 厂家： | 
Intersil |
| 描述： | Synchronizing Current Mode PWM for Subscriber Line Interface Circuits (SLICs) |
| 文件： | 总11页 (文件大小：330K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
ISL6401
®
Data Sheet
April 13, 2005
FN9007.7
Synchronizing Current Mode PWM for
Subscriber Line Interface Circuits (SLICs)
Features
• Positive and Negative Output Regulation
The ISL6401 adjustable frequency, low power, pulse width
modulating (PWM) current mode controller is designed for a
wide range of DC/DC conversion applications including
boost, flyback, and isolated output configurations. The
device is optimized to provide a high performance, low-cost
solution for Ringing SLIC (RSLIC) ring and talk power
supplies. An integrated inverter allows for easy design of
negative voltage regulation circuits with a minimal amount of
external components. Internal soft-start minimizes start-up
stress without any external components. Peak current mode
control effectively handles Ring trip transients and provides
inherent overcurrent protection.
• Starting Supply Current . . . . . . . . . . . . . . . . . 100µA (typ.)
• Quiescent Current . . . . . . . . . . . . . . . . . . . . . . 55µA (typ.)
• Output Frequency Range. . . . . . . . . . . . 50kHz to 600kHz
• External Clock Synchronization
• Fast Transient Response with Peak Current Mode Control
• Internal Soft-Start
• Drives N-Channel MOSFET
• Logic Level Shutdown
• Leading Edge Blanking
This advanced BiCMOS design features low operating
current, adjustable operating output frequency (50kHz to
600kHz), internal soft-start and a SYNC input that allows the
oscillator to be locked to an external clock for noise sensitive
applications. DC/DC conversion efficiency is optimized by
use of a low current sense voltage. A logic level shutdown
input is included, which reduces supply current to 55µA in
the shutdown mode.
• 1% Tolerance Voltage Reference Over Line and
Temperature
• Pb-Free Available (RoHS Compliant)
Applications
• VoIP/VoDSL Ringer and Off-Hook Voltage Generators
• Multi-Output Flyback Supplies
• Cable and DSL Modems
• Set-Top Boxes
Pinouts
ISL6401 (SOIC)
TOP VIEW
SD
SYNC
CT
1
2
3
4
5
6
7
14 V
CC
• Wireless Local Loops
• LMDH and FTTH Supplies
• Boost Regulators
13 PV
CC
12 GATE
11 PGND
10 GND
COMP
FB
• Routers
Ordering Information
NFB OUT
NFB IN
9
8
CS
NC
TEMP.
PKG.
PART NUMBER RANGE (°C)
PACKAGE
DWG. #
ISL6401CB
ISL6401CBZ (Note) -40 to 85 14 Ld SOIC (Pb-free)
ISL6401CR -40 to 85 16 Ld 4x4 QFN
-40 to 85 14 Ld SOIC
M14.15
M14.15
L16.4x4
ISL6401 (QFN)
TOP VIEW
ISL6401CRZ (Note) -40 to 85 16 Ld 4x4 QFN (Pb-free) L16.4x4
Add -T suffix for tape and reel packaging.
NOTES:
1. The parts with suffix ‘C’ are being tested to the industrial temperature
grade. This has been updated in the table above. The IB and IR parts are
being discontinued. The affected parts are: ISL6401IB, ISL6401IBZ,
ISL6401IB-T, ISL6401IBZ-T, ISL6401IR, ISL6401IRZ, ISL6401IR-T,
ISL6401IRZ-T.
16 15 14 13
PGND
SYNC
1
2
3
4
12
11
10
9
PGND
GND
CS
CT
COMP
FB
2. Intersil Pb-free products employ special Pb-free material sets; molding
compounds/die attach materials and 100% matte tin plate termination
finish, which are RoHS compliant and compatible with both SnPb and Pb-
free soldering operations. Intersil Pb-free products are MSL classified at
Pb-free peak reflow temperatures that meet or exceed the Pb-free
requirements of IPC/JEDEC J STD-020.
5
6
7
8
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-352-6832 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2001-2005. All Rights Reserved
1
All other trademarks mentioned are the property of their respective owners.
ISL6401
Functional Block Diagram
10
14
1
SD
3
CT
2
GND
VCC
SYNC
200K
VREF
VOLTAGE
OSC
4
5
COMP
ERROR
REFERENCE
IBIAS
13
PVCC
GATE
AMP
ICSCOMP
+
-
ILIM
-
S
R
+
FB
CS
12
11
Q
SOFT-START
TFF
CLK
R
Q
CLK
POR
SOFT-
PWRGND
UVLO
ENPWM
START
VREF
V
CC
LVC
LEADING
EDGE
BLANKING
-
9
+
-
CS
+
ICLCOMP
NFB
AMP
NFB_IN
7
NFB-OUT
6
NC
8
Typical Application Schematic for 4 Line VoIP
VIN
C2
C1A-E
2.2µF
35V
9V TO
20V
1µF
50V
8
T1
2
VCC
GND
0
0
0
0
0
0
SEC
0 7
3
C10A
C11
1µF
PRI
1
+
+5V
330µF
±10%
D1
35V
VOUT1
VOUT2
-24V,
120mA
C3
MUR5120T3
1µF
16V
0
6
4
GND
C4
D2
560pF
MUR5160T3
5
-72V,
0
SD
V
CC
1
2
3
4
14
13
12
11
120mA
C5
IFLY0012
R7
220
C12
E
SYNC PV
CC
220pF
0.1µF
C9
Q1
100µF
CT
GATE
C6
+
R1
20K
IRLR2905
C8
100V
COMP GNDP
330pF
0.027µF
R6
5
6
7
10
9
FB
GND
CS
100
NFB
R2
8
NFB_IN
NC
10K
R5
R3
ISL6401
0.025
1W
R4B
R4A
43.5K
1.24K
C7
1000pF
143K
NOTES:
3. C2 fit as close as possible to transformer.
4. T1 = IFLY0012 contacts: Coilcraft: (847) 516-7377 GCI Technology: (972) 423-8411 ext. 245
5. For custom specific designs or questions please contact Intersil at 1-888-INTERSIL or 321-724-7143.
FN9007.7
2
April 13, 2005
ISL6401
Absolute Maximum Ratings
Thermal Information
Thermal Resistance (Typical)
14 Lead SOIC (Note 6) . . . . . . . . . . . .
16 Lead QFN (Note 7) . . . . . . . . . . . . .
Supply Voltage, V
PV
. . . . . . . . . . . . . . . .GND -0.3V to +7.0V
θ
(°C/W)
90
46
θ
(°C/W)
JC
NA
9
CC, CC
JA
PGND to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±0.3V
Peak GATE Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .1A
ESD Classification . . . . . . . . . . . . . . . . . . . . . Class 1 (HBM, 2500V)
NFB Pin Voltage. . . . . . . . . . . . . . . . . . . . . . ±10V (Transient, 10ms)
Maximum Junction Temperature (Plastic Package) .-55°C to 150°C
Maximum Storage Temperature Range. . . . . . . . . . .-65°C to 150°C
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . . 300°C
(SOIC - Lead Tips Only)
Operating Conditions
Temperature Range
ISL6401C . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to 85°C
Supply Voltage Range (Typical). . . . . . . . . . . . . . . . . . . . . 5V ±10%
For Recommended soldering conditions see Tech Brief TB389.
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTES:
6. θ is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
JA
7. θ is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. θ
the
JC,
JA
“case temp” is measured at the center of the exposed metal pad on the package underside. See Tech Brief TB379.
8. All voltages are with respect to GND.
Electrical Specifications Recommended operating conditions unless otherwise noted. Refer to Block Diagram and Typical Application
schematic. V
= +5.0V ±10%, T = -40 to 85°C (Note 9), Typical values are at T = 25°C
CC
A
A
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
V
SUPPLY
CC
Supply Voltage Range
Shutdown Supply Current
Start-Up Current
4.5
5.0
55
5.5
100
0.20
6.0
V
SHDN = GND
< 3.7V
-
-
-
µA
mA
mA
V
0.1
3.7
CC
(Note 10)
Operating Supply Current
REFERENCE VOLTAGE
Output Voltage
1.237
-
1.25
5
1.262
-
V
Long Term Stability
NEGATIVE FEEDBACK
Source Current
T
= 125°C, 1000 hours
mV
A
-
1.0
2.0
mA
CURRENT SENSE
Maximum Input Signal
Input Bias Current
0.2
-2.0
0.4
0.260
0.0
0.3
2.0
0.6
V
µA
V
Overcurrent Threshold
ERROR AMPLIFIER
Open Loop Voltage Gain
Gain-Bandwidth Product
Input Voltage
0.52
-
10
78
-
-
dB
MHz
V
-
1.275
-
1.225
-
1.25
1.0
Input Bias Current
µA
PWM
Maximum Duty Cycle
Minimum Duty Cycle
UNDERVOLTAGE LOCKOUT
Start Threshold
47
-
48
0
50
-
%
%
COMP = 0V
3.7
3.2
4.1
3.6
4.3
4.0
V
V
Stop Threshold
FN9007.7
April 13, 2005
3
ISL6401
Electrical Specifications Recommended operating conditions unless otherwise noted. Refer to Block Diagram and Typical Application
schematic. V
= +5.0V ±10%, T = -40 to 85°C (Note 9), Typical values are at T = 25°C (Continued)
CC
A
A
PARAMETER
Start to Stop Hysteresis
TEST CONDITIONS
MIN
TYP
MAX
UNITS
0.2
0.5
0.8
V
SOFT-START - DIGITAL
COMP Rise Time
Rise from 0.5V to REF -1V
Ct = 560pF
-
2048 clk
-
OSCILLATOR
Gate Output Frequency
Gate Output Frequency Range
Temperature Stability
Sync. Frequency Range
90
50
-
100
108
600
-
kHz
kHz
%
-
5
-
1.1 Times the natural switching
frequency.
-
1.2
MHz
Sync Input HIGH
Sync Input LOW
3.5
-
-
-
1.5
-
V
V
-
-
Minimum Sync. Input Pulse Duty Cycle
OUTPUT
20
%
GATE Low Level
-
4.4
-
0.2
4.9
35
0.5
5.5
-
V
V
GATE High V
Rise Time
Fall Time
SAT
C load = 1500pF
C load = 1500pF
ns
ns
-
40
-
NOTES:
9. Specifications at -40°C are guaranteed by design, not production tested.
10. This is the V current consumed when the device is active but not switching. Does not include gate drive current.
CC
Typical Performance Curves
2.0
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
120
100
80
60
40
20
0
-40
-20
25
60
85
-40
-20
25
TEMPERATURE (°C)
60
85
TEMPERATURE (°C)
FIGURE 1. FREQUENCY vs TEMPERATURE
FIGURE 2. REFERENCE VOLTAGE vs TEMPERATURE
FN9007.7
April 13, 2005
4
ISL6401
Typical Performance Curves (Continued)
5.0
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
600
500
400
300
200
100
0
82
120 180 250 300 390 510 610 820 1200
-40
-20
25
60
85
TEMPERATURE (°C)
CAPACITANCE (pF)
FIGURE 3. SUPPLY CURRENT vs TEMPERATURE
FIGURE 4. CAPACITANCE vs FREQUENCY
series resistor for negative output voltage regulation
applications.
Pin Descriptions
SD - This pin is logic level compatible and can be pulled
high, tied to V or left open for normal operation. Logic low
IN
CS - This is the input of the current sense comparator. The
IC has two different comparators: The PWM comparator and
an overcurrent comparator.
on the SD activates shutdown, reducing the part’s supply
current to approximately 55µA.
SYNC - This pin is the input pin for external frequency
synchronization. The switching frequency of the device can
be synchronized by an external clock signal inserted at this
The overcurrent comparator is only intended for fault
sensing, and exceeding the overcurrent threshold will cause
a soft-start cycle.
pin. The oscillator timing capacitor, C , is still required, even
T
GND - GND is a small signal reference ground for all analog
functions on this part.
if an external clock is used. Program the free-running
frequency to be a minimum of 10% slower than the SYNC
input frequency.
PGND - This pin provides a dedicated ground for the output
gate driver. The GND and PGND pins should be connected
externally using a short printed circuit board trace close to
the IC. This is imperative to prevent large, high frequency
switching currents flowing through the ground metallization
CT - This is the oscillator timing pin. The free-running
frequency can be set by connecting a timing capacitor to this
pin. The oscillator produces a sawtooth waveform with a
programmable frequency range of 100kHz to 1.2MHz.
Figure 4 may be used as a guideline in selecting the
capacitor value required for a given frequency.
inside the IC. (Decouple PV
0.1µF capacitor.)
to PGND with a low ESR
CC
GATE - This is the device output. It is a high current power
driver capable of driving the gate of a power MOSFET with
peak currents exceeding 1.0A. This GATE output is actively
COMP - COMP is the output of the error amplifier and input
of the current comparator.
The ISL6401 features built-in full cycle soft-start. Soft-start is
implemented as a clamp on the maximum COMP voltage.
held low when V
is below the UVLO threshold (3.7V typ).
CC
The high-current power driver consists of FET output
FB - Feedback pin that is used for positive output voltage
sensing. It is the inverting input of the error amplifier. The
non-inverting input of the error amplifier is internally tied to a
reference voltage.
devices, which can switch all the way to GND and all the way
to V . The output stage also provides very low impedance
CC
to overshoot and undershoot.
PV
- This pin is for separate collector supply to the output
and PGnd helps decouple the
IC’s analog circuitry from the high power gate drive noise.
CC
NFB-IN - Negative feedback pin that is used for negative
output voltage sensing. It is connected to the inverting input
of the negative feedback amplifier through a 100K source
resistor.
gate drive. Separate PV
CC
Connect this pin to V
circuit board.
with external short trace on printed
CC
NFB OUT - This pin is the output of the negative feedback
inverter. This pin should be connected the FB pin with a 10K
V
- V is the power connection for the device. Although
CC
CC
quiescent V
current is very low, total supply current will be
CC
FN9007.7
5
April 13, 2005
ISL6401
higher, depending on the output current. Total V
current is
the error amplifier output (COMP pin) and the reference
CC
current and the average
the sum of the quiescent V
input (non-inverting terminal of the error amplifier) to the
internally generated soft-start voltage. The oscillator
sawtooth waveform is compared to the ramping error
amplifier voltage. This generates GATE pulses of increasing
width that charge the output capacitor(s). With sufficient
output voltage, the clamp on the reference input controls the
output voltage. When the internally generated soft-start
voltage exceeds the FB pin voltage, the output voltage is in
regulation. This method provides a rapid, controlled output
voltage rise. Soft-start is implemented during start-up, after
an overcurrent has cleared, or when exiting shutdown or
undervoltage lock-out (UVLO).
CC
output current. Knowing the operating frequency and the
MOSFET gate charge (Qg), average output current can be
calculated from:
I
= Qg × F
OUT
To prevent noise problems, bypass V
to GND with a
CC
pin as possible. An
ceramic capacitor as close to the V
CC
electrolytic capacitor may also be used in addition to the
ceramic capacitor.
Functional Description
Features
Gate Drive
The ISL6401 current mode, synchronizable PWM, makes an
ideal choice for low-cost, low-power, multi-output flyback
topology applications with low input-output ripple current
requirements. When configured in a multi-winding flyback
topology, the IC is capable of generating the negative Talk
and Ring voltages required for Ringing Subscriber Line
Interface (RSLIC) power supplies. This approach provides
dual outputs from a single power switch and control IC. Low
current sense voltage and shutdown mode leads to high
efficiency operation. Other features include peak current
mode control, internal soft-start, adjustable current limit,
adjustable frequency and external frequency
The ISL6401 is capable of sourcing 1A of peak-drive current.
Separate collector supply (PV ) and power ground (PGnd)
CC
pins help isolate the IC’s analog circuitry from the high power
gate drive noise. To limit the peak current through the IC, an
external resistor is placed between the totem-pole output of
the IC and the gate of the MOSFET. The minimum value of
this resistor is determined by:
Rgate = (Vdd(min) - Vsat) / Igate(peak)
This small series resistor also damps any oscillations
caused by the resonant tank of the parasitic inductances in
the traces of the board and the FET’s input capacitance. A
pull-down resistor is sometimes added to the gate drive to
insure the MOSFET gate does not get charged to its turn-on
threshold during device start-up. Adding a fast-switching
diode and smaller value resistor in parallel with the gate
resistor helps to control the current the IC needs to sink
during turn-off and protects the output stage of the device.
These components also help to reduce turn-off losses, which
tend to dominate the switching losses in discontinuous
current-mode (DCM) converters.
synchronization.
Oscillator
The ISL6401 has an internal sawtooth oscillator with a
programmable frequency range of 100kHz to 1MHz, which
can be programmed with a capacitor on the CT pin. (Please
refer to Figure 4 for the capacitance required for a given
frequency.) With a maximum 50% duty cycle operation, the
output switching frequency is half the oscillator frequency.
Implementing Synchronization
Ground Plane Requirements
The oscillator can be synchronized by an external clock
inserted at the SYNC pin. Program the free running
frequency of the oscillator to be 10% slower than the desired
synchronous frequency. The external clock signal should
have a minimum pulse width of 20ns.
Careful layout is essential for correct operation of the device.
A good ground plane must be employed. A unique section of
the ground plane must be designated for high di/dt currents
associated with the output stage. Power ground (PGND) can
be separated from the analog ground (GND) and connected
at a single point. V
should be bypassed directly to PGND
Soft-Start Operation
CC
with good high frequency capacitors. The return connection
for input power to the system and the bulk input capacitor
should be connected to the PGND ground plane.
The ISL6401 features an internal digital soft-start with no
external capacitor required. Soft-start is used to reduce
transformer and output capacitor stress and to reduce the
surge on the input circuits, when the converter action starts.
The considerable capacitance on the output lines should be
charged slowly, so as not to reflect an excessive transient. A
very wide initial pulse could result in saturation of the core
and voltage overshoot on the output, if the inductor current is
allowed to rise to a high value during start-up.
Application Information
Subscriber Line Interface Circuit Requirements
As worldwide demand for inexpensive Voice over Internet
Protocol telephony grows, so will the need for ICs that
enable compatibility between new telephony systems and
older telephones based on analog standards. Old style
telephones require signal and power inputs that are not
generally available on purely digital systems. Analog ring
Upon start-up, the peak primary current increments from
1/5th of the value set by R
to the full current limit value in
CS
steps, over 2048 cycles of Fosc or Fsync. Soft-start clamps
FN9007.7
6
April 13, 2005
ISL6401
signal generation and off-hook loop current supply are two
analog functions that are performed by Subscriber Line
Interface Circuits (SLICs). A SLIC is the primary interface
between the 4-wire (ground referenced) low voltage switch
environment and the 2 wire (floating) high voltage loop
environment. It performs a number of important functions
including battery feed, overvoltage protection, ringing,
signaling, coding, hybrid balancing and testing.
sensed by the primary side controller feedback
comparator.
• Capacitors C9 to C12 filter out high frequency noise on the
output bus directly at the output diode.
• R7 and C8 provide secondary side snubbing.
• R6 and C7 filter out the leading edge voltage spikes
resulting from the leakage inductance of the transformer.
• C4 sets the switching frequency of the converter.
The Ringing SLIC (RSLIC) typically requires two high
voltage power supply inputs. The first is a tightly regulated
voltage around -24V or -48V for off-hook voice transmission.
The second is a loosely regulated -70 to -100V for ring tone
generation. When the switch hook is released the phone
puts approximately 200Ω of resistance across the phone
terminals. Once voice transmission begins, the SLIC
requires a lower voltage input to establish a current loop of
approximately 25mA. The loop feeds the 200Ω, protection
resistors, and line resistances within the phone.
• C3 is a decoupling capacitor, which should always be a
good quality low-ESR/ESL type capacitor, placed as close
to the IC pins as possible and returned directly to the IC
ground reference.
• The gate drive circuitry can be composed of a small gate
drive resistor, necessary for damping any oscillations
resulting from the input capacitance of Q1 and any
parasitic stray inductance.
• The voltage sense feedback loop is comprised of R4 and
R3. Feedback components R1, C6, and C5 provide the
necessary gain and pole to stabilize the control loop.
ISL6401 Flyback Reference Design
The Typical Application Schematic shows a current mode
power supply using the Intersil ISL6401 in a standard
flyback topology. The IC requires +5V Bias. The application
circuit is intended for wall adapters that power home
gateway/router boxes. This circuit input voltage can be 9V
to 20V with the selected transformer and external
components.
Component Selection Guidelines
Power MOSFET
The MOSFET switch is selected to meet the drain to source
voltage stress resulting from the maximum input voltage
(V
), the reflected secondary voltages, equal to the
IN(max)
output voltage (V
), plus the output diode voltage drop
OUT
(V ), and the voltage spike due to the leakage inductance,
F
assumed to be 30% of the input voltage.
The output voltages are -24V at 120mA and -72V at
120mA. The circuit uses inexpensive transformers to
generate both outputs using a single controller. The
transformer turns ratio is such that 24V appear across each
secondary winding and the primary during the switch off-
time. The remaining secondary windings are stacked in
series to develop -48V. The -48V section is then stacked on
the -24V section to get the -72V. This technique provides
good cross regulation, lowers the voltage rating required
for the output capacitors, and lowers the RMS current,
allowing the use of less expensive output capacitors. Also,
the selection of a transformer with multifilar winding lowers
the leakage inductance and cost. The -24V output is
precisely regulated by feeding back this output to the
controller. The -72V output is derived from the third pair of
windings. Regulation of this output is obtained by the turn’s
ratio of the transformer with -24V output, as well as with
split feedback.
Vds (stress) = [(V
) + (N)(Vout +Vf)] + (0.3)(V )
IN(max) IN(max)
The switch must also be able to conduct the repetitive peak
primary current as determined by:
Ipeak (primary) = (Vin
- Vds) (t
) / Lp
ON(max)
min
The primary current waveform of a discontinuous mode
flyback converter is triangular in shape, therefore, its root
mean square(rms) current is calculated by:
Irms(prim)= (IPEAKprim ⁄ 3)( (TONmax) ⁄ T)
The chosen device should also have a low R
DS(ON)
value,
because the conduction losses of the device are proportional
to the square of the primary rms current through the device.
Selection of a device that has a peak current rating of at
least three times the peak current usually insures acceptably
low conduction losses.
Circuit Element Descriptions
2
Pconduction = (I
) (R
)
DS(on)
prms
• Transformers T1, MOSFET Q1, Schottky diode D1, D2,
and input capacitor C1 and C2 form the power stage of
the converter. Power resistor R5 senses the switch
current and converts this current into a voltage to be
FN9007.7
7
April 13, 2005
ISL6401
Switching losses are the result of overlapping drain current
and source voltage at turn-off. The drain voltage begins to
rise only after the miller capacitance of the device begins to
discharge. This discharging time is a function of the external
gate resistance, Rgate and the gate-to-drain miller charge
Qgd, as shown in the following equation,
where Ipeak (sec) is the peak-secondary current and t
RESET
is equal to the off-time of the switch. The same selection
criteria is used for the input capacitor, keeping in mind these
capacitors must also be rated to handle the maximum input
voltage.
Output Voltage
T miller = (Qgd)(R
)/(Vdd-Vth),
GATE
The output voltage can be set by a feedback resistor divider
network. The output is resistively divided and compared to
the reference voltage. For negative flyback output
applications, the sensed output will be fed to the NFB IN pin.
The sensed voltage in inverted, and this positive voltage is
fed to the FB- inverting input pin of the error amplifier. The
non-inverting input of the error amplifier will be a reference
voltage. So, when FB- is higher than REF voltage, the output
drivers are turned off. The opposite happens when the
resistively divided output voltage falls below the 1.24V
reference voltage.
where Vth is the turn ON threshold voltage of the gate.
The power loss due to the external capacitance of the
MOSFET also contributes to the total switching losses,
which can be calculated as shown.
C
× V
2
oss
DS(stress)
P
= F
------------------------------------------------------- + V
switching
sw
DS(stress)
2
+ I
+ t
miller
peak(primary)
During turn on there is no overlap of drain voltage and
current because there is no current in a discontinuous
current mode converter at turn-on. Minimal losses also occur
during the off-time of the FET due to the leakage current.
Output Diode
The output diode in a flyback converter is subject to large
peak and rms current stresses. Schottky diodes are
recommended, because of their low forward-voltage drop and
the virtual absence of minority carrier reverse recovery. The
secondary-side Schottky rectifier was selected to meet the
working peak-reverse voltage, the peak repetitive forward-
current and the average forward-current of the application.
The working peak-reverse voltage Vrev, or blocking voltage, is
calculated according to the following equation:
P
= (1 - D )
)(I
)(V
off (time)
max leak ds(stress)
Output and Input Capacitors
Output capacitors are selected based upon their value,
equivalent series resistance (ESR), equivalent series
inductance and capacitor ripple current rating. The capacitor
value controls the peak-to-peak output ripple voltage at the
switching frequency. Assuming a linear decay of the
capacitor voltage during the off time, during which the
capacitor must supply the load current, the minimum value of
the output capacitor can be calculated as follows,
V
= [(V
INmax
+ V
) / N ]+ V
OUT
R
RDSon
The reflected peak primary current constitutes the peak
repetitive forward-current through the diode. Because all
current to the output capacitors and load must flow through
the diode, the average forward diode current is equal to the
steady-state load current. Power loss in the Schottky is the
sum of the conduction losses and reverse leakage losses.
Conduction losses are calculated using the forward voltage
drop across the diode and the average forward-current.
Reverse leakage losses are dependent upon the reverse
leakage-current, the blocking voltage, and the on-time of
the FET.
Cout = [(T- T
ON(max)
)(Iout)] / Vripple),
where Vripple is the acceptable peak-to peak output voltage
ripple. However, there are practical limitations to how low a
single stage output filter can reduce the ripple voltage and
sometimes an extra LC filter stage is necessary. This second
stage filter would also reduce the output high frequency noise.
Parasitic resistance and inductance in the output capacitors
tend to make the ripple voltage much greater than expected,
based upon the above equation. Using capacitors with the
lowest possible ESR and ESL helps reduce high frequency
ripple. The rms ripple current that the output capacitors
experience is not the same as the secondary side rms output
current; it is the AC portion of it. The secondary side rms
current is in the shape of a clipped sawtooth, or trapezoid,
where the output capacitor’s current waveform is in the shape
of right triangle. Therefore, the typical capacitor ripple current
rating the output capacitor must meet is equal to,
Determining the Turns Ratio of the Flyback
Transformer
The turns ratio of the flyback transformer can be calculated
by using the this steady-state volt-second approach:
n = [(V
- V )(D
DS max
)(T)] / [(V
OUT
+ V )(0.8 - D
)(T)]
max
INmin
F
(3)(Treset)
4 – --------------------------------
T
Treset
------------------ -----------------------------------------------
Irms = (Ipeak)
T
12
FN9007.7
April 13, 2005
8
ISL6401
This limits the peak primary current in the event of an output
Primary Inductance
short circuit. This resistor must have a power rating to meet
The flyback transformer is actually a coupled inductor, acting
as an energy storage unit, as well as performing the usual
transformer functions. Crucial considerations include primary
inductance, working flux density swing, gap length, the
winding scheme and wire diameter. The primary inductance,
LP, for a discontinuous mode flyback converter can be
calculated according to the following relationship:
2
the (I
)(R) requirement, where I
is the root mean
rms
rms
square (rms) primary current. Because this resistor defines
the maximum peak primary current, the input energy to the
2
transformer is defined and equal to (L )(I
) / 2. This
PEAK
P
defined energy in a fixed frequency discontinuous-mode
flyback results in a fixed output power.
2
The advantage of current-mode control is that the output
voltage is held constant despite changes in the input voltage,
because the peak-primary current remains constant; the
slope of this inductor current and its pulse width are
adjusted. Leading edge spikes or noise are caused by the
reverse recovery of the rectifier, equivalent capacitive
loading on the secondary, and parasitic circuit inductances.
A small low pass RC filter is added to the current-sense
signal to filter out these spikes, so the comparator does not
assume an overload condition is present during switch turn-
on. To avoid excessive phase lag on the current-sense
signal, the low pass filter corner frequency is selected to be
at least a decade above the switching frequency.
Lp = n [(V
- V )(T
)] / (2)(T)(V
)(I )
INmin
DS ONmax
OUT OUT
Where n is the assumed efficiency of the converter and Iout
is the output current. The ferrite core should have high
saturation, low residual flux density, and low losses. An
EFD15 core material proved to be suitable for this
application.
Current Sense
The ground referenced sense resistor is selected such that
the maximum peak primary current trips the CS pin threshold
when this current is 10% higher than its normal operating
peak value at the minimum input voltage.
FN9007.7
9
April 13, 2005
ISL6401
Small Outline Plastic Packages (SOIC)
M14.15 (JEDEC MS-012-AB ISSUE C)
14 LEAD NARROW BODY SMALL OUTLINE PLASTIC
PACKAGE
N
INDEX
0.25(0.010)
M
B M
H
AREA
E
INCHES
MILLIMETERS
-B-
SYMBOL
MIN
MAX
MIN
1.35
0.10
0.33
0.19
8.55
3.80
MAX
1.75
0.25
0.51
0.25
8.75
4.00
NOTES
A
A1
B
C
D
E
e
0.0532
0.0040
0.013
0.0688
0.0098
0.020
-
1
2
3
L
-
SEATING PLANE
A
9
0.0075
0.3367
0.1497
0.0098
0.3444
0.1574
-
-A-
o
h x 45
D
3
4
-C-
α
µ
0.050 BSC
1.27 BSC
-
e
A1
C
H
h
0.2284
0.0099
0.016
0.2440
0.0196
0.050
5.80
0.25
0.40
6.20
0.50
1.27
-
B
0.10(0.004)
5
0.25(0.010) M
C A M B S
L
6
N
α
14
14
7
NOTES:
o
o
o
o
0
8
0
8
-
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of
Rev. 0 12/93
Publication Number 95.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate burrs.
Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006
inch) per side.
4. Dimension“E”doesnotincludeinterleadflashorprotrusions. Interlead
flash and protrusions shall not exceed 0.25mm (0.010 inch) per side.
5. The chamfer on the body is optional. If it is not present, a visual index
feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater
above the seating plane, shall not exceed a maximum value of
0.61mm (0.024 inch).
10. Controlling dimension: MILLIMETER. Converted inch dimensions
are not necessarily exact.
FN9007.7
10
April 13, 2005
ISL6401
Quad Flat No-Lead Plas tic Package (QFN)
Micro Lead Frame Plas tic Package (MLFP)
L16.4x4
16 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220-VGGC ISSUE C)
MILLIMETERS
SYMBOL
MIN
0.80
NOMINAL
MAX
1.00
0.05
1.00
NOTES
A
A1
A2
A3
b
0.90
-
-
-
-
-
-
9
0.20 REF
9
0.23
1.95
1.95
0.28
0.35
2.25
2.25
5, 8
D
4.00 BSC
-
D1
D2
E
3.75 BSC
9
2.10
7, 8
4.00 BSC
-
E1
E2
e
3.75 BSC
9
2.10
7, 8
0.65 BSC
-
k
0.25
0.50
-
-
-
-
L
0.60
0.75
0.15
8
L1
N
-
16
4
4
-
10
2
Nd
Ne
P
3
3
-
-
0.60
12
9
θ
-
9
Rev. 5 5/04
NOTES:
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
2. N is the number of terminals.
3. Nd and Ne refer to the number of terminals on each D and E.
4. All dimensions are in millimeters. Angles are in degrees.
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
8. Nominal dimensionsare provided toassistwith PCBLandPattern
Design efforts, see Intersil Technical Brief TB389.
9. Features and dimensions A2, A3, D1, E1, P & θ are present when
Anvil singulation method is used and not present for saw
singulation.
10. Depending on the method of lead termination at the edge of the
package, a maximum 0.15mm pull back (L1) maybe present. L
minus L1 to be equal to or greater than 0.3mm.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
FN9007.7
11
April 13, 2005
相关型号：
ISL6401IB
1 A SWITCHING CONTROLLER, 600 kHz SWITCHING FREQ-MAX, PDSO14, PLASTIC, MS-012AB, SOIC-14
ROCHESTER
ISL6401IB-T
1A SWITCHING CONTROLLER, 600kHz SWITCHING FREQ-MAX, PDSO14, PLASTIC, MS-012AB, SOIC-14
RENESAS
ISL6401IB-T
1 A SWITCHING CONTROLLER, 600 kHz SWITCHING FREQ-MAX, PDSO14, PLASTIC, MS-012AB, SOIC-14
ROCHESTER
ISL6401IBZ-T
1A SWITCHING CONTROLLER, 600kHz SWITCHING FREQ-MAX, PDSO14, PLASTIC, LEAD FREE, MS-012AB, SOIC-14
RENESAS
©2020 ICPDF网 联系我们和版权申明