ISL70001SRHF_技术文档

Rad Hard and SEE Hard 6A Synchronous Buck Regulator

ISL70001SEH

Features

The ISL70001SEH is a radiation hardened and SEE hardened

high efficiency monolithic synchronous buck regulator with

integrated MOSFETs. This single chip power solution operates

over an input voltage range of 3V to 5.5V and provides a tightly

regulated output voltage that is externally adjustable from 0.8V

to ~85% of the input voltage. Output load current capacity is 6A

• ±1% Reference Voltage Over Line, Load, Temperature and

Radiation

• Current Mode Control for Excellent Dynamic Response

• Full Mil-Temp Range Operation (T = -55°C to +125°C)

A

• High Efficiency > 90%

for T < +145°C.

J

• Fixed 1MHz Operating Frequency

• Operates from 3V to 5.5V Supply

• Adjustable Output Voltage

High integration and class leading radiation tolerance makes

the ISL70001SEH an ideal choice to power many of today’s

small form factor applications. Two devices can be

- Two External Resistors Set V

OUT

• Bi-directional SYNC Pin Allows Two Devices to be Synchronized

180° Out-of-Phase

from 0.8V to ~85% of V

IN

synchronized to provide a complete power solution for large

scale digital ICs, like field programmable gate arrays (FPGAs),

that require separate core and I/O voltages.

• Device Enable with Comparator Type Input

• Power-Good Output Voltage Monitor

• Adjustable Analog Soft-Start

Applications

• FPGA, CPLD, DSP, CPU Core or I/O Voltages

• Low-Voltage, High-Density Distributed Power Systems

• Input Undervoltage, Output Undervoltage and Output

Overcurrent Protection

Related Literature

• ISL70001SRHEVAL1Z Evaluation Board, AN1518

• Starts Into Pre-Biased Load

• Electrically Screened to DLA SMD 5962-09225

• QML Qualified per MIL-PRF-38535 Requirements

Specifications for Rad Hard QML devices are controlled by the

Defense Logistics Agency Land and Maritime (DLA). The SMD

numbers listed in the Ordering Information table on page 2 must

be used when ordering.

• Radiation Hardness

- Total Dose [50-300rad(Si)/s] . . . . . . . . . . .100krad(Si) min

- Total Dose [<10mrad(Si)/s] . . . . . . . . . . . . .50krad(Si) min

Detailed Electrical Specifications for these devices are contained

in SMD 5962-09225. This link is also available on the

ISL70001SEH device information page on the Intersil web site.

• SEE Hardness

- SEL and SEB LET . . . . . . . . . . . . 86.4MeV/mg/cm min

2

-6

eff

2

- SEFI X-section (LET = 86.4MeV/mg/cm ) 1.4 x 10 cm

eff

max

2

- SET LET (< 1 Pulse Perturbation) 86.4MeV/mg/cm min

eff

95

90

85

80

75

70

5V SUPPLY

ISL70001SEH

CORE

AUX

I/O

SYNCH

RAD TOLERANT

FPGA

RAD HARD LDO

ISL70001SEH

0

1

2

3

4

5

6

LOAD CURRENT (A)

FIGURE 1. TYPICAL APPLICATION

FIGURE 2. EFFICIENCY 5V INPUT TO 3.3V OUTPUT, T = +25°C

A

November 30, 2011

FN7956.0

CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.

Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.

All other trademarks mentioned are the property of their respective owners.

1

ISL70001SEH

Functional Block Diagram

POWER-ON

RESET (POR)

PORSEL

PVINx

CURRENT

SENSE

SLOPE

COMPENSATION

SOFT

START

SS

PWM

CONTROL

LOGIC

GATE

DRIVE

LXx

EA

GM

FB

COMPENSATION

PGNDx

UV

POWER-GOOD

PGOOD

REF

PWM

REFERENCE

0.6V

TDI

BIT

TDO

TRIM

ZAP

SYNC

M/S

Ordering Information

PART NUMBER

(Note 2)

TEMP. RANGE

(°C)

ORDERING NUMBER

5962R0922502VXC

5962R0922502V9A

ISL70001SRHF/PROTO

ISL70001SRHX/SAMPLE

ISL70001SRHEVAL1Z

NOTES:

PACKAGE

ISL70001SEHVF (Note 1)

ISL70001SEHVX

-55 to +125

48 Ld CQFP (Pb-Free)

Die

ISL70001SRHF/PROTO (Note 1)

ISL70001SRHX/SAMPLE

Evaluation Board

48 Ld CQFP (Pb-Free)

Die

1. These Intersil Pb-free Hermetic packaged products employ 100% Au plate - e4 termination finish, which is RoHS compliant and compatible with both

SnPb and Pb-free soldering operations.

2. For Moisture Sensitivity Level (MSL), please see device information page for ISL70001SEH. For more information on MSL, please see Tech Brief

TB363.

FN7956.0

November 30, 2011

2

ISL70001SEH

Pin Configuration

ISL70001SEH

(48 LD CQFP)

TOP VIEW

6

5

4

3

2

1

48 47 46 45 44 43

42

PVIN3

LX3

7

M/S

ZAP

TDI

41

40

39

38

37

8

PGND3

PGND4

LX4

9

10

11

12

13

14

15

16

17

18

TDO

PGOOD

SS

36

35

34

33

32

31

DVDD

DGND

AGND

PVIN4

PVIN5

LX5

19 20 21 22 23 24 25 26 27 28 29 30

Pin Descriptions

PIN NUMBER

PIN NAME

DESCRIPTION

1, 2, 27, 28, 29, 30,

37, 38, 39, 40, 47,

48

PGNDx

These pins are the power grounds associated with the corresponding internal power blocks. Connect these pins

directly to the ground plane. These pins should also connect to the negative terminals of the input and output

capacitors. Locate the input and output capacitors as close as possible to the IC.

3, 26, 31, 36, 41,

46

LXx

These pins are the outputs of the corresponding internal power blocks and should be connected to the output

filter inductor. Internally, these pins are connected to the synchronous MOSFET power switches. To minimize

voltage undershoot, it is recommended that a Schottky diode be connected from these pins to PGNDx. The

Schottky diode should be located as close as possible to the IC.

4, 5, 24, 25, 32, 33,

34, 35, 42, 43, 44,

45

PVINx

SYNC

These pins are the power supply inputs to the corresponding internal power blocks. These pins must be

connected to a common power supply rail, which must fall in the range of 3V to 5.5V. Bypass these pins directly

to PGNDx with ceramic capacitors located as close as possible to the IC.

6

This pin is the synchronization I/O for the IC. When configured as an output (Master Mode), this pin drives the

SYNC input of another ISL70001SEH. When configured as an input (Slave Mode), this pin accepts the SYNC

output from another ISL70001SEH or an external clock. Synchronization of the slave unit is 180° out-of-phase

with respect to the master unit. If synchronizing to an external clock, the clock must be SEE hardened and the

frequency must be within the range of 1MHz ±20%.

7

M/S

This pin is the Master/Slave input for selecting the direction of the bi-directional SYNC pin. For SYNC = Output

(Master Mode), connect this pin to DVDD. For SYNC = Input (Slave Mode), connect this pin to DGND.

8

9

ZAP

TDI

This pin is a trim input and is used to adjust various internal circuitry. Connect this pin to DGND.

This pin is the test data input of the internal BIT circuitry. Connect this pin to DGND.

This pin is the test data output of the internal BIT circuitry. Connect this pin to DGND.

10

11

TDO

PGOOD

This pin is the power-good output. This pin is an open drain logic output that is pulled to DGND when the output

voltage is outside a ±11% typical regulation window. This pin can be pulled up to any voltage from 0V to 5.5V,

independent of the supply voltage. A nominal 1kΩ to 10kΩ pull-up resistor is recommended. Bypass this pin

to DGND with a 10nF ceramic capacitor to mitigate SEE.

FN7956.0

November 30, 2011

3

ISL70001SEH

Pin Descriptions(Continued)

PIN NUMBER

PIN NAME

DESCRIPTION

12

SS

This pin is the soft-start input. Connect a ceramic capacitor from this pin to DGND to set the soft-start output

ramp time in accordance with Equation 1:

t

= C ⋅ V ⁄ I

REF SS

(EQ. 1)

SS

where:

t

C

= Soft-start output ramp time

= Soft-start capacitor

SS

V

= Reference voltage (0.6V typical)

REF

I

= Soft-start charging current (23µA typical)

SS

Soft-start time is adjustable from approximately 2ms to 200ms.

The range of the soft-start capacitor should be 82nF to 8.2µF, inclusive.

13, 14

DVDD

These pins are the bias supply inputs to the internal digital control circuitry. Connect these pins together at the

IC and locally filter them to DGND using a 1Ω resistor and a 1µF ceramic capacitor. Locate both filter

components as close as possible to the IC.

15, 16

17, 18

19

DGND

AGND

AVDD

REF

These pins are the digital ground associated with the internal digital control circuitry. Connect these pins

directly to the ground plane.

These pins are the analog ground associated with the internal analog control circuitry. Connect these pins

directly to the ground plane.

This pin is the bias supply input to the internal analog control circuitry. Locally filter this pin to AGND using a 1Ω

resistor and a 1µF ceramic capacitor. Locate both filter components as close as possible to the IC.

20

This pin is the internal reference voltage output. Bypass this pin to AGND with a 220nF ceramic capacitor

located as close as possible to the IC. The bypass capacitor is needed to mitigate SEE. No current (sourcing or

sinking) is available from this pin.

21

FB

This pin is the voltage feedback input to the internal error amplifier. Connect a resistor from FB to VOUT and

from FB to AGND to adjust the output voltage in accordance with Equation 2:

V

= V

⋅ [1 + (R ⁄ R )]

REF T B

(EQ. 2)

OUT

where:

= Output voltage

V

OUT

V

= Reference voltage (0.6V typical)

REF

R = Top divider resistor (Must be 1kΩ)

T

R

= Bottom divider resistor

B

The top divider resistor must be 1kΩ to mitigate SEE. Connect a 4.7nF ceramic capacitor across RT to mitigate SEE

and to improve stability margins.

22

23

EN

This pin is the enable input to the IC. This is a comparator type input with a rising threshold of 0.6V and

programmable hysteresis. Driving this pin above 0.6V enables the IC. Bypass this pin to AGND with a 10nF

ceramic capacitor to mitigate SEE.

PORSEL

This pin is the input for selecting the rising and falling POR (Power-On-Reset) thresholds. For a nominal 5V

supply, connect this pin to DVDD. For a nominal 3.3V supply, connect this pin to DGND. For nominal supply

voltages between 5V and 3.3V, connect this pin to DGND.

FN7956.0

November 30, 2011

4

ISL70001SEH

Typical Application Schematic

LX1

LX2

LX3

LX4

LX5

LX6

PVIN1

PVIN2

PVIN3

PVIN4

PVIN5

PVIN6

5V

100µF

1µF

1µH

1.8V

6A

470µF

20V

3A

1kΩ

1

AVDD

FB

1µF

ISL70001SEH

0V TO 5.5V

499Ω

AGND

DVDD

1

PGOOD

10nF

1µF

VSENSE

SYNC

REF

DGND

EN

220nF

10nF

M/S

PORSEL

TDI

SS

TDO

ZAP

100nF

FIGURE 3. 5V INPUT SUPPLY VOLTAGE WITH MASTER MODE SYNCHRONIZATION

FN7956.0

November 30, 2011

5

ISL70001SEH

Typical Application Schematic(Continued)

LX1

LX2

LX3

LX4

LX5

LX6

PVIN1

PVIN2

PVIN3

PVIN4

PVIN5

PVIN6

3.3V

100µF

1µF

1µH

1.8V

6A

470µF

20V

3A

4.7nF

1kΩ

1

DVDD

FB

1µF

ISL70001SEH

0V TO 5.5V

499Ω

DGND

1

PGOOD

AVDD

10nF

1µF

VSENSE

SYNC

REF

AGND

EN

220nF

M/S

10nF

PORSEL

TDI

SS

TDO

ZAP

100nF

FIGURE 4. 3.3V INPUT SUPPLY VOLTAGE WITH SLAVE MODE SYNCHRONIZATION

FN7956.0

November 30, 2011

6

ISL70001SEH

Electrical Specifications Unless otherwise noted, VIN = AVDD = DVDD = PVINx = EN = M/S = 3V or 5.5V;

GND = AGND = DGND = PGNDx = TDI = TDO = ZAP = 0V; FB = 0.65V; PORSEL = VIN for 4.5V ≤ VIN ≤ 5.5V and GND for V < 4.5V,

IN

SYNC = LXx = Open Circuit; PGOOD is pulled up to V with a 1k resistor; REF is bypassed to GND with a 220nF capacitor; SS is bypassed to GND with a

IN

100nF capacitor; I

OUT

= 0A; T = T = +25°C. (Note 3). Boldface limits apply over the operating temperature range, -55°C to +125°C; over a total iodizing

A J

dose of 100krad(Si) with exposure at a high dose rate of 50 - 300krad(Si)/s; and over a total iodizing dose of 50krad(Si) with exposure at a low dose rate

of <10mrad(Si)/s.

MIN

MAX

PARAMETER

POWER SUPPLY

TEST CONDITIONS

(Note 4)

TYP

(Note 4)

UNITS

Operating Supply Current

V

= 5.5V

= 3.6V

40

25

6

65

45

12

6

mA

IN

Shutdown Supply Current

= 5.5V, EN = GND

= 3.6V, EN = GND

3

OUTPUT VOLTAGE

Reference Voltage Tolerance

Output Voltage Tolerance

0.594

-2

0.6

0

0.606

2

V

I

= 0.8V to 2.5V for V = 4.5V to 5.5V,

IN

%

OUT

= 0.8V to 2.5V for V = 3V to 3.6V,

IN

= 0A to 6A (Notes 5, 6)

Feedback (FB) Input Leakage Current

PWM CONTROL LOGIC

Oscillator Accuracy

V

= 5.5V, V = 0.6V

FB

-1

0

1

µA

IN

0.85

.8

1

1.15

1.2

MHz

ns

V

External Oscillator Range

Minimum LXx On Time

V

= 5.5V, Test Mode

= 3V, Test Mode

110

40

150

50

1.3

1.2

0

150

100

210

100

IN

Minimum LXx Off Time

Minimum LXx On Time

Minimum LXx Off Time

Master/Slave (M/S) Input Voltage

Input High Threshold

Input Low Threshold

V

- 0.5

IN

0.5

1

V

Master/Slave (M/S) Input Leakage Current

Synchronization (SYNC) Input Voltage

V

= 5.5V, M/S = GND or V

-1

µA

V

IN

Input High Threshold, M/S = GND

Input Low Threshold, M/S = GND

2.3

1.7

1.5

0

1

V

Synchronization (SYNC) Input Leakage

Current

V

= 5.5V, M/S = GND, SYNC = GND or V

-1

µA

IN

Synchronization (SYNC) Output Voltage

V

- V @ I = -1mA

IN OH OH

0.15

0.4

V

@ I = 1mA

OL OL

POWER BLOCKS

Upper Device r

Lower Device r

V

= 3V, 0.4A Per Power Block, Test Mode (Note 6)

= 5.5V, EN = LXx = GND, Single LXx Output

122

77

-1

215

146

0

346

236

mΩ

µA

DS(ON)

IN

DS(ON)

LXx Output Leakage

= 5.5V, EN = GND, LXx = V , Single LXx Output

IN

0

15

µA

Deadtime

Efficiency

Within a Single Power Block or between Power

Blocks (Note 6)

1.7

5

ns

V

= 3.3V, V

OUT

= 1.8V, I

= 3A

90

92

%

IN

OUT

= 5V, V

OUT

= 2.5V, I

OUT

= 3A

FN7956.0

November 30, 2011

7

ISL70001SEH

Electrical Specifications Unless otherwise noted, VIN = AVDD = DVDD = PVINx = EN = M/S = 3V or 5.5V;

GND = AGND = DGND = PGNDx = TDI = TDO = ZAP = 0V; FB = 0.65V; PORSEL = VIN for 4.5V ≤ VIN ≤ 5.5V and GND for V < 4.5V,

IN

SYNC = LXx = Open Circuit; PGOOD is pulled up to V with a 1k resistor; REF is bypassed to GND with a 220nF capacitor; SS is bypassed to GND with a

IN

100nF capacitor; I

OUT

= 0A; T = T = +25°C. (Note 3). Boldface limits apply over the operating temperature range, -55°C to +125°C; over a total iodizing

A J

dose of 100krad(Si) with exposure at a high dose rate of 50 - 300krad(Si)/s; and over a total iodizing dose of 50krad(Si) with exposure at a low dose rate

of <10mrad(Si)/s. (Continued)

MIN

MAX

PARAMETER

POWER-ON RESET

TEST CONDITIONS

(Note 4)

TYP

(Note 4)

UNITS

POR Select (PORSEL)

Input High Threshold

Input Low Threshold

= 5.5V, PORSEL = GND or V

V

- 0.5

1.4

1.2

0

V

IN

0.5

1

POR Select (PORSEL) Input Leakage

Current

V

-1

µA

IN

VIN POR

Rising Threshold, PORSEL = V

4.1

225

2.65

90

4.25

325

2.8

175

0.6

0

4.45

425

2.95

260

0.64

3

V

mV

V

IN

Hysteresis, PORSEL = V

IN

Rising Threshold, PORSEL = GND

Hysteresis, PORSEL = GND

Rising/Falling Threshold

mV

V

Enable (EN) Input Voltage

Enable (EN) Input Leakage Current

Enable (EN) Sink Current

SOFT-START

0.56

-3

V

= 5.5V, EN = GND or V

µA

IN

EN = 0.3V

6.4

11

16.6

Soft-Start Source Current

Soft-Start Discharge ON-Resistance

Soft-Start Discharge Time

POWER-GOOD SIGNAL

Rising Threshold

SS = GND

20

23

2.2

256

27

µA

Ω

4.7

Clock Cycles

V

as a % of V , Test Mode

REF

107

2

111

3.5

115

5

%

FB

IN

Rising Hysteresis

as a % of V , Test Mode

REF

Falling Threshold

as a % of V , Test Mode

REF

85

2

89

93

5

%

Falling Hysteresis

as a % of V , Test Mode

REF

3.5

%

Power-Good Drive

= 3V, PGOOD = 0.4V, EN = GND

= PGOOD = 5.5V

7.3

8.2

mA

µA

Power-Good Leakage

0.001

1

IN

PROTECTION FEATURES

Undervoltage Monitor

Undervoltage Trip Threshold

Undervoltage Recovery Threshold

Overcurrent Monitor

V

= 3V, V as a % of V , Test mode

FB REF

71

84

75

88

79

92

%

IN

= 3V, V as a % of V , Test mode

FB REF

IN

Overcurrent Trip Level

LX4 Power Block, Test Mode, (Note 7)

= 3V, SS interval = 200µs, Test Mode, Fault

1.3

1.9

0.8

2.5

5

A

Overcurrent or Short-Circuit Duty-Cycle

V

%

IN

interval divided by hiccup interval

NOTES:

3. Typical values shown are not guaranteed. Guaranteed min/max values are provided in the SMD.

4. Compliance to datasheet limits is assured by one or more methods: production test, characterization and/or design.

5. Limits do not include tolerance of external feedback resistors. The 0A to 6A output current range may be reduced by Minimum LXx On Time and

Minimum LXx Off Time specifications.

6. Limits established by characterization or analysis and are not production tested.

7. During an output short-circuit, peak current through the power block(s) can continue to build beyond the overcurrent trip level by up to 3A. With all six

power blocks connected, peak current through the power blocks and output inductor could reach (6 x 2.5A) + 3A = 18A. The output inductor must

support this peak current without saturating.

FN7956.0

November 30, 2011

8

ISL70001SEH

output current. See the “Typical Application Schematic” on

page 5 for pin connection guidance.

Functional Description

The ISL70001SEH is a monolithic, fixed frequency, current-mode

synchronous buck regulator with user configurable power blocks.

Two ISL70001SEH devices can be used to provide a total DC/DC

solution for FPGAs, CPLDs, DSPs and CPUs.

A scaled pilot device associated with each power block provides

current feedback. Power block 4 contains the master pilot device

and this is why it must be connected to the output inductor.

The ISL70001SEH utilizes peak current-mode control with

integrated compensation and switches at a fixed frequency of

1MHz. Two ISL70001SEH devices can be synchronized 180°

out-of-phase to reduce input RMS ripple current. These attributes

reduce the number and size of external components required,

while providing excellent output transient response. The

internal synchronous power switches are optimized for high

efficiency and good thermal performance.

Main Control Loop

During normal operation, the internal top power switch is turned

on at the beginning of each clock cycle. Current in the output

inductor ramps up until the current comparator trips and turns

off the top power MOSFET. The bottom power MOSFET turns on

and the inductor current ramps down for the rest of the cycle.

The current comparator compares the output current at the

ripple current peak to a current pilot. The error amplifier monitors

The chip features a comparator type enable input that provides

flexibility. It can be used for simple digital on/off control or,

alternately, can provide undervoltage lockout capability by

using two external resistors to precisely sense the level of an

external supply voltage. A power-good signal indicates when the

output voltage is within ±11% typical of the nominal output

voltage.

V

and compares it with an internal reference voltage. The

OUT

output voltage of the error amplifier drives a proportional current

to the pilot. If V is low, the current level of the pilot is

OUT

increased and the trip off current level of the output is increased.

The increased output current raises V

with the reference voltage.

until it is in agreement

OUT

Output Voltage Selection

The output voltage of the ISL70001SEH can be adjusted using an

external resistor divider as shown in Figure 6.

Regulator start-up is controlled by an analog soft-start circuit,

which can be adjusted from approximately 2ms to 200ms by

using an external capacitor.

The ISL70001SEH incorporates fault protection for the

regulator. The protection circuits include input undervoltage,

output undervoltage, and output overcurrent.

V

= 0.6V

= 220nF

= 1k

= 4.7nF

REF

T

C

L

OUT

C

R

C

V

OUT

LXx

Power Blocks

C

OUT

The power output stage of the regulator consists of six 1A

capable power blocks that are paralleled to provide full 6A

output current capability. The block diagram in Figure 5 shows a

top level view of the individual power blocks.

C

R

C

T

ERROR

AMPLIFIER

FB

-

V

REF

R

B

+

C

REF

PVIN1

LX1

PGND1

PVIN6

LX6

PGND6

POWER BLOCK 1

POWER BLOCK 2

POWER BLOCK 3

POWER BLOCK 6

POWER BLOCK 5

POWER BLOCK 4

FIGURE 6. OUTPUT VOLTAGE SELECTION

R should be selected as 1kΩ to mitigate SEE. R should be

shunted by a 4.7nF ceramic capacitor, C , to mitigate SEE and to

improve loop stability margins. The REF pin should be bypassed

to AGND with a 220nF ceramic capacitor to mitigate SEE. It

should be noted that no current (sourcing or sinking) is available

from the REF pin. R can be determined from Equation 3. The

designer can configure the output voltage from 0.8V to 85% of

the input voltage.

PVIN2

LX2

PGND2

T

PVIN5

LX5

PGND5

C

PVIN3

LX3

PGND3

PVIN4

LX4

PGND4

B

V

(EQ. 3)

FIGURE 5. POWER BLOCK DIAGRAM

REF

– V

REF

-------------------------------

= R ⋅

T

R

B

V

OUT

Each power block has a power supply input pin, PVINx, a phase

output pin, LXx, and a power supply ground pin, PGNDx. All PVINx

pins must be connected to a common power supply rail and all

PGNDx pins must be connected to a common ground. LXx pins

should be connected to the output inductor based on the

required load current, but must include the LX4 pin. For example,

if 3A of output current is needed, any three LXx pins can be

connected to the inductor as long as one of them is the LX4 pin.

The unused LXx pins should be left unconnected. Connecting all

six LXx pins to the output inductor provides a maximum 6A of

Switching Frequency/Synchronization

The ISL70001SEH features an internal oscillator running at a

fixed frequency of 1MHz ±15% over recommended operating

conditions. The regulator can be configured to run from the

internal oscillator or can be synchronized to another

ISL70001SEH or an SEE hardened external clock with a

frequency range of 1MHz ±20%.

FN7956.0

November 30, 2011

9

ISL70001SEH

To run the regulator from the internal oscillator, connect the M/S

pin to DVDD. In this case, the output of the internal oscillator

appears on the SYNC pin. To synchronize the regulator to the

SYNC output of another ISL70001SEH regulator or to an SEE

hardened external clock, connect the M/S pin to DGND. In this

case, the SYNC pin is an input that accepts an external

V

I

C

= 0.6V

= 11µA

R

EN

V

IN1

= 10nF

EN

PVINx

synchronizing signal. When synchronizing multiple devices, Slave

regulators are synchronized 180° out-of-phase with respect to

the SYNC output of a Master regulator or to an external clock.

V

IN2

ENABLE

COMPARATOR

R1

R2

Operation Initialization

EN

+

The ISL70001SEH initializes based on the state of the power-on

reset (POR) monitor of the PVINx inputs and the state of the EN

input. Successful initialization prompts a soft-start interval, and

the regulator begins slowly ramping the output voltage. Once the

commanded output voltage is within the proper window of

operation, the power-good signal changes state from low to high,

indicating proper regulator operation.

C

V

R

EN

-

POR

LOGIC

I

EN

FIGURE 7. ENABLE CIRCUIT

Power-On Reset

The POR circuitry prevents the controller from attempting to

soft-start before sufficient bias is present at the PVINx pins.

With the part enabled and the I current sink off, the disable

EN

level is set by the resistor divider. The disable level is defined by

Equation 5.

The POR threshold of the PVINx pins is controlled by the PORSEL

pin. For a nominal 5V supply voltage, PORSEL should be

connected to DVDD. For a nominal 3.3V supply voltage, PORSEL

should be connected to DGND. For nominal supply voltages

between 5V and 3.3V, PORSEL should be connected to DGND.

The POR rising and falling thresholds are shown in the “Electrical

Specifications” table on page 8.

R1

R2

(EQ. 5)

-------

= V ⋅ 1 +

R

V

DISABLE

The difference between the enable and disable levels provides

adjustable hysteresis so that noise on V does not interfere

IN2

with the enabling or disabling of the regulator.

To mitigate SEE, the EN pin should be bypassed to the AGND pin

with a 10nF ceramic capacitor.

Hysteresis between the rising and falling thresholds ensures that

small perturbations on PVINx seen during turn-on/turn-off of the

regulator do not cause inadvertent turn-off/turn-on of the

regulator. When the PVINx pins are below the POR rising

threshold, the internal synchronous power MOSFET switches are

turned off, and the LXx pins are held in a high-impedance state.

Soft-Start

Once the POR and enable circuits are satisfied, the regulator

initiates a soft-start. Figure 8 shows that the soft-start circuit

clamps the error amplifier reference voltage to the voltage on an

external soft-start capacitor connected to the SS pin.

Enable and Disable

After the POR input requirement is met, the ISL70001SEH

remains in shutdown until the voltage at the enable input rises

above the enable threshold. As shown in Figure 7, the enable

circuit features a comparator type input. In addition to simple

logic on/off control, the enable circuit allows the level of an

external voltage to precisely gate the turn-on/turn-off of the

V

I

R

= 0.6V

= 23µA

= 2.2Ω

REF

SS

V

OUT

D

R

T

FB

regulator. An internal I current sink with a typical value of

11µA is only active when the voltage on the EN pin is below the

R

EN

B

ERROR

AMPLIFIER

SS

enable threshold. The current sink pulls the EN pin low. As V

IN2

rises, the enable level is not set exclusively by the resistor divider

from V

-

C

SS

+

REF

.

IN2

V

REF

PWM

LOGIC

C

R

REF

D

With the current sink active, the enable level is defined by

I

SS

Equation 4. R1 is the resistor from the EN pin to V

the resistor from the EN pin to the AGND pin.

and R2 is

IN2

R1

R2

-------

+ I ⋅ R1

EN

V

= V

⋅

1 +

(EQ. 4)

ENABLE

R

Once the voltage at the EN pin reaches the enable threshold, the

current sink turns off.

I

EN

FIGURE 8. SOFT-START CIRCUIT

FN7956.0

November 30, 2011

10

ISL70001SEH

The soft-start capacitor is charged by an internal I current

is a fixed percentage of the reference voltage. Once the

SS

source. As the soft-start capacitor is charged, the output voltage

comparator trips, indicating a valid undervoltage condition, a

3-bit undervoltage counter increments. The counter is reset if the

feedback voltage rises back above the undervoltage threshold,

plus a specified amount of hysteresis outlined in the “Electrical

Specifications” table on page 8. If the 3-bit counter overflows, the

undervoltage protection logic shuts down the regulator.

slowly ramps to the set point determined by the reference

voltage and the feedback network. Once the voltage on the SS

pin is equal to the internal reference voltage, the soft-start

interval is complete. The controlled ramp of the output voltage

reduces the inrush current during start-up. The soft-start output

ramp interval is defined in Equation 6 and is adjustable from

approximately 2ms to 200ms. The value of the soft-start

After the regulator shuts down, it enters a delay interval

equivalent to the soft-start interval, which allows the device to

cool. The undervoltage counter is reset when the device enters

the delay interval. The protection logic initiates a normal

soft-start once the delay interval ends. If the output successfully

soft-starts, the power-good signal goes high, and normal

operation continues. If undervoltage conditions continue to exist

during the soft-start interval, the undervoltage counter must

overflow before the regulator shuts down again. This hiccup

mode continues indefinitely until the output soft-starts

successfully.

capacitor, C , should range from 8.2nF to 8.2µF, inclusive. The

SS

peak inrush current can be computed from Equation 7. The soft-

start interval should be long enough to ensure that the peak

inrush current plus the peak output load current does not exceed

the overcurrent trip level of the regulator.

V

REF

------------

t

= C

⋅

(EQ. 6)

SS

I

SS

V

OUT

------------

I

= C

⋅

(EQ. 7)

INRUSH

OUT

t

SS

Overcurrent Protection

The soft-start capacitor is immediately discharged by a 2.2Ω

resistor whenever POR conditions are not met or EN is pulled low.

The soft-start discharge time is equal to 256 clock cycles.

A pilot device integrated into the PMOS transistor of Power Block 4

samples current each cycle. This current feedback is scaled and

compared to an overcurrent threshold based on the number of

power blocks connected. Each additional power block connected

beyond Power Block 4 increases the overcurrent limit by 2A. For

example, if three power blocks are connected, the typical current

limit threshold would be 3 x 2A = 6A.

Power-Good

The power-good (PGOOD) pin is an open-drain logic output that

indicates when the output voltage of the regulator is within

regulation limits. The power-good pin pulls low during shutdown

and remains low when the controller is enabled. After a

successful soft-start, the PGOOD pin releases, and the voltage

rises with an external pull-up resistor. The power-good signal

transitions low immediately when the EN pin is pulled low.

If the sampled current exceeds the overcurrent threshold, a 3-bit

overcurrent counter increments by one LSB. If the sampled current

falls below the threshold before the counter overflows, the counter

is reset. Once the overcurrent counter reaches 111, the regulator

shuts down.

The power-good circuitry monitors the FB pin and compares it to

the rising and falling thresholds shown in the “Electrical

Specifications” table on page 8. If the feedback voltage exceeds

the typical rising limit of 111% of the reference voltage, the

PGOOD pin pulls low. The PGOOD pin continues to pull low until

the feedback voltage falls to a typical of 107.5% of the reference

voltage. If the feedback voltage drops below a typical of 89% of

the reference voltage, the PGOOD pin pulls low. The PGOOD pin

continues to pull low until the feedback voltage rises to a typical

92.5% of the reference voltage. The PGOOD pin then releases

and signals the return of the output voltage to within the

power-good window.

After the regulator shuts down, it enters a delay interval,

equivalent to the soft-start interval, which allows the device to

cool. The overcurrent counter is reset when the device enters the

delay interval. The protection logic initiates a normal soft-start

once the delay interval ends. If the output successfully

soft-starts, the power-good signal goes high, and normal

operation continues. If overcurrent conditions continue to exist

during the soft-start interval, the overcurrent counter must

overflow before the regulator shut downs the output again. This

hiccup mode continues indefinitely until the output soft-starts

successfully.

The PGOOD pin can be pulled up to any voltage from 0V to 5.5V,

independently from the supply voltage. The pull-up resistor

should have a nominal value from 1kΩ to 10kΩ. The PGOOD pin

should be bypassed to DGND, with a 10nF ceramic capacitor to

mitigate SEE.

Note: To prevent severe negative ringing that can disturb the

overcurrent counter, it is recommended that a Schottky diode of

appropriate rating be added from the LXx pins to the PGNDx pins.

Feedback Loop Compensation

To reduce the number of external components and to simplify the

process of determining compensation components, the

ISL70001SEH PWM controller has an internally compensated

error amplifier.

Fault Monitoring and Protection

The ISL70001SEH actively monitors output voltage and current

to detect fault conditions. Fault conditions trigger protective

measures to prevent damage to the regulator and external load

device.

Due to the current loop feedback in peak current mode control, the

modulator has a single pole response with -20dB slope at a

frequency determined by the load (Equation 8):

1

Undervoltage Protection

A hysteretic comparator monitors the FB pin of the regulator. The

feedback voltage is compared to an undervoltage threshold that

------------------------------------

=

F

(EQ. 8)

PO

2π ⋅ R ⋅ C

O

OUT

FN7956.0

November 30, 2011

11

ISL70001SEH

where R is load resistance and C

OUT

capacitance. For this type of modulator, a Type 2 compensation

circuit is usually sufficient.

is the output load

Output Filter Design

O

The output inductor and the output capacitor bank together form

a low-pass filter responsible for smoothing the pulsating voltage

at the phase node. The filter must also provide the transient

energy until the regulator can respond. Since the filter has low

bandwidth relative to the switching frequency, it limits the

system transient response. The output capacitors must supply or

sink current while the current in the output inductor increases or

decreases to meet the load demand.

Figure 9 shows a Type 2 amplifier and its response, along with

the responses of the current mode modulator and the converter.

C2

C1

R2

CONVERTER

R1

EA

OUTPUT CAPACITOR SELECTION

TYPE 2 EA

The critical load parameters in choosing the output capacitors are

the maximum size of the load step (ΔI

), the load-current slew

STEP

rate (di/dt), and the maximum allowable output voltage deviation

G

= 25.1dB

EA

MODULATOR

under transient loading (ΔV ). Capacitors are characterized

MAX

F

P

Z

according to their capacitance, ESR (Equivalent Series Resistance)

and ESL (Equivalent Series Inductance).

F

PO

F

C

At the beginning of a load transient, the output capacitors supply all

of the transient current. The output voltage initially deviates by an

amount approximated by the voltage drop across the ESL. As the

load current increases, the voltage drop across the ESR increases

linearly until the load current reaches its final value. Neglecting the

contribution of inductor current and regulator response, the output

voltage initially deviates by an amount shown in Equation 11.

FIGURE 9. FEEDBACK LOOP COMPENSATION

The Type 2 amplifier, in addition to the pole at origin, has a

zero-pole pair that causes a flat gain region at frequencies

between the zero and the pole (Equations 9 and 10).

1

di

-----

ΔV

≈

ESL ×

+ [ESR × ΔI

]

STEP

(EQ. 11)

MAX

-----------------------------

F

=

= 8.6kHz

dt

Z

(EQ. 9)

2π ⋅ R ⋅ C

2

1

The filter capacitors selected must have sufficiently low ESL and

ESR such that the total output voltage deviation is less than the

maximum allowable ripple.

1

-----------------------------

2π ⋅ R ⋅ C

F

=

= 546kHz

(EQ. 10)

P

1

2

Most capacitor solutions rely on a mixture of high frequency

capacitors with relatively low capacitance in combination with

bulk capacitors having high capacitance but larger ESR.

Minimizing the ESL of the high-frequency capacitors allows them

to support the output voltage as the current increases.

Minimizing the ESR of the bulk capacitors allows them to supply

the increased current with less output voltage deviation.

Zero frequency and amplifier high-frequency gain were chosen to

satisfy typical applications. The crossover frequency will appear

at the point where the modulator attenuation equals the

amplifier high frequency gain. The only task that the system

designer has to complete is to specify the output filter capacitors

to position the load main pole somewhere within one decade

lower than the amplifier zero frequency. Equation 13 on page 12

approximates the amount of capacitance needed to achieve an

optimal pole location depending on the number of LXx pins

connected. With this type of compensation, plenty of phase

margin is easily achieved due to zero-pole pair phase ‘boost’.

Ceramic capacitors with X7R dielectric are recommended.

Alternately, a combination of low ESR solid tantalum capacitors

and ceramic capacitors with X7R dielectric may be used.

The ESR of the bulk capacitors is responsible for most of the

output voltage ripple. As the bulk capacitors sink and source the

Conditional stability may occur only when the main load pole is

positioned too much to the left side on the frequency axis due to

excessive output filter capacitance. In this case, the ESR zero

placed within the 1.2kHz to 30kHz range gives some additional

phase ‘boost’. Some phase boost is also be achieved by

inductor AC ripple current, a voltage, V

, develops across

P-P(MAX)

the bulk capacitor according to Equation 12.

(V – V

)V

OUT OUT

IN

----------------------------------------------

= ESR ×

V

(EQ. 12)

P-P(MAX)

L

× f × V

s IN

OUT

connecting the recommended capacitor C in parallel with the

C

upper resistor R of the divider that sets the output voltage value,

as demonstrated in Figure 6.

T

Another consideration in selecting the output capacitors is loop

stability. The total output capacitance sets the dominant pole of

the PWM. Because the ISL70001SEH uses integrated

compensation techniques, it is necessary to restrict the output

capacitance in order to optimize loop stability. The

recommended load capacitance can be estimated using

Equation 13.

Component Selection Guide

This design guide is intended to provide a high-level explanation

of the steps necessary to create a power converter. It is assumed

the reader is familiar with many of the basic skills and

techniques referenced below. In addition to this guide, Intersil

provides a complete evaluation board that includes schematic,

BOM, and an example PCB layout (see Ordering Information

table on page 2).

1.8V

------------

C

= 75μF × NumberofLXxPinsConnected ×

OUT

V

OUT

(EQ. 13)

FN7956.0

November 30, 2011

12

ISL70001SEH

Another stability requirement on the selection of the output

capacitor is that the ‘ESR zero’ (f ) be placed at 60kHz to

90kHz. This range is set by an internal, single compensation zero

at 8.6kHz. This ESR zero location contributes to increased phase

margin of the control loop; therefore (Equation 14):

Input Capacitor Selection

ZESR

Input capacitors are responsible for sourcing the AC component

of the input current flowing into the switching power devices.

Their RMS current capacity must be sufficient to handle the AC

component of the current drawn by the switching power devices,

which is related to duty cycle. The maximum RMS current

required by the regulator is closely approximated by Equation 19.

1

----------------------------------------------

ESR =

(EQ. 14)

2π(f

)(C

)

OUT

ZESR

2

V

– V

V

⎛

⎜

⎝

⎛

⎜

⎝

⎞

⎟

⎠

⎞

2

OUT

1

12

IN

OUT

-----------------

------

--------------------------------- -----------------

I

=

×

I

+

×

⎟

In conclusion, the output capacitors must meet three criteria:

RMS

OUT

V

L

× f

⎠

MAX

IN

OUT

s

1. They must have sufficient bulk capacitance to sustain the

output voltage during a load transient while the output

inductor current is slewing to the value of the load transient.

(EQ. 19)

The important parameters to consider when selecting an input

capacitor are the voltage rating and the RMS ripple current

rating. For reliable operation, select capacitors with voltage

ratings at least 1.5x greater than the maximum input voltage.

The capacitor RMS ripple current rating should be higher than

the largest RMS ripple current required by the circuit.

2. The ESR must be sufficiently low to meet the desired output

voltage ripple due to the output inductor current.

3. The ESR zero should be placed, in a rather large range, to

provide additional phase margin.

Ceramic capacitors with X7R dielectric are recommended.

Alternately, a combination of low ESR solid tantalum capacitors

and ceramic capacitors with X7R dielectric may be used. The

ISL70001SEH requires a minimum effective input capacitance of

100µF for stable operation.

OUTPUT INDUCTOR SELECTION

Once the output capacitors are selected, the maximum allowable

ripple voltage, V

, determines the lower limit on the

inductance as shown in Equation 15.

P-P(MAX)

(V – V

)V

OUT OUT

IN

-------------------------------------------------

L

≥ ESR ×

(EQ. 15)

OUT

Derating Current Capability

f × V × V

P-P(MAX)

s

IN

Most space programs issue specific derating guidelines for parts,

but these guidelines take the pedigree of the part into account.

For instance, a device built to MIL-PRF-38535, such as the

ISL70001, is already heavily derated from a current density

standpoint. However, a mil-temp or commercial IC that is

up-screened for use in space applications may need additional

current derating to ensure reliable operation because it was not

built to the same standards as the ISL70001.

Since the output capacitors are supplying a decreasing portion of

the load current while the regulator recovers from the transient,

the capacitor voltage becomes slightly depleted. The output

inductor must be capable of assuming the entire load current

before the output voltage decreases more than ΔV

. This

MAX

places an upper limit on inductance.

Equation 16 gives the upper limit on output inductance for the

case when the trailing edge of the current transient causes a

greater output voltage deviation than the leading edge.

Equation 17 addresses the leading edge. Normally, the trailing

edge dictates the inductance selection because duty cycles are

usually <50%. Nevertheless, both inequalities should be

evaluated, and inductance should be governed based on the

Figure 10 shows the maximum average output current of the

ISL70001 with respect to junction temperature. These plots take

into account the worst-case current share mismatch in the power

blocks and the current density requirement of MIL-PRF-38535

2

(< 2 x 105 A/cm ). The plot clearly shows that the ISL70001 can

handle 12.1A at +125°C from a worst-case current density

standpoint, but the part is limited to 7.8A because that is the

lower limit of the current limit threshold with all six power blocks

connected.

lower of the two results. In each equation, L

is the output

OUT

inductance, C

OUT

is the total output capacitance, and ΔI is

L(P-P)

the peak-to-peak ripple current in the output inductor.

2 ⋅ C

⋅ V

OUT

(EQ. 16)

13

L

≤ --------------------------------------- ΔV

– (ΔI ⋅ ESR)

L(P-P)

OUT

MAX

2

(ΔI

)

12.14

STEP

12

11

2 ⋅ C

10.18

OUT

10

⎛

⎞

MINIMUM OCP

≤ -------------------------- ΔV

– (ΔI

⋅ ESR)

V

– V

IN OUT

(EQ. 17)

OUT

MAX

L(P-P)

2

⎝

⎠

(ΔI

)

LEVEL = 7.8A

9

8

7

6

5

4

STEP

8.57

The other concern when selecting an output inductor is to ensure

there is adequate slope compensation when the regulator is

operated above 50% duty cycle. Since the internal slope

compensation is fixed, output inductance should satisfy

Equation 18 to ensure this requirement is met.

7.25

6.16

6A @ +146°C

5.3

120

125

130

135

140

145

150

155

4.32μH

JUNCTION TEMPERATURE (°C)

(EQ. 18)

-----------------------------------------------------------------------------------

≥

L

OUT

NumberofLXxPinsConnected

FIGURE 10. CURRENT vs TEMPERATURE

FN7956.0

November 30, 2011

13

ISL70001SEH

PCB Design

L

V

OUT

LXx

3A

PCB design is critical to high-frequency switching regulator

performance. Careful component placement and trace routing

are necessary to reduce voltage spikes and minimize

undesirable voltage drops. Selection of a suitable thermal

interface material is also required for optimum heat dissipation

and to provide lead strain relief. See Table 1 on page 16 for

layout x-y coordinates.

C

OUT

R

C

S

R

C

T

PGNDx

FB

ERROR

AMPLIFIER

-

PCB Plane Allocation

V

R

REF

B

+

Four layers of 2-ounce copper are recommended. Layer 2 should

be a dedicated ground plane with all critical component ground

connections made with vias to this layer. Layer 3 should be a

dedicated power plane split between the input and output power

rails. Layers 1 and 4 should be used primarily for signals but can

also provide additional power and ground islands, as required.

C

REF

FIGURE 11. SCHOTTKY DIODE AND R-C SNUBBER

PCB Layout

Use a small island of copper to connect the LXx pins of the IC to

the output inductor on Layers 1 and 4. To minimize capacitive

coupling to the power and ground planes, void the copper on

Layers 2 and 3 adjacent to the island. Place most of the island of

Layer 4 to minimize the amount of copper that must be voided

from the ground plane (Layer 2).

PCB Component Placement

Components should be placed as close as possible to the IC to

minimize stray inductance and resistance. Prioritize the

placement of bypass capacitors on the pins of the IC in the order

shown: REF, SS, AVDD, DVDD, PVINx (high frequency capacitors),

EN, PGOOD, PVINx (bulk capacitors).

Keep all other signal traces as short as possible.

For an example layout, see AN1518.

Locate the output voltage resistive divider as close as possible to

the FB pin of the IC. The top leg of the divider should connect

directly to the POL (Point of Load), and the bottom leg of the

divider should connect directly to AGND. The junction of the

resistive divider should connect directly to the FB pin.

Thermal Management

For optimum thermal performance, place a pattern of vias on the

top layer of the PCB directly underneath the IC. Connect the vias

to the ground plane on Layer 2, which serves as a heat sink. To

ensure good thermal contact, thermal interface material such as

a Sil-Pad or thermally conductive epoxy should be used to fill the

gap between the vias and the bottom of the IC.

Locate a Schottky clamp diode as close as possible to the LXx

and PGNDx pins of the IC. A small series R-C snubber connected

from the LXx pins to the PGNDx pins may be used to damp high

frequency ringing on the LXx pins, if desired, see Figure 11.

Lead Strain Relief

For strain relief, a Sil-Pad or a thin layer of thermally conductive

epoxy can be used to raise the bottom of the IC from the PCB

surface so that a slight bend can be added to the leads of the IC.

FN7956.0

November 30, 2011

14

ISL70001SEH

BACKSIDE FINISH

Die Characteristics

Die Dimensions

Silicon

ASSEMBLY RELATED INFORMATION

5720µm x 5830µm (225.2 mils x 229.5 mils)

Thickness: 483µm ± 25.4µm (19.0 mils ± 1 mil)

Substrate Potential

PGND

Interface Materials

ADDITIONAL INFORMATION

Worst Case Current Density

GLASSIVATION

Type: Silicon Oxide and Silicon Nitride

Thickness: 0.3µm ± 0.03µm to 1.2µm ± 0.12µm

5

2

< 2 x 10 A/cm

Transistor Count

TOP METALLIZATION

25030

Type: AlCu (0.5%)

Thickness: 2.7µm ±0.4µm

Layout Characteristics

SUBSTRATE

Step and Repeat

Type: Silicon

Isolation: Junction

5720µm x 5830µm

Connect PGND to PGNDx

Metallization Mask Layout

ISL70001SEH

LX1

PGND1

LX2

PVIN2

PVIN3

SYNC PVIN1

PGND2

M/S

ZAP

TDI

LX3

TDO

PGND3

PGND4

PGOOD

SS

LX4

DVDD

PVIN4

PVIN5

LX5

DGND

PGND

AGND

AVDD REF

FB

EN PORSEL

PVIN6

LX6

PGND6

PGND5

FN7956.0

November 30, 2011

15

ISL70001SEH

TABLE 1. LAYOUT X-Y COORDINATES

X

(µm)

Y

(µm)

dX

(µm)

dY

(µm)

BOND WIRES

PER PAD

PAD NAME

PAD NUMBER

AVDD

REF

15

16

17

18

19

20

21

22

23

24

25

26

27

28

29

30

31

32

33

34

1

478

865

263

135

521

135

521

135

521

135

333

135

1

3

1

2

1

FB

1295

1751

2151

2838

3449

4060

4845

5449

4941

4137

3449

2838

2227

1578

962

263

EN

263

PORSEL

PVIN6

LX6

263

188

PGND6

PGND5

LX5

188

925

PVIN5

PVIN4

LX4

1651

2263

2874

3485

4096

4745

5559

5280

4910

4540

4170

3777

3425

2566

1538

1018

654

PGND4

PGND3

LX3

PVIN3

PVIN2

LX2

PGND2

PGND1

LX1

2

PVIN1

SYNC

M/S

3

4

544

5

226

ZAP

6

226

TDI

7

226

TDO

8

226

PGOOD

SS

9

226

10

11

12

13

14

226

DVDD

DGND

PGND

AGND

226

FN7956.0

November 30, 2011

16

ISL70001SEH

Revision History

The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make

sure you have the latest Rev.

DATE

REVISION

FN7956.0

CHANGE

November 30, 2011

Initial Release

Products

Intersil Corporation is a leader in the design and manufacture of high-performance analog semiconductors. The Company's products

address some of the industry's fastest growing markets, such as, flat panel displays, cell phones, handheld products, and notebooks.

Intersil's product families address power management and analog signal processing functions. Go to www.intersil.com/products for a

complete list of Intersil product families.

For a complete listing of Applications, Related Documentation and Related Parts, please see the respective device information page on

intersil.com: ISL70001SEH

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For information regarding Intersil Corporation and its products, see www.intersil.com

FN7956.0

November 30, 2011

17


型号：	ISL70001SRHF
厂家：	Intersil
描述：	Rad Hard and SEE Hard 6A Synchronous Buck Regulator 抗辐射，并请参阅硬盘6A同步降压稳压器稳压器
文件：	总17页 (文件大小：583K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

ISL70001SRHF [INTERSIL]

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