KB3302
更新时间：2024-09-18 06:05:32
品牌：KINGBOR
描述：2Amp, 2MHz Step-up Switching regulator with Soft-Start
概述
数据手册KB3302 概述
2Amp, 2MHz Step-up Switching regulator with Soft-Start 2Amp ， 2MHz降压型开关调节器具有软启动
KB3302 数据手册
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PDF下载Kingbor Technology Co.,Ltd
TEL:(86)0755-26508846 FAX:(86)0755-26509052
KB3302
2Amp, 2MHz Step-up Switching
regulator with Soft-Start
DESCRIPTION
FEATURES
The KB3302 is a high-frequency current-mode step-up
switching regulator with an integrated 2A power transis-
tor. Its high switching frequency (programmable up to
2MHz) allows the use of tiny surface-mount external pas-
sive components. Programmable soft-start eliminates high
inrush current during start-up. The internal switch is rated
at 32V making the converter suitable for high voltage ap-
plications such as Boost, SEPIC and Flyback.
■
■
■
■
■
■
■
■
■
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Up to 95% Efficiency
TDB uA No Load Current
1000mA Output Current
1.5V to 16V Input Voltage Range
Programmable switching frequency up to 2MHz
Output voltage up to 32V
Constant switching frequency current-mode control
1.23V Reference Allows Low Output Voltages
Shutdown Mode Draws)10µA Supply Current
Low saturation voltage switch: 220mV at 2A
The operating frequency of the KB3302 can be set with an
external resistor. The ability to set the operating frequency
gives the KB3302 design flexibilities. A dedicated COMP
pin allows optimization of the loop response. The KB3302
is available in thermally enhanced 8-Pin MSOP packages.
■
■
Overtemperature Protected,Soft-Start function
8-Pin MSOP Packages
APPLICATIONS
■
Flat screen LCD bias supplies
■
TFT bias supplies
■
XDSL power supplies
■
Medical equipment
■
Digital video cameras
■
Portables devices
■
White LED power supplies
TYPICAL APPLICATION
KB3302 Efficiency
L1
3.3µH
ꢀꢁ
V
OUT
V
= 3.3V
IN
TO 4.2V
5.0V
95
90
85
80
75
70
65
60
55
50
1000mA
VOUT = 5V
1.2MHz
1N5819
C1
4.7µF
6
5
SW
R1
VIN
300k
VIN = 4.2V
2
3
ꢂꢃꢃꢄꢄꢂꢅ
FB
SHDN
SS
C3
22 µF
KB3302
MSOP8
7
1
COMP
R2
100k
R3
GND
4
ROSC
8
17.4k
100nF
R3
10.7k
1nF
VIN = 3.6V
VIN = 2.6V
L1: Sumida CR43
Figure 1. 1.2MHzAll Ceramic Capacitor Single Li-ion Cell
to 5V Boost Converter.
0.001
0.010
0.100
1.000
Load Current (A)
1
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KB3302
(Note 1)
ABSOLUTE MAXIMUM RATINGS
Peak SW Sink and Source Current ........................ 2A
Operating Temperature Range (Note 2) .. –40°C to 85°C
Junction Temperature (Note 3)............................ 125°C
Storage Temperature Range ................ –65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
Input Supply Voltage .................................. –0.3V to 18V
SHDN, VFB Voltages .................................. –0.3V to 5V
SW Voltage ................................................ –0.3V to 32V
PACkAGE/ORDER INFORMATION
TOP VIEW
ORDER PART
NUMBER
ORDER PART
NUMBER
TOP VIEW
COMP
FB
1
2
3
4
5
10 SS
1
2
3
4
COMP
FB
8 SS
9
8
7
6
ROSC
KB3302DD
KB3302EMS
7
6
5
ROSC
GND
SHDN
GND
GND
VIN
SW
SW
GND
SHDN
GND
VIN
SW
3000 Units on Tape and Reel
2500 Units on Tape and Reel
DD PART MARKING
EMS PART MARKING
DD PACKAGE
8-LEAD PLASTIC MSOP
EXPOSED PAD IS PGND
10-LEAD (3mm × 3mm) PLASTIC DFN
EXPOSED PAD IS PGND (PIN 11)
MUST BE CONNECTED TO GND
MUST BE CONNECTED TO GND
TJMAX = 125°C, θJA = 45°C/W, θJC = 10°C/W
TJMAX = 125°C, θJA = 45°C/W, θJC = 10°C/W
ELECTRICAL CHARACTERISTICS
Unless specified: VIN = 2V, SHDN = 1.5V, ROSC = 7.68kΩ, -40°C < TA = TJ < 85°C
Parameter
Test Conditions
Min
Typ
Max
1.4
Unit
V
Undervoltage Lockout Threshold
Maximum Operating Voltage
1.3
16
V
TA = 25°C
1.224
1.217
1.242
1.260
1.267
V
Feedback Voltage
-40°C < TA < 85°C
1.5V < VIN < 16V
V
Feedback Voltage Line
Regulation
0.01
40
60
49
5
%
FB Pin Bias Current
80
nA
µΩ−1
dB
µA
µA
mA
µA
MHz
%
Error Amplifier Transconductance
Error Amplifier Open-Loop Gain
COMP Source Current
V
V
FB = 1.1V
FB = 1.4V
COMP Sink Current
5
VIN Quiescent Supply Current
VIN Supply Current in Shutdown
Switching Frequency
VSHDN = 1.5V, VCOMP = 0 ( Not Switching )
VSHDN = 0
1.1
10
1.5
90
1.6
18
1.3
85
1.7
Maximum Duty Cycle
Minimum Duty Cycle
0
%
Switch Current Limit
2
2.8
A
Switch Saturation Voltage
ISW = 2A
220
350
mV
2
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KB3302
ELECTRICAL CHARACTERISTICS
Unless specified: VIN = 2V, SHDN = 1.5V, ROSC = 7.68kΩ, -40°C < TA = TJ < 85°C
Parameter
Test Conditions
SW = 5V
Min
Typ
0.01
1.1
-4.6
0
Max
1
Unit
µA
V
Switch Leakage Current
Shutdown Threshold Voltage
V
1.02
1.18
VSHDN = 1.2V
SHDN = 0
µA
µA
µA
°C
°C
Shutdown Pin Current
V
0.1
Soft-Start Charging Current
Thermal Shutdown Temperature
Thermal Shutdown Hysteresis
VSS = 0.3V
1.5
160
10
TYPICAL PERFORMANCE CHARACTERISTICS
Shutdown Pin Current
vs Temperature
VIN Current vs SHDN Pin Voltage
VIN Current vs SHDN Pin Voltage
-3
-4
-5
-6
1.2
1
0.1
0.08
0.06
0.04
0.02
0
VIN = 2V
V
IN = 2V
VSHDN = 1.25V
125ºC
25ºC
0.8
0.6
0.4
0.2
0
VIN = 2V
V
IN = 12V
125ºC
-40ºC
-40ºC
0
0.5
1
1.5
0
0.2
0.4
0.6
0.8
1
1.2
-50
-25
0
25
50
75
100 125
SHDN Voltage (V)
SHDN Voltage (V)
Temperature (ºC)
Soft-Start Charging Current
vs Temperature
Transconductance vs Temperature
2
1.8
1.6
1.4
1.2
1
80
VSS = 0.3V
VIN = 2V
70
60
50
40
30
-50 -25
0
25
50
75
100 125
-50
-25
0
25
50
75
100 125
Temperature (ºC)
Temperature (ºC)
3
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KB3302
TYPICAL PERFORMANCE CHARACTERISTICS
Switching Frequency
vs Temperature
Feedback Voltage vs Temperature
ROSC vs Switching Frequency
1.7
1.6
1.5
1.4
1.3
1.3
100
10
1
ROSC = 7.68KΩ
VIN = 2V
25ºC
1.25
1.2
VIN = 12V
VIN = 2V
1.15
-50 -25
0
25
50
75 100 125
0.0
0.5
1.0
1.5
2.0
2.5
3.0
-50 -25
0
25
50
75
100 125
Temperature (ºC)
Frequency (MHz)
Temperature (ºC)
Switch Current Limit
vs Temperature
Switch Saturation Voltage
vs Switch Current
Minimum VIN vs Temperature
400
300
200
100
0
3
2.8
2.6
2.4
2.2
2
1.5
1.4
1.3
1.2
1.1
1
25ºC
85ºC
-40ºC
-50
-25
0
25
50
75
100
0
0.5
1
1.5
2
2.5
3
-50 -25
0
25
50
75 100 125
Switch Current (A)
Temperature (ºC)
Temperature (ºC)
VIN Current in Shutdown
vs Input Voltage
Shutdown Threshold
vs Temperature
VIN Quiescent Current vs Temperature
1.3
50
40
30
20
10
0
1.20
1.15
1.10
1.05
1.00
Not Switching
VIN = 2V
1.2
VIN = 16V
-40ºC
125ºC
1.1
1
VIN = 2V
0.9
0.8
VSHDN = 0
-50 -25
0
25
50
75
100 125
-50 -25
0
25
50
75
100 125
0
5
10
15
20
Temperature (ºC)
Temperature (ºC)
Input Voltage (V)
4
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KB3302
PIN FUNCTIONS
Pin
Pin Name
Pin Function
1
COMP
The output of the internal transconductance error amplifier. This pin is used for loop compensation.
The inverting input of the error amplifier. Tie to an external resistive divider to set the output voltage.
2
3
FB
Shutdown Pin. The accurate 1.1V shutdown threshold and the 4.6uA shutdown pin current
hysteresis allow the user to set the undervoltage lockout threshold and hysteresis for the switching
regulator. Pulling this pin below 0.1V causes the converter to shut down to low quiescent current.
Tie this pin to IN if the UVLO and the shutdown features are not used. This pin should not be left
floating.
SHDN
4
5
GND
SW
Ground. Tie to the ground plane.
Collector of the internal power transistor. Connect to the boost inductor and the rectifying diode.
6
7
IN
Power Supply Pin. Bypassed with capacitors close to the pin.
A resistor from this pin to the ground sets the switching frequency.
ROSC
Soft-Start Pin. A capacitor from this pin to the ground lengthens the start-up time and reduces start-
up current.
8
SS
Exposed Pad The exposed pad must be soldered to the ground plane on the PCB for good thermal conduction.
SIMPLIFIED BLOC DIAGRAM
IN
SW
5
6
4.6µA
SHDN
3
+
CMP
INTERNAL
SUPPLY
REG
-
1.1V
ENABLE
CLK
VOLTAGE
THERMAL
REFERENCE
SHUTDOWN
1.242V
+
R
FB
2
EA
-
Q
-
PWM
REG
S
+
COMP
1
1.5µA
SS
8
+
-
I-LIMIT
ILIM
REG_GOOD
ENABLE
R
SENSE
+
Σ
+
CLK
SLOPE COMP
4
ROSC
7
OSCILLATOR
GND
5
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KB3302
APPLICATIONS INFORMATION
Setting the Output Voltage
OPERATION
The KB3302 is a programmable constant-frequency peak
current-mode step-up switching regulator with an
integrated 2A power transistor. Referring to the block
diagrams in Figures 2 and 3, the power transistor is
switched on at the trailing edge of the clock. Switch
current is sensed with an integrated sense resistor. The
sensed current is summed with the slope-compensating
ramp before compared to the output of the error
amplifier EA. The PWM comparator trip point determines
the switch turn-on pulse width. The current-limit
comparator ILIM turns off the power switch when the
switch current exceeds the 2.8A current-limit threshold.
ILIM therefore provides cycle-by-cycle current limit.
Current-limit is not affected by slope compensation
because the current comparator ILIM is not in the PWM
signal path.
An external resistive divider R1 and R2 with its center tap
tied to the FB pin (Figure 4) sets the output voltage.
VOUT
1.242V
⎛
⎜
⎝
⎞
⎟
⎠
R1 = R2
−1
(1)
VOUT
KB3302
FB
R1
40nA
2
R2
Current-mode switching regulators utilize a dual-loop
feedback control system. In the KB3302 the amplifier
output COMP controls the peak inductor current. This is
the inner current loop. The double reactive poles of the
output LC filter are reduced to a single real pole by the
inner current loop, easing loop compensation. Fast
transient response can be obtained with a simple Type-2
compensation network. In the outer loop, the error
amplifier regulates the output voltage.
Figure 4. The Output Voltage is set with a Resistive Divider
The input bias current of the error amplifier will introduce
an error of:
∆VOUT 40nA
VOUT
(
R1 //R2
1.242V
)
100
=
%
(2)
The percentage error of a VOUT = 5V converter with R1 =
100KΩ and R2 = 301KΩ is
The switching frequency of the KB3302 can be programmed
up to 2MHz with an external resistor from the ROSC pin
to the ground. For converters requiring extreme duty
cycles, the operating frequency can be lowered to
maintain the necessary minimum on time or the minimum
off time.
∆VOUT 40nA
VOUT
(
100K //301K
)
100
=
= 0.24%
1.242V
Operating Frequency and Efficiency
The KB3302 requires a minimum input of 1.4V to operate.
A voltage higher than 1.1V at the shutdown pin enables
the internal linear regulator REG in the KB3302. After VREG
becomes valid, the soft-start capacitor is charged with a
1.5µA current source. A PNP transistor clamps the output
of the error amplifier as the soft-start capacitor voltage
rises. Since the COMP voltage controls the peak inductor
current, the inductor current is ramped gradually during
soft-start, preventing high input start-up current. Under
fault conditions (VIN<1.4V or over temperature) or when
the shutdown pin is pulled below 1.1V, the soft-start
capacitor is discharged to ground. Pulling the shutdown
pin below 0.1V reduces the total supply current to 10µA.
Switching frequency of KB3302 is set with an external
resistor from the ROSC pin to the ground. A graph showing
the relationship between ROSC and switching frequency is
given in the “Typical Characteristics”.
High frequency operation reduces the size of passive
components but switching losses are higher. The efficiencies
of 5V to 12V converters operating at 700KHz, 1.35MHz
and 2MHz are shown in Figure 1(b). The peak efficiency
of the KB3302 appears to decrease 0.5% for every
100KHz increase in frequency.
6
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KB3302
APPLICATIONS INFORMATION
It is worth noting that IOUTMAX is directly proportional to the
Duty Cycle
V
IN
ratio
. Equation (4) over-estimates the maximum
VOUT
The duty cycle D of a boost converter is:
output current at high frequencies (>1MHz) since
switching losses are neglected in its derivation.
Nevertheless it is a useful first-order approximation.
V
IN
1 −
VOUT + VD
VCESAT
VOUT + VD
D =
(3)
1 −
Using VCESAT = 0.3V, VD = 0.5V and ILIM = 2A in (3) and (4),
the maximum output currents for three VIN and VOUT
combinations are shown in Table 1.
where VCESAT is the switch saturation voltage and VD is
voltage drop across the rectifying diode.
Maximum Output Current
D
VIN ( V )
2.5
VOUT ( V )
IOUTMAX ( A )
0.35
In a boost switching regulator the inductor is connected
to the input. The DC inductor current is the input current.
When the power switch is turned on, the inductor current
flows into the switch. When the power switch is off, the
inductor current flows through the rectifying diode to the
output. The output current is the average diode current.
The diode current waveform is trapezoidal with pulse width
(1 – D)T (Figure 5). The output current available from a
boost converter therefore depends on the converter
operating duty cycle. The power switch current in the
KB3302 is internally limited to 2A. This is also the maximum
inductor or the input current. By estimating the conduction
losses in both the switch and the diode, an expression of
the maximum available output current of a boost converter
can be derived:
12
5
0.820
0.423
0.615
3.3
1.14
5
12
0.76
Table 1. Calculated Maximum Output Current [ Equation (4)]
Considerations for High Frequency Operation
The operating duty cycle of a boost converter decreases as
VIN approaches VOUT. The PWM modulating ramp in a
current-mode switching regulator is the sensed current ramp
of the control switch. This current ramp is absent unless
the switch is turned on. The intersection of this ramp with
the output of the voltage feedback error amplifier
determines the switch pulse width. The propagation delay
time required to immediately turn off the switch after it
is turned on is the minimum switch on time. Regulator
closed-loop measurement shows that the KB3302 has
a minimum on time of about 150ns at room temperature.
The power switch in the KB3302 is either not turned on
at all or for at least 150ns. If the required switch on time
is shorter than the minimum on time, the regulator will
either skip cycles or it will start to jitter.
⎡
⎤
ILIMV
VOUT
D
45
VD −D
(
VD − VCESAT
)
IN
IOUTMAX
=
1 −
−
⎢
⎣
⎥
⎦
(4)
V
IN
where ILIM is the switch current limit.
I
IN
Inductor Current
Switch Current
Diode Current
ON
OFF
ON
Example: Determine the maximum operating frequency
of a Li-ion cell to 5V converter using the KB3302.
Assuming that VD=0.5V, VCESAT=0.3V and VIN=2.6 - 4.2V,
the minimum duty ratio can be found using (3).
DT
(1-D)T
ON
I
OUT
4.2
ON
OFF
OFF
ON
1 −
5 + 0.5
DMIN
=
= 0.25
0.3
1 −
Figure 5. Current Waveforms in a Boost Regulator
5 + 0.5
7
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KB3302
APPLICATIONS INFORMATION
D
(
V − VCESAT
)
The absolute maximum operating frequency of the
IN
∆IL =
(5)
fL
DMIN
150ns 150ns
0.25
=
= 1.67MHz
. The
converter is therefore
where f is the switching frequency and L is the inductance.
Substituting (3) into (5) and neglecting VCESAT
actual operating frequency needs to be lower to allow for
modulating headroom.
,
⎛
⎞
V
fL
V
IN
IN
⎜
⎜
⎟
⎟
∆IL =
1 −
(6)
The power transistor in the KB3302 is turned off every
switching period for an interval determined by the
discharge time of the oscillator ramp and the propagation
delay of the power switch. This minimum off time limits
the maximum duty cycle of the regulator at a given
VOUT + VD
⎝
⎠
In current-mode control, the slope of the modulating
(sensed switch current) ramp should be steep enough to
lessen jittery tendency but not so steep that large flux swing
decreases efficiency. Inductor ripple current ∆IL between
25-40% of the peak inductor current limit is a good
compromise. Inductors so chosen are optimized in size
and DCR. Setting ∆IL = 0.3•(2) = 0.6A, VD=0.5V in (6),
VOUT
switching frequency. A boost converter with high
ratio
V
In
requires long switch on time and high duty cycle. If the
required duty cycle is higher than the attainable maximum,
then the converter will operate in dropout. (Dropout is a
condition in which the regulator cannot attain its set
output voltage below current limit.)
⎛
⎜
⎞
⎟
⎛
⎞
V
V
V
0.6f
V
IN
IN
IN
IN
⎜
⎜
⎟
⎟
L =
1 −
=
1 −
(7)
⎜
⎝
⎟
⎠
f∆IL
VOUT + VD
VOUT + 0.5
⎝
⎠
where L is in µH and f is in MHz.
The minimum off times of closed-loop boost converters set
to various output voltages were measured by lowering their
input voltages until dropout occurs. It was found that the
minimum off time of the KB3302 ranged from 80 to 110ns
at room temperature.
Equation (6) shows that for a given VOUT, ∆IL is the highest
(
VOUT + VD
)
V =
when
. If VIN varies over a wide range, then
IN
2
choose L based on the nominal input voltage.
Beware of dropout when operating at very low input voltages
(1.5-2V) and with off times approaching 110ns. Shorten
the PCB trace between the power source and the device
input pin, as line drop may be a significant percentage of
the input voltage. A regulator in dropout may appear as if
it is in current limit. The cycle-by-cycle current limit of the
KB3302 is duty-cycle and input voltage invariant and is
typically 2.8A. If the switch current limit is not at least 2A,
then the converter is likely in dropout. The switching
frequency should then be lowered to improve controllability.
The saturation current of the inductor should be 20-30%
higher than the peak current limit (2.8A). Low-cost powder
iron cores are not suitable for high-frequency switching
power supplies due to their high core losses. Inductors
with ferrite cores should be used.
Input Capacitor
The input current in a boost converter is the inductor
current, which is continuous with low RMS current ripples.
A 2.2-4.7µF ceramic input capacitor is adequate for most
applications.
Both the minimum on time and the minimum off time
reduce control range of the PWM regulator. Bench
measurement showed that reduced modulating range
started to be a problem at frequencies over 2MHz. Although
the oscillator is capable of running well above 2MHz,
controllability limits the maximum operating frequency.
Output Capacitor
Both ceramic and low ESR tantalum capacitors can be
used as output filtering capacitors. Multi-layer ceramic
capacitors, due to their extremely low ESR (<5mΩ), are
the best choice. Use ceramic capacitors with stable
temperature and voltage characteristics. One may be
tempted to use Z5U and Y5V ceramic capacitors for
output filtering because of their high capacitance and
Inductor Selection
The inductor ripple current ∆IL of a boost converter
operating in continuous-conduction mode is
8
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KB3302
APPLICATIONS INFORMATION
forward voltages). This is because the diode conduction
interval is much longer than that of the transistor.
Converter efficiency will be improved if the voltage drop
across the diode is lower.
small sizes. However these types of capacitors have high
temperature and high voltage coefficients. For example,
the capacitance of a Z5U capacitor can drop below 60%
of its room temperature value at –25°C and 90°C. X5R
ceramic capacitors, which have stable temperature and
voltage coefficients, are the preferred type.
The rectifying diodes should be placed close to the SW
pins of the KB3302 to minimize ringing due to trace
inductance. Surface-mount equivalents of 1N5817,
1N5819, MBRM120 (ON Semi) and 10BQ015 (IRF) are
all suitable.
The diode current waveform in Figure 5 is discontinuous
with high ripple-content. In a buck converter the inductor
ripple current ∆IL determines the output ripple voltage.
The output ripple voltage of a boost regulator is however
much higher and is determined by the absolute inductor
current. Decreasing the inductor ripple current does not
appreciably reduce the output ripple voltage. The current
flowing in the output filter capacitor is the difference
between the diode current and the output current. This
capacitor current has a RMS value of:
Soft-Start
Soft-start prevents a DC-DC converter from drawing
excessive current (equal to the switch current limit) from
the power source during start up. If the soft-start time is
made sufficiently long, then the output will enter regulation
without overshoot. An external capacitor from the SS pin
to the ground and an internal 1.5µA charging current
source set the soft-start time. The soft-start voltage ramp
at the SS pin clamps the error amplifier output. During
regulator start-up, COMP voltage follows the SS voltage.
The converter starts to switch when its COMP voltage
exceeds 0.7V. The peak inductor current is gradually
increased until the converter output comes into regulation.
If the shutdown pin is forced below 1.1V or if fault is
detected, then the soft-start capacitor will be discharged
to ground immediately.
VOUT
IOUT
−1
(8)
V
IN
If a tantalum capacitor is used, then its ripple current rating
in addition to its ESR will need to be considered.
When the switch is turned on, the output capacitor supplies
the load current IOUT (Figure 5). The output ripple voltage
due to charging and discharging of the output capacitor is
therefore:
The SS pin can be left open if soft-start is not required.
IOUTDT
COUT
Shutdown
∆VOUT
=
(9)
The input voltage and shutdown pin voltage must be greater
than 1.4V and 1.1V respectively to enable the KB3302.
Forcing the shutdown pin below 1.1V stops switching.
Pulling this pin below 0.1V completely shuts off the KB3302.
The total VIN current decreases to 10µA at 2V. Figure 6
shows several ways of interfacing the control logic to the
shutdown pin. Beware that the shutdown pin is a high
impedance pin. It should always be driven from a low-
impedance source or tied to a resistive divider. Floating
the shutdown pin will result in undefined voltage. In Figure
6(c) the shutdown pin is driven from a logic gate whose
VOH is higher than the supply voltage of the KB3302. The
diode clamps the maximum shutdown pin voltage to one
diode voltage above the input power supply.
For most applications, a 10-22µF ceramic capacitor is
sufficient for output filtering. It is worth noting that the
output ripple voltage due to discharging of a 10µF ceramic
capacitor (9) is higher than that due to its ESR.
Rectifying Diode
For high efficiency, Schottky barrier diodes should be used
as rectifying diodes for the KB3302. These diodes should
have a RMS current rating of at least 1A and a reverse
blocking voltage of at least a few Volts higher than the
output voltage. For switching regulators operating at low
duty cycles (i.e. low output voltage to input voltage
conversion ratios), it is beneficial to use rectifying diodes
with somewhat higher RMS current ratings (thus lower
9
Kingbor Technology Co.,Ltd
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KB3302
APPLICATIONS INFORMATION
IN
IN
KB3302
KB3302
SHDN
SHDN
(a)
(b)
V
IN
IN
IN
KB3302
KB3302
1N4148
SHDN
SHDN
(c)
(d)
Figure 6. Methods of Driving the Shutdown Pin
(a) Directly Driven from a Logic Gate
(b) Driven from an Open-drain N-channel MOSFET or an Open-Collector NPN Transistor (VOL < 0.1V)
(c) Driven from a Logic Gate with V > VIN
(d) Combining Shutdown with ProgOraH mmed UVLO (See Section Below).
Programming Undervoltage Lockout
VH and VL are therefore:
The KB3302 has an internal V undervoltage lockout
(UVLO) threshold of 1.4V. The INtransition from idle to
switching is abrupt but there is no hysteresis. If the input
voltage ramp rate is slow and the input bypass is limited,
then sudden turn on of the power transistor will cause a
dip in the line voltage. Switching will stop if VIN falls below
the internal UVLO threshold. The resulting output voltage
rise may be non-monotonic. The 1.1V disable threshold of
the KB3302 can be used in conjunction with a resistive
voltage divider to raise the UVLO threshold and to add an
UVLO hysteresis. Figure 7 shows the scheme. Both VH and
VL (the desired upper and the lower UVLO threshold
voltages) are determined by the 1.1V threshold crossings,
⎛
⎞
R3
R4
⎜
⎟
⎟
VH = 1 +
(1.1V
)
⎜
⎝
⎠
(10)
V = VH − VHYS = VH −IHYSR3
L
Re-arranging,
VHYS
R3 =
(11)
(12)
IHYS
R3
R4 =
VH
−1
1.1
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KB3302
APPLICATIONS INFORMATION
The turn off voltage is:
V = VH − VHYS = 2.75 − 0.69 = 2.06V > 1.4V
.
L
IN
6/8
Frequency Compensation
I
HYS
4.6µA
Figure 8 shows the equivalent circuit of a boost converter
using the KB3302. The output filter capacitor and the load
form an output pole at frequency:
R3
SWITCH CLOSED
WHEN Y = “1”
SHDN
2IOUT
VOUTC2
2
3
+
-
ωp2 = −
= −
(13)
ROUTC2
Y
1.1V
R4
COMPARATOR
VOUT
IOUT
ROUT
=
where C2 is the output capacitor and
equivalent load resistance.
is the
KB3302
The zero formed by C2 and its equivalent series resistance
(ESR) is neglected due to low ESR of the ceramic output
capacitor.
Figure 7. Programmable Hysteretic UVLO Circuit
with VL > 1.4V
.
There is also a right half plane (RHP) zero at angular
frequency:
Example: Increase the turn on voltage of a VIN = 3.3V boost
converter from 1.4V to 2.75V.
2
ROUT 1 −D
( )
ωZ2
=
(14)
L
Using VH = 2.75V and R4 = 100KΩ in (12),
ωz2 decreases with increasing duty cycle D and increasing
IOUT. Using the 5V to 12V boost regulator (1.35MHz) in
Figure 1(a) as an example,
R3 = 150KΩ
.
The resulting UVLO hysteresis is:
5V
0.74A
ROUT
≥
= 6.8Ω
VHYS = IHYSR3 = 4.6µA •150KΩ = 0.69V
.
I
OUT
V
IN
POWER
STAGE
V
OUT
ESR
C2
R
R1
C5
OUT
FB
-
COMP
Gm
+
R3
C6
1.242V
RO
R2
VOLTAGE
REFERENCE
C4
Figure 8. Simplified Block Diagram of a Boost Converter
11
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KB3302
APPLICATIONS INFORMATION
1
ROC4
1
ωp1 = −
= −
4.7MΩ •820pF
5
1 −
12 + 0.5
D =
= 0.62
= −260 rads−1 = −41Hz
C4 and R3 also forms a zero with angular frequency:
0.3
1 −
12 + 0.5
Therefore
1
R3C4
1
ωZ1 = −
= −
2
30.9KΩ•820pF
ωp2
and
≤
= 29.4Krads−1 = 4.68KHz
= −39.5Krads−1 = −6.3KHz
(6.8Ω
)• (10µF)
2
6.8Ω •
(
1 − 0.62
)
The poles p1, p2 and the RHP zero z2 all increase phase
shift in the loop response. For stable operation, the overall
loop gain should cross 0dB with -20dB/decade slope. Due
to the presence of the RHP zero, the 0dB crossover frequency
ωZ2
≥
= 209Krads−1 = 33.3KHz
4.7µH
The spacing between p2 and z2 is the closest when the
converter is delivering the maximum output current from
the lowest VIN. This represents the worst-case compensation
condition. Ignoring C5 and C6 for the moment, C4 forms a
low frequency pole with the equivalent output resistance
RO of the error amplifier:
z2
3
should not be higher than
. Placing z1 near p2 nulls its
effect and maximizes loop bandwidth. Thus
VOUTC2
2IOUT(MAX)
R3C4 ≈
(15)
AmplifierOpenLoop Gain
Transconductance
49dB
RO =
=
= 4.7MΩ
60µΩ−1
R3 determines the mid-band loop gain of the converter.
Increasing R3 increases the mid-band gain and the crossover
GND
C3
R4
R3
C4
C6
R2
U1
C1
L1
SHDN
R1
C5
C2
D1
VOUT
VIN
Figure 9. Suggested PCB Layout for the KB3302. Notice that there is no via
directly under the device. All vias are 12mil in diameter.
12
Kingbor Technology Co.,Ltd
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KB3302
APPLICATIONS INFORMATION
frequency. However it reduces the phase margin. The the size of the loop formed by these components should
values of R3 and C4 can be determined empirically by be minimized. Since the power switch is integrated inside
observing the inductor current and the output voltage the KB3302, grounding the output filter capacitor next to
during load transient. Compensation is optimized when the KB3302 ground pin minimizes size of the high di/dt
the largest R3 and the smallest C4without producing current loop. The input bypass capacitors should also be
ringing or excessive overshoot in its inductor current and placed close to the input pins. Shortening the trace at the
output voltage are found.
SW node reduces the parasitic trace inductance. This not
only reduces EMI but also decreases the sizes of the
switching voltage spikes and glitches.
C5 adds a feedforward zero to the loop response. In some
cases it improves the transient speed of the converter. C6
rolls off the gain at high frequency. This helps to stabilize
Figure 9 shows how various external components are placed
around the KB3302. The frequency-setting resistor should
be placed near the ROSC pin with a short ground trace
on the PC board. These precautions reduce switching
noise pickup at the ROSC pin.
the loop. C5 and C6 are often not needed.
Board Layout Considerations
In a step-up switching regulator, the output filter capacitor,
the main power switch and the rectifying diode carry
switched currents with high di/dt. For jitter-free operation,
To achieve a junction to ambient thermal resistance (θJA)
of 40°C/W, the exposed pad of the KB3302 should be
properly soldered to a large ground plane. Use only 12mil
diameter vias in the ground plane if necessary. Avoid using
larger vias under the device. Molten solder may seep
through large vias during reflow, resulting in poor adhesion,
poor thermal conductivity and low reliability.
Typical Application Circuits
D1
L1
VOUT
VIN
12V, 0.7A
5V
10BQ015
R1
174K
6
5
Efficiency
IN
SHDN
KB3302
SW
FB
95
90
85
80
75
70
65
60
55
50
3
8
2
1
OFF ON
10.5µH, 700KHz
4.7µH, 1.4MHz
C2
10µF
C1
2.2µF
SS
GND
COMP
ROSC
R2
20K
R3
C4
C3
47nF
4
7
C6
R4
3.3µH, 2MHz
All Capacitors are Ceramic.
MSOP-8 Pinout
f / MHz R3 / KΩ R4 / KΩ C4 / pF C6 / pF
L1 / µH
VIN = 5V
VOUT = 12V
0.7
1.35
2
22.1
30.9
63.4
22.1
9.31
4.75
2200
820
-
-
10.5 (Falco D08019)
4.7 (Falco D08017)
0.0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
Load Current (A)
470
22
3.3 (Coilcraft DO1813P)
Figure 10(a). 1.35 MHz All Ceramic Capacitor 5V to 12V Boost
Converter. Pinout Shown is for MSOP-8
13
Kingbor Technology Co.,Ltd
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KB3302
PACAGE DESCRIPTION
Efficiency
95
90
85
80
75
70
65
60
55
50
VOUT = 5V
1.2MHz
D1
L1
VOUT
2.6 - 4.2V
1.8µH
5V, 0.8A
10BQ015
V
IN = 4.2V
R1
301K
6
5
IN
SHDN
KB3302
SW
FB
3
8
2
OFF ON
1-CELL
LI-ION
C2
10µF
C1
2.2µF
1
SS
GND
COMP
ROSC
R2
100K
R3
17.4K
C3
47nF
7
VIN = 3.6V
VIN = 2.6V
4
R4
10.7K
C4
1nF
0.001
0.010
0.100
1.000
Load Current (A)
L1: Sumida CR43
Figure 11(a). 1.2 MHz All Ceramic Capacitor Single Li-ion Cell
to 5V Boost Converter.
Figure 11(b). Efficiency of the Single Li-ion Cell to 5V Boost
Converter in Figure 11(a).
4-CELL
3.6 - 6V
VOUT
5V
C6
L1
D1
4.9µH
2.2µF
10BQ015
R1
60.4K
C5
47pF
6
5
IN
SW
FB
3
2
1
OFF ON
SHDN
C2
10µF
C1
2.2µF
KB3302
8
SS
GND
COMP
ROSC
L2
4.9µH
R2
20K
R3
20K
C3
47nF
4
7
R4
7.68K
C4
560pF
L1 and L2: Coiltronics CTX5-1
Figure 12(a). 1.5 MHz All Ceramic Capacitor 4-Cell to 5V SEPIC Converter. Pinout Shown is for MSOP-8.
14
Kingbor Technology Co.,Ltd
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KB3302
D2
D3
D4
D5
OUT2
23V (10mA)
C5
0.1µF
C6
0.1µF
C7
0.1µF
C8
1µF
L1
D1
3.3V
OUT1
2.2µH
8V (0.55A)
10BQ015
R5
150K
R1
274K
6
5
IN
SHDN
KB3302
SW
3
8
2
1
FB
C2
10µF
C1
2.2µF
C9
0.1µF
SS
GND
COMP
ROSC
R6
100K
R2
49.9K
R3
40.2K
C3
47nF
4
7
R4
7.68K
C4
820pF
D7
OUT3
-8V (10mA)
C10
1µF
L1 : Cooper-Bussmann SD25-2R2
D2 - D7 : BAT54S
D6
Figure 13(a). 1.5MHz Triple-Output TFT Power Supply.
-
3.4V to 3.8V +
0.7A (FLASH)
0.2A (TORCH)
D2
R6
0.1Ω
LXCL-PWF1
R1
698
D1
L1
2.6 - 4.2V
2.2µH
SUMIDA
CR43
10BQ015
1/2
LM358
Q1
MMBT3904T
6
5
1-CELL
LI-ION
IN
SHDN
KB3302
SW
2
OFF ON
3
8
FB
C2
4.7µF
C1
2.2µF
C5
0.1µF
R6
17.4K
1
SS
GND
COMP
ROSC
R2
43.2K
C4 R5
10nF 10K
M1
MMBF2201NT1
4
7
C3
10nF
R4
8.06K
TORCH FLASH
Figure 14(a). 1.4MHz LuxeonTM Flash White LED Driver for Camera Phones
15
Kingbor Technology Co.,Ltd
TEL:(86)0755-26508846 FAX:(86)0755-26509052
KB3302
PACAGE DESCRIPTION - MSOP8
e/2
DIMENSIONS
INCHES MILLIMETERS
MIN NOM MAX MIN NOM MAX
A
D
DIM
A
N
-
-
-
-
-
-
-
-
-
-
-
-
.043
1.10
0.15
0.95
0.38
0.23
A1 .000
A2 .030
.006 0.00
.037 0.75
.015 0.22
.009 0.08
E/2
2X
b
c
.009
.003
E1
E
PIN 1
INDICATOR
D .114 .118 .122 2.90 3.00 3.10
E1 .114 .118 .122 2.90 3.00 3.10
E
e
.193 BSC
.026 BSC
4.90 BSC
0.65 BSC
ccc C
2X N/2 TIPS
1 2
F
L
L1
N
01
aaa
.068 .076 .080 1.73 1.93 2.03
.016 .024 .032 0.40 0.60 0.80
e
B
(.037)
(0.95)
8
-
8
-
0°
8°
0°
8°
D
aaa
C
C
.004
.005
.010
0.10
0.13
0.25
bbb
ccc
A2
A
SEATING
PLANE
A1
bxN
H
bbb
C A-B D
c
GAGE
PLANE
F
EXPOSED PAD
L
01
0.25
(L1)
F
DETAIL A
BOTTOM VIEW
SEE DETAIL A
SIDE VIEW
NOTES:
1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES).
2. DATUMS -A- AND -B- TO BE DETERMINED AT DATUM PLANE-H-
3. DIMENSIONS "E1" AND "D" DO NOT INCLUDE MOLD FLASH, PROTRUSIONS
OR GATE BURRS.
REFERENCE JEDEC STD MO-187, VARIATION AA-T.
4.
Land Pattern - MSOP-8L-EDP
F
DIMENSIONS
INCHES MILLIMETERS
DIM
(.161)
(4.10)
C
F
.081
.098
.026
.016
.063
.224
2.08
2.50
0.65
0.40
1.60
5.70
F
Z
(C)
G
P
G
P
X
Y
Z
X
NOTES:
1. THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY.
CONSULT YOUR MANUFACTURING GROUP TO ENSURE YOUR
COMPANY'S MANUFACTURING GUIDELINES ARE MET.
16
Kingbor Technology Co.,Ltd
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KB3511
PACAGE DESCRIPTION - DFN33
DIMENSIONS
INCHES MILLIMETERS
MIN NOM MAX MIN NOM MAX
A
E
B
E
DIM
-
-
A
.031
.039 0.80
1.00
0.05
-
-
-
A1 .000
-
.002 0.00
-
-
(.008)
(0.20)
A2
b
C
D
E
e
.007 .009 .011 0.18 0.23 0.30
.074 .079 .083 1.87 2.02 2.12
.042 .048 .052 1.06 1.21 1.31
.114 .118 .122 2.90 3.00 3.10
PIN 1
INDICATOR
(LASER MARK)
.020 BSC
0.50 BSC
L
N
.012 .016 .020 0.30 0.40 0.50
10
10
aaa
.003
.004
0.08
0.10
bbb
A
C
SEATING
PLANE
aaa C
LxN
A1
A2
C
1
2
D
N
bxN
bbb
e
C
A B
NOTES:
1.
CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES).
2. COPLANARITY APPLIES TO THE EXPOSED PAD AS WELL AS TERMINALS.
Land Pattern - DFN33-10
DIMENSIONS
K
DIM
INCHES
MILLIMETERS
(.112)
.075
.055
.087
.020
.012
.037
.150
(2.85)
1.90
1.40
2.20
0.50
0.30
0.95
3.80
C
G
H
K
P
X
Y
Z
H
G
Y
(C)
Z
X
P
NOTES:
1. THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY.
CONSULT YOUR MANUFACTURING GROUP TO ENSURE YOUR
COMPANY'S MANUFACTURING GUIDELINES ARE MET.
Kingbor Technology
TEL:(86)0755-26508846 FAX:(86)0755-26509052 www.kingbor.com
17
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