BAV21W [Linear]
100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch; 100VIN微功率隔离型反激式转换器， 150V / 260毫安开关
| 型号： | BAV21W |
| 厂家： | 
Linear |
| 描述： | 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch |
| 文件： | 总24页 (文件大小：313K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LT8300
100V Micropower Isolated
IN
Flyback Converter with
150V/260mA Switch
DescripTion
FeaTures
TheLT®8300isamicropowerhighvoltageisolatedflyback
converter. By sampling the isolated output voltage directly
from the primary-side flyback waveform, the part requires
nothirdwindingoropto-isolatorforregulation.Theoutput
voltage is programmed with a single external resistor. In-
ternalcompensationandsoft-startfurtherreduceexternal
component count. Boundary mode operation provides a
small magnetic solution with excellent load regulation.
LowrippleBurstModeoperationmaintainshighefficiency
at light load while minimizing the output voltage ripple.
A 260mA, 150V DMOS power switch is integrated along
withallhighvoltagecircuitryandcontrollogicintoa5-lead
ThinSOT™ package.
n
6V to 100V Input Voltage Range
n
260mA, 150V Internal DMOS Power Switch
n
Low Quiescent Current:
70µA in Sleep Mode
330µA in Active Mode
n
Boundary Mode Operation at Heavy Load
Low-Ripple Burst Mode® Operation at Light Load
Minimum Load <0.5% (Typ) of Full Output
n
n
n
V
Set with a Single External Resistor
OUT
n
No Transformer Third Winding or Opto-Isolator
Required for Regulation
n
n
n
Accurate EN/UVLO Threshold and Hysteresis
Internal Compensation and Soft-Start
5-Lead TSOT-23 Package
The LT8300 operates from an input voltages range of 6V
to 100V and can deliver up to 2W of isolated output power.
The high level of integration and the use of boundary
and low ripple burst modes result in a simple to use, low
componentcount, andhighefficiencyapplicationsolution
for isolated power delivery.
applicaTions
n
Isolated Telecom, Automotive, Industrial, Medical
Power Supplies
n
Isolated Auxiliary/Housekeeping Power Supplies
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Protected by U.S. Patents, including 5438499, 7463497,
and 7471522.
Typical applicaTion
5V Micropower Isolated Flyback Converter
Efficiency vs Load Current
100
+
V
OUT
90
80
70
60
50
40
30
20
10
0
V
V
= 36V
IN
IN
5V
4:1
36V TO 72V
1mA TO 300mA
•
2.2µF
300µH
19µH
V
IN
V
IN
= 48V
47µF
1M
•
V
= 72V
IN
LT8300
EN/UVLO
–
SW
V
OUT
40.2k
210k
R
FB
GND
8300 TA01a
0
50
100
150
200
250
300
LOAD CURRENT (mA)
8300 TA01b
8300f
1
LT8300
absoluTe MaxiMuM raTings
pin conFiguraTion
(Note 1)
TOP VIEW
SW (Note 2)........................................................... 150V
EN/UVLO 1
GND 2
5 V
IN
V ......................................................................... 100V
IN
EN/UVLO................................................................... V
IN
IN
R
FB
3
4 SW
R ...................................................... V – 0.5V to V
FB
IN
S5 PACKAGE
5-LEAD PLASTIC TSOT-23
= 150°C, θ = 150°C/W
Current into R ................................................... 200µA
FB
T
Operating Junction Temperature Range (Notes 3, 4)
LT8300E, LT8300I ............................. –40°C to 125°C
LT8300H ............................................ –40°C to 150°C
LT8300MP......................................... –55°C to 150°C
Storage Temperature Range .................. –65°C to 150°C
JMAX
JA
orDer inForMaTion
LEAD FREE FINISH
LT8300ES5#PBF
LT8300IS5#PBF
LT8300HS5#PBF
LT8300MPS5#PBF
TAPE AND REEL
PART MARKING*
LTGFF
PACKAGE DESCRIPTION
5-Lead Plastic TSOT-23
5-Lead Plastic TSOT-23
5-Lead Plastic TSOT-23
5-Lead Plastic TSOT-23
TEMPERATURE RANGE
LT8300ES5#TRPBF
LT8300IS5#TRPBF
LT8300HS5#TRPBF
LT8300MPS5#TRPBF
–40°C to 125°C
–40°C to 125°C
–40°C to 150°C
–55°C to 150°C
LTGFF
LTGFF
LTGFF
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
8300f
2
LT8300
elecTrical characTerisTics The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, VEN/UVLO = VIN unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
100
6
UNIT
V
IN
Input Voltage Range
6
V
V
IN
UVLO Threshold
Rising
Falling
5.8
3.2
V
V
I
V
Quiescent Current
V
V
= 0.3V
= 1.1V
1.2
200
70
2
µA
µA
µA
µA
Q
IN
EN/UVLO
EN/UVLO
Sleep Mode (Switch Off)
Active Mode (Switch On)
330
l
l
EN/UVLO Shutdown Threshold
EN/UVLO Enable Threshold
For Lowest Off I
0.3
0.75
V
Q
Falling
Hysteresis
1.199
1.223
0.016
1.270
V
V
I
EN/UVLO Hysteresis Current
V
V
V
= 0.3V
= 1.1V
= 1.3V
–0.1
2.2
–0.1
0
2.5
0
0.1
2.8
0.1
µA
µA
µA
HYS
EN/UVLO
EN/UVLO
EN/UVLO
f
f
t
t
t
I
I
Maximum Switching Frequency
Minimum Switching Frequency
Minimum Switch-On Time
Minimum Switch-Off Time
Maximum Switch-Off Time
Maximum SW Current Limit
Minimum SW Current Limit
SW Over Current Limit
720
6
750
7.5
780
9
kHz
kHz
ns
MAX
MIN
160
350
200
260
52
ON(MIN)
OFF(MIN)
OFF(MAX)
SW(MAX)
SW(MIN)
ns
Backup Timer
µs
l
l
228
34
292
70
mA
mA
mA
Ω
To Initiate Soft-Start
I = 100mA
SW
520
10
R
Switch On-Resistance
DS(ON)
I
I
Switch Leakage Current
V
= 100V, V = 150V
0.1
0.5
102
0.01
µA
LKG
IN
SW
l
R
FB
R
FB
Regulation Current
98
100
0.001
2.7
µA
RFB
Regulation Current Line Regulation
6V ≤ V ≤ 100V
%/V
ms
IN
t
Soft-Start Timer
SS
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The SW pin is rated to 150V for transients. Depending on the
leakage inductance voltage spike, operating waveforms of the SW pin
should be derated to keep the flyback voltage spike below 150V as shown
in Figure 5.
Note 3: The LT8300E is guaranteed to meet performance specifications
from 0°C to 125°C operating junction temperature. Specifications over
the –40°C to 125°C operating junction temperature range are assured by
design, characterization and correlation with statistical process controls.
The LT8300I is guaranteed over the full –40°C to 125°C operating junction
temperature range. The LT8300H is guaranteed over the full –40°C to
150°C operating junction temperature range. The LT8300MP is guaranteed
over the full –55°C to 150°C operating junction temperature range. High
junction temperatures degrade operating lifetimes. Operating lifetime is
derated at junction temperature greater than 125°C.
Note 4: The LT8300 includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 150°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
8300f
3
LT8300
TA = 25°C, unless otherwise noted.
Typical perForMance characTerisTics
Switching Frequency
vs Load Current
Output Load and Line Regulation
Output Temperature Variation
5.20
5.15
5.10
5.05
5.00
4.95
4.90
4.85
4.80
5.5
5.4
5.3
5.2
5.1
5.0
4.9
4.8
4.7
4.6
4.5
500
FRONT PAGE APPLICATION
FRONT PAGE APPLICATION
FRONT PAGE APPLICATION
V
= 48V, I
= 200mA
V
= 48V
IN
OUT
IN
400
300
200
100
0
V
V
V
= 36V
= 48V
= 72V
IN
IN
IN
0
50
100
150
200
250
300
–50 –25
0
25 50 75 100 125 150
0
50
100
150
200
250
300
LOAD CURRENT (mA)
AMBIENT TEMPERATURE (°C)
LOAD CURRENT (mA)
8300 G01
8300 G02
8300 G03
Boundary Mode Waveforms
Discontinuous Mode Waveforms
Burst Mode Waveforms
I
LPRI
100mA/DIV
I
I
LPRI
100mA/DIV
LPRI
100mA/DIV
V
SW
V
V
SW
50V/DIV
SW
50V/DIV
50V/DIV
V
OUT
V
V
OUT
50mV/DIV
OUT
50mV/DIV
50mV/DIV
8300 G04
8300 G05
8300 G06
2µs/DIV
2µs/DIV
20µs/DIV
FRONT PAGE APPLICATION
FRONT PAGE APPLICATION
FRONT PAGE APPLICATION
V
= 48V, I
= 300mA
V
= 48V, I
= 60mA
V
= 48V, I
= 1mA
IN
OUT
IN
OUT
IN
OUT
VIN Quiescent Current,
Sleep Mode
VIN Quiescent Current,
Active Mode
VIN Shutdown Current
100
90
80
70
60
50
380
360
340
320
300
280
10
8
T
= 150°C
= 25°C
J
T
= 150°C
J
6
T
T
= 25°C
J
J
4
T
= –55°C
J
T
= –55°C
J
T
= 25°C
J
T
= 150°C
40
J
2
T
= –55°C
80
J
40
0
0
20
40
60
80
100
0
20
40
60
80
100
0
20
60
(V)
100
V
(V)
V
(V)
IN
V
IN
IN
8300 G08
8300 G09
8300 G07
8300f
4
LT8300
TA = 25°C, unless otherwise noted.
Typical perForMance characTerisTics
EN/UVLO Enable Threshold
EN/UVLO Hysteresis Current
RFB Regulation Current
1.240
1.235
1.230
1.225
1.220
1.215
1.210
1.205
1.200
5
4
3
2
1
0
105
104
103
102
101
100
99
98
97
96
95
–50 –25
0
25 50 75 100 125 150
–50 –25
0
25 50 75 100 125 150
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
8300 G10
8300 G11
8300 G12
RDS(ON)
Switch Current Limit
Maximum Switching Frequency
300
250
200
150
100
50
25
20
15
10
5
1000
800
600
400
200
0
MAXIMUM CURRENT LIMIT
MINIMUM CURRENT LIMIT
0
0
–50 –25
0
25 50 75 100 125 150
–50 –25
0
25 50 75 100 125 150
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
8300 G14
8300 G13
8300 G15
Minimum Switching Frequency
Minimum Switch-On Time
Minimum Switch-Off Time
400
300
200
100
0
400
300
200
100
0
20
16
12
8
4
0
–50 –25
0
25 50 75 100 125 150
–50 –25
0
25 50 75 100 125 150
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
8300 G17
8300 G18
8300 G16
8300f
5
LT8300
pin FuncTions
EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The
EN/UVLO pin is used to enable the LT8300. Pull the pin
below 0.3V to shut down the LT8300. This pin has an ac-
mary SW pin. The ratio of the R resistor to the internal
FB
trimmed 12.23k resistor, times the internal bandgap
reference, determines the output voltage (plus the effect
of any non-unity transformer turns ratio). Minimize trace
area at this pin.
curate 1.223V threshold and can be used to program a V
IN
undervoltage lockout (UVLO) threshold using a resistor
divider from V to ground. A 2.5µA current hysteresis
IN
SW (Pin 4): Drain of the 150V Internal DMOS Power
Switch. Minimize trace area at this pin to reduce EMI and
voltage spikes.
allowstheprogrammingofV UVLOhysteresis. Ifneither
IN
function is used, tie this pin directly to V .
IN
GND (Pin 2): Ground. Tie this pin directly to local ground
plane.
V
(Pin 5): Input Supply. The V pin supplies current
IN
IN
to internal circuitry and serves as a reference voltage for
R
(Pin 3): Input Pin for External Feedback Resistor.
the feedback circuitry connected to the R pin. Locally
FB
FB
Connect a resistor from this pin to the transformer pri-
bypass this pin to ground with a capacitor.
8300f
6
LT8300
block DiagraM
T1
:1
D
OUT
N
V
PS
IN
+
–
V
OUT
•
C
IN
L
L
SEC
PRI
C
OUT
•
R
FB
V
OUT
5
3
4
V
R
FB
SW
IN
BOUNDARY
DETECTOR
1:4
M3
M2
OSCILLATOR
–
+
–
+
g
m
S
R
REF
25µA
12.23kΩ
A3
R
Q
DRIVER
1.223V
M1
R1
R2
EN/UVLO
–
+
–
1
R
SENSE
A2
0.3Ω
A1
2.5µA
+
V
1.223V
IN
GND
2
REFERENCE
REGULATORS
M4
8300 BD
8300f
7
LT8300
operaTion
The LT8300 is a current mode switching regulator IC de-
signed specially for the isolated flyback topology. The key
problem in isolated topologies is how to communicate the
output voltage information from the isolated secondary
side of the transformer to the primary side for regulation.
Historically, opto-isolators or extra transformer windings
communicatethisinformationacrosstheisolationbound-
ary. Opto-isolator circuits waste output power, and the
extra components increase the cost and physical size of
the power supply. Opto-isolators can also cause system
issuesduetolimiteddynamicresponse,nonlinearity,unit-
to-unitvariationandagingoverlifetime.Circuitsemploying
extra transformer windings also exhibit deficiencies, as
using an extra winding adds to the transformer’s physical
size and cost, and dynamic response is often mediocre.
conduction mode is a variable frequency, variable peak-
current switching scheme. The power switch turns on
and the transformer primary current increases until an
internally controlled peak current limit. After the power
switch turns off, the voltage on the SW pin rises to the
output voltage multiplied by the primary-to-secondary
transformer turns ratio plus the input voltage. When the
secondary current through the output diode falls to zero,
. A
the SW pin voltage collapses and rings around V
IN
boundary mode detector senses this event and turns the
power switch back on.
Boundaryconductionmodereturnsthesecondarycurrent
to zero every cycle, so parasitic resistive voltage drops
do not cause load regulation errors. Boundary conduc-
tion mode also allows the use of smaller transformers
compared to continuous conduction mode and does not
exhibit sub-harmonic oscillation.
The LT8300 samples the isolated output voltage through
the primary-side flyback pulse waveform. In this manner,
neither opto-isolator nor extra transformer winding is re-
quired for regulation. Since the LT8300 operates in either
boundary conduction mode or discontinuous conduction
mode, the output voltage is always sampled on the SW
pin when the secondary current is zero. This method im-
proves load regulation without the need of external load
compensation components.
Discontinuous Conduction Mode Operation
As the load gets lighter, boundary conduction mode in-
creases the switching frequency and decreases the switch
peakcurrentatthesameratio.Runningatahigherswitching
frequency up to several MHz increases switching and gate
charge losses. To avoid this scenario, the LT8300 has an
additional internal oscillator, which clamps the maximum
switching frequency to be less than 750kHz. Once the
switching frequency hits the internal frequency clamp,
the part starts to delay the switch turn-on and operates
in discontinuous conduction mode.
TheLT8300is asimple to use micropowerisolatedflyback
converterhousedina5-leadTSOT-23package.Theoutput
voltage is programmed with a single external resistor. By
integratingtheloopcompensationandsoft-startinside,the
part further reduces the number of external components.
As shown in the Block Diagram, many of the blocks are
similar to those found in traditional switching regulators
including reference, regulators, oscillator, logic, current
amplifier, current comparator, driver, and power switch.
The novel sections include a flyback pulse sense circuit,
a sample-and-hold error amplifier, and a boundary mode
detector, as well as the additional logic for boundary
conduction mode, discontinuous conduction mode, and
low ripple Burst Mode operation.
Low Ripple Burst Mode Operation
Unlike traditional flyback converters, the LT8300 has to
turn on and off at least for a minimum amount of time
andwithaminimumfrequencytoallowaccuratesampling
of the output voltage. The inherent minimum switch cur-
rent limit and minimum switch-off time are necessary to
guarantee the correct operation of specific applications.
As the load gets very light, the LT8300 starts to fold back
theswitchingfrequencywhilekeepingtheminimumswitch
current limit. So the load current is able to decrease while
still allowing minimum switch-off time for the sample-
and-hold error amplifier. Meanwhile, the part switches
betweensleepmodeandactivemode,therebyreducingthe
8300f
Boundary Conduction Mode Operation
TheLT8300featuresboundaryconductionmodeoperation
at heavy load, where the chip turns on the primary power
switch when the secondary current is zero. Boundary
8
LT8300
operaTion
effectivequiescentcurrenttoimprovelightloadefficiency.
In this condition, the LT8300 operates in low ripple Burst
Mode. The typical 7.5kHz minimum switching frequency
determines how often the output voltage is sampled and
also the minimum load requirement.
applicaTions inForMaTion
Output Voltage
bandgap reference voltage V . The resulting relationship
BG
between V
and V can be expressed as:
FLBK
BG
The R resistor as depicted in the Block Diagram is the
FB
only external resistor used to program the output voltage.
The LT8300 operates similar to traditional current mode
switchers, except in the use of a unique flyback pulse
sensecircuitandasample-and-holderroramplifier, which
sample and therefore regulate the isolated output voltage
from the flyback pulse.
V
FLBK
• R
= V
BG
REF
R
FB
or
V
BG
V
=
• R = I
• R
RFB FB
FLBK
FB
R
REF
Operation is as follows: when the power switch M1 turns
V
= Bandgap reference voltage
BG
off, the SW pin voltage rises above the V supply. The
IN
amplitude of the flyback pulse, i.e., the difference between
I
= R regulation current = 100µA
FB
RFB
the SW pin voltage and V supply, is given as:
IN
Combination with the previous V
equation yields an
FLBK
V
FLBK
= (V
+ V + I
• ESR) • N
SEC PS
equationforV , intermsoftheR resistor, transformer
OUT
F
OUT
FB
turns ratio, and diode forward voltage:
V = Output diode forward voltage
F
RFB
I
= Transformer secondary current
SEC
VOUT = 100µA •
− V
F
N
PS
ESR = Total impedance of secondary circuit
N
= Transformer effective primary-to-secondary
turns ratio
Output Temperature Coefficient
ThefirsttermintheV equationdoesnothavetempera-
PS
OUT
The flyback voltage is then converted to a current I
the flyback pulse sense circuit (M2 and M3). This cur-
rent I
REF
resultingvoltagefeedstotheinvertinginputofthesample-
and-hold error amplifier. Since the sample-and-hold error
amplifier samples the voltage when the secondary current
is zero, the (I • ESR) term in the V
assumed to be zero.
by
ture dependence, but the output diode forward voltage V
RFB
F
hasasignificantnegativetemperaturecoefficient(–1mV/°C
to–2mV/°C). Suchanegativetemperaturecoefficientpro-
duces approximately 200mV to 300mV voltage variation
on the output voltage across temperature.
also flows through the internal trimmed 12.23k
RFB
R
resistor to generate a ground-referred voltage. The
Forhighervoltageoutputs,suchas12Vand24V,theoutput
diodetemperaturecoefficienthasanegligibleeffectonthe
output voltage regulation. For lower voltage outputs, such
as 3.3V and 5V, however, the output diode temperature
coefficientdoescountforanextra2%to5%outputvoltage
regulation. For customers requiring tight output voltage
regulation across temperature, please refer to other LTC
parts with integratedtemperaturecompensation features.
equation can be
SEC
FLBK
The bandgap reference voltage V , 1.223V, feeds to the
BG
non-inverting input of the sample-and-hold error ampli-
fier. The relatively high gain in the overall loop causes
the voltage across R resistor to be nearly equal to the
REF
8300f
9
LT8300
applicaTions inForMaTion
Selecting Actual R Resistor Value
Output Power
FB
The LT8300 uses a unique sampling scheme to regulate
the isolated output voltage. Due to the sampling nature,
the scheme contains repeatable delays and error sources,
whichwillaffecttheoutputvoltageandforceare-evaluation
Aflybackconverterhasacomplicatedrelationshipbetween
the input and output currents compared to a buck or a
boostconverter.Aboostconverterhasarelativelyconstant
maximum input current regardless of input voltage and a
buck converter has a relatively constant maximum output
current regardless of input voltage. This is due to the
continuous non-switching behavior of the two currents. A
flybackconverterhasbothdiscontinuousinputandoutput
currentswhichmakeitsimilartoanon-isolatedbuck-boost
converter. The duty cycle will affect the input and output
currents, making it hard to predict output power. In ad-
dition, the winding ratio can be changed to multiply the
output current at the expense of a higher switch voltage.
of the R resistor value. Therefore, a simple two-step
FB
process is required to choose feedback resistor R .
FB
Rearrangement of the expression for V
in the Output
FB
OUT
Voltage section yields the starting value for R :
NPS • V
+ VF
(
)
OUT
RFB =
100µA
V
OUT
= Output voltage
V = Output diode forward voltage = ~0.3V
F
The graphs in Figures 1 to 4 show the typical maximum
output power possible for the output voltages 3.3V, 5V,
12V, and 24V. The maximum output power curve is the
calculated output power if the switch voltage is 120V dur-
ing the switch-off time. 30V of margin is left for leakage
inductance voltage spike. To achieve this power level at
a given input, a winding ratio value must be calculated
to stress the switch to 120V, resulting in some odd ratio
values. The curves below the maximum output power
curve are examples of common winding ratio values and
the amount of output power at given input voltages.
N
PS
= Transformer effective primary-to-secondary
turns ratio
Power up the application with the starting R value and
FB
other components connected, and measure the regulated
output voltage, V
adjusted to:
. The final R value can be
OUT(MEAS)
FB
VOUT
VOUT(MEAS)
RFB(FINAL)
=
•RFB
OncethefinalR valueisselected,theregulationaccuracy
FB
One design example would be a 5V output converter with
a minimum input voltage of 36V and a maximum input
voltage of 72V. A six-to-one winding ratio fits this design
example perfectly and outputs equal to 2.44W at 72V but
lowers to 1.87W at 36V.
from board to board for a given application will be very
consistent, typically under 5% when including device
variation of all the components in the system (assuming
resistor tolerances and transformer windings matching
within 1%). However, if the transformer or the output
diode is changed, or the layout is dramatically altered,
The following equations calculate output power:
there may be some change in V
.
OUT
POUT = η • VIN •D•ISW(MAX) • 0.5
η = Efficiency = 85%
V
+ V •N
F
PS
(
)
OUT
D = DutyCycle =
V
+ V •N + V
F IN
PS
(
)
OUT
I
= Maximum switch current limit = 260mA
SW(MAX)
8300f
10
LT8300
applicaTions inForMaTion
3.5
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
MAXIMUM
OUTPUT
POWER
MAXIMUM
OUTPUT
POWER
3.0
2.5
2.0
1.5
1.0
0.5
0
N = 12:1
N = 8:1
N = 6:1
N = 4:1
N = 8:1
N = 6:1
N = 4:1
N = 2:1
0
20
40
60
80
100
0
20
40
60
80
100
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
8300 F01
8300 F02
Figure 1. Output Power for 3.3V Output
Figure 2. Output Power for 5V Output
3.5
3.5
N = 4:1
MAXIMUM
OUTPUT
POWER
N = 2:1
3.0
2.5
2.0
1.5
1.0
0.5
0
3.0
2.5
2.0
1.5
1.0
0.5
0
N = 3:1
N = 3:2
N = 1:1
MAXIMUM
OUTPUT
POWER
N = 2:1
N = 1:1
N = 1:2
0
20
40
60
80
100
0
20
40
60
80
100
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
8300 F03
8300 F04
Figure 3. Output Power for 12V Output
Figure 4. Output Power for 24V Output
Primary Inductance Requirement
In addition to the primary inductance requirement for
the minimum switch-off time, the LT8300 has minimum
switch-on time that prevents the chip from turning on
the power switch shorter than approximately 160ns. This
minimumswitch-ontimeismainlyforleading-edgeblank-
ing the initial switch turn-on current spike. If the inductor
current exceeds the desired current limit during that time,
oscillation may occur at the output as the current control
loopwillloseitsabilitytoregulate.Therefore,thefollowing
equation relating to maximum input voltage must also be
followedinselectingprimary-sidemagnetizinginductance:
The LT8300 obtains output voltage information from the
reflected output voltage on the SW pin. The conduction
of secondary current reflects the output voltage on the
primarySWpin.Thesample-and-holderroramplifierneeds
aminimum350nstosettleandsamplethereflectedoutput
voltage. In order to ensure proper sampling, the second-
ary winding needs to conduct current for a minimum of
350ns. The following equation gives the minimum value
for primary-side magnetizing inductance:
tOFF(MIN) •NPS • V
+ VF
(
)
OUT
tON(MIN) • VIN(MAX)
LPRI
≥
LPRI
≥
ISW(MIN)
ISW(MIN)
t
= Minimum switch-off time = 350ns
OFF(MIN)
t
= Minimum Switch-On Time = 160ns
ON(MIN)
I
= Minimum switch current limit = 52mA
SW(MIN)
8300f
11
LT8300
applicaTions inForMaTion
In general, choose a transformer with its primary mag-
netizing inductance about 20% to 40% larger than the
minimum values calculated above. A transformer with
much larger inductance will have a bigger physical size
and may cause instability at light load.
Linear Technology has worked with several leading mag-
netic component manufacturers to produce pre-designed
flyback transformers for use with the LT8300. Table 1
shows the details of these transformers.
Turns Ratio
Selecting a Transformer
Note that when choosing the R resistor to set output
FB
Transformer specification and design is perhaps the most
criticalpartofsuccessfullyapplyingtheLT8300.Inaddition
to the usual list of guidelines dealing with high frequency
isolated power supply transformer design, the following
information should be carefully considered.
voltage, the user has relative freedom in selecting a trans-
former turns ratio to suit a given application. In contrast,
the use of simple ratios of small integers, e.g., 4:1, 2:1,
1:1, provides more freedom in settling total turns and
mutual inductance.
Table 1. Predesigned Transformers — Typical Specifications
TRANSFORMER
PART NUMBER
L
L
LEAKAGE
(µH)
PRI
(µH)
400
300
NP:NS:NB
8:1
VENDOR
TARGET APPLICATIONS
750312367
750312557
4.5
2.5
Würth Elektronik
Würth Elektronik
48V to 3.3V/0.51A, 24V to 3.3V/0.37A, 12V to 3.3V/0.24A
6:1
48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A
48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A
750312365
750312558
300
300
1.8
4:1
Würth Elektronik
Würth Elektronik
48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
1.75
2:1:1
48V to 12V/67mA, 24V to 12V/50mA, 12V to 12V/33mA
48V to 15V/62mA, 24V to 15V/44mA, 12V to 15V/28mA
750312559
750311019
300
400
2
5
1:1
Würth Elektronik
Würth Elektronik
48V to 24V/67mA, 24V to 24V/50mA, 12V to 24V/33mA
6:1:2
48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A
48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A
750311558
750311660
300
350
1.5
3
4:1:1
Würth Elektronik
Würth Elektronik
48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
2:1:0.33
48V to 12V/0.134A, 24V to 12V/0.1A, 12V to 12V/0.066A
48V to 15V/0.124A, 24V to 15V/0.088A, 12V to 15V/0.056A
750311838
350
3
2:1:1
Würth Elektronik
48V to 12V/67mA, 24V to 12V/50mA, 12V to 12V/33mA
48V to 15V/62mA, 24V to 15V/44mA, 12V to 15V/28mA
750311659
10396-T026
300
300
2
1:1:0.2
6:1:2
Würth Elektronik
Sumida
48V to 24V/67mA, 24V to 24V/50mA, 12V to 24V/33mA
2.5
48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A
48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A
10396-T024
10396-T022
300
300
2
2
4:1:1
Sumida
Sumida
48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
2:1:0.33
48V to 12V/0.134A, 24V to 12V/0.1A, 12V to 12V/0.066A
48V to 15V/0.124A, 24V to 15V/0.088A, 12V to 15V/0.056A
10396-T028
L10-0116
300
500
2.5
7.3
2:1:1
6:1
Sumida
48V to 12V/67mA, 24V to 12V/50mA, 12V to 12V/33mA
48V to 15V/62mA, 24V to 15V/44mA, 12V to 15V/28mA
BH Electronics
48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A
48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A
L10-0112
L11-0067
230
230
3.38
2.16
4:1
4:1
BH Electronics
BH Electronics
48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
* All the transformers are rated for 1.5kV Isolation.
8300f
12
LT8300
applicaTions inForMaTion
Typically, choose the transformer turns ratio to maximize
available output power. For low output voltages (3.3V or
5V), a larger N:1 turns ratio can be used with multiple
primarywindingsrelativetothesecondarytomaximizethe
transformer’s current gain (and output power). However,
remember that the SW pin sees a voltage that is equal
to the maximum input supply voltage plus the output
voltage multiplied by the turns ratio. In addition, leakage
Saturation Current
Thecurrentinthetransformerwindingsshouldnotexceed
itsratedsaturationcurrent.Energyinjectedoncethecoreis
saturated will not be transferred to the secondary and will
instead be dissipated in the core. When designing custom
transformers to be used with the LT8300, the saturation
current should always be specified by the transformer
manufacturers.
inductance will cause a voltage spike (V
) on top of
LEAKAGE
this reflected voltage. This total quantity needs to remain
below the 150V absolute maximum rating of the SW pin
to prevent breakdown of the internal power switch. To-
gether these conditions place an upper limit on the turns
Winding Resistance
Resistance in either the primary or secondary windings
will reduce overall power efficiency. Good output voltage
regulation will be maintained independent of winding re-
sistance due to the boundary/discontinuous conduction
mode operation of the LT8300.
ratio, N , for a given application. Choose a turns ratio
PS
low enough to ensure:
150V − VIN(MAX) − VLEAKAGE
NPS <
Leakage Inductance and Snubbers
VOUT + V
F
Transformer leakage inductance on either the primary or
secondarycausesavoltagespiketoappearontheprimary
after the power switch turns off. This spike is increasingly
prominent at higher load currents where more stored en-
ergy must be dissipated. It is very important to minimize
transformer leakage inductance.
For lower output power levels, choose a smaller N:1 turns
ratio to alleviate the SW pin voltage stress. Although a
1:N turns ratio makes it possible to have very high output
voltages without exceeding the breakdown voltage of the
internalpowerswitch, themultipliedparasiticcapacitance
through turns ratio coupled with the relatively resistive
150V internal power switch may cause the switch turn-on
currentspikeringingbeyond160nsleading-edgeblanking,
thereby producing light load instability in certain applica-
tions. So any 1:N turns ratio should be fully evaluated
before its use with the LT8300.
When designing an application, adequate margin should
be kept for the worst-case leakage voltage spikes even
under overload conditions. In most cases shown in Figure
5, the reflected output voltage on the primary plus V
IN
shouldbekeptbelow120V. Thisleavesatleast30Vmargin
for the leakage spike across line and load conditions. A
larger voltage margin will be required for poorly wound
transformers or for excessive leakage inductance.
The turns ratio is an important element in the isolated
feedback scheme, and directly affects the output voltage
accuracy. Make sure the transformer manufacturer speci-
fies turns ratio accuracy within 1%.
In addition to the voltage spikes, the leakage inductance
also causes the SW pin ringing for a while after the power
switchturnsoff. Topreventthevoltageringingfalselytrig-
ger boundary mode detector, the LT8300 internally blanks
theboundarymodedetectorforapproximately250ns.Any
remaining voltage ringing after 250ns may turn the power
switch back on again before the secondary current falls
to zero. So the leakage inductance spike ringing should
be limited to less than 250ns.
8300f
13
LT8300
applicaTions inForMaTion
V
SW
V
V
SW
SW
<150V
<120V
<150V
<120V
<150V
<120V
V
V
V
LEAKAGE
LEAKAGE
LEAKAGE
t
> 350ns
t
> 350ns
t
> 350ns
OFF
OFF
OFF
t
SP
< 250ns
t
SP
< 250ns
t
SP
< 250ns
TIME
TIME
TIME
8300 F05
No Snubber
with DZ Snubber
with RC Snubber
Figure 5. Maximum Voltages for SW Pin Flyback Waveform
L
L
ℓ
ℓ
Z
•
•
C
•
•
R
D
8300 F06a
8300 F06b
DZ Snubber
RC Snubber
Figure 6. Snubber Circuits
A snubber circuit is recommended for most applications.
Two types of snubber circuits shown in Figure 6 that can
protect the internal power switch include the DZ (diode-
Zener)snubberandtheRC(resistor-capacitor)snubber.The
DZ snubber ensures well defined and consistent clamping
voltage and has slightly higher power efficiency, while the
RC snubber quickly damps the voltage spike ringing and
provides better load regulation and EMI performance.
Figure 5 shows the flyback waveforms with the DZ and
RC snubbers.
The Zener diode breakdown voltage should be chosen to
balancepowerlossandswitchvoltageprotection.Thebest
compromise is to choose the largest voltage breakdown.
Use the following equation to make the proper choice:
V
≤ 150V – V
IN(MAX)
ZENER(MAX)
For an application with a maximum input voltage of 72V,
choose a 68V Zener diode, the V of which is
ZENER(MAX)
around 72V and below the 78V maximum.
The power loss in the clamp will determine the power rat-
ing of the Zener diode. Power loss in the clamp is highest
at maximum load and minimum input voltage. The switch
currentishighestatthispointalongwiththeenergystored
in the leakage inductance. A 0.5W Zener will satisfy most
For the DZ snubber, proper care must be taken when
choosing both the diode and the Zener diode. Schottky
diodes are typically the best choice, but some PN diodes
can be used if they turn on fast enough to limit the leak-
age inductance spike. Choose a diode that has a reverse-
voltage rating higher than the maximum SW pin voltage.
applications when the highest V
is chosen.
ZENER
8300f
14
LT8300
applicaTions inForMaTion
Tables 2 and 3 show some recommended diodes and
Zener diodes.
Note that energy absorbed by the RC snubber will be
converted to heat and will not be delivered to the load.
In high voltage or high current applications, the snubber
may need to be sized for thermal dissipation.
Table 2. Recommended Zener Diodes
V
ZENER
(V)
POWER
(W)
PART
CASE
VENDOR
Undervoltage Lockout (UVLO)
MMSZ5266BT1G
MMSZ5270BT1G
CMHZ5266B
CMHZ5267B
BZX84J-68
BZX100A
68
91
0.5
0.5
0.5
0.5
0.5
0.5
SOD-123 On Semi
SOD-123
A resistive divider from V to the EN/UVLO pin imple-
IN
ments undervoltage lockout (UVLO). The EN/UVLO pin
falling threshold is set at 1.223V with 16mV hysteresis.
In addition, the EN/UVLO pin sinks 2.5µA when the volt-
age at the pin is below 1.223V. This current provides user
programmable hysteresis based on the value of R1. The
programmable UVLO thresholds are:
68
SOD-123 Central
Semiconductor
75
SOD-123
68
SOD323F NXP
SOD323F
100
Table 3. Recommended Diodes
V
REVERSE
1.239V •(R1+ R2)
PART
I (A)
0.625
0.625
(V)
CASE
VENDOR
V
=
=
+ 2.5µA •R1
IN(UVLO+ )
R2
1.223V •(R1+ R2)
R2
BAV21W
BAV20W
200
150
SOD-123 Diodes Inc.
SOD-123
V
IN(UVLO− )
The recommended approach for designing an RC snubber
is to measure the period of the ringing on the SW pin when
the power switch turns off without the snubber and then
add capacitance (starting with 100pF) until the period of
the ringing is 1.5 to 2 times longer. The change in period
will determine the value of the parasitic capacitance, from
which the parasitic inductance can be determined from
the initial period, as well. Once the value of the SW node
capacitanceandinductanceisknown,aseriesresistorcan
be added to the snubber capacitance to dissipate power
andcriticallydampentheringing.Theequationforderiving
the optimal series resistance using the observed periods
Figure 7 shows the implementation of external shutdown
control while still using the UVLO function. The NMOS
grounds the EN/UVLO pin when turned on, and puts the
LT8300 in shutdown with quiescent current less than 2µA.
V
IN
R1
R2
EN/UVLO
LT8300
RUN/STOP
CONTROL
(OPTIONAL)
GND
( t
and t
) is:
) and snubber capacitance
PERIOD(SNUBBED)
PERIOD
(C
8300 F07
SNUBBER
CSNUBBER
Figure 7. Undervoltage Lockout (UVLO)
CPAR
=
2
t
PERIOD(SNUBBED)
− 1
tPERIOD
2
tPERIOD
LPAR
=
CPAR • 4π2
LPAR
CPAR
RSNUBBER
=
8300f
15
LT8300
applicaTions inForMaTion
Minimum Load Requirement
Design Example
The LT8300 samples the isolated output voltage from the
primary-side flyback pulse waveform. The flyback pulse
occursoncetheprimaryswitchturnsoffandthesecondary
winding conducts current. In order to sample the output
voltage, the LT8300 has to turn on and off at least for a
minimum amount of time and with a minimum frequency.
The LT8300 delivers a minimum amount of energy even
duringlightloadconditionstoensureaccurateoutputvolt-
age information. The minimum energy delivery creates a
minimum load requirement, which can be approximately
Use the following design example as a guide to design
applications for the LT8300. The design example involves
designing a 12V output with a 120mA load current and an
input range from 36V to 72V.
V
V
= 36V, V
= 48V, V
= 72V,
IN(MIN)
OUT
IN(NOM)
= 120mA
IN(MAX)
= 12V, I
OUT
Step 1: Select the Transformer Turns Ratio.
150V − VIN(MAX) − VLEAKAGE
NPS <
VOUT + VF
estimated as:
2
L
•I
• f
PRI
MIN
SW(MIN)
V
= Margin for transformer leakage spike = 30V
I
=
LEAKAGE
LOAD(MIN)
2 • V
OUT
V = Output diode forward voltage = ~0.3V
F
L
PRI
= Transformer primary inductance
Example:
I
f
= Minimum switch current limit = 52mA
SW(MIN)
150V − 72V − 30V
12V + 0.3V
NPS <
= 3.9
= Minimum switching frequency = 7.5kHz
MIN
The LT8300 typically needs less than 0.5% of its full output
power as minimum load. Alternatively, a Zener diode with its
breakdown of 20% higher than the output voltage can serve
as a minimum load if pre-loading is not acceptable. For a 5V
output,usea6VZenerwithcathodeconnectedtotheoutput.
The choice of transformer turns ratio is critical in deter-
mining output current capability of the converter. Table 4
shows the switch voltage stress and output current capa-
bility at different transformer turns ratio.
Table 4. Switch Voltage Stress and Output Current Capability
vs Turns Ratio
Output Short Protection
V
V
at
(V)
I
at
OUT(MAX)
SW(MAX)
When the output is heavily overloaded or shorted, the
reflected SW pin waveform rings longer than the internal
blanking time. After the 350ns minimum switch-off time,
the excessive ring falsely trigger the boundary mode
detector and turn the power switch back on again before
the secondary current falls to zero. Under this condition,
the LT8300 runs into continuous conduction mode at
750kHz maximum switching frequency. Depending on the
N
V
(mA)
DUTY CYCLE (%)
15-25
PS
IN(MAX)
IN(MIN)
1:1
2:1
3:1
84.3
84
96.6
135
168
25-41
108.9
34-51
SincebothN =2andN =3canmeetthe120mAoutput
PS
PS
current requirement, N = 2 is chosen in this example
PS
to allow more margin for transformer leakage inductance
V supply voltage, the switch current may run away and
IN
voltage spike.
exceed 260mA maximum current limit. Once the switch
current hits 520mA over current limit, a soft-start cycle
initiates and throttles back both switch current limit and
switchfrequency.Thisoutputshortprotectionpreventsthe
switch current from running away and limits the average
output diode current.
8300f
16
LT8300
applicaTions inForMaTion
Step 2: Determine the Primary Inductance.
Example:
Primary inductance for the transformer must be set above
a minimum value to satisfy the minimum switch-off and
switch-on time requirements:
(12V + 0.3V)• 2
(12V + 0.3V)• 2+ 48V
12V • 0.12A • 2
D =
= 0.34
ISW
=
= 0.21A
0.85 • 48V • 0.34
fSW = 260kHz
tOFF(MIN) •NPS • V
+ VF
(
)
OUT
LPRI
≥
≥
ISW(MIN)
tON(MIN) • VIN(MAX)
The transformer also needs to be rated for the correct
saturation current level across line and load conditions. A
saturation current rating larger than 400mA is necessary
to work with the LT8300. The 10396-T022 from Sumida
is chosen as the flyback transformer.
LPRI
ISW(MIN)
t
= 350ns
= 160ns
= 52mA
OFF(MIN)
ON(MIN)
SW(MIN)
t
I
Step 3: Choose the Output Diode.
Two main criteria for choosing the output diode include
forward current rating and reverse voltage rating. The
maximum load requirement is a good first-order guess
as the average current requirement for the output diode.
Aconservativemetricisthemaximumswitchcurrentlimit
multiplied by the turns ratio,
Example:
350ns • 2 •(12V + 0.3V)
LPRI
LPRI
≥
≥
= 166µH
52mA
160ns • 72V
= 222µH
52mA
I
= I
• N
SW(MAX) PS
DIODE(MAX)
Mosttransformersspecifyprimaryinductancewithatoler-
anceof 20%.Withothercomponenttoleranceconsidered,
choose a transformer with its primary inductance 20% to
40% larger than the minimum values calculated above.
Example:
I
= 0.52A
DIODE(MAX)
Next calculate reverse voltage requirement using maxi-
mum V :
L
PRI
= 300µH is then chosen in this example.
IN
Once the primary inductance has been determined, the
maximum load switching frequency can be calculated as:
VIN(MAX)
VREVERSE = VOUT
Example:
VREVERSE = 12V +
+
NPS
1
1
fSW
=
=
LPRI •ISW
LPRI •ISW
tON + tOFF
+
VIN
NPS •(VOUT + VF )
72V
2
= 48V
VOUT •IOUT • 2
η • VIN •D
ISW
=
The SBR0560S1 (0.5A, 60V diode) from Diodes Inc. is
chosen.
8300f
17
LT8300
applicaTions inForMaTion
Step 4: Choose the Output Capacitor.
A 68V Zener with a maximum of 72V will provide optimal
protection and minimize power loss. So a 68V, 0.5W Zener
from On Semiconductor (MMSZ5266BT1G) is chosen.
The output capacitor should be chosen to minimize the
outputvoltageripplewhileconsideringtheincreaseinsize
and cost of a larger capacitor. Use the equation below to
calculate the output capacitance:
Choose a diode that is fast and has sufficient reverse
voltage breakdown:
2
V
V
> V
SW(MAX)
REVERSE
SW(MAX)
L
•I
SW
PRI
C
=
OUT
= V
+ V
ZENER(MAX)
IN(MAX)
2 • V
• ∆V
OUT
OUT
Example:
Example:
V
> 144V
REVERSE
Design for output voltage ripple less than 1% of V
i.e., 120mV.
,
OUT
A 150V, 0.6A diode from Diodes Inc. (BAV20W) is chosen.
Step 6: Select the R Resistor.
300µH•(0.21A)2
2 •12V • 0.12V
FB
COUT
=
= 4.6µF
Use the following equation to calculate the starting value
for R :
FB
Remember ceramic capacitors lose capacitance with ap-
plied voltage. The capacitance can drop to 40% of quoted
capacitanceatthemaximumvoltagerating.Soa10uF,16V
rating ceramic capacitor is chosen.
NPS •(VOUT + VF )
RFB =
100µA
Example:
Step 5: Design Snubber Circuit.
2 •(12V + 0.3V)
RFB =
= 246k
Thesnubbercircuitprotectsthepowerswitchfromleakage
inductance voltage spike. A DZ snubber is recommended
for this application because of lower leakage inductance
and larger voltage margin. The Zener and the diode need
to be selected.
100µA
Depending on the tolerance of standard resistor values,
the precise resistor value may not exist. For 1% standard
values, a 243k resistor in series with a 3.01k resistor
should be close enough. As discussed in the Application
Informationsection, thefinalR valueshouldbeadjusted
on the measured output voltage.
The maximum Zener breakdown voltage is set according
to the maximum V :
FB
IN
V
≤ 150V – V
IN(MAX)
ZENER(MAX)
Example:
V
≤ 150V – 72V = 78V
ZENER(MAX)
8300f
18
LT8300
applicaTions inForMaTion
Step 7: Select the EN/UVLO Resistors.
Step 8: Ensure minimum load.
Determinetheamountofhysteresisrequiredandcalculate
R1 resistor value:
The theoretical minimum load can be approximately
estimated as:
300µH•(52mA)2 • 7.5kHz
V
= 2.5µA • R1
IN(HYS)
ILOAD(MIN)
=
= 0.25mA
Example:
2 •12V
Choose 2.5V of hysteresis,
R1 = 1M
Remembertochecktheminimumloadrequirementinreal
application. The minimum load occurs at the point where
the output voltage begins to climb up as the converter
delivers more energy than what is consumed at the out-
put. The real minimum load for this application is about
0.6mA, 0.5% of 120mA maximum load. In this example,
a 20k resistor is selected as the minimum load.
Determine the UVLO thresholds and calculate R2 resistor
value:
1.239V •(R1+ R2)
V
=
+ 2.5µA •R1
IN(UVLO+ )
R2
Example:
Set V UVLO rising threshold to 34.5V,
IN
R2 = 40.2k
V
V
= 34.1V
= 31.6V
IN(UVLO+)
IN(UVLO–)
8300f
19
LT8300
Typical applicaTions
5V Micropower Isolated Flyback Converter
D1
T1
+
V
OUT
V
IN
5V
1mA TO 330mA
6:1
36V TO 72V
•
2.2µF
300µH
8µH
V
IN
47µF
1M
•
LT8300
EN/UVLO
–
SW
V
OUT
40.2k
316k
R
FB
GND
T1: WÜRTH 750312557
D1: DIODES INC. SBR2A30P1
8300 TA02
12V Micropower Isolated Flyback Converter
T1
2:1
D1
+
V
OUT
V
IN
12V
0.6mA TO 120mA
36V TO 72V
•
2.2µF
300µH
75µH
V
IN
10µF
1M
•
LT8300
EN/UVLO
–
SW
V
OUT
40.2k
243k
R
FB
GND
T1: SUMIDA 10396-TO22
D1: DIODES INC. SBR0560S1
8300 TA03
8300f
20
LT8300
Typical applicaTions
24V Micropower Isolated Flyback Converter
T1
1:1
D1
+
V
OUT
24V
0.3mA TO 60mA
V
IN
36V TO 72V
•
2.2µF
300µH
300µH
V
IN
LT8300
EN/UVLO
4.7µF
1M
•
–
SW
V
OUT
40.2k
243k
R
FB
GND
T1: WÜRTH 750311559
D1: DIODES DFLS 1200-7
8300 TA04
3.3V Micropower Isolated Flyback Converter
T1
8:1
D1
+
V
OUT
3.3V
2mA TO 440mA
V
IN
36V TO 72V
•
2.2µF
400µH
6µH
V
IN
LT8300
EN/UVLO
100µF
1M
•
–
SW
V
OUT
40.2k
287k
R
FB
GND
T1: WÜRTH 750312367
D1: NXP PMEG2020EH
8300 TA05
8300f
21
LT8300
Typical applicaTions
VIN to (VIN + 10V) Micropower Converter
+
V
OUT
10V
50mA
–
4.7µF
Z1
V
OUT
V
IN
15V TO 80V
1µF
L1
330µH
V
IN
1M
LT8300
EN/UVLO
D1
SW
118k
102k
R
FB
GND
8300 TA06
L1: COILTRONICS DR73-331-R
D1: DIODES INC. SBR1U150SA
Z1: CENTRAL CMDZ12L
VIN to (VIN – 10V) Micropower Converter
V
IN
+
15V TO 80V
V
OUT
10V
1µF
4.7µF
Z1
100mA
–
V
OUT
L1
330µH
V
IN
1M
LT8300
EN/UVLO
D1
SW
118k
102k
R
FB
GND
8300 TA07
L1: COILTRONICS DR73-331-R
D1: DIODES INC. SBR1U150SA
Z1: CENTRAL CMDZ12L
8300f
22
LT8300
package DescripTion
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
S5 Package
5-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1635 Rev B)
0.62
MAX
0.95
REF
2.90 BSC
(NOTE 4)
1.22 REF
1.4 MIN
1.50 – 1.75
(NOTE 4)
2.80 BSC
3.85 MAX 2.62 REF
PIN ONE
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45 TYP
5 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
DATUM ‘A’
0.01 – 0.10
1.00 MAX
0.30 – 0.50 REF
1.90 BSC
0.09 – 0.20
(NOTE 3)
NOTE:
S5 TSOT-23 0302 REV B
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
8300f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT8300
Typical applicaTion
3.3V Isolated Converter (Conforming to DEF-STAN61-5)
L1
1:1
D1
+
V
OUT
V
IN
IN
OUT
LT3009-3.3
SHDN
3.3V
0mA TO 20mA
18V TO 32V
•
1µF
150µH
150µH
V
IN
Z1
1µF
1µF
1M
•
GND
LT8300
EN/UVLO
–
SW
V
OUT
93.1k
42.2k
D1: DIODES INC. SBR0560S1-7
L1: DRQ73-151-R
Z1: CENTRAL CMDZ4L7
R
FB
GND
8300 TA08a
Input Current with No Load
400
300
200
100
0
18
20
22
24
V
26
(V)
28
30
32
IN
8300 TA08b
relaTeD parTs
PART NUMBER
DESCRIPTION
COMMENTS
LT3511/LT3512
100V Isolated Flyback Converters
Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch,
MSOP-16(12)
LT3748
LT3798
100V Isolated Flyback Controller
5V ≤ V ≤ 100V, No Opto Flyback , MSOP-16 with High Voltage Spacing
IN
Off-Line Isolated No Opto-Coupler Flyback Controller
with Active PFC
V
IN
and V
Limited Only by External Components
OUT
LT3573/LT3574/LT3575 40V Isolated Flyback Converters
LT3757/LT3759/LT3758 40V/100V Flyback/Boost Controllers
Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch
Universal Controllers with Small Package and Powerful Gate Drive
Monolithic with Integrated 5A/3.3A Switch
LT3957/LT3958
40V/100V Flyback/Boost Converters
LTC3803/LTC3803-3/
LTC3803-5
200kHz/300kHz Flyback Controllers in SOT-23
V
and V
Limited by External Components
IN
OUT
LTC3805/LTC3805-5
Adjustable Frequency Flyback Controllers
V
and V
Limited by External Components
IN
OUT
8300f
LT 0812 • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
24
●
●
LINEAR TECHNOLOGY CORPORATION 2012
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
相关型号：
BAV21W-E3-08
Rectifier Diode, 1 Element, 0.2A, 250V V(RRM), Silicon, ROHS COMPLIANT PACKAGE-2
VISHAY
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