LT1076HV [Linear]
High Efficiency, Synchronous, 4-Switch Buck-Boost Controller; 高英法fi效率，同步，四开关降压 - 升压型控制器
| 型号： | LT1076HV |
| 厂家： | 
Linear |
| 描述： | High Efficiency, Synchronous, 4-Switch Buck-Boost Controller |
| 文件： | 总28页 (文件大小：421K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC3780
High Efficiency, Synchronous,
4-Switch Buck-Boost Controller
U
FEATURES
DESCRIPTIO
The LTC®3780 is a high performance buck-boost switch-
ing regulator controller that operates from input voltages
above, below or equal to the output voltage. The constant
frequency current mode architecture allows a phase-
lockable frequency of up to 400kHz. With a wide 4V to 30V
(36V maximum) input and output range and seamless
transfers between operating modes, the LTC3780 is ideal
for automotive, telecom and battery-powered systems.
■
Single Inductor Architecture Allows VIN Above,
Below or Equal to VOUT
■
Wide VIN Range: 4V to 36V Operation
■
Synchronous Rectification: Up to 98% Efficiency
■
Current Mode Control
■
±1% Output Voltage Accuracy: 0.8V < VOUT < 30V
■
Phase-Lockable Fixed Frequency: 200kHz to 400kHz
■
Power Good Output Voltage Monitor
■
Internal LDO for MOSFET Supply
Theoperatingmodeofthecontrollerisdeterminedthrough
the FCB pin. For boost operation, the FCB mode pin can
selectamongBurstMode® operation,Discontinuousmode
and Forced Continuous mode. During buck operation, the
FCBmodepincanselectamongSkip-Cyclemode,Discon-
tinuous mode and Forced Continuous mode. Burst Mode
operation and Skip-Cycle mode provide high efficiency
operation at light loads while Forced Continuous mode
and Discontinuous mode operate at a constant frequency.
■
Quad N-Channel MOSFET Synchronous Drive
■
VOUT Disconnected from VIN During Shutdown
■
Adjustable Soft-Start Current Ramping
■
Foldback Output Current Limiting
■
Selectable Low Current Modes
■
Output Overvoltage Protection
■
Available in 24-Lead SSOP and Exposed Pad
(5mm × 5mm) 3U2-Lead QFN Packages
Fault protection is provided by an output overvoltage
comparatorandinternalfoldbackcurrentlimiting.APower
Good output pin indicates when the output is within 7.5%
of its designed set point.
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a registered trademark of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
APPLICATIO S
■
Automotive Systems
■
Telecom Systems
■
DC Power Distribution Systems
■
High Power Battery-Operated Devices
■
Industrial Control
Protected by U.S. Patents, including 5481178, 6304066, 5929620, 5408150, 6580258,
patent pending on current mode architecture and protection
U
TYPICAL APPLICATIO
High Efficiency Buck-Boost Converter
V
12V
5A
OUT
V
IN
4V TO 36V
100µF
16V
CER
22µF
50V
CER
+
Efficiency and Power Loss
OUT = 12V, ILOAD = 5A
4.7µF
1µF
CER
V
V
PGOOD INTV
IN
CC
TG2
TG1
100
95
10
9
8
7
6
5
4
3
2
1
0
0.1µF
0.1µF
BOOST2
SW2
BOOST1
SW1
LTC3780
BG2
BG1
PLLIN
RUN
90
I
TH
105k
1%
2200pF
ON/OFF
85
SS
V
20k
0.1µF
OSENSE
80
75
70
SGND
SENSE SENSE PGND
FCB
7.5k
+
–
1000pF
0.010Ω
2µH
0
5
10
15
V
20
25
30
35
(V)
IN
3780 TA01b
3780 TA01
3780f
1
LTC3780
ABSOLUTE MAXIMUM RATINGS
W W U W
(Note 1)
Input Supply Voltage (VIN)........................ –0.3V to 36V
Topside Driver Voltages
(BOOST1, BOOST2) .................................. –0.3V to 42V
Switch Voltage (SW1, SW2) ........................ –5V to 36V
INTVCC, EXTVCC, RUN, SS, (BOOST – SW1),
(BOOST2 – SW2), PGOOD.......................... –0.3V to 7V
PLLIN Voltage.......................................... –0.3V to 5.5V
PLLFLTR Voltage ......................................–0.3V to 2.7V
FCB, STBYMD Voltages ....................... –0.3V to INTVCC
ITH, VOSENSE Voltages .............................. –0.3V to 2.4V
Peak Output Current <10ms (TG1, TG2, BG1, BG2) .. 3A
INTVCC Peak Output Current ................................ 40mA
Operating Temperature Range (Note 7)
LTC3780E........................................... – 40°C to 85°C
LTC3780I............................................ – 40°C to 85°C
Junction Temperature (Note 2)............................ 125°C
Storage Temperature Range .................. –65°C to 125°C
Lead Temperature (Soldering, 10 sec)
SSOP Only........................................................ 300°C
U
W U
PACKAGE/ORDER INFORMATION
TOP VIEW
ORDER PART
NUMBER
ORDER PART
NUMBER
TOP VIEW
1
2
BOOST1
TG1
24
23
22
21
20
19
18
17
16
15
14
13
PGOOD
SS
32 31 30 29 28 27 26 25
LTC3780EG
LTC3780IG
LTC3780EUH
+
SENSE
SENSE
I
1
2
3
4
5
6
7
8
24 SW1
LTC3780IUH
+
3
SW1
SENSE
–
23
V
IN
–
4
V
SENSE
IN
EXTV
INTV
22
TH
CC
5
EXTV
CC
I
TH
V
21
OSENSE
SGND
CC
33
6
INTV
CC
V
OSENSE
SGND
20 BG1
7
BG1
RUN
FCB
PGND
19
UH PART
MARKING
8
PGND
BG2
RUN
FCB
18 BG2
17 SW2
9
PLLFTR
10
11
12
SW2
9
10 11 12 13 14 15 16
PLLFLTR
PLLIN
3780
3780I
TG2
BOOST2
STBYMD
UH PACKAGE
G PACKAGE
24-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 130°C/W
32-LEAD (5mm × 5mm) PLASTIC QFN
TJMAX = 125°C, θJA = 34°C/W
EXPOSED PAD (PIN 33) IS SGND
(MUST BE SOLDERED TO PCB)
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS The ● indicates specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Main Control Loop
V
Feedback Reference Voltage
Feedback Pin Input Current
Output Voltage Load Regulation
I
= 1.2V (Note 3)
TH
●
0.792
0.800
–5
0.808
–50
V
OSENSE
I
(Note 3)
nA
VOSENSE
V
(Note 3)
LOADREG
∆I = 1.2V to 0.7V
∆I = 1.2V to 1.8V
TH
●
●
0.1
–0.1
0.5
–0.5
%
%
TH
3780f
2
LTC3780
ELECTRICAL CHARACTERISTICS The ● indicates specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
0.002
0.32
0.6
MAX
UNITS
%/V
V
Reference Voltage Line Regulation
Error Amplifier Transconductance
Error Amplifier GBW
V
= 4V to 30V, I = 1.2V (Note 3)
0.02
REF(LINEREG)
m(EA)
IN
TH
g
g
I
= 1.2V, Sink/Source = 3µA (Note 3)
mS
TH
MHz
m(GBW)
I
Input DC Supply Current
Normal
Standby
(Note 4)
Q
2400
1500
55
µA
µA
µA
V
RUN
V
RUN
= 0V, V
= 0V, V
> 2V
= Open
STBYMD
STBYMD
Shutdown Supply Current
70
V
Forced Continuous Threshold
Forced Continuous Pin Current
0.76
0.800
–0.18
5.3
0.84
–0.1
5.5
V
µA
V
FCB
I
V
= 0.85V
–0.30
FCB
FCB
V
Burst Inhibit (Constant Frequency)
Threshold
Measured at FCB Pin
BINHIBIT
UVLO
Undervoltage Reset
V
Falling
●
3.8
0.86
–380
0.7
4
V
V
IN
V
Feedback Overvoltage Lockout
Sense Pins Total Source Current
Start-Up Threshold
Measured at V
Pin
0.84
0.4
0.88
OVL
OSENSE
+
–
I
V
V
V
= V = 0V
SENSE
µA
V
SENSE
SENSE
V
V
Rising
STBYMD(START)
STBYMD(KA)
STBYMD
STBYMD
Keep-Alive Power-On Threshold
Rising, V
= 0V
1.25
99
V
RUN
DF MAX, BOOST Maximum Duty Factor
DF MAX, BUCK Maximum Duty Factor
% Switch C On
% Switch A On (in Dropout)
%
%
V
99
V
RUN Pin On Threshold
V
V
Rising
= 2V
1
1.5
2
RUN(ON)
RUN
RUN
I
Soft-Start Charge Current
Maximum Current Sense Threshold
0.5
1.2
µA
SS
V
Boost: V
Buck: V
= V – 50mV
●
●
160
–130
185
–150
mV
mV
SENSE(MAX)
OSENSE
OSENSE
REF
= V – 50mV
–95
REF
V
Minimum Current Sense Threshold
TG Rise Time
Discontinuous Mode
–6
50
45
45
55
80
mV
ns
ns
ns
ns
ns
SENSE(MIN,BUCK)
TG1, TG2 t
TG1, TG2 t
C
C
C
C
C
= 3300pF (Note 5)
= 3300pF (Note 5)
= 3300pF (Note 5)
= 3300pF (Note 5)
= 3300pF Each Driver
r
f
LOAD
LOAD
LOAD
LOAD
LOAD
TG Fall Time
BG1, BG2 t
BG1, BG2 t
BG Rise Time
r
f
BG Fall Time
TG1/BG1 t
BG1/TG1 t
TG2/BG2 t
BG2/TG2 t
Mode
TG1 Off to BG1 On Delay,
Switch C On Delay
1D
BG1 Off to TG1 On Delay,
Synchronous Switch D On Delay
C
C
C
C
C
= 3300pF Each Driver
= 3300pF Each Driver
= 3300pF Each Driver
= 3300pF Each Driver
= 3300pF Each Driver
80
80
80
90
90
ns
ns
ns
ns
ns
2D
3D
4D
LOAD
LOAD
LOAD
LOAD
LOAD
TG2 Off to BG2 On Delay,
Synchronous Switch B On Delay
BG2 Off to TG2 On Delay,
Switch A On Delay
BG1 Off to BG2 On Delay,
Switch A On Delay
Transition 1
Mode
BG2 Off to BG1 On Delay,
Transition 2
Synchronous Switch D On Delay
3780f
3
LTC3780
ELECTRICAL CHARACTERISTICS The ● indicates specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
t
Minimum On-Time for Main Switch in
Boost Operation
Switch C (Note 6)
200
240
ns
ON(MIN,BOOST)
ON(MIN,BUCK)
t
Minimum On-Time for Synchronous
Switch in Buck Operation
Switch B (Note 6)
180
220
ns
Internal V Regulator
CC
V
Internal V Voltage
7V < V < 30V, V = 5V
EXTVCC
●
●
5.7
5.4
6
6.3
2
V
%
INTVCC
CC
IN
∆V
Internal V Load Regulation
I
I
= 0mA to 20mA, V = 5V
EXTVCC
0.2
5.7
200
150
LDO(LOADREG)
EXTVCC
CC
CC
CC
V
EXTV Switchover Voltage
= 20mA, V
Rising
V
CC
EXTVCC
EXTVCC
∆V
∆V
EXTV Switchover Hysteresis
mV
mV
EXTVCC(HYS)
CC
EXTV Switch Drop Voltage
I
= 20mA, V
= 6V
300
EXTVCC
CC
CC
Oscillator and Phase-Locked Loop
f
f
f
Nominal Frequency
V
V
V
= 1.2V
PLLFLTR
260
170
340
300
200
400
50
330
220
440
kHz
kHz
kHz
kΩ
NOM
LOW
HIGH
Lowest Frequency
= 0V
PLLFLTR
PLLFLTR
Highest Frequency
= 2.4V
R
PLLIN Input Resistance
Phase Detector Output Current
PLLIN
I
f
f
< f
OSC
> f
OSC
–15
15
µA
µA
PLLLPF
PLLIN
PLLIN
PGOOD Output
∆V
∆V
∆V
PGOOD Upper Threshold
PGOOD Lower Threshold
PGOOD Hysteresis
V
V
V
Rising
OSENSE
5.5
7.5
–7.5
2.5
10
%
%
%
V
FBH
Falling
–5.5
–10
FBL
OSENSE
OSENSE
Returning
FB(HYST)
V
PGOOD Low Voltage
I
= 2mA
= 5V
0.1
0.3
PGL
PGOOD
I
PGOOD Leakage Current
V
±1
µA
PGOOD
PGOOD
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 5: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 2: T for the QFN package is calculated from the temperature T and
Note 6: The minimum on-time condition is specified for an inductor peak-
J
A
power dissipation P according to the following formula:
to-peak ripple current ≥ 40% of I
(see minimum on-time
D
MAX
considerations in the Applications Information section).
T = T + (P • 34°C/W)
J
A
D
Note 7: The LTC3780E is guaranteed to meet performance specifications
from 0°C to 85°C. Performance over the –40°C to 85°C operating
temperature range is assured by design, characterization and correlation
with statistical process controls. The LTC3780I is guaranteed and tested
over the – 40°C to 85°C operating temperature range.
Note 3: The IC is tested in a feedback loop that servos V to a specified
ITH
voltage and measures the resultant V
.
OSENSE
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
3780f
4
LTC3780
U W
TYPICAL PERFOR A CE CHARACTERISTICS TA = 25°C unless otherwise noted.
Efficiency vs Output Current
(Boost Operation)
Efficiency vs Output Current
Efficiency vs Output Current
(Buck Operation)
100
90
80
70
60
50
40
100
90
80
70
60
50
40
100
90
80
70
60
50
40
BURST
BURST
DCM
SC
DCM
CCM
CCM
DCM
CCM
V
V
= 12V
V
IN
V
OUT
= 18V
V
V
= 6V
IN
OUT
IN
OUT
= 12V
= 12V
= 12V
0.01
0.1
1
10
0.01
0.1
1
10
0.01
0.1
1
10
I
(A)
I
(A)
LOAD
I
(A)
LOAD
LOAD
3780 G02
3780 G03
3780 G01
Internal 6V LDO Line Regulation
EXTVCC Voltage Drop
Supply Current vs Input Voltage
2500
2000
1500
1000
500
120
100
6.5
6.0
5.5
5.0
V
FCB
= 0V
80
60
STANDBY
40
20
0
4.5
4.0
3.5
SHUTDOWN
0
1
10
20
30
40
50
20
INPUT VOLTAGE (V)
30
35
0
5
10
15
25
0
5
10
15
20
25
30
35
CURRENT (mA)
INPUT VOLTAGE (V)
3780 G06
3780 G05
3780 G04
INTVCC and EXTVCC Switch
Voltage vs Temperature
EXTVCC Switch Resistance
vs Temperature
Load Regulation
5
4
3
2
1
0
6.05
6.00
5.95
5.90
5.85
5.80
5.75
5.70
5.65
5.60
5.55
0
–0.1
–0.2
–0.3
–0.4
–0.5
V
IN
= 18V
INTV VOLTAGE
CC
V
= 12V
IN
V
= 6V
IN
EXTV SWITCHOVER THRESHOLD
CC
FCB = 0V
V
= 12V
OUT
–50 –25
0
25
50
75 100 125
–50
0
25
50
75 100 125
–25
0
1
2
3
4
5
TEMPERATURE (°C)
TEMPERATURE (°C)
LOAD CURRENT (A)
3780 G08
3780 G07
3780 G09
3780f
5
LTC3780
TYPICAL PERFOR A CE CHARACTERISTICS TA = 25°C unless otherwise noted.
U W
Discontinuous Current Mode
(DCM, VIN = 6V, VOUT = 12V)
Continuous Current Mode
(CCM, VIN = 12V, VOUT = 12V)
Continuous Current Mode
(CCM, VIN = 18V, VOUT = 12V)
SW2
10V/DIV
SW2
10V/DIV
SW2
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
V
OUT
V
V
OUT
OUT
100mV/DIV
100mV/DIV
100mV/DIV
I
I
I
L
L
L
2A/DIV
2A/DIV
2A/DIV
3780 G10
3780 G11
3780 G12
V
V
= 6V
5µs/DIV
V
V
= 12V
5µs/DIV
V
V
= 18V
5µs/DIV
IN
OUT
IN
OUT
IN
OUT
= 12V
= 12V
= 12V
Burst Mode Operation
(VIN = 6V, VOUT = 12V)
Burst Mode Operation
(VIN = 12V, VOUT = 12V)
Skip Cycle Mode
(VIN = 18V, VOUT = 12V)
SW2
10V/DIV
SW2
10V/DIV
SW2
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
V
OUT
V
V
OUT
OUT
500mV/DIV
200mV/DIV
100mV/DIV
I
I
L
L
2A/DIV
2A/DIV
I
L
1A/DIV
3780 G14
3780 G13
3780 G15
V
V
= 12V
10µs/DIV
V
V
= 6V
25µs/DIV
V
V
= 18V
2.5µs/DIV
IN
OUT
IN
OUT
IN
OUT
= 12V
= 12V
= 12V
Discontinuous Current Mode
(DCM, VIN = 6V, VOUT = 12V)
Discontinuous Current Mode
(DCM, VIN = 12V, VOUT = 12V)
Discontinuous Current Mode
(DCM, VIN = 18V, VOUT = 12V)
SW2
10V/DIV
SW2
10V/DIV
SW2
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
V
V
OUT
OUT
V
OUT
100mV/DIV
100mV/DIV
100mV/DIV
I
L
I
I
L
L
2A/DIV
1A/DIV
1A/DIV
3780 G16
3780 G17
3780 G18
V
V
= 6V
5µs/DIV
V
V
= 12V
5µs/DIV
V
V
= 18V
2.5µs/DIV
IN
OUT
IN
OUT
IN
OUT
= 12V
= 12V
= 12V
3780f
6
LTC3780
U W
TYPICAL PERFOR A CE CHARACTERISTICS TA = 25°C unless otherwise noted.
Oscillator Frequency
vs Temperature
Undervoltage Reset
vs Temperature
Minimum Current Sense
Threshold vs Duty Factor (Buck)
4.0
3.8
3.6
3.4
3.2
3.0
450
400
350
300
250
200
150
100
50
–20
–40
–60
–80
V
V
= 2.4V
= 1.2V
PLLFLTR
PLLFLTR
V
= 0V
PLLFLTR
0
–50 –25
0
25
50
75 100 125
–50 –25
0
25
125
50
75 100
100
80
60
40
20
0
TEMPERATURE (°C)
TEMPERATURE (°C)
DUTY FACTOR (%)
3780 G20
3780 G19
3780 G21
Maximum Current Sense
Threshold vs Duty Factor (Boost)
Maximum Current Sense
Threshold vs Duty Factor (Buck)
Minimum Current Sense
Threshold vs Temperature
140
130
120
110
200
150
180
160
140
120
100
BOOST
100
50
0
–50
–100
BUCK
50
–150
100 125
0
20
40
60
80
100
–50 –25
0
25
75
0
20
40
60
80
100
DUTY FACTOR (%)
TEMPERATURE (°C)
DUTY FACTOR (%)
3780 G23
3780 G24
3780 G22
Peak Current Threshold
vs VITH (Boost)
Valley Current Threshold
vs VITH (Buck)
200
150
100
50
100
50
0
–50
0
–100
–150
–50
–100
0
0.8
1.2
(V)
1.6
1.8
2.4
0
0.8
1.2
(V)
1.6
2.0
2.4
0.4
0.4
V
V
ITH
ITH
3780 G25
3780 G26
3780f
7
LTC3780
TYPICAL PERFOR A CE CHARACTERISTICS TA = 25°C unless otherwise noted.
U W
Load Step
Load Step
Load Step
V
V
V
OUT
OUT
OUT
500mV/DIV
500mV/DIV
500mV/DIV
I
L
I
I
L
5A/DIV
L
5A/DIV
5A/DIV
3780 G27
3780 G28
3780 G29
V
V
= 18V
200µs/DIV
V
V
= 12V
200µs/DIV
IN
OUT
IN
OUT
V
V
= 6V
200µs/DIV
IN
OUT
= 12V
= 12V
= 12V
LOAD STEP: 0A TO 5A
CONTINUOUS MODE
LOAD STEP: 0A TO 5A
CONTINUOUS MODE
LOAD STEP: 0A TO 5A
CONTINUOUS MODE
Line Transient
Line Transient
V
IN
V
IN
10V/DIV
10V/DIV
V
OUT
V
OUT
500mV/DIV
500mV/DIV
I
I
L
1A/DIV
L
1A/DIV
3780 G30
3780 G31
V
= 12V
= 1A
500µs/DIV
V
= 12V
= 1A
500µs/DIV
OUT
OUT
I
I
LOAD
LOAD
V
STEP: 7V TO 20V
V
IN
STEP: 20V TO 7V
IN
CONTINUOUS MODE
CONTINUOUS MODE
U
U
U
(SSOP/QFN)
PI FU CTIO S
PGOOD (Pin 1/Pin 30): Open-Drain Logic Output. PGOOD
is pulled to ground when the output voltage is not within
±7.5% of the regulation point.
voltage and built-in offsets between SENSE– and SENSE+
pins, in conjunction with RSENSE, set the current trip
threshold.
SS (Pin 2/Pin 31): Soft-start reduces the input power
sources’ surge currents by gradually increasing the
controller’s current limit. A minimum value of 6.8nF is
recommended on this pin.
SENSE+ (Pin 3/Pin 1): The (+) Input to the Current Sense
and Reverse Current Detect Comparators. The ITH pin
SENSE– (Pin 4/Pin 2): The (–) Input to the Current Sense
and Reverse Current Detect Comparators.
ITH(Pin5/Pin3):CurrentControlThresholdandErrorAm-
plifierCompensationPoint.Thecurrentcomparatorthresh-
old increases with this control voltage. The voltage ranges
from 0V to 2.4V.
3780f
8
LTC3780
U
U
U
(SSOP/QFN)
PI FU CTIO S
VOSENSE(Pin6/Pin4):ErrorAmplifierFeedbackInput.This
BOOST2, BOOST1 (Pins 13, 24/Pins 14, 27): Boosted
FloatingDriverSupply.The(+)terminalofthebootstrapca-
pacitor CA and CB (Figure 11) connects here. The BOOST2
pin swings from a diode voltage below INTVCC up to VIN +
INTVCC. TheBOOST1pinswingsfromadiodevoltagebelow
INTVCC up to VOUT + INTVCC.
pin connects the error amplifier input to an external resis-
tor divider from VOUT
.
SGND (Pin 7/Pin 5): Signal Ground. All small-signal com-
ponentsandcompensationcomponentsshouldconnectto
thisground,whichshouldbeconnectedtoPGNDatasingle
point.
TG2,TG1(Pins14,23/Pins15,26):TopGateDrive.Drives
the top N-channel MOSFET with a voltage swing equal to
INTVCC superimposed on the switch node voltage SW.
RUN(Pin8/Pin6):RunControlInput.ForcingtheRUNpin
below1.5VcausestheICtoshutdowntheswitchingregu-
latorcircuitry.Thereisa100kresistorbetweentheRUNpin
and SGND in the IC. Do not apply >6V to this pin.
SW2, SW1 (Pins 15, 22/Pins 17, 24): Switch Node. The
(–)terminalofthebootstrapcapacitorCAandCB(Figure 11)
connectshere. TheSW2pinswingsfromaSchottkydiode
(external) voltage drop below ground up to VIN. The SW1
pin swings from a Schottky diode (external) voltage drop
FCB (Pin 9/Pin 7): Forced Continuous Control Input. The
voltage applied to this pin sets the operating mode of the
controller. When the applied voltage is less than 0.8V, the
forced continuous current mode is active. When this pin
is allowed to float, the burst mode is active in boost
operation and the skip cycle mode is active in buck
operation. When the pin is tied to INTVCC, the constant
frequency discontinuous current mode is active in buck or
boost operation.
below ground up to VOUT
.
BG2, BG1 (Pins 16, 18/Pins 18, 20): Bottom Gate Drive.
Drives the gate of the bottom N-channel MOSFET between
ground and INTVCC.
PGND (Pin 17/Pin 19): Power Ground. Connect this pin
closelytothesourceofthebottomN-channelMOSFET,the
(–)terminalofCVCC andthe(–)terminalofCIN (Figure11).
PLLFLTR (Pin 10/Pin 8): The Phase-Locked Loop’s Low-
pass Filter is Tied to This Pin. Alternatively, this pin can be
drivenwithanACorDCvoltagesourcetovarythefrequency
of the internal oscillator.
INTVCC (Pin19/Pin21):Internal6VRegulatorOutput. The
driver and control circuits are powered from this voltage.
Decouple this pin to ground with a minimum of 4.7µF low
ESR tantalum or ceramic capacitor.
PLLIN (Pin 11/Pin 10): External Synchronization Input to
Phase Detector. This pin is internally terminated to SGND
with 50kΩ. The phase-locked loop will force the rising
bottomgatesignalofthecontrollertobesynchronizedwith
the rising edge of the PLLIN signal.
EXTVCC(Pin20/Pin22):ExternalVCC Input.WhenEXTVCC
exceeds5.7V,aninternalswitchconnectsthispintoINTVCC
andshutsdowntheinternalregulatorsothatthecontroller
andgatedrivepowerisdrawnfromEXTVCC.Donotexceed
7V at this pin and ensure that EXTVCC < VIN.
STBYMD (Pin 12/Pin 11): LDO Control Pin. Determines
whethertheinternalLDOremainsactivewhenthecontrol-
ler is shut down. See Operation section for details. If the
STBYMD pin is pulled to ground, the SS pin is internally
pulled to ground, preventing start-up and thereby provid-
ing a single control pin for turning off the controller.
Decouple this pin with 0.1µF if not tied to a DC potential.
VIN (Pin 21/Pin 23): Main Input Supply. Decouple this pin
to SGND with an RC filter (1Ω, 0.1µF).
Exposed Pad (Pin 33, QFN Only): This pin is SGND and
must be soldered to PCB ground.
3780f
9
LTC3780
W
BLOCK DIAGRA
INTV
CC
V
IN
BOOST2
TG2
STBYMD
FCB
FCB
I
LIM
SW2
+
–
BUCK
LOGIC
INTV
CC
BG2
R
SENSE
PGND
BG1
I
REV
+
–
FCB
INTV
CC
BOOST
LOGIC
SW1
TG1
1.2V
4(V
)
FB
I
CMP
+
–
BOOST1
0.86V
1.2µA
OV
EA
SS
–
+
INTV
CC
V
OUT
RUN
SLOPE
V
OSENSE
100k
–
+
V
FB
0.80V
I
TH
SHDN
RST
FB
RUN/
SS
4(V
)
+
SENSE
–
SENSE
PLLIN
50k
V
REF
F
IN
PHASE DET
V
IN
V
IN
+
–
5.7V
PLLFLTR
CLK
R
LP
OSCILLATOR
6V
LDO
REG
C
LP
EXTV
INTV
CC
–
+
0.86V
6V
+
CC
PGOOD
INTERNAL
SUPPLY
SGND
V
OSENSE
–
+
0.74V
3780 BD
3780f
10
LTC3780
U
OPERATIO
MAIN CONTROL LOOP
SS voltage while CSS is slowly charged during start-up.
This “soft-start” clamping prevents abrupt current from
being drawn from the input power supply.
TheLTC3780isacurrentmodecontrollerthatprovidesan
output voltage above, equal to or below the input voltage.
The LTC proprietary topology and control architecture
employsacurrent-sensingresistorinBuckorBoostmodes.
Thesensedinductorcurrentiscontrolledbythevoltageon
the ITH pin, which is the output of the amplifier EA. The
POWER SWITCH CONTROL
Figure1showsasimplifieddiagramofhowthefourpower
switches are connected to the inductor, VIN, VOUT and
GND. Figure 2 shows the regions of operation for the
LTC3780asafunctionofdutycycleD.Thepowerswitches
are properly controlled so the transfer between modes is
continuous. When VIN approaches VOUT, the Buck-Boost
region is reached; the mode-to-mode transition time is
typically 200ns.
V
OSENSE pin receives the voltage feedback signal, which is
compared to the internal reference voltage by the EA.
ThetopMOSFETdriversarebiasedfromfloatingbooststrap
capacitors CA and CB (Figure 11), which are normally
rechargedthroughanexternaldiodewhenthetopMOSFET
is turned off. Schottky diodes across the synchronous
switch D and synchronous switch B are not required, but
provide a lower drop during the dead time. The addition of
the Schottky diodes will typically improve peak efficiency
by 1% to 2% at 400kHz.
Buck Region (VIN > VOUT
)
Switch D is always on and Switch C is always off during
thismode. Atthestartofeverycycle, SynchronousSwitch
B is turned on first. Inductor current is sensed when
Synchronous Switch B is turned on. After the sensed
inductorcurrentfallsbelowthereferencevoltage,whichis
proportional to VITH, Synchronous Switch B is turned off
The main control loop is shut down by pulling the RUN pin
low. When the RUN pin voltage is higher than 1.5V, an
internal 1.2µA current source charges soft-start capacitor
CSS at the SS pin. The ITH voltage is then clamped to the
V
IN
V
OUT
TG2
BG2
A
D
TG1
BG1
L
SW2
SW1
B
C
R
SENSE
3780 F01
Figure 1. Simplified Diagram of the Output Switches
98%
D
MAX
BOOST
A ON, B OFF
BOOST REGION
BUCK/BOOST REGION
BUCK REGION
PWM C, D SWITCHES
D
MIN
BOOST
FOUR SWITCH PWM
D
MAX
BUCK
D ON, C OFF
PWM A, B SWITCHES
3%
D
MIN
BUCK
3780 F02
Figure 2. Operating Mode vs Duty Cycle
3780f
11
LTC3780
U
OPERATIO
and Switch A is turned on for the remainder of the cycle.
Switches A and B will alternate, behaving like a typical
synchronous buck regulator. The duty cycle of switch A
increasesuntilthemaximumdutycycleoftheconverterin
Buck mode reaches DMAX_BUCK, given by:
CLOCK
SWITCH A
SWITCH B
SWITCH C
SWITCH D
DMAX_BUCK = (1 – DBUCK-BOOST) • 100%
where DBUCK-BOOST = duty cycle of the Buck-Boost switch
range:
I
L
3780 F04a
(4a) Buck-Boost Mode (VIN ≥ VOUT
)
DBUCK-BOOST = (200ns • f) • 100%
and f is the operating frequency in Hz.
CLOCK
Figure 3 shows typical Buck mode waveforms. If VIN
approaches VOUT, the Buck-Boost region is reached.
SWITCH A
SWITCH B
CLOCK
SWITCH C
SWITCH D
SWITCH A
SWITCH B
I
L
0V
3780 F04b
SWITCH C
2.4V
(4b) Buck-Boost Mode (VIN ≤ VOUT
)
SWITCH D
I
Figure 4. Buck-Boost Mode
3780 F03
Figure 3. Buck Mode (VIN > VOUT
)
Buck-Boost (VIN ≅ VOUT
)
Boost Region (VIN < VOUT
)
When VIN is close to VOUT, the controller is in Buck-Boost
mode. Figure 4 shows typical waveforms in this mode.
Every cycle, if the controller starts with Switches B and D
turned on, Switches A and C are then turned on. Finally,
Switches A and D are turned on for the remainder of the
time. If the controller starts with Switches A and C turned
on, Switches B and D are then turned on. Finally, Switches
A and D are turned on for the remainder of the time.
SwitchAisalwaysonandSynchronousSwitchBisalways
off in Boost mode. Every cycle, Switch C is turned on first.
Inductor current is sensed when Synchronous Switch C is
turned on. After the sensed inductor current exceeds the
reference voltage which is proportional to VITH, Switch C
is turned off and Synchronous Switch D is turned on for
the remainder of the cycle. Switches C and D will alternate,
behaving like a typical synchronous boost regulator.
3780f
12
LTC3780
U
OPERATIO
The duty cycle of Switch C decreases until the minimum
duty cycle of the converter in Buck mode reaches
DMIN_BOOST, given by:
When the FCB pin voltage is lower than 0.8V, the control-
ler behaves as a continuous, PWM current mode syn-
chronousswitchingregulator. InBoostmode, SwitchAis
always on. Switch C and Synchronous Switch D are
alternately turned on to maintain the output voltage
independent of direction of inductor current. Every ten
cycles, Switch A is forced off for about 300ns to allow CA
to recharge. In Buck mode, Synchronous Switch D is
always on. Switch A and Synchronous Switch B are
alternately turned on to maintain the output voltage inde-
pendent of direction of inductor current. Every ten cycles,
Synchronous Switch D is forced off for about 300ns to
allow CB to recharge. This is the least efficient operating
mode at light load, but may be desirable in certain appli-
cations. In this mode, the output can source or sink
current. The sunk current will be forced back into the main
power supply potentially boosting the input supply to
dangerous voltage levels—BEWARE!
DMIN_BOOST = (DBUCK-BOOST) • 100%
where DBUCK-BOOST is the duty cycle of the Buck-Boost
switch range:
DBUCK-BOOST = (200ns • f) • 100%
and f is the operating frequency in Hz.
Figure 5 shows typical boost mode waveforms. If VIN
approaches VOUT, the Buck-Boost region is reached.
CLOCK
2.4V
SWITCH A
0V
SWITCH B
SWITCH C
SWITCH D
When the FCB pin voltage is below VINTVCC – 1V, but
greater than 0.8V, the controller enters Burst Mode opera-
tion in Boost operation or enters Skip-Cycle mode in Buck
operation. During Boost operation, Burst Mode operation
sets a minimum output current level before inhibiting the
switch C and turns off Synchronous Switch D when the
inductor current goes negative. This combination of re-
quirements will, at low currents, force the ITH pin below a
voltage threshold that will temporarily inhibit turn-on of
power switches C and D until the output voltage drops.
There is 100mV of hysteresis in the burst comparator tied
to the ITH pin. This hysteresis produces output signals to
the MOSFETs C and D that turn them on for several cycles,
followed by a variable “sleep” interval depending upon the
loadcurrent.Themaximumoutputvoltagerippleislimited
to 3% of the nominal DC output voltage as determined by
a resistive feedback divider. During buck operation, Skip-
Cycle mode sets a minimum positive inductor current
level. When inductor current is lower than this level,
Synchronous Switch B is kept off. In every cycle, the body
I
3780 F05
Figure 5. Boost Mode (VIN < VOUT
)
LOW CURRENT OPERATION
TheFCBpinisamultifunctionpinprovidingtwofunctions:
1) to provide regulation for a secondary winding by
temporarily forcing continuous PWM operation in Buck
mode and 2) to select among three modes for both buck
and boost operations by accepting a logic input. Figure 6
shows the different modes.
FCB PIN
0V to 0.75V
0.85V to 5V
>5.3V
BUCK MODE
BOOST MODE
Force Continuous Mode
Skip-Cycle Mode
Force Continuous Mode
Burst Mode Operation
DCM with Constant Freq
DCM with Constant Freq
Figure 6. Different Operating Modes
3780f
13
LTC3780
U
OPERATIO
diode of Synchronous Switch B or the Schottky diode,
which is in parallel in with Synchronous Switch B, is used
todischargeinductorcurrent. Asaresult, somecycleswill
be skipped when the output load current drops below 1%
of the maximum designed load in order to maintain the
output voltage.
INTVCC/EXTVCC POWER
Power for all power MOSFET drivers and most internal
circuitry is derived from the INTVCC pin. When the EXTVCC
pin is left open, an internal 6V low dropout linear regulator
supplies INTVCC power. If EXTVCC is taken above 5.7V, the
6V regulator is turned off and an internal switch is turned
on, connecting EXTVCC to INTVCC. This allows the INTVCC
powertobederivedfromahighefficiencyexternalsource.
When the FCB pin voltage is tied to the INTVCC pin, the
controller enters constant frequency Discontinuous Cur-
rent mode (DCM). For Boost operation, Synchronous
Switch D is held off whenever the ITH pin is below a
threshold voltage. In every cycle, Switch C is used to
charge inductor current. After the output voltage is high
enough, the controller will enter continuous current Buck
mode for one cycle to discharge inductor current. In the
following cycle, the controller will resume DCM Boost
operation. For Buck operation, constant frequency Dis-
continuousCurrentmodesetsaminimumnegativeinduc-
tor current level. Synchronous Switch B is turned off
whenever inductor current is lower than this level. At very
light loads, this constant frequency operation is not as
efficient as Burst Mode operation or Skip-Cycle, but does
provide lower noise, constant frequency operation.
POWER GOOD (PGOOD) PIN
ThePGOODpinisconnectedtoanopendrainofaninternal
MOSFET.TheMOSFETturnsonandpullsthepinlowwhen
the output is not within ±7.5% of the nominal output level
as determined by the resistive feedback divider. When the
output meets the ±7.5% requirement, the MOSFET is
turned off and the pin is allowed to be pulled up by an
external resistor to a source of up to 7V.
FOLDBACK CURRENT
Foldback current limiting is activated when the output
voltagefallsbelow70%ofitsnominallevel,reducingpower
waste.Duringstart-up,foldbackcurrentlimitingisdisabled.
FREQUENCY SYNCHRONIZATION AND
FREQUENCY SETUP
INPUT UNDERVOLTAGE RESET
The phase-locked loop allows the internal oscillator to be
synchronized to an external source via the PLLIN pin. The
phase detector output at the PLLFLTR pin is also the DC
frequency control input of the oscillator. The frequency
ranges from 200kHz to 400kHz, corresponding to a DC
voltage input from 0V to 2.4V at PLLFLTR. When locked,
the PLL aligns the turn on of the top MOSFET to the rising
edgeofthesynchronizingsignal. WhenPLLINisleftopen,
the PLLFLTR pin goes low, forcing the oscillator to its
minimum frequency.
TheSScapacitorwillberesetiftheinputvoltageisallowed
to fall below approximately 4V. The SS capacitor will
attempt to charge through a normal soft-start ramp after
the input voltage rises above 4V.
3780f
14
LTC3780
U
OPERATIO
OUTPUT OVERVOLTAGE PROTECTION
In every Boost mode cycle, current is limited by a voltage
reference, which is proportional to the ITH pin voltage. The
maximum sensed current is limited to 160mV. In every
Buck mode cycle, the maximum sensed current is limited
to 130mV.
Anovervoltagecomparatorguardsagainsttransientover-
shoots (>7.5%) as well as other more serious conditions
that may overvoltage the output. In this case, Synchro-
nous Switch B and Synchronous Switch D are turned on
until the overvoltage condition is cleared or the maximum
negative current limit is reached. When inductor current is
lower than the maximum negative current limit, Synchro-
nous Switch B and Synchronous Switch D are turned off,
and Switch A and Switch C are turned on until the inductor
current reaches another negative current limit. If the
comparator still detects an overvoltage condition, Switch
A and Switch C are turned off, and Synchronous Switch B
and Synchronous Switch D are turned on again.
STANDBY MODE PIN
The STBYMD pin is a three-state input that controls
circuitry within the IC as follows: When the STBYMD pin
is held at ground, the SS pin is pulled to ground. When the
pin is left open, the internal SS current source charges the
SS capacitor, allowing turn-on of the controller and acti-
vating necessary internal biasing. When the STBYMD pin
is taken above 2V, the internal linear regulator is turned on
independent of the state on the RUN and SS pins, provid-
ing an output power source for “wake-up” circuitry.
Decouple the pin with a small capacitor (0.1µF) to ground
if the pin is not connected to a DC potential.
SHORT-CIRCUIT PROTECTION AND CURRENT LIMIT
Switch A on-time is limited by output voltage. When
output voltage is reduced and is lower than its nominal
level, Switch A on-time will be reduced.
3780f
15
LTC3780
W U U
U
APPLICATIO S I FOR ATIO
Figure 11 is a basic LTC3780 application circuit. External
component selection is driven by the load requirement,
and begins with the selection of RSENSE and the inductor
value. Next, the power MOSFETs are selected. Finally, CIN
and COUT are selected. This circuit can be configured for
operation up to an input voltage of 36V.
Allowing a margin for variations in LTC3780 and external
component values yields:
2 •160mV • V
IN
RSENSE
=
2 •IOUT(MAX,BOOST) • VOUT + ∆IL(BOOST)
Selection of Operation Frequency
RSENSE Selection and Maximum Output Current
The LTC3780 uses a constant frequency architecture and
hasaninternalvoltagecontrolledoscillator. Theswitching
frequency is determined by the internal oscillator capaci-
tor. This internal capacitor is charged by a fixed current
plus an additional current that is proportional to the
voltage applied to the PLLFLTR pin. The frequency of this
oscillator can be varied over a 2-to-1 range. The PLLFLTR
pin can be grounded to lower the frequency to 200kHz or
tiedto2.4Vtoyieldapproximately400kHz. WhenPLLINis
left open, the PLLFLTR pin goes low, forcing the oscillator
to minimum frequency.
R
SENSE is chosen based on the required output current.
The current comparator threshold sets the peak of the
inductor current in Boost mode and the maximum induc-
tor valley current in Buck mode. In Boost mode, the
maximum average load current is:
160mV • V
∆IL
IN
IOUT(MAX,BOOST)
=
–
RSENSE • VOUT
2
where ∆IL is peak-to-peak inductor ripple current. In Buck
mode, the maximum average load current is:
A graph for the voltage applied to the PLLFLTR pin vs
frequency is given in Figure 8. As the operating frequency
isincreasedthegatechargelosseswillbehigher,reducing
efficiency. The maximum switching frequency is approxi-
mately 400kHz.
130mV ∆IL
IOUT(MAX,BUCK)
=
+
RSENSE
2
Figure 7 shows how the load current (IMAXLOAD • RSENSE
varies with input and output voltage
)
450
400
350
300
250
200
150
100
50
160
150
140
130
120
110
100
0
0.1
1
10
0
2
2.5
0.5
1
1.5
V
/V
IN OUT
(V)
PLLFLTR PIN VOLTAGE (V)
3780 F07
3780 F08
Figure 8. Frequency vs PLLFLTR Pin Voltage
Figure 7. Load Current vs VIN/VOUT
3780f
16
LTC3780
W U U
APPLICATIO S I FOR ATIO
U
Inductor Selection
This formula has a maximum at VIN = 2VOUT, where
RMS = IOUT(MAX)/2. This simple worst-case condition is
commonly used for design because even significant
deviations do not offer much relief. Note that ripple
current ratings from capacitor manufacturers are often
based on only 2000 hours of life which makes it advisable
to derate the capacitor.
I
The operating frequency and inductor selection are inter-
related in that higher operating frequencies allow the use
of smaller inductor and capacitor values. The inductor
value has a direct effect on ripple current. The inductor
current ripple ∆IL is typically set to 20% to 40% of the
maximum inductor current. For a given ripple the induc-
tance terms are as follows:
In Boost mode, the discontinuous current shifts from the
input to the output, so COUT must be capable of reducing
the output voltage ripple. The effects of ESR (equivalent
series resistance) and the bulk capacitance must be con-
sidered when choosing the right capacitor for a given
output ripple voltage. The steady ripple due to charging
and discharging the bulk capacitance is given by:
V
2 • VOUT – V
•100
IN(MIN)
(
)
IN(MIN)
LBOOST
>
H,
2
ƒ •IOUT(MAX) •%Ripple • VOUT
VOUT • VIN(MAX) – VOUT •100
(
)
LBUCK
>
H
ƒ •IOUT(MAX) •%Ripple • V
IN(MAX)
IOUT(MAX) • VOUT – V
(
)
)
IN(MIN)
where:
f is operating frequency, Hz
Ripple(Boost,Cap) =
Ripple(Buck,Cap) =
V
V
COUT • VOUT • f
IOUT(MAX) • VIN(MAX) – VOUT
% Ripple is allowable inductor current ripple, %
VIN(MIN) is minimum input voltage, V
VIN(MAX) is maximum input voltage, V
VOUT is output voltage, V
(
COUT • VIN(MAX) • f
where COUT is the output filter capacitor.
The steady ripple due to the voltage drop across the ESR
is given by:
IOUT(MAX) is maximum output load current
For high efficiency, choose an inductor with low core loss,
such as ferrite and molypermalloy (from Magnetics, Inc.).
Also,theinductorshouldhavelowDCresistancetoreduce
the I2R losses, and must be able to handle the peak
inductor current without saturating. To minimize radiated
noise, use a toroid, pot core or shielded bobbin inductor.
∆VBOOST,ESR = IL(MAX,BOOST) • ESR
∆VBUCK,ESR = IL(MAX,BUCK) • ESR
Multiple capacitors placed in parallel may be needed to
meet the ESR and RMS current handling requirements.
Dry tantalum, special polymer, aluminum electrolytic and
ceramic capacitors are all available in surface mount
packages. Ceramic capacitors have excellent low ESR
characteristics but can have a high voltage coefficient.
Capacitors are now available with low ESR and high ripple
current ratings such as OS-CON and POSCAP.
CIN and COUT Selection
InBoostmode, inputcurrentiscontinuous. InBuckmode,
input current is discontinuous. In Buck mode, the selec-
tion of input capacitor CIN is driven by the need to filter the
input square wave current. Use a low ESR capacitor sized
to handle the maximum RMS current. For Buck operation,
the input RMS current is given by:
Power MOSFET Selection and
Efficiency Considerations
The LTC3780 requires four external N-channel power
MOSFETs, two for the top switches (Switch A and D,
shown in Figure 1) and two for the bottom switches
VOUT
V
IN
V
IN
VOUT
IRMS ≈IOUT(MAX)
•
•
– 1
3780f
17
LTC3780
W U U
U
APPLICATIO S I FOR ATIO
(Switch B and C shown in Figure 1). Important parameters
Switch C operates in Boost mode as the control switch. Its
power dissipation at maximum current is given by:
forthepowerMOSFETsarethebreakdownvoltageVBR,DSS
,
threshold voltage VGS,TH, on-resistance RDS(ON), reverse
transfercapacitanceCRSS andmaximumcurrentIDS(MAX)
.
V
– V V
IN OUT
(
)
OUT
PC,BOOST
=
•IOUT(MAX)2 • ρT •RDS(ON)
2
The drive voltage is set by the 6V INTVCC supply. Conse-
quently, logic-level threshold MOSFETs must be used in
LTC3780 applications. If the input voltage is expected to
drop below 5V, then the sub-logic threshold MOSFETs
should be considered.
V
IN
IOUT(MAX)
3
+ k • VOUT
•
•CRSS • f
V
IN
where CRSS is usually specified by the MOSFET manufac-
turers. Theconstantk, whichaccountsforthe losscaused
by reverse recovery current, is inversely proportional to
the gate drive current and has an empirical value of 1.7.
In order to select the power MOSFETs, the power dissi-
pated by the device must be known. For Switch A, the
maximum power dissipation happens in Boost mode,
when it remains on all the time. Its maximum power
dissipation at maximum output current is given by:
For Switch D, the maximum power dissipation happens in
Boost mode, when its duty cycle is higher than 50%. Its
maximum power dissipation at maximum output current
is given by:
2
⎛ VOUT
⎞
PA,BOOST
=
•IOUT(MAX) • ρT •RDS(ON)
⎟
⎜
⎝ V
⎠
IN
2
V
IN
⎛ VOUT
⎞
PD,BUCK
=
•
•IOUT(MAX) • ρT •RDS(ON)
⎟
⎜
whereρT isanormalizationfactor(unityat25°C)account-
ing for the significant variation in on-resistance with
temperature, typically about 0.4%/°C as shown in Fig-
ure 9. For a maximum junction temperature of 125°C,
using a value ρT = 1.5 is reasonable.
VOUT ⎝ V
⎠
IN
For the same output voltage and current, Switch A has the
highest power dissipation and Switch B has the lowest
power dissipation unless a short occurs at the output.
From a known power dissipated in the power MOSFET, its
junction temperature can be obtained using the following
formula:
Switch B operates in Buck mode as the synchronous
rectifier. Its power dissipation at maximum output current
is given by:
TJ = TA + P • RTH(JA)
V – VOUT
P
=
•IOUT(MAX)2 • ρT •RDS(ON)
IN
B,BUCK
V
IN
2.0
1.5
1.0
0.5
0
50
100
–50
150
0
JUNCTION TEMPERATURE (°C)
3780 F09
Figure 9. Normalized RDS(ON) vs Temperature
3780f
18
LTC3780
W U U
APPLICATIO S I FOR ATIO
U
The RTH(JA) to be used in the equation normally includes
the RTH(JC) for the device plus the thermal resistance from
the case to the ambient temperature (RTH(JC)). This value
of TJ can then be compared to the original, assumed value
used in the iterative calculation process.
to approximately 20% of the maximum inductor current.
The output voltage ripple can increase during Burst Mode
operation.
INTVCC Regulator
An internal P-channel low dropout regulator produces 6V
at the INTVCC pin from the VIN supply pin. INTVCC powers
the drivers and internal circuitry within the LTC3780. The
INTVCC pin regulator can supply a peak current of 40mA
andmustbebypassedtogroundwithaminimumof4.7µF
tantalum, 10µF special polymer or low ESR type electro-
lytic capacitor. A 1µF ceramic capacitor placed directly
adjacent to the INTVCC and PGND IC pins is highly
recommended. Good bypassing is necessary to supply
the high transient current required by MOSFET gate
drivers.
Schottky Diode (D1, D2) Selection
and Light Load Operation
The Schottky diodes D1 and D2 shown in Figure 1 conduct
during the dead time between the conduction of the power
MOSFET switches. They are intended to prevent the body
diode of Synchronous Switches B and D from turning on
and storing charge during the dead time. In particular, D2
significantly reduces reverse recovery current between
Switch D turn-off and Switch C turn-on, which improves
converter efficiency and reduces Switch C voltage stress.
In order for the diode to be effective, the inductance
between it and the synchronous switch must be as small
as possible, mandating that these components be placed
adjacently.
HigherinputvoltageapplicationsinwhichlargeMOSFETs
are being driven at high frequencies may cause the
maximumjunctiontemperatureratingfortheLTC3780to
be exceeded. The system supply current is normally
dominated by the gate charge current. Additional external
loading of the INTVCC also needs to be taken into account
for the power dissipation calculations. The total INTVCC
current can be supplied by either the 6V internal linear
regulator or by the EXTVCC input pin. When the voltage
applied to the EXTVCC pin is less than 5.7V, all of the
INTVCC current is supplied by the internal 6V linear
regulator. Power dissipation for the IC in this case is
VIN • IINTVCC, and overall efficiency is lowered. The junc-
tion temperature can be estimated by using the equations
given in Note 2 of the Electrical Characteristics. For
example, LTC3780 VIN current is limited to less than
24mA from a 24V supply when not using the EXTVCC pin
as:
In Buck mode, when the FCB pin voltage is 0.85 < VFCB
<
5V, the converter operates in Skip-Cycle mode. In this
mode, Synchronous Switch B remains off until the induc-
tor peak current exceeds one-fifth of its maximum peak
current. As a result, D1 should be rated for about one-half
to one-third of the full load current.
In Boost mode, when the FCB pin voltage is higher than
5.3V, the converter operates in Discontinuous Current
mode. In this mode, Synchronous Switch D remains off
until the inductor peak current exceeds one-fifth of its
maximum peak current. As a result, D2 should be rated for
about one-third to one-fourth of the full load current.
In Buck mode, when the FCB pin voltage is higher than
5.3V, the converter operates in constant frequency Dis-
continuous Current mode. In this mode, Synchronous
Switch B remains on until the inductor valley current is
lower than the sense voltage representing the minimum
negative inductor current level (VSENSE = –5mV). Both
Switch A and B are off until next clock signal.
TJ = 70°C + 24mV • 24V • 95°C/W = 125°C
UseoftheEXTVCC inputpinreducesthejunctiontempera-
ture to:
TJ = 70°C + 24mV • 6V • 95°C/W = 84°C
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked
operating in continuous mode at maximum VIN.
In Boost mode, when the FCB pin voltage is 0.85 < VFCB
5.3V, the converter operates in Burst Mode operation. In
this mode, the controller clamps the peak inductor current
<
3780f
19
LTC3780
W U U
U
APPLICATIO S I FOR ATIO
EXTVCC Connection
Topside MOSFET Driver Supply (CA, DA, CB, DB)
The LTC3780 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
When the voltage applied to EXTVCC rises above 5.7V, the
internal regulator is turned off and a switch connects the
EXTVCC pin to the INTVCC pin thereby supplying internal
power. The switch remains closed as long as the voltage
applied to EXTVCC remains above 5.5V. This allows the
MOSFET driver and control power to be derived from the
output when (5.7V < VOUT < 7V) and from the internal
regulator when the output is out of regulation (start-up,
short-circuit). If more current is required through the
EXTVCC switch than is specified, an external Schottky
diode can be interposed between the EXTVCC and INTVCC
pins. Ensure that EXTVCC ≤ VIN.
Referring to Figure 11, the external bootstrap capacitors
CA and CB connected to the BOOST1 and BOOST2 pins
supply the gate drive voltage for the topside MOSFET
Switches A and D. When the top MOSFET Switch A turns
on, the switch node SW2 rises to VIN and the BOOST2 pin
rises to approximately VIN + INTVcc. When the bottom
MOSFET Switch B turns on, the switch node SW2 drops to
lowandtheboostcapacitorCB ischargedthroughDB from
INTVCC. When the top MOSFET Switch D turns on, the
switch node SW1 rises to VOUT and the BOOST1 pin rises
to approximately VOUT + INTVCC. When the bottom MOS-
FET Switch C turns on, the switch node SW1 drops to low
and the boost capacitor CA is charged through DA from
INTVCC. The boost capacitors CA and CB need to store
about 100 times the gate charge required by the top
MOSFET Switch A and D. In most applications a 0.1µF to
0.47µF, X5R or X7R dielectric capacitor is adequate.
The following list summarizes the three possible connec-
tions for EXTVCC:
1. EXTVCC left open (or grounded). This will cause INTVCC
to be powered from the internal 6V regulator at the cost
of a small efficiency penalty.
Run Function
The RUN pin provides simple ON/OFF control for the
LTC3780. Driving the RUN pin above 1.5V permits the
controller to start operating. Pulling RUN below 1.5V puts
the LTC3780 into low current shutdown. Do not apply
more than 6V to the RUN pin.
2. EXTVCC connected directly to VOUT (5.7V < VOUT < 7V).
This is the normal connection for a 6V regulator and
provides the highest efficiency.
3. EXTVCC connected to an external supply. If an external
supply is available in the 5.5V to 7V range, it may be
used to power EXTVCC provided it is compatible with
the MOSFET gate drive requirements.
Soft-Start Function
Soft-start reduces the input power sources’ surge cur-
rents by gradually increasing the controller’s current limit
(proportionaltoaninternallybufferedandclampedequiva-
lent of VITH).
Output Voltage
The LTC3780 output voltage is set by an external feedback
resistive divider carefully placed across the output capaci-
tor. The resultant feedback signal is compared with the
internal precision 0.800V voltage reference by the error
amplifier. The output voltage is given by the equation:
An internal 1.2µA current source charges up the CSS
capacitor. As the voltage on SS increases from 0V to 2.4V,
the internal current limit rises from 0V/RSENSE to
150mV/RSENSE. The output current limit ramps up slowly,
taking 1.5s/µF to reach full current. The output current
thus ramps up slowly, eliminating the starting surge
current required from the input power supply.
R2
R1
⎛
⎝
⎞
⎟
⎠
VOUT = 0.8V • 1+
⎜
2.4V
T
IRMP
=
•CSS = 1.5s/µF •C
SS
(
)
1.2µA
Do not apply more than 6V to the SS pin.
3780f
20
LTC3780
W U U
APPLICATIO S I FOR ATIO
U
The Standby Mode (STBYMD) Pin Function
The secondary output voltage VSEC is normally set as
shown in Figure 10 by turns ratio N of the transformer:
TheStandbymode(STBYMD)pinprovidesseveralchoices
for start-up and standby operational modes. If the pin is
pulled to ground, the SS pin is internally pulled to ground,
preventing start-up and thereby providing a single control
pin for turning off the controller. If the pin is left open or
decoupled with a capacitor to ground, the SS pin is
internally provided with a starting current, permitting
external control for turning on the controller. If the pin is
connected to a voltage greater than 1.25V, the internal
regulator (INTVCC) will be on even when the controller is
shut down (RUN pin voltage < 1.5V). In this mode, the
onboard 6V linear regulator can provide power to keep-
alive functions such as a keyboard controller.
V
SEC ≈ (N + 1) • VOUT
However, if the controller goes into Burst Mode operation
and halts switching due to a light primary load current,
thenVSEC willdrop. AnexternalresistivedividerfromVSEC
to the FCB pin sets a minimum voltage VSEC(MIN)
:
R6
R5
⎛
⎞
⎟
⎠
VSEC(MIN) ≈ 0.8 • 1+
⎜
⎝
If the VSEC drops below this level, the FCB voltage forces
temporary continuous switching operation until VSEC is
again above its minimum.
In order to prevent erratic operation if no external connec-
tions are made to FCB pin, the FCB pin has a 0.18µA
internal current source pulling the pin high. Include this
current when choosing resistor values R5 and R6.
FCB Pin Regulates Secondary Winding in Buck Mode
In Buck mode, the FCB pin can be used to regulate a
secondary winding or as a logic level input. Continuous
operation is forced when the FCB pin drops below 0.8V.
During continuous mode, current flows continuously in
the transformer primary. The secondary winding(s) draw
current only when Switch B and Switch D are on in Buck
mode. When primary load currents are low and/or the
VIN/VOUT ratio is low, the Synchronous Switch B may not
be on for a sufficient amount of time to transfer power
from the output capacitor to the secondary load. Forced
continuous operation will support secondary windings if
there is sufficient synchronous switch duty factor. Thus,
the FCB input pin removes the requirement that power
must be drawn from the auxiliary windings. With the loop
in continuous mode, the auxiliary outputs may nominally
be loaded without regard to the primary output load.
Fault Conditions: Current Limit and Current Foldback
The maximum inductor current is inherently limited in a
currentmodecontrollerbythemaximumsensevoltage.In
Boost mode, maximum sense voltage and the sense
resistance determines the maximum allowed inductor
peak current, which is:
160mV
RSENSE
IL(MAX,BOOST)
=
V
SEC
V
OUT
V
IN
C
OUT
A
SW2
D
SW1
TG1
BG1
TG2
BG2
R6
R5
•
LTC3780
FCB
•
T1
1:N
B
C
SGND
R
SENSE
3780 F10
Figure 10. Secondary Output Loop
3780f
21
LTC3780
W U U
U
APPLICATIO S I FOR ATIO
In Buck mode, maximum sense voltage and the sense
resistance determines the maximum allowed inductor
valley current, which is:
2. Transition loss. This loss arises from the brief amount
of time Switch A or Switch C spends in the saturated
region during switch node transitions. It depends upon
the input voltage, load current, driver strength and
MOSFET capacitance, among other factors. The loss is
significant at input voltages above 20V and can be
estimated from:
130mV
RSENSE
IL(MAX,BUCK)
=
To further limit current in the event of a short circuit to
ground, the LTC3780 includes foldback current limiting. If
the output falls by more than 30%, then the maximum
sense voltage is progressively lowered to about one third
of its full value.
Transition Loss ≈ 1.7A–1 • VIN2 • IOUT • CRSS • f
where CRSS is the reverse transfer capacitance.
3. INTVCC current. This is the sum of the MOSFET driver
and control currents. This loss can be reduced by
supplying INTVCC current through the EXTVCC pin from
a high efficiency source, such as an output derived
boost network or alternate supply if available.
Fault Conditions: Overvoltage Protection
A comparator monitors the output for overvoltage condi-
tions. The comparator (OV) detects overvoltage faults
greater than 7.5% above the nominal output voltage.
Whentheconditionissensed,SwitchesAandCareturned
off, and Switches B and D are turned on until the overvolt-
age condition is cleared. During an overvoltage condition,
a negative current limit (VSENSE = –60mV) is set to limit
negative inductor current. When the sensed current in-
ductor current is lower than –60mV, Switch A and C are
turned on, and Switch B and D are turned off until the
sensed current is higher than –20mV. If the output is still
in overvoltage condition, Switch A and C are turned off,
and Switch B and D are turned on again.
4. CIN and COUT loss. The input capacitor has the difficult
job of filtering the large RMS input current to the regu-
lator in Buck mode. The output capacitor has the more
difficult job of filtering the large RMS output current in
Boostmode.BothCIN andCOUTarerequiredtohavelow
ESR to minimize the AC I2R loss and sufficient capaci-
tance to prevent the RMS current from causing addi-
tional upstream losses in fuses or batteries.
5. Otherlosses.SchottkydiodeD1andD2areresponsible
for conduction losses during dead time and light load
conduction periods. Inductor core loss occurs pre-
dominately at light loads. Switch C causes reverse
recovery current loss in Boost mode.
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Although all dissipative
elements in circuit produce losses, four main sources
account for most of the losses in LTC3780 circuits:
1. DC I2R losses. These arise from the resistances of the
MOSFETs, sensing resistor, inductor and PC board
traces and cause the efficiency to drop at high output
currents.
Whenmakingadjustmentstoimproveefficiency,theinput
current is the best indicator of changes in efficiency. If you
make a change and the input current decreases, then the
efficiency has increased. If there is no change in input
current, then there is no change in efficiency.
Design Example
As a design example, assume VIN = 5V to 18V (12V nomi-
nal), VOUT = 12V (5%), IOUT(MAX) = 5A and f = 400kHz.
3780f
22
LTC3780
W U U
APPLICATIO S I FOR ATIO
U
Tie the PLLFLTR pin to INTVCC for 400kHz operation. The
inductance value is chosen first based on a 30% ripple
current assumption. In Buck mode, the ripple current is:
Double-check the TJ in the MOSFET with 70°C ambient
temperature:
TJ = 70°C + 1.94W • 40°C/W = 147.6°C
The maximum power dissipation of Switch B occurs in
Buckmode. AssumingajunctiontemperatureofTJ=80°C
with ρ80°C = 1.2, the power dissipation at VIN = 18V is:
VOUT
⎛
VOUT
V
IN
⎞
∆IL,BUCK
=
• 1–
⎜
⎟
f •L ⎝
⎠
Thehighestvalueofripplecurrentoccursatthemaximum
input voltage. In Boost mode, the ripple current is:
18 – 12
PB,BUCK
=
• 52 •1.2 • 0.009 = 135mW
12
V
⎛
V
IN
⎞
IN
∆IL,BOOST
=
• 1–
⎜
Double-check the TJ in the MOSFET at 70°C ambient
temperature:
⎟
f •L ⎝ VOUT
⎠
The highest value of ripple current occurs at VIN = VOUT/2.
TJ = 70°C + 0.135W • 40°C/W = 75.4°C
A 6.8µH inductor will produce 13% ripple in Boost mode
The maximum power dissipation of Switch C occurs in
Boostmode.AssumingajunctiontemperatureofTJ=110°C
with ρ110°C = 1.4, the power dissipation at VIN = 5V is:
(VIN = 6V) and 29% ripple in Buck mode (VIN = 18V).
The RSENSE resistor value can be calculated by using the
maximum current sense voltage specification with some
accommodation for tolerances.
12 – 5 •12
(
)
PC,BOOST
=
• 52 •1.4 • 0.009
52
2 •160mV • V
2 •IOUT(MAX,BOOST) + ∆IL,BOOST • V
IN
5
5
RSENSE
=
+ 2 •123 • •150p • 400k = 1.08W
(
)
OUT
Select an RSENSE of 10mΩ.
Double-check the TJ in the MOSFET at 70°C ambient
temperature:
Output voltage is 12V. Select R1 as 20k. R2 is:
TJ = 70°C + 1.08W • 40°C/W = 113°C
VOUT •R1
R2 =
–R1
The maximum power dissipation of Switch D occurs in
Boost mode when its duty cycle is higher than 50%.
Assuming a junction temperature of TJ = 100°C with
ρ100°C = 1.35, the power dissipation at VIN = 5V is:
0.8
SelectR2as280k.BothR1andR2shouldhaveatolerance
of no more than 1%.
Next, choose the MOSFET switches. A suitable choice is
theSiliconixSi4840(RDS(ON) =0.009Ω(atVGS=6V),CRSS
= 150pF, θJA = 40°C/W).
2
5
12
12
5
⎛
⎜
⎝
⎞
⎠
PD,BUCK
=
•
• 5 •1.35 • 0.009 = 0.73W
⎟
The maximum power dissipation of Switch A occurs
in Boost mode when Switch A stays on all the time.
Assuming a junction temperature of TJ = 150°C with
ρ150°C = 1.5, the power dissipation at VIN = 5V is:
Double-check the TJ in the MOSFET at 70°C ambient
temperature:
TJ = 70°C + 0.73W • 40°C/W = 99°C
CIN is chosen to filter the square current in Buck mode. In
this mode, the maximum input current peak is:
2
12
5
⎛
⎜
⎝
⎞
⎠
PA,BOOST
=
• 5 •1.5 • 0.009 = 1.94W
⎟
IIN,PEAK(MAX,BUCK) = 5 • (1 + 29%) = 6.5A
3780f
23
LTC3780
W U U
U
APPLICATIO S I FOR ATIO
A low ESR (10mΩ) capacitor is selected. Input voltage
• Place Switch B and Switch C as close to the controller
as possible, keeping the PGND, BG and SW traces
short.
ripple is 65mV.
COUT is chosen to filter the square current in Boost mode.
In this mode, the maximum output current peak is:
• KeepthehighdV/dTSW1,SW2,BOOST1,BOOST2,TG1
andTG2nodesawayfromsensitivesmall-signalnodes.
12
5
IOUT,PEAK(MAX,BUCK)
=
• 5 • 1+ 13% = 13.6A
(
)
• The path formed by Switch A, Switch B, D2 and the CIN
capacitor should have short leads and PC trace lengths.
ThepathformedbySwitchC,SwitchD,D1andtheCOUT
capacitor also should have short leads and PC trace
lengths.
A low ESR (5mΩ) capacitor is suggested. This capacitor
will limit output voltage ripple to 68mV.
PC Board Layout Checklist
• Theoutputcapacitor(–)terminalsshouldbeconnected
as close as possible the (–) terminals of the input
capacitor.
The basic PC board layout requires a dedicated ground
plane layer. Also, for high current, a multilayer board
provides heat sinking for power components.
• Connect the INTVCC decoupling capacitor CVCC closely
to the INTVCC and PGND pins.
• The ground plane layer should not have any traces and
it should be as close as possible to the layer with power
MOSFETs.
• Connect the top driver boost capacitor CA closely to the
BOOST1 and SW1 pins. Connect the top driver boost
capacitor CB closely to the BOOST2 and SW2 pins.
• Place CIN, Switch A, Switch B and D2 in one compact
area. Place COUT, Switch C, Switch D and D1 in one
compact area.
• Connect the input capacitors CIN and output capacitors
COUT close to the power MOSFETs. These capacitors
carry the MOSFET AC current in Boost and Buck mode.
• Use immediate vias to connect the components (in-
cluding the LTC3780’s SGND and PGND pins) to the
ground plane. Use several large vias for each power
component.
• Connect VOSENSE pin resistive dividers to the (+) termi-
nals of COUT and signal ground. A small VOSENSE
decoupling capacitor should be as close as possible to
the LTC3780 SGND pin. The R2 connection should not
be along the high current or noise paths, such as the
input capacitors.
• Route SENSE– and SENSE+ leads together with mini-
mum PC trace spacing. The filter capacitor between
SENSE+ and SENSE– should be as close as possible to
the IC. Ensure accurate current sensing with Kelvin
connections at the SENSE resistor.
• Use planes for VIN and VOUT to maintain good voltage
filtering and to keep power losses low.
• Flood all unused areas on all layers with copper. Flood-
ing with copper will reduce the temperature rise of
power components. Connect the copper areas to any
DC net (VIN or GND).
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC3780. These items are also illustrated in Figure 11.
• Connect the ITH pin compensation network close to the
IC, between ITH and the signal ground pins. The capaci-
tor helps to filter the effects of PCB noise and output
voltage ripple voltage from the compensation loop.
• Segregate the signal and power grounds. All small
signal components should return to the SGND pin at
onepointwhichisthentiedtothePGNDpinclosetothe
sources of Switch B and Switch C.
3780f
24
LTC3780
W U U
APPLICATIO S I FOR ATIO
U
• Connect the INTVCC decoupling capacitor close to the
IC, between the INTVCC and the power ground pins.
This capacitor carries the MOSFET drivers’ current
peaks. An additional 1µF ceramic capacitor placed
immediatelynexttotheINTVccandPGNDpinscanhelp
improve noise performance substantially.
V
OUT
R
PU
V
C
PULLUP
OUT
C
A
1
2
24
23
C
SS
PGOOD BOOST1
D2
SS
TG1
D
C
LTC3780
+
D
A
C
3
4
22
21
20
19
18
17
16
15
C2
SENSE
SENSE
SW1
C
C
F
–
C
V
C1
IN
R
C
5
I
EXTV
TH
CC
CC
6
R2
VCC
L
R1
V
INTV
OSENSE
7
SGND
RUN
BG1
PGND
BG2
R
SENSE
8
9
FCB
B
A
D1
10
PLLFLTR
SW2
D
B
11
12
14
13
f
PLLIN
TG2
IN
C
B
C
IN
STBYMD BOOST2
R
IN
V
IN
3780 F11
Figure 11. LTC3780 Layout Diagram
3780f
25
LTC3780
U
PACKAGE DESCRIPTIO
G Package
24-Lead Plastic SSOP (5.3mm)
(Reference LTC DWG # 05-08-1640)
7.90 – 8.50*
(.311 – .335)
1.25 ±0.12
24 23 22 21 20 19 18 17 16 15 14
13
7.8 – 8.2
5.3 – 5.7
7.40 – 8.20
(.291 – .323)
0.42 ±0.03
0.65 BSC
RECOMMENDED SOLDER PAD LAYOUT
5
7
8
1
2
3
4
6
9 10 11 12
2.0
5.00 – 5.60**
(.197 – .221)
(.079)
MAX
0° – 8°
0.65
(.0256)
BSC
0.09 – 0.25
0.55 – 0.95
(.0035 – .010)
(.022 – .037)
0.05
0.22 – 0.38
(.009 – .015)
TYP
(.002)
NOTE:
MIN
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
G24 SSOP 0204
3. DRAWING NOT TO SCALE
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED .152mm (.006") PER SIDE
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED .254mm (.010") PER SIDE
3780f
26
LTC3780
U
PACKAGE DESCRIPTIO
UH Package
32-Lead Plastic QFN (5mm × 5mm)
(Reference LTC DWG # 05-08-1693)
0.70 ±0.05
5.50 ±0.05
4.10 ±0.05
3.45 ±0.05
(4 SIDES)
PACKAGE
OUTLINE
0.25 ± 0.05
0.50 BSC
RECOMMENDED SOLDER PAD LAYOUT
BOTTOM VIEW—EXPOSED PAD
PIN 1 NOTCH R = 0.30 TYP
OR 0.35 × 45° CHAMFER
R = 0.115
TYP
0.75 ± 0.05
5.00 ± 0.10
(4 SIDES)
31 32
0.00 – 0.05
0.40 ± 0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
3.45 ± 0.10
(4-SIDES)
(UH32) QFN 1004
0.200 REF
0.25 ± 0.05
0.50 BSC
1. DRAWING PROPOSED TO BE A JEDEC PACKAGE OUTLINE 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
NOTE:
M0-220 VARIATION WHHD-(X) (TO BE APPROVED)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
3780f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
27
LTC3780
U
TYPICAL APPLICATIO
V
12V
5A
OUT
R
PU
C
C
A
OUT
V
PULLUP
D2
0.22µF
200µF
C
SS
6.8nF
B320A
1
2
24
23
PGOOD BOOST1
D
SS
TG1
Si7884DP
C
C2
47pF
LTC3780
+
D
A
1N5819HW
68pF
3
4
22
21
20
19
18
17
16
15
SENSE
SENSE
SW1
C
C1
3300pF
C 0.1µF
F
R
–
C
C
V
IN
5.6k
Si7884DP
5
L
I
EXTV
TH
CC
CC
C
4.7µF
R1
20k
VCC
6.8µF
6
R2 280k
V
INTV
OSENSE
D1
B320A
7
SGND
RUN
BG1
PGND
BG2
10mΩ
8
ON/OFF
9
B
FCB
Si7884DP
10
PLLFLTR
SW2
INTV
CC
D
B
1N5819HW
A
11
12
14
13
PLLIN
TG2
Si7884DP
C
IN
STBYMD BOOST2
47µF
C
C
0.22µF
STBYMD
0.01µF
10Ω
B
V
IN
100Ω
3780 TA02
100Ω
Figure 12. LTC3780 12V/3A, Buck-Boost Regulator
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1074HV/LT1076HV Monolithic 5A/2A Step-Down DC/DC Converters
V
V
up to 60V, TO-220 and DD Packages
IN
IN
LT1339
High Power Synchronous DC/DC Controller
Dual, 2-Phase Synchronous DC/DC Controller
Synchronous Step-Down DC/DC Controller
up to 60V, Drivers 10,000pF Gate Capacitance, I
≤ 20A
OUT
LTC1702A
LTC1735
LTC1778
LT1956
550kHz Operation, No R
, 3V ≤ V ≤ 7V, I
≤ 20A
SENSE
IN
OUT
3.5V ≤ V ≤ 36V, 0.8V ≤ V
≤6V, Current Mode, I
≤ 20A
IN
OUT
OUT
No R
Synchronous DC/DC Controller
4V ≤ V ≤ 36V, Fast Transient Response, Current Mode, I
≤ 20A
OUT
SENSE
IN
Monolithic 1.5A, 500kHz Step-Down Regulator
50mA, 3V to 80V Linear Regulator
5.5V ≤ V ≤ 60V, 2.5mA Supply Current, 16-Pin SSOP
IN
LT3010
1.275V ≤ V
≤ 60V, No Protection Diode Required, 8-Lead MSOP
OUT
LT3430/LT3431
LT3433
Monolithic 3A, 200kHz/500kHz Step-Down Regulator
Monolithic Step-Up/Step-Down DC/DC Converter
5.5V ≤ V ≤ 60V, 0.1Ω Saturation Switch, 16-Pin SSOP
IN
4V ≤ V ≤ 60V, 500mA Switch, Automatic Step-Up/Step-Down,
IN
Single Inductor
LTC®3443
LTC3703
Monolithic Buck-Boost Converter
2.4V ≤ V ≤ 5.5V, 96% Efficiency, 600kHz Operation, 2A Switch
IN
100V Synchronous DC/DC Controller
V up to 100V, 9.3V to 15V Gate Drive Supply
IN
3780f
LT/TP 0305 500 • PRINTED IN THE USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
28
●
●
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
©LINEAR TECHNOLOGY CORPORATION 2005
相关型号：
LT1076HVCT#PBF
LT1076 - Step-Down Switching Regulator; Package: TO-220; Pins: 5; Temperature Range: 0°C to 70°C
Linear
LT1076HVCT#TRPBF
IC 3.2 A SWITCHING REGULATOR, 125 kHz SWITCHING FREQ-MAX, PZFM7, PLASTIC, TO-220, 7 PIN, Switching Regulator or Controller
Linear
LT1076HVCT-5#PBF
LT1076-5 - 5V Step-Down Switching Regulator; Package: TO-220; Pins: 5; Temperature Range: 0°C to 70°C
Linear
LT1076HVCT-5#TRPBF
IC 3.2 A SWITCHING REGULATOR, 120 kHz SWITCHING FREQ-MAX, PZFM5, PLASTIC, TO-220, 5 PIN, Switching Regulator or Controller
Linear
©2020 ICPDF网 联系我们和版权申明