LT1228CS8 [Linear]
100MHz Current Feedback Amplifier with DC Gain Control; 100MHz的电流反馈放大器的直流增益控制型号: | LT1228CS8 |
厂家: | Linear |
描述: | 100MHz Current Feedback Amplifier with DC Gain Control |
文件: | 总20页 (文件大小:434K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
LT1228
100MHz Current Feedback
Amplifier with DC Gain Control
U
DESCRIPTIO
EATURE
S
F
■
Very Fast Transconductance Amplifier
Bandwidth: 75MHz
The LT1228 makes it easy to electronically control the gain
of signals from DC to video frequencies. The LT1228
implements gain control with a transconductance amplifier
(voltage to current) whose gain is proportional to an exter-
nally controlled current. A resistor is typically used to
convert the output current to a voltage, which is then
amplified with a current feedback amplifier. The LT1228
combines both amplifiers into an 8-pin package, and oper-
ates on any supply voltage from 4V (±2V) to 30V (±15V). A
complete differential input, gain controlled amplifier can be
implemented with the LT1228 and just a few resistors.
gm = 10 × ISET
Low THD: 0.2% at 30mVRMS Input
Wide ISET Range: 1µA to 1mA
Very Fast Current Feedback Amplifier
Bandwidth: 100MHz
■
Slew Rate: 1000V/µs
Output Drive Current: 30mA
Differential Gain: 0.04%
Differential Phase: 0.1°
High Input Impedance: 25MΩ, 6pF
Wide Supply Range: ±2V to ±15V
Inputs Common Mode to Within 1.5V of Supplies
Outputs Swing Within 0.8V of Supplies
Supply Current: 7mA
The LT1228 transconductance amplifier has a high imped-
ancedifferentialinputandacurrentsourceoutputwithwide
output voltage compliance. The transconductance, gm, is
setbythecurrentthatflowsintopin5, ISET. Thesmallsignal
gm isequaltotentimesthevalueofISETandthisrelationship
holdsoverseveraldecadesofsetcurrent. Thevoltageatpin
5 is two diode drops above the negative supply, pin 4.
■
■
■
■
O U
PPLICATI
S
A
■
■
■
■
■
■
Video DC Restore (Clamp) Circuits
Video Differential Input Amplifiers
Video Keyer/Fader Amplifiers
AGC Amplifiers
The LT1228 current feedback amplifier has very high input
impedance and therefore it is an excellent buffer for the
output of the transconductance amplifier. The current feed-
back amplifier maintains its wide bandwidth over a wide
range of voltage gains making it easy to interface the
transconductance amplifier output to other circuitry. The
current feedback amplifier is designed to drive low imped-
ance loads, such as cables, with excellent linearity at high
frequencies.
Tunable Filters
Oscillators
U
O
TYPICAL APPLICATI
Frequency Response
6
Differential Input Variable Gain Amp
V
= ±15V
= 100Ω
S
L
R
3
0
15V
4.7µF
+
I
= 1mA
SET
R3A
10k
–3
7
4
3
2
+
–
+
–6
1
8
g
R2A
10k
V
m
+
–
IN
–9
6
I
I
= 300µA
SET
–
5
CFA
V
OUT
–12
–15
–18
I
SET
R
–15V
R2
F
470Ω
R4
R5
10k
R3
100Ω
= 100µA
4.7µF
1.24k
SET
+
100Ω
R1
270Ω
R
G
–21
–24
HIGH INPUT RESISTANCE
EVEN WHEN POWER IS OFF
–18dB < GAIN < 2dB
10Ω
R6
100k
1M
10M
100M
6.19Ω
V
≤ 3V
IN
RMS
LT1228 • TA01
FREQUENCY (Hz)
LT1228 • TA02
1
LT1228
W W W
U
/O
ABSOLUTE AXI U RATI GS
PACKAGE RDER I FOR ATIO
Supply Voltage ...................................................... ±18V
Input Current, Pins 1, 2, 3, 5, 8 (Note 7) ............ ±15mA
Output Short Circuit Duration (Note 1) .........Continuous
Operating Temperature Range
LT1228C................................................ 0°C to 70°C
LT1228M........................................ –55°C to 125°C
Storage Temperature Range ................. –65°C to 150°C
Junction Temperature
TOP VIEW
ORDER PART
I
1
2
3
4
8
7
6
5
GAIN
NUMBER
OUT
+
–
+
V
–IN
g
m
LT1228MJ8
LT1228CJ8
LT1228CN8
LT1228CS8
+IN
V
OUT
–
V
I
SET
J8 PACKAGE
N8 PACKAGE
8-LEAD CERAMIC DIP 8-LEAD PLASTIC DIP
S8 PACKAGE
8-LEAD PLASTIC SOIC
S8 PART MARKING
1228
Plastic Package .............................................. 150°C
Ceramic Package ............................................ 175°C
Lead Temperature (Soldering, 10 sec).................. 300°C
TJ MAX = 175°C, θJA = 100°C/W (J)
TJ MAX = 150°C, θJA = 100°C/W (N)
TJ MAX = 150°C, θJA = 150°C/W (S)
Consult Factory for Industrial grade parts.
ELECTRICAL CHARACTERISTICS
Current Feedback Amplifier, Pins 1, 6, 8. ±5V ≤ VS ≤ ±15V, ISET = 0µA, VCM = 0V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
T = 25°C
MIN
TYP MAX
UNITS
V
OS
Input Offset Voltage
±3
±10
±15
mV
mV
A
●
●
Input Offset Voltage Drift
Noninverting Input Current
10
µV/°C
+
I
I
T = 25°C
A
±0.3
±3
±10
µA
µA
IN
●
●
–
Inverting Input Current
T = 25°C
A
±10
±65
±100
µA
µA
IN
e
Input Noise Voltage Density
Input Noise Current Density
Input Resistance
f = 1kHz, R = 1k, R = 10Ω, R = 0Ω
6
nV/√Hz
pV/√Hz
n
F
G
S
i
n
f = 1kHz, R = 1k, R = 10Ω, R = 10k
1.4
F
G
S
R
V
V
= ±13V, V = ±15V
●
●
2
2
25
25
MΩ
MΩ
IN
IN
IN
S
= ±3V, V = ±5V
S
C
IN
Input Capacitance (Note 2)
Input Voltage Range
V = ±5V
6
pF
S
V = ±15V, T = 25°C
S
±13 ±13.5
±12
V
V
V
V
A
●
●
V = ±5V, T = 25°C
S
±3
±2
±3.5
A
CMRR
PSRR
Common-Mode Rejection Ratio
V = ±15V, V = ±13V, T = 25°C
55
55
55
55
69
69
dB
dB
dB
dB
S
CM
A
V = ±15V, V = ±12V
●
●
S
CM
V = ±5V, V = ±3V, T = 25°C
S
CM
A
V = ±5V, V = ±2V
S
CM
Inverting Input Current
Common-Mode Rejection
V = ±15V, V = ±13V, T = 25°C
2.5
2.5
10
10
10
10
µA/V
µA/V
µA/V
µA/V
S
CM
A
V = ±15V, V = ±12V
●
●
S
CM
V = ±5V, V = ±3V, T = 25°C
S
CM
A
V = ±5V, V = ±2V
S
CM
Power Supply Rejection Ratio
V = ±2V to ±15V, T = 25°C
60
60
80
10
dB
dB
S
A
V = ±3V to ±15V
S
●
●
●
Noninverting Input Current
Power Supply Rejection
V = ±2V to ±15V, T = 25°C
50
50
nA/V
nA/V
S
A
V = ±3V to ±15V
S
Inverting Input Current
Power Supply Rejection
V = ±2V to ±15V, T = 25°C
0.1
5
5
µA/V
µA/V
S
A
V = ±3V to ±15V
S
2
LT1228
ELECTRICAL CHARACTERISTICS
Current Feedback Amplifier, Pins 1, 6, 8. ±5V ≤ VS ≤ ±15V, ISET = 0µA, VCM = 0V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
V = ±15V, V
MIN
TYP
MAX
UNITS
A
V
Large-Signal Voltage Gain
= ±10V, R = 1k
LOAD
●
●
55
55
65
65
dB
dB
S
OUT
V = ±5V, V
= ±2V, R
= 150Ω
S
OUT
LOAD
–
R
OL
Transresistance, ∆V /∆I
V = ±15V, V
= ±10V, R = 1k
LOAD
●
●
100
100
200
200
kΩ
kΩ
OUT IN
S
OUT
V = ±5V, V
= ±2V, R
= 150Ω
S
OUT
LOAD
V
OUT
Maximum Output Voltage Swing
V = ±15V, R
= 400Ω, T = 25°C
±12
±10
±3
±13.5
V
V
V
V
S
LOAD
A
●
●
V = ±5V, R
= 150Ω, T = 25°C
A
±3.7
S
LOAD
±2.5
I
I
Maximum Output Current
R
= 0Ω, T = 25°C
30
25
65
125
125
mA
mA
OUT
s
LOAD
A
●
●
Supply Current
V
= 0V, I
= 0V
6
11
mA
V/µs
V/µs
ns
OUT
SET
SR
SR
Slew Rate (Notes 3 and 5)
Slew Rate
T = 25°C
A
300
500
3500
10
V = ±15V, R = 750Ω, R = 750Ω, R = 400Ω
S
F
G
L
t
Rise Time (Notes 4 and 5)
Small-Signal Bandwidth
Small-Signal Rise Time
Propagation Delay
T = 25°C
A
20
r
BW
V = ±15V, R = 750Ω, R = 750Ω, R = 100Ω
S
100
3.5
MHz
ns
F
G
L
t
t
V = ±15V, R = 750Ω, R = 750Ω, R = 100Ω
S F G L
r
V = ±15V, R = 750Ω, R = 750Ω, R = 100Ω
S
3.5
ns
F
G
L
Small-Signal Overshoot
Settling Time
V = ±15V, R = 750Ω, R = 750Ω, R = 100Ω
S
15
%
F
G
L
0.1%, V
= 10V, R =1k, R = 1k, R =1k
45
ns
s
OUT
F
G
L
Differential Gain (Note 6)
Differential Phase (Note 6)
Differential Gain (Note 6)
Differential Phase (Note 6)
V = ±15V, R = 750Ω, R = 750Ω, R = 1k
0.01
0.01
0.04
0.1
%
S
F
G
L
V = ±15V, R = 750Ω, R = 750Ω, R = 1k
DEG
%
S
F
G
L
V = ±15V, R = 750Ω, R = 750Ω, R = 150Ω
S
F
G
L
V = ±15V, R = 750Ω, R = 750Ω, R = 150Ω
DEG
S
F
G
L
ELECTRICAL CHARACTERISTICS
Transconductance Amplifier, Pins 1, 2, 3, 5. ±5V ≤ VS ≤ ±15V, ISET = 100µA, VCM = 0V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
= 1mA, T = 25°C
MIN
TYP
MAX
UNITS
V
OS
Input Offset Voltage
I
±0.5
±5
±10
mV
mV
SET
A
●
●
Input Offset Voltage Drift
Input Offset Current
10
40
µV/°C
I
I
T = 25°C
A
200
500
nA
nA
OS
●
●
Input Bias Current
T = 25°C
A
0.4
1
5
µA
µA
B
e
Input Noise Voltage Density
f = 1kHz
20
nV/√Hz
kΩ
n
R
Input Resistance-Differential Mode
Input Resistance-Common Mode
V
≈ ±30mV
IN
●
30
200
IN
V = ±15V, V = ±12V
●
●
50
50
1000
1000
MΩ
MΩ
S
CM
V = ±5V, V = ±2V
S
CM
C
IN
Input Capacitance
3
pF
Input Voltage Range
V = ±15V, T = 25°C
±13
±12
±3
±14
V
V
V
V
S
A
V = ±15V
●
●
S
V = ±5V, T = 25°C
±4
S
A
V = ±5V
S
±2
3
LT1228
ELECTRICAL CHARACTERISTICS
Transconductance Amplifier, Pins 1, 2, 3, 5. ±5V ≤ VS ≤ ±15V, ISET = 100µA, VCM = 0V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
CMRR
Common-Mode Rejection Ratio
V = ±15V, V = ±13V, T = 25°C
60
60
60
60
100
dB
dB
dB
dB
S
CM
A
V = ±15V, V = ±12V
●
●
S
CM
V = ±5V, V = ±3V, T = 25°C
100
100
S
CM
A
V = ±5V, V = ±2V
S
CM
PSRR
Power Supply Rejection Ratio
V = ±2V to ±15V, T = 25°C
60
60
dB
dB
S
A
V = ±3V to ±15V
S
●
g
Transconductance
I
= 100µA, I
= ±30µA, T = 25°C
0.75
1.00
–0.33
100
1.25
130
µA/mV
%/°C
µA
m
SET
OUT
A
Transconductance Drift
Maximum Output Current
Output Leakage Current
●
●
I
I
I
I
= 100µA
70
OUT
OL
SET
SET
= 0µA (+I of CFA), T = 25°C
0.3
3
10
µA
µA
IN
A
●
V
Maximum Output Voltage Swing
Output Resistance
V = ±15V , R1 = ∞
V = ±5V , R1 = ∞
S
●
●
±13
±3
±14
±4
V
V
OUT
S
R
V = ±15V, V = ±13V
OUT
●
●
2
2
8
8
MΩ
MΩ
O
S
V = ±5V, V
S
= ±3V
OUT
Output Capacitance (Note 2)
Supply Current, Both Amps
Total Harmonic Distortion
Small-Signal Bandwidth
Small-Signal Rise Time
Propagation Delay
V = ±5V
6
pF
mA
%
S
I
I
= 1mA
SET
●
9
15
S
THD
BW
V
= 30mV
at 1kHz, R1 = 100k
0.2
80
5
IN
RMS
R1 = 50Ω, I = 500µA
MHz
ns
SET
t
R1 = 50Ω, I = 500µA, 10% to 90%
SET
r
R1 = 50Ω, I = 500µA, 50% to 50%
5
ns
SET
The
range.
● denotes specifications which apply over the operating temperature
Note 4: Rise time is measured from 10% to 90% on a ±500mV output
signal while operating on ±15V supplies with R = 1k, R = 110Ω and
F
G
R = 100Ω. This condition is not the fastest possible, however, it does
guarantee the internal capacitances are correct and it makes automatic
testing practical.
Note 5: AC parameters are 100% tested on the ceramic and plastic DIP
packaged parts (J and N suffix) and are sample tested on every lot of
the SO packaged parts (S suffix).
Note 1: A heat sink may be required depending on the power supply
voltage.
Note 2: This is the total capacitance at pin 1. It includes the input
capacitance of the current feedback amplifier and the output capacitance
of the transconductance amplifier.
L
Note 3: Slew rate is measured at ±5V on a ±10V output signal while
Note 6: NTSC composite video with an output level of 2V.
operating on ±15V supplies with R = 1k, R = 110Ω and R = 400Ω. The
F
G
L
slew rate is much higher when the input is overdriven, see the applications
section.
Note 7: Back to back 6V Zener diodes are connected between pins 2 and
3 for ESD protection.
4
LT1228
TYPICAL PERFOR A CE CHARACTERISTICS Transconductance Amplifier, Pins 1, 2, 3 & 5
U W
Small-Signal Bandwidth vs
Set Current
Small-Signal Transconductance
and Set Current vs Bias Voltage
Small-Signal Transconductance
vs DC Input Voltage
100
10
1
2.0
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
100
10
10000
1000
100
10
R1 = 100Ω
V
= ±2V TO ±15V
= 25°C
V
= ±15V
S
A
S
V
SET
= ±2V TO ±15V
S
T
I
= 100µA
R1 = 1k
–55°C
1
R1 = 10k
25°C
0.1
125°C
0.01
0.001
1.0
R1 = 100k
0.2
0
0.1
0.1
1.5
10
100
SET CURRENT (µA)
1000
–200 –150 –100 –50
0
50 100 150 200
0.9
1.0
1.1
1.2
1.3
1.4
INPUT VOLTAGE (mVDC)
BIAS VOLTAGE, PIN 5 TO 4, (V)
LT1228 • TPC01
LT1228 • TPC03
LT1228 • TPC02
Total Harmonic Distortion vs
Input Voltage
Spot Output Noise Current vs
Frequency
Input Common-Mode Limit vs
Temperature
+
1000
100
10
V
10
1
V
= ±2V TO ±15V
= 25°C
+
S
A
V
= ±15V
V
= 2V TO 15V
S
–0.5
–1.0
–1.5
–2.0
T
I
I
= 1mA
SET
I
= 100µA
SET
2.0
1.5
1.0
0.5
0.1
0.01
–
V
= –2V TO –15V
= 100µA
SET
I
= 1mA
10
SET
–
V
10
100
1k
FREQUENCY (Hz)
10k
100k
–50 –25
0
25
50
75 100 125
1
100
1000
INPUT VOLTAGE (mV
)
LT1228 • TPC04
TEMPERATURE (°C)
P–P
LT1228 • TPC05
LT1228 • TPC06
Small-Signal Control Path
Bandwidth vs Set Current
Small-Signal Control Path
Gain vs Input Voltage
Output Saturation Voltage vs
Temperature
100
10
1
+
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
V
V
V
= ±2V TO ±15V
= 200mV
S
IN
–0.5
–1.0
(PIN 2 TO 3)
±2V ≤ V ≤ ±15V
S
∆I
∆I
OUT
OUT
R1 = ∞
∆I
∆I
SET
SET
+1.0
+0.5
–
V
10
100
SET CURRENT (µA)
1000
0
40
80
120
160
200
–50 –25
0
25
50
75 100 125
INPUT VOLTAGE, PIN 2 TO 3, (mVDC)
TEMPERATURE (°C)
LT1228 • TPC07
LT1228 • TPC08
LT1228 • TPC09
5
LT1228
TYPICAL PERFOR A CE CHARACTERISTICS Current Feedback Amplifier, Pins 1, 6, 8
U W
Voltage Gain and Phase vs
Frequency, Gain = 6dB
–3dB Bandwidth vs Supply
Voltage, Gain = 2, RL = 100Ω
–3dB Bandwidth vs Supply
Voltage, Gain = 2, RL = 1k
180
160
140
180
160
140
8
7
6
0
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
PHASE
GAIN
45
90
R
= 500Ω
F
R
R
= 500Ω
= 750Ω
R
= 750Ω
F
F
135
180
225
120
100
80
60
40
20
0
120
100
80
60
40
20
0
5
4
F
3
2
1
R
= 1k
F
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
V
= ±15V
S
L
R
= 100Ω
0
–1
–2
R = 2k
F
R
= 1k
6
R
= 2k
F
R = 750Ω
F
F
0.1
1
10
100
0
0
0
2
4
6
8
10 12 14 16 18
0
0
0
2
4
8
10 12 14 16 18
FREQUENCY (MHz)
SUPPLY VOLTAGE (±V)
SUPPLY VOLTAGE (±V)
LT1228 • TPC10
LT1228 • TPC11
LT1228 • TPC12
Voltage Gain and Phase vs
Frequency, Gain = 20dB
–3dB Bandwidth vs Supply
Voltage, Gain = 10, RL = 100Ω
–3dB Bandwidth vs Supply
Voltage, Gain = 10, RL = 1kΩ
180
160
140
22
21
20
0
180
160
140
PHASE
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
45
90
GAIN
135
180
225
19
18
120
100
80
60
40
20
0
120
100
80
60
40
20
0
R
= 250Ω
R = 500Ω
F
F
R
= 250Ω
F
17
16
15
R
= 500Ω
F
R
= 750Ω
F
R
= 750Ω
F
R
F
= 1k
= 2k
V
= ±15V
S
L
R
= 1k
= 2k
F
R
= 100Ω
14
13
12
R = 750Ω
R
F
R
F
F
0.1
1
10
100
2
4
6
8
10 12 14 16 18
2
4
6
8
10 12 14 16 18
FREQUENCY (MHz)
SUPPLY VOLTAGE (±V)
SUPPLY VOLTAGE (±V)
LT1228 • TPC13
LT1228 • TPC14
LT1228 • TPC15
Voltage Gain and Phase vs
Frequency, Gain = 40dB
–3dB Bandwidth vs Supply
Voltage, Gain = 100, RL = 100Ω
–3dB Bandwidth vs Supply
Voltage, Gain = 100, RL = 1kΩ
18
16
14
42
41
40
0
18
16
14
PHASE
45
90
R
= 500Ω
GAIN
F
135
180
225
39
38
12
10
8
12
10
8
R = 500Ω
F
R
= 1k
F
37
36
35
R = 1k
F
R
= 2k
F
6
R = 2k
F
6
V
= ±15V
S
L
R
= 100Ω
4
4
34
33
32
R = 750Ω
F
2
2
0
0
0.1
1
10
100
2
4
6
8
10 12 14 16 18
2
4
6
8
10 12 14 16 18
FREQUENCY (MHz)
SUPPLY VOLTAGE (±V)
SUPPLY VOLTAGE (±V)
LT1228 • TPC16
LT1228 • TPC17
LT1228 • TPC18
6
LT1228
U W
TYPICAL PERFOR A CE CHARACTERISTICS Current Feedback Amplifier, Pins 1, 6, 8
Maximum Capacitive Load vs
Feedback Resistor
Total Harmonic Distortion vs
Frequency
2nd and 3rd Harmonic
Distortion vs Frequency
0.10
10k
1k
–20
–30
V
V
= ±15V
P–P
= 100Ω
S
O
L
V
= ±15V
S
L
= 2V
R
= 400Ω
R
V
= ±5V
R = R = 750Ω
S
F
G
R = 750Ω
A
F
= 10dB
2nd
V
–40
–50
–60
–70
V
= ±15V
S
100
10
1
0.01
V
V
= 7V
= 1V
3rd
O
O
RMS
RMS
R
= 1k
L
PEAKING ≤ 5dB
GAIN = 2
0.001
0
1
2
3
10
100
1k
FREQUENCY (Hz)
10k
100k
1
10
FREQUENCY (MHz)
100
FEEDBACK RESISTOR (kΩ)
LT1228 • TPC19
LT1228 • TPC20
LT1228 • TPC21
Input Common-Mode Limit vs
Temperature
Output Saturation Voltage vs
Temperature
Output Short-Circuit Current vs
Temperature
+
+
70
60
50
40
30
V
V
–0.5
–1.0
–1.5
–2.0
–0.5
–1.0
+
V
V
= 2V TO 15V
R
= ∞
L
±2V ≤ V ≤ ±15V
S
2.0
1.5
1.0
0.5
–
= –2V TO –15V
1.0
0.5
–
–
V
V
–50 –25
0
25
50
75 100 125
–50 –25
0
25
50
75 100 125
–50 –25
0
25 50 75 100 125 150 175
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
LT1228 • TPC22
LT1228 • TPC23
LT1228 • TPC24
Spot Noise Voltage and Current vs
Frequency
Power Supply Rejection vs
Frequency
Output Impedance vs
Frequency
100
80
60
100
10
V
= ±15V
S
V
= ±15V
S
L
R
= 100Ω
R = R = 750Ω
F
G
POSITIVE
1.0
–i
n
R = R = 2k
F
G
10
40
20
0
R = R = 750Ω
F
G
e
0.1
0.01
n
NEGATIVE
+i
n
0.001
1
10k
100k
1M
10M
100M
10
100
1k
FREQUENCY (Hz)
10k
100k
10k
100k
1M
10M
100M
FREQUENCY (Hz)
FREQUENCY (Hz)
LT1228 • TPC27
LT1228 • TPC25
LT1228 • TPC26
7
LT1228
TYPICAL PERFOR A CE CHARACTERISTICS Current Feedback Amplifier, Pins 1, 6 & 8
U W
Settling Time to 10mV vs
Output Step
Settling Time to 1mV vs
Output Step
Supply Current vs Supply Voltage
10
9
10
8
10
8
NONINVERTING
NONINVERTING
INVERTING
8
6
6
–55°C
INVERTING
4
7
4
25°C
2
6
2
V
= ±15V
G
V
= ±15V
G
S
F
S
F
0
0
5
125°C
R
= R = 1k
R = R = 1k
–2
–4
–2
–4
4
3
2
1
0
175°C
INVERTING
–6
–6
NONINVERTING
–8
–8
NONINVERTING
20
INVERTING
60 80
–10
–10
0
4
8
12
16
20
0
40
100
0
2
4
6
8
10 12 14 16 18
SETTLING TIME (µs)
SETTLING TIME (ns)
SUPPLY VOLTAGE (±V)
LT1228 • TPC29
LT1228 • TPC28
LT1228 • TPC30
W
W
SI PLIFIED SCHE ATIC
+
V
V
V
7
6
4
BIAS
+IN
3
–IN
2
I
OUT
8
GAIN
OUT
1
I
SET
5
–
LT1228 • TA03
8
LT1228
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PPLICATI
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The LT1228 contains two amplifiers, a transconductance
amplifier (voltage-to-current) and a current feedback am-
plifier (voltage-to-voltage). The gain of the transconduc-
tance amplifier is proportional to the current that is exter-
nally programmed into pin 5. Both amplifiers are designed
to operate on almost any available supply voltage from 4V
(±2V) to 30V (±15V). The output of the transconductance
amplifier is connected to the noninverting input of the
current feedback amplifier so that both fit into an eight pin
package.
Resistance Controlled Gain
If the set current is to be set or varied with a resistor or
potentiometer it is possible to use the negative tempera-
ture coefficient at pin 5 (with respect to pin 4) to compen-
sateforthenegativetemperaturecoefficientofthetranscon-
ductance. The easiest way is to use an LT1004-2.5, a 2.5V
reference diode, as shown below:
Temperature Compensation of gm with a 2.5V Reference
R
TRANSCONDUCTANCE AMPLIFIER
I
SET
TheLT1228transconductanceamplifierhasahighimped-
ance differential input (pins 2 and 3) and a current source
output (pin 1) with wide output voltage compliance. The
voltage to current gain or transconductance (gm) is set by
the current that flows into pin 5, ISET. The voltage at pin 5
is two forward biased diode drops above the negative
supply, pin 4. Therefore the voltage at pin 5 (with
respect to V–) is about 1.2V and changes with the log of
the set current (120mV/decade), see the characteristic
curves. The temperature coefficient of this voltage is
about –4mV/°C (–3300ppm/°C) and the temperature co-
efficient of the logging characteristic is 3300ppm/°C. It is
important that the current into pin 5 be limited to less than
15mA. THE LT1228 WILL BE DESTROYED IF PIN 5 IS
SHORTED TO GROUND OR TO THE POSITIVE SUPPLY. A
limiting resistor (2k or so) should be used to prevent more
than 15mA from flowing into pin 5.
g
m
I
V
V
be
4
2.5V
2E
g
5
R
SET
be
LT1004-2.5
–
V
LT1228 • TA04
The current flowing into pin 5 has a positive temperature
coefficient that cancels the negative coefficient of the
transconductance. The following derivation shows why a
2.5V reference results in zero gain change with tempera-
ture:
q
ISET
kT 3.87
Since gm =
×
= 10 × ISET
akT
q
cTn
Ic
and Vbe = Eg –
where a = In
≈ 19.4 at 27°C
The small-signal transconductance (gm) is equal to ten
times the value of ISET (in mA/mV) and this relationship
holdsovermanydecadesofsetcurrent(seethecharacter-
istic curves). The transconductance is inversely propor-
tional to absolute temperature (–3300ppm/°C). The input
stage of the transconductance amplifier has been de-
signedtooperatewithmuchlargersignalsthanispossible
with an ordinary diff-amp. The transconductance of the
input stage varies much less than 1% for differential input
signals over a ±30 mV range (see the characteristic curve
Small-Signal Transconductance vs DC Input Voltage).
c = 0.001,n = 3,Ic = 100µA
(
)
Eg is about 1.25V so the 2.5V reference is 2Eg. Solving
the loop for the set current gives:
akT
2Eg – 2 Eg –
q
2akT
Rq
ISET
=
or ISET =
R
9
LT1228
PPLICATI
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A
S I FOR ATIO
Substitutingintotheequationfortransconductancegives:
is two diode drops above the negative supply, a single
resistorfromthecontrolvoltagesourcetopin5willsuffice
in many applications. The control voltage is referenced to
thenegativesupplyandhasanoffsetofabout900mV. The
conversion will be monotonic, but the linearity is deter-
mined by the change in the voltage at pin 5 (120mV per
decade of current). The characteristic is very repeatable
since the voltage at pin 5 will vary less than ±5% from part
to part. The voltage at pin 5 also has a negative tempera-
turecoefficientasdescribedintheprevioussection.When
the gain of several LT1228s are to be varied together, the
current can be split equally by using equal value resistors
to each pin 5.
a
10
R
gm =
=
1.94R
The temperature variation in the term “a” can be ignored
since it is much less than that of the term “T” in the
equation for Vbe. Using a 2.5V source this way will main-
tain the gain constant within 1% over the full temperature
range of –55°C to 125°C. If the 2.5V source is off by 10%,
the gain will vary only about ±6% over the same tempera-
ture range.
We can also temperature compensate the transconduc-
tance without using a 2.5V reference if the negative power
supply is regulated. A Thevenin equivalent of 2.5V is
generated from two resistors to replace the reference. The
two resistors also determine the maximum set current,
approximately 1.1V/RTH. By rearranging the Thevenin
equations to solve for R4 and R6 we get the following
equations in terms of RTH and the negative supply, VEE.
For more accurate (and linear) control, a voltage-to-
current converter circuit using one op amp can be used.
The following circuit has several advantages. The input no
longer has to be referenced to the negative supply and the
input can be either polarity (or differential). This circuit
works on both single and split supplies since the input
voltage and the pin 5 voltage are independent of each
other. The temperature coefficient of the output current is
set by R5.
RTH
2.5V
VEE
RTHVEE
2.5V
R4 =
andR6 =
1–
R3
1M
Temperature Compensation of gm with a Thevenin Voltage
R1
1M
1.03k
R'
R5
1k
V1
V2
+
–
I
SET
I
OUT
R2
1M
LT1006
50pF
TO PIN 5
g
OF LT1228
m
I
V
V
be
4
R4
1M
R6
V
= 2.5V
TH
6.19kΩ
5
R'
SET
be
R1 = R2
R3 = R4
R4
1.24kΩ
(V1 – V2) R3
–15V
I
=
×
= 1mA/V
OUT
LT1228 • TA05
R5
R1
LT1228 • TA19
Voltage Controlled Gain
Digital control of the transconductance amplifier gain is
donebyconvertingtheoutputofaDACtoacurrentflowing
into pin 5. Unfortunately most current output DACs
sink rather than source current and do not have output
To use a voltage to control the gain of the transconduc-
tance amplifier requires converting the voltage into a
current that flows into pin 5. Because the voltage at pin 5
10
LT1228
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PPLICATI
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Transconductance Amp Small-Signal Response
compliance compatible with pin 5 of the LT1228. There-
fore, the easiest way to digitally control the set current is
to use a voltage output DAC and a voltage-to-current
circuit.Thepreviousvoltage-to-currentconverterwilltake
the output of any voltage output DAC and drive pin 5 with
a proportional current. The R, 2R CMOS multiplying DACs
operatinginthevoltageswitchingmodeworkwellonboth
single and split supplies with the above circuit.
ISET = 500µA, R1 = 50Ω
Logarithmic control is often easier to use than linear
control. A simple circuit that doubles the set current for
each additional volt of input is shown in the voltage
controlled state variable filter application near the end of
this data sheet.
Transconductance Amplifier Frequency Response
CURRENT FEEDBACK AMPLIFIER
The bandwidth of the transconductance amplifier is a
function of the set current as shown in the characteristic
curves. At set currents below 100µA, the bandwidth is
approximately:
The LT1228 current feedback amplifier has very high
noninverting input impedance and is therefore an excel-
lent buffer for the output of the transconductance ampli-
fier. The noninverting input is at pin 1, the inverting input
at pin 8 and the output at pin 6. The current feedback
amplifier maintains its wide bandwidth for almost all
voltage gains making it easy to interface the output levels
of the transconductance amplifier to other circuitry. The
current feedback amplifier is designed to drive low imped-
ance loads such as cables with excellent linearity at high
frequencies.
–3dB bandwidth = 3 × 1011 ISET
The peak bandwidth is about 80MHz at 500µA. When a
resistor is used to convert the output current to a voltage,
the capacitance at the output forms a pole with the
resistor. The best case output capacitance is about 5pF
with ±15V supplies and 6pF with ±5V supplies. You must
add any PC board or socket capacitance to these values to
get the total output capacitance. When using a 1k resistor
at the output of the transconductance amp, the output
capacitance limits the bandwidth to about 25MHz.
Feedback Resistor Selection
The small-signal bandwidth of the LT1228 current feed-
backamplifierisset bythe external feedbackresistors and
the internal junction capacitors. As a result, the bandwidth
is a function of the supply voltage, the value of the
feedback resistor, the closed-loop gain and load resistor.
The characteristic curves of bandwidth versus supply
voltage are done with a heavy load (100Ω) and a light load
(1k) to show the effect of loading. These graphs also show
The output slew rate of the transconductance amplifier is
the set current divided by the output capacitance, which is
6pF plus board and socket capacitance. For example with
the set current at 1mA, the slew rate would be over
100V/µs.
11
LT1228
PPLICATI
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the family of curves that result from various values of the
feedback resistor. These curves use a solid line when the
response has less than 0.5dB of peaking and a dashed line
for the response with 0.5dB to 5dB of peaking. The curves
stop where the response has more than 5dB of peaking.
Capacitance on the Inverting Input
Current feedback amplifiers want resistive feedback from
the output to the inverting input for stable operation. Take
care to minimize the stray capacitance between the output
and the inverting input. Capacitance on the inverting input
to ground will cause peaking in the frequency response
(and overshoot in the transient response), but it does not
degrade the stability of the amplifier. The amount of
capacitance that is necessary to cause peaking is a func-
tion of the closed-loop gain taken. The higher the gain, the
more capacitance is required to cause peaking. For ex-
ample, in a gain of 100 application, the bandwidth can be
increased from 10MHz to 17MHz by adding a 2200pF
capacitor, as shown below. CG must have very low series
resistance, such as silver mica.
Current Feedback Amp Small-Signal Response
VS = ±15V, RF = RG = 750Ω, RL = 100Ω
1
+
V
IN
6
CFA
V
OUT
8
–
R
F
510Ω
At a gain of two, on ±15V supplies with a 750Ω feedback
resistor, the bandwidth into a light load is over 160MHz
without peaking, but into a heavy load the bandwidth
reduces to 100MHz. The loading has so much effect
because there is a mild resonance in the output stage that
enhances the bandwidth at light loads but has its Q
reduced by the heavy load. This enhancement is only
usefulatlowgainsettings, atagainoftenitdoesnotboost
the bandwidth. At unity gain, the enhancement is so
effective the value of the feedback resistor has very little
effect on the bandwidth. At very high closed-loop gains,
the bandwidth is limited by the gain-bandwidth product of
about 1GHz. The curves show that the bandwidth at a
closed-loop gain of 100 is 10MHz, only one tenth what it
is at a gain of two.
R
G
5.1Ω
C
G
LT1228 • TA08
Boosting Bandwidth of High Gain Amplifier
with Capacitance On Inverting Input
49
46
C
= 4700pF
G
43
40
37
34
31
28
25
22
19
C
= 2200pF
G
C
= 0
G
1
10
FREQUENCY (MHz)
100
LT1228 • TA09
12
LT1228
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PPLICATI
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Capacitive Loads
The output slew rate is set by the value of the feedback
resistors and the internal capacitance. At a gain of ten with
a 1k feedback resistor and ±15V supplies, the output slew
rate is typically 500V/µs and –850V/µs. There is no input
stage enhancement because of the high gain. Larger
feedback resistors will reduce the slew rate as will lower
supply voltages, similar to the way the bandwidth is
reduced.
The LT1228 current feedback amplifier can drive capaci-
tive loads directly when the proper value of feedback
resistor is used. The graph of Maximum Capacitive Load
vs Feedback Resistor should be used to select the appro-
priate value. The value shown is for 5dB peaking when
driving a 1k load, at a gain of 2. This is a worst case
condition, the amplifier is more stable at higher gains, and
driving heavier loads. Alternatively, a small resistor (10Ω
to 20Ω) can be put in series with the output to isolate the
capacitive load from the amplifier output. This has the
advantage that the amplifier bandwidth is only reduced
when the capacitive load is present and the disadvantage
that the gain is a function of the load resistance.
Current Feedback Amp Large-Signal Response
VS = ±15V, RF = 1k, RG = 110Ω, RL = 400Ω
Slew Rate
The slew rate of the current feedback amplifier is not
independent of the amplifier gain configuration the way it
is in a traditional op amp. This is because the input stage
and the output stage both have slew rate limitations. The
inputstageoftheLT1228currentfeedbackamplifierslews
at about 100V/µs before it becomes nonlinear. Faster
input signals will turn on the normally reverse biased
emittersontheinputtransistorsandenhancetheslewrate
significantly. This enhanced slew rate can be as much as
3500V/µs!
Settling Time
The characteristic curves show that the LT1228 current
feedback amplifier settles to within 10mV of final value in
40ns to 55ns for any output step less than 10V. The curve
ofsettlingto1mVoffinalvalueshowsthatthereisaslower
thermal contribution up to 20µs. The thermal settling
component comes from the output and the input stage.
Theoutputcontributesjustunder1mV/Vofoutputchange
and the input contributes 300µV/V of input change.
Fortunately the input thermal tends to cancel the output
thermal. For this reason the noninverting gain of two
configuration settles faster than the inverting gain of one.
Current Feedback Amp Large-Signal Response
VS = ±15V, RF = RG = 750Ω Slew Rate Enhanced
13
LT1228
PPLICATI
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Power Supplies
For example, let’s calculate the worst case power dissipa-
tion in a variable gain video cable driver operating on ±12V
supplies that delivers a maximum of 2V into 150Ω. The
maximum set current is 1mA.
The LT1228 amplifiers will operate from single or split
supplies from ±2V (4V total) to ±18V (36V total). It is not
necessary to use equal value split supplies, however the
offset voltage and inverting input bias current of the
current feedback amplifier will degrade. The offset voltage
changesabout350µV/Vofsupplymismatch,theinverting
bias current changes about 2.5µA/V of supply mismatch.
VOMAX
PD = 2V I
+ 3.5ISET + V – V
S OMAX
(
)
(
)
RL
S
SMAX
2V
150Ω
PD = 2 × 12V × 7mA + 3.5 × 1mA + 12V – 2V
(
)
]
(
)
[
= 0.252 + 0.133 = 0.385W
Power Dissipation
Thetotalpowerdissipationtimesthethermalresistanceof
the package gives the temperature rise of the die above
ambient. The above example in SO-8 surface mount
package (thermal resistance is 150°C/W) gives:
The worst case amplifier power dissipation is the total of
the quiescent current times the total power supply voltage
plus the power in the IC due to the load. The quiescent
supply current of the LT1228 transconductance amplifier
isequalto3.5timesthesetcurrentatalltemperatures.The
quiescent supply current of the LT1228 current feedback
amplifier has a strong negative temperature coefficient
and at 150°C is less than 7mA, typically only 4.5mA. The
power in the IC due to the load is a function of the output
voltage, the supply voltage and load resistance. The worst
case occurs when the output voltage is at half supply, if it
can go that far, or its maximum value if it cannot reach half
supply.
Temperature Rise = PDθJA = 0.385W × 150°C/W
= 57.75°C
Therefore the maximum junction temperature is 70°C
+57.75°C or 127.75°C, well under the absolute maximum
junction temperature for plastic packages of 150°C.
U
O
TYPICAL APPLICATI S
Basic Gain Control
gm × R1= 10 × ISET × R1
The basic gain controlled amplifier is shown on the front
page of the data sheet. The gain is directly proportional to
the set current. The signal passes through three stages
from the input to the output.
Lastly the signal is buffered and amplified by the current
feedback amplifier (CFA). The voltage gain of the current
feedback amplifier is:
RF
1+
RG
First the input signal is attenuated to match the dynamic
range of the transconductance amplifier. The attenuator
should reduce the signal down to less than 100mV peak.
The characteristic curves can be used to estimate how
muchdistortiontherewillbeatmaximuminputsignal. For
single ended inputs eliminate R2A or R3A.
The overall gain of the gain controlled amplifier is the
product of all three stages:
R3
R3 + R3A
RF
RG
A V =
× 10 × ISET × R1× 1+
The signal is then amplified by the transconductance
amplifier (gm) and referred to ground. The voltage gain of
the transconductance amplifier is:
More than oneoutput can be summedinto R1 becausethe
output of the transconductance amplifier is a current. This
is the simplest way to make a video mixer.
14
LT1228
U
O
TYPICAL APPLICATI S
Video Fader
Video DC Restore (Clamp) Circuit
NOT NECESSARY IF THE SOURCE RESISTANCE IS LESS THAN 50Ω
200Ω
1k
3
+
V
V
1000pF
+
IN1
1
g
3
2
7
m
+
–
+
2
LT1223
CFA
1
8
–
V
OUT
g
+
–
m
5
6
V
0.01µF
–
CFA
R
5
OUT
1k
100Ω
4
10k
10k
10k
5.1k
F
–
V
V
= ±5V
R
10k
5.1k
–5V
1k
S
G
5V
3k
3k
VIDEO
INPUT
1k
3
2
5
V
+
IN2
LOGIC
INPUT
1
g
2N3906
m
100Ω
–
RESTORE
LT1228 • TA13
LT1228 • TA12
The video fader uses the transconductance amplifiers
from two LT1228s in the feedback loop of another current
feedback amplifier, the LT1223. The amount of signal
from each input at the output is set by the ratio of the
set currents of the two LT1228s, not by their absolute
value. The bandwidth of the current feedback amplifier
is inversely proportional to the set current in this
configuration. Therefore, the set currents remain high
over most of the pot’s range, keeping the bandwidth over
15MHz even when the signal is attenuated 20dB. The pot
is set up to completely turn off one LT1228 at each end of
the rotation.
Thevideorestore(clamp)circuitrestorestheblacklevelof
the composite video to zero volts at the beginning of every
line. This is necessary because AC coupled video changes
DC level as a function of the average brightness of the
picture. DC restoration also rejects low frequency noise
such as hum.
The circuit has two inputs: composite video and a logic
signal. The logic signal is high except during the back
porch time right after the horizontal sync pulse. While the
logicishigh, thePNPisoffandISET iszero. WithISET equal
to zero the feedback to pin 2 has no affect. The video input
drives the noninverting input of the current feedback
amplifier whose gain is set by RF and RG. When the logic
signalislow,thePNPturnsonandISET goestoabout1mA.
Then the transconductance amplifier charges the capaci-
tor to force the output to match the voltage at pin 3, in this
case zero volts.
ThiscircuitcanbemodifiedsothatthevideoisDCcoupled
by operating the amplifier in an inverting configuration.
Just ground the video input shown and connect RG to the
video input instead of to ground.
15
LT1228
U
O
TYPICAL APPLICATI S
Single Supply Wien Bridge Oscillator
3 at resonance; therefore the attenuation of the 1.8k resis-
tor and the transconductance amplifier must be about 11,
resulting in a set current of about 600µA at oscillation. At
start-upthereisnosetcurrentandthereforenoattenuation
for a net gain of about 11 around the loop. As the output
oscillation builds up it turns on the PNP transistor which
generates the set current to regulate the output voltage.
100Ω
2N3906
+
V
6V TO 30V
+
V
+
470Ω
10µF
10kΩ
10kΩ
7
3
+
–
5
1
12MHz Negative Resistance LC Oscillator
g
+
m
4
0.1µF
51Ω
2
6
CFA
R
8
+
V
–
9.1k
F
3
2
7
680Ω
+
–
V
O
1
8
g
+
–
m
1k
R
G
V
51Ω
1.8k
O
6
20Ω
5
CFA
+
160Ω
1000pF
4
10µF
1000pF
+
–
50Ω
V
750Ω
10µF
1k
160Ω
50Ω
4.7µH
30pF
4.3k
330Ω
2N3906
f = 1MHz
= 6dBm (450mV
V
)
RMS
O
2nd HARMONIC = –38dBc
3rd HARMONIC = –54 dBc
2N3904
LT1228 • TA14
FOR 5V OPERATION SHORT OUT 100Ω RESISTOR
0.1µF
10k
In this application the LT1228 is biased for operation from
a single supply. An artificial signal ground at half supply
voltage is generated with two 10k resistors and bypassed
with a capacitor. A capacitor is used in series withRG to set
the DC gain of the current feedback amplifier to unity.
–
V
V
= 10dB
O
AT V = ±5V ALL HARMONICS 40dB DOWN
AT V = ±12V ALL HARMONICS 50dB DOWN
S
S
LT1228 • TA15
This oscillator uses the transconductance amplifier as a
negative resistor to cause oscillation. A negative resistor
results when the positive input of the transconductance
amplifier is driven and the output is returned to it. In this
example a voltage divider is used to lower the signal level at
the positive input for less distortion. The negative resistor
will not DC bias correctly unless the output of the transcon-
ductanceamplifierdrivesaverylowresistance. Hereitsees
an inductor to ground so the gain at DC is zero. The
oscillator needs negative resistance to start and that is
provided by the 4.3k resistor to pin 5. As the output level
rises it turns on the PNP transistor and in turn the NPN
which steals current from the transconductance amplifier
bias input.
The transconductance amplifier is used as a variable resis-
tor to control gain. A variable resistor is formed by driving
theinvertinginputandconnectingtheoutputbacktoit. The
equivalent resistor value is the inverse of the gm. This
works with the 1.8k resistor to make a variable attenuator.
The 1MHz oscillation frequency is set by the Wien bridge
network made up of two 1000pF capacitors and two 160Ω
resistors.
For clean sine wave oscillation, the circuit needs a net gain
of one around the loop. The current feedback amplifier has
a gain of 34 to keep the voltage at the transconductance
amplifier input low. The Wien bridge has an attenuation of
16
LT1228
U
O
TYPICAL APPLICATI S
Filters
Single Pole Low/High/Allpass Filter
R3A
1k
V
IN
3
LOWPASS
INPUT
+
1
g
+
m
R3
120Ω
2
C
6
V
–
CFA
R
OUT
330pF
5
8
–
I
SET
R
G
F
V
1k
1k
IN
HIGHPASS
INPUT
R2A
1k
10
2π
9
I
R + 1
R2
R2 + R2A
R2
120Ω
SET
C
F
f
=
×
×
×
C
R
G
LT1228 • TA16
f
= 10 I FOR THE VALUES SHOWN
C
SET
Allpass Filter Phase Response
0
1mA SET CURRENT
–45
–90
–135
–180
100µA SET CURRENT
10k
100k
1M
10M
FREQUENCY (Hz)
LT1228 • TA17
Using the variable transconductance of the LT1228 to
make variable filters is easy and predictable. The most
straight forward way is to make an integrator by putting a
capacitor at the output of the transconductance amp and
buffering it with the current feedback amplifier. Because
the input bias current of the current feedback amplifier
must be supplied by the transconductance amplifier, the
setcurrentshouldnotbeoperatedbelow10µA.Thislimits
the filters to about a 100:1 tuning range.
values shown give a 100kHz corner frequency for 100µA
set current. The circuit has two inputs, a lowpass filter
input and a highpass filter input. To make a lowpass filter,
ground the highpass input and drive the lowpass input.
Conversely for a highpass filter, ground the lowpass input
and drive the highpass input. If both inputs are driven, the
result is an allpass filter or phase shifter. The allpass has
flat amplitude response and 0° phase shift at low frequen-
cies, going to –180° at high frequencies. The allpass filter
has –90° phase shift at the corner frequency.
The Single Pole circuit realizes a single pole filter with a
corner frequency (fC) proportional to the set current. The
17
LT1228
U
O
TYPICAL APPLICATI S
Voltage Controlled State Variable Filter
+
–
1k
LT1006
10k
2N3906
V
C
100pF
180Ω
51k
5V
3k
–5V
3k
7
3.3k
100Ω
3
2
+
V
5
IN
1
8
g
+
m
6
BANDPASS
OUTPUT
–
CFA
4
18pF
–
–5V
1k
3.3k
3.3k
5
100Ω
5V
7
3
2
+
–
1
8
g
100Ω
+
m
LOWPASS
OUTPUT
6
CFA
4
18pF
3.3k
–
–5V
1k
f
f
f
f
f
= 100kHz AT V = 0V
C
O
O
O
O
O
= 200kHz AT V = 1V
C
= 400kHz AT V = 2V
C
= 800kHz AT V = 3V
C
= 1.6MHz AT V = 4V
LT1228 • TA18
C
The state variable filter has both lowpass and bandpass
outputs. Each LT1228 is configured as a variable integra-
tor whose frequency is set by the attenuators, the capaci-
torsandthesetcurrent. Becausetheintegratorshaveboth
positive and negative inputs, the additional op amp nor-
mally required is not needed. The input attenuators set the
circuit up to handle 3VP–P signals.
best accuracy. If discrete transistors are used, the 51k
resistor should be trimmed to give proper frequency
response with VC equal zero. The circuit generates 100µA
for VC equal zero volts and doubles the current for every
additional volt. The two 3k resistors divide the current
between the two LT1228s. Therefore the set current of
each amplifier goes from 50µA to 800µA for a control
voltage of 0V to 4V. The resulting filter is at 100kHz for VC
equal zero, and changes it one octave/V of control input.
Thesetcurrentisgeneratedwithasimplecircuitthatgives
logarithmic voltage to current control. The two PNP tran-
sistors should be a matched pair in the same package for
18
LT1228
U
O
TYPICAL APPLICATI S
RF AGC Amplifier (Leveling Loop)
15V
10k
RF INPUT
to 1.3V
7
3
2
+
–
0.6V
RMS
RMS
25MHz
1
g
+
100Ω
m
OUTPUT
P–P
300Ω
CFA
2V
5
8
–
4
470Ω
0.01µF
10k
–15V
10k
4pF
10
Ω
0.01µF
15V
10k
10k
100k
4.7k
–
–15V
A3
LT1006
AMPLITUDE
ADJUST
1N4148’s
COUPLE THERMALLY
LT1004
1.2V
+
LT1228 • TA20
–15V
Inverting Amplifier with DC Output Less Than 5mV
+
V
2
3
7
–
+
1
8
g
+
–
m
+
6
5
CFA
V
O
100µF
4
R5
–
V
R
1k
F
V
V
V
= ±5V, R5 = 3.6k
S
S
R
G
= ±15V, R5 = 13.6k
1k
MUST BE LESS THAN
OUT
200mV
FOR LOW OUTPUT OFFSET
P–P
V
IN
BW = 30Hz TO 20MHz
LT1228 • TA21
INCLUDES DC
Amplitude Modulator
5V
4.7µF
+
3
7
+
1
8
g
+
–
m
V
OUT
2
6
5
CFA
0dBm(230mV) AT
MODULATION = 0V
–
CARRIER
INPUT
30mV
4
10k
1k
R
750Ω
F
4.7µF
+
R
–5V
G
750Ω
MODULATION
INPUT ≤ 8V
LT1228 • TA22
P–P
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
19
LT1228
U
PACKAGE DESCRIPTIO
Dimensions in inches (millimeters) unless otherwise noted.
J8 Package
8-Lead Ceramic DIP
0.405
(10.287)
MAX
0.005
(0.127)
MIN
0.200
(5.080)
MAX
0.300 BSC
(0.762 BSC)
6
5
4
8
7
0.015 – 0.060
(0.381 – 1.524)
0.025
(0.635)
RAD TYP
0.220 – 0.310
(5.588 – 7.874)
0.008 – 0.018
0° – 15°
(0.203 – 0.457)
1
2
3
0.045 – 0.068
0.385 ± 0.025
0.125
3.175
MIN
(1.143 – 1.727)
(9.779 ± 0.635)
0.100 ± 0.010
(2.540 ± 0.254)
0.014 – 0.026
CORNER LEADS OPTION
(0.360 – 0.660)
(4 PLCS)
NOTE: LEAD DIMENSIONS APPLY TO SOLDER DIP OR TIN PLATE LEADS.
J8 0293
0.023 – 0.045
(0.584 – 1.143)
HALF LEAD
OPTION
0.045 – 0.068
(1.143 – 1.727)
FULL LEAD
OPTION
N8 Package
8-Lead Plastic DIP
0.130 ± 0.005
0.045 – 0.065
0.300 – 0.320
0.400*
(3.302 ± 0.127)
(1.143 – 1.651)
(7.620 – 8.128)
(10.160)
MAX
8
1
7
6
5
4
0.065
(1.651)
TYP
0.009 – 0.015
(0.229 – 0.381)
0.250 ± 0.010*
(6.350 ± 0.254)
0.125
(3.175)
MIN
0.020
(0.508)
MIN
+0.025
–0.015
0.045 ± 0.015
(1.143 ± 0.381)
0.325
+0.635
8.255
2
3
(
)
–0.381
0.100 ± 0.010
(2.540 ± 0.254)
0.018 ± 0.003
(0.457 ± 0.076)
N8 0594
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTURSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm).
S8 Package
8-Lead Plastic SOIC
0.189 – 0.197*
(4.801 – 5.004)
0.010 – 0.020
(0.254 – 0.508)
7
5
8
6
× 45°
0.053 – 0.069
(1.346 – 1.752)
0.004 – 0.010
(0.101 – 0.254)
0.008 – 0.010
(0.203 – 0.254)
0°– 8° TYP
0.150 – 0.157*
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
0.016 – 0.050
0.406 – 1.270
0.050
(1.270)
BSC
0.014 – 0.019
(0.355 – 0.483)
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006 INCH (0.15mm).
1
3
4
2
SO8 0294
LT/GP 0694 5K REV A • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1994
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7487
20
●
●
(408) 432-1900 FAX: (408) 434-0507 TELEX: 499-3977
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