LT1302CS8-5#PBF [Linear]
LT1302 - Micropower High Output Current Step-Up Adjustable and Fixed 5V DC/DC Converters; Package: SO; Pins: 8; Temperature Range: 0°C to 70°C;型号: | LT1302CS8-5#PBF |
厂家: | Linear |
描述: | LT1302 - Micropower High Output Current Step-Up Adjustable and Fixed 5V DC/DC Converters; Package: SO; Pins: 8; Temperature Range: 0°C to 70°C 开关 光电二极管 |
文件: | 总16页 (文件大小:313K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
LT1302/LT1302-5
Micropower
High Output Current
Step-Up Adjustable and
Fixed 5V DC/DC Converters
U
DESCRIPTIO
EATURE
S
F
The LT®1302/LT1302-5 are micropower step-up DC/DC
converters that maintain high efficiency over a wide
range of output current. They operate from a supply
voltage as low as 2V and feature automatic shifting
between Burst Mode operation at light load, and current
mode operation at heavy load.
■
■
■
■
■
■
5V at 600mA or 12V at 120mA from 2-Cell Supply
200µA Quiescent Current
Logic Controlled Shutdown to 15µA
Low VCESAT Switch: 310mV at 2A Typical
Burst ModeTM Operation at Light Load
Current Mode Operation for Excellent
Line and Load Transient Response
Available in 8-Lead SO or PDIP
The internal low loss NPN power switch can handle
current in excess of 2A and switch at frequencies up to
400kHz. Quiescent current is just 200µA and can be
further reduced to 15µA in shutdown.
■
■
Operates with Supply Voltage as Low as 2V
O U
PPLICATI
S
A
Availablein8-pinPDIPor8-pinSOpackaging,theLT1302/
LT1302-5 have the highest switch current rating of any
similarly packaged switching regulators presently on the
market.
■
■
■
■
■
Notebook and Palmtop Computers
Portable Instruments
Personal Digital Assistants
Cellular Telephones
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
Flash Memory
U
O
TYPICAL APPLICATI
2-Cell to 5V Converter Efficiency
NC
5
L1
6
90
10µH
V
I
T
IN
3
4
7
88
86
84
SW
SHDN
LT1302-5
SHUTDOWN
C3
0.1µF
V
= 3V
IN
8
+
C1
100µF
PGND
GND
1
SENSE
V
C
2 CELLS
D1
V
= 2.5V
IN
82
80
2
V
= 2V
IN
R
C
20k
C
0.01µF
+
78
76
74
72
70
C2
C
100µF
OUTPUT
5V
LT1302 • F01
600mA
1
10
100
1000
C1 = C2 = SANYO OS-CON
L1 = COILTRONICS CTX10-3
COILCRAFT DO3316-103
D1 = MOTOROLA MBRS130LT3
LOAD CURRENT (mA)
LT1302 • TA02
Figure 1. 2-Cell to 5V/600mA DC/DC Converter
1
LT1302/LT1302-5
W W W
U
/O
ABSOLUTE AXI U RATI GS
PACKAGE RDER I FOR ATIO
TOP VIEW
VIN Voltage ............................................................. 10V
SW Voltage ............................................................. 25V
FB Voltage .............................................................. 10V
SHDN Voltage ......................................................... 10V
VC Voltage ................................................................ 4V
IT Voltage.................................................................. 4V
Maximum Power Dissipation ............................ 700mW
Operating Temperature Range .................... 0°C to 70°C
Storage Temperature Range ............... – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
ORDER PART
NUMBER
GND
1
2
3
4
PGND
SW
8
7
6
5
V
C
LT1302CN8
LT1302CS8
LT1302CN8-5
LT1302CS8-5
SHDN
V
IN
(SENSE*)FB
I
T
N8 PACKAGE
8-LEAD PDIP
S8 PACKAGE
8-LEAD PLASTIC SO
*FIXED VERSION
PINS 1 AND 8 ARE INTERNALLY
CONNECTED IN SOIC PACKAGE
S8 PART MARKING
1302
13025
TJMAX = 125°C, θJA = 100°C/W (N8)
JMAX = 125°C, θJA = 80°C/W (S8)
T
Consult factory for Industrial and Military grade parts.
DC ELECTRICAL CHARACTERISTICS
TA = 25°C, VIN = 2.5V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
I
Quiescent Current
V
SHDN
V
SHDN
= 0.5V, V = 1.3V
●
●
200
15
300
25
µA
µA
Q
FB
= 1.8V
V
V
Input Voltage Range
2.0
2.2
1.22
V
V
V
nA
IN
●
●
8
1.26
Feedback Voltage (LT1302)
Feedback Pin Bias Current (LT1302)
Output Sense Voltage (LT1302-5)
Output Ripple Voltage (LT1302-5)
Sense Pin Resistance to Ground (LT1302-5)
Offset Voltage
V = 0.4V
1.24
100
5.05
50
420
15
FB
C
V
FB
= 1V
V = 0.4V
●
4.85
5.25
V
C
V = 0.4V
C
mV
kΩ
mV
mV
V
OS
See Block Diagram
(Note 1)
Comparator Hysteresis
5
Oscillator Frequency
Current Limit Not Asserted (Note 2)
175
160
75
220
265
310
95
kHz
kHZ
%
µs
µs
%/V
mV
mV
µA
A
A
V/V
V
●
●
DC
Maximum Duty Cycle
Switch On Time
Switch Off Time
Output Line Regulation
Switch Saturation Voltage
86
3.9
0.7
0.06
310
t
t
Current Limit Not Asserted
ON
OFF
2 < V < 8V
0.15
400
475
IN
V
CESAT
I
= 2A
SW
●
●
Switch Leakage Current
Switch Current Limit
V
SW
= 5V, Switch Off
0.1
1
2.8
10
V = 0.4V (Burst Mode Operation)
C
V = 1.25V (Full Power) (Note 3)
●
2.0
50
1.8
3.9
C
Error Amplifier Voltage Gain
Shutdown Pin High
Shutdown Pin Low
0.9V ≤ V ≤ 1.2V, ∆V /∆V
75
C
C
FB
V
V
●
●
●
SHDNH
SHDNL
SHDN
0.5
20
V
I
Shutdown Pin Bias Current
V
SHDN
V
SHDN
V
SHDN
= 5V
= 2V
= 0V
8
3
0.1
µA
µA
µA
●
1
I Pin Resistance to Ground
T
3.9
kΩ
The
● denotes specifications which apply over the 0°C to 70°C
Note 2: The LT1302 operates in a variable frequency mode. Switching
frequency depends on load inductance and operating conditions and may
be above specified limits.
temperature range.
Note 1: Hysteresis is specified at DC. Output ripple depends on capacitor
size and ESR.
Note 3: Minimum switch current 100% tested. Maximum switch current
guaranteed by design.
2
LT1302/LT1302-5
U W
TYPICAL PERFORMANCE CHARACTERISTICS
No-Load Quiescent Current
Circuit of Figure 1
Switch Saturation Voltage
Switch Saturation Voltage
600
500
400
300
200
100
0
400
350
300
250
200
150
100
500
450
400
350
300
250
200
150
100
50
T
A
= 25°C
T
A
= 25°C
I
= 2A
SW
0
0
1
2
3
4
–50
0
25
50
75
100
–25
2.0
2.5
3.0
3.5
4.0
4.5
5.0
SWITCH CURRENT (A)
TEMPERATURE (°C)
SUPPLY VOLTAGE (V)
1302 G02
1302 G03
1302 G01
LT1302 Feedback Voltage
LT1302-5 Sense Pin Resistance
Quiescent Current
300
250
200
150
100
50
1.250
1.245
1.240
1.235
1.230
1.225
1.220
1.215
1.210
1.205
1.200
600
500
400
300
200
100
0
V
= 2.5V
IN
SWITCH OFF
0
–50
0
25
50
75
100
–50
0
25
50
75
100
–25
–25
–50
0
25
50
75
100
–25
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
1302 G06
1302 G04
1302 G05
Error Amplifier Offset Voltage
LT1302-5 Output Voltage
Maximum On-Time
5.0
4.5
4.0
3.5
3.0
2.5
2.0
5.100
5.075
5.050
5.025
5.000
4.975
4.950
4.925
4.900
30
25
20
15
10
5
0
–50
0
25
50
75
100
–50
0
25
50
75
100
–25
–25
–50
0
25
50
75
100
–25
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
1302 G09
1302 G08
1302 G07
3
LT1302/LT1302-5
TYPICAL PERFORMANCE CHARACTERISTICS
U W
Shutdown Pin Bias Current
Oscillator Frequency
Maximum Duty Cycle
100
90
80
70
60
50
300
275
250
225
200
175
150
20
18
16
14
12
10
8
T
A
= 25°C
6
4
2
0
–50
0
25
50
75
100
–50
0
25
50
75
100
–25
–25
0
4
6
7
1
2
3
5
8
TEMPERATURE (°C)
TEMPERATURE (°C)
SHUTDOWN VOLTAGE (V)
1302 G10
1302 G11
1302 G12
LT1302-5 Output Voltage vs
Load Current
Maximum Output Power*
Boost Mode
5.20
5.15
5.10
5.05
5.00
4.95
4.90
4.85
4.80
20
16
12
8
V
= 4V
IN
V
IN
= 2.2V
V
IN
= 3V
4
0
0
0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
LOAD CURRENT (A)
0
2
4
6
8
10
0.1
INPUT VOLTAGE (V)
1302 G13
1302 G14
* APPROXIMATE
U
U
U
PI FU CTIO S
IT (Pin5):Normallyleftfloating. Additionofa3.3kresistor
toGNDforcestheLT1302intocurrentmodeatlightloads.
Efficiency drops at light load but increases at medium
loads. See Applications Information section.
GND (Pin 1): Signal Ground. Feedback resistor and 0.1µF
ceramic bypass capacitor from VIN should be connected
directly to this pin.
VC (Pin 2): Frequency Compensation Pin. Connect series
RC to GND. Keep trace short.
VIN (Pin6):SupplyPin.Mustbebypassedwith:(1)a0.1µF
ceramictoGND, and(2)alargevalueelectrolytictoPGND.
When VIN is greater than 5V, a low value resistor (2Ω to
10Ω) is recommended to isolate the VIN pin from input
supply noise.
SHDN (Pin 3): Shutdown. Pull high to effect shutdown; tie
to ground for normal operation.
FB/Sense (Pin 4): Feedback/Sense. On the LT1302 this
pin connects to CMP1 input. On the LT1302-5 this pin
connects to the output resistor string.
4
LT1302/LT1302-5
U
U
U
PI FU CTIO S
SW (Pin 7): Switch Pin. Connect inductor and diode here.
Keep layout short and direct.
and 8 are thermally connected to the die. One square inch
of PCB copper provides an adequate heat sink for the
device.
PGND (Pin 8): Power Ground. Pins 8 and 1 should be
connected under the package. In the SO package, pins 1
W
BLOCK DIAGRA SM
D1
L1
V
IN
V
OUT
+
+
C2
0.1µF
C1
C3
6
7
V
SW
IN
36mV
R4
+
–
1.75Ω
R5
730Ω
A2
CMP1
1.24V
REFERENCE
OFF
+
–
ENABLE
Q3
220kHz
OSCILLATOR
Q4
160X
A3
R1
R2
HYSTERETIC
DRIVER
FB
COMPARATOR
2
µA
4
3
V
OS
C5
100pF
V
IN
15mV
Q5
V
IN
–
Q1
Q2
A1
BIAS
SHDN
SHUTDOWN
+
ERROR
AMPLIFIER
300Ω
3.6k
GND
V
C
I
PGND
T
1
2
5
8
R3
22k
1302 F02
C4
0.01µF
Figure 2. LT1302 Block Diagram
5
LT1302/LT1302-5
W
BLOCK DIAGRA SM
SENSE
4
V
IN
6
SW
7
36mV
R4
1.75Ω
+
–
R5
A2
730Ω
R1
315k
CMP1
1.24V
OFF
220kHz
OSCILLATOR
+
–
REFERENCE
ENABLE
Q3
Q4
A3
160X
HYSTERETIC
COMPARATOR
DRIVER
2
µA
V
OS
V
IN
15mV
Q5
R2
105k
V
IN
–
Q1
Q2
A1
BIAS
SHDN
3
SHUTDOWN
+
ERROR
AMPLIFIER
300Ω
3.6k
1
2
5
I
8
1302 F03
GND
V
C
PGND
T
Figure 3. LT1302-5 Block Diagram
U
OPERATIO
CMP1’s hysteresis (about 5mV) CMP1 turns the oscilla-
tor off. In this mode, peak switch current is limited to
approximately 1A by A2, Q2, and Q3. Q2’s current, set at
34µA, flows through R5, causing A2’s negative input to
be 25mV lower than VIN. This node must fall more than
36mV below VIN for A2 to trip and turn off the oscillator.
The remaining 11mV is generated by Q3’s current flow-
ing through R4. Emitter-area scaling sets Q3’s collector
current to 0.625% of switch Q4’s current. When Q4’s
current is 1A, Q3’s current is 6.25mA, creating an 11mV
drop across R4 which, added to R5’s 25mV drop, is
enough to trip A2.
The LT1302’s operation can best be understood by
examining the block diagram in Figure 2. The LT1302
operates in one of two modes, depending on load. With
light loads, comparator CMP1 controls the output; with
heavy loads, control is passed to error amplifier A1.
Burst Mode operation consists of monitoring the FB pin
voltage with hysteretic comparator CMP1. When the FB
voltage, related to the output voltage by external attenu-
ator R1 and R2, falls below the 1.24V reference voltage,
the oscillator is enabled. Switch Q4 alternately turns on,
causing current buildup in inductor L1, then turns off,
allowing the built-up current to flow into output capaci-
tor C3 via D1. As the output voltage increases, so does
the FB voltage; when it exceeds the reference plus
When the output load is increased to the point where the
1A peak current cannot support the output voltage,
6
LT1302/LT1302-5
U
OPERATIO
CMP1 stays on and the peak switch current is regulated
by the voltage on the VC pin (A1’s output). VC drives the
base of Q1. As the VC voltage rises, Q2 conducts less
current, resulting in less drop across R5. Q4’s peak
current must then increase in order for A2 to trip. This
currentmodecontrolresultsingoodstabilityandimmu-
nity to input voltage variations. Because this is a linear,
closed-loopsystem,frequencycompensationisrequired.
A series RC from VC to ground provides the necessary
pole-zero combination.
The LT1302-5 incorporates feedback resistors R1 and
R2 into the device. Output voltage is set at 5.05V in Burst
Mode, dropping to 4.97V in current mode.
U
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APPLICATIONS INFORMATION
Inductor Selection
t
×
ON
V − V
(
)
IN
SW
L ≤
Inductors used with the LT1302 must fulfill two require-
ments. First, the inductor must be able to handle current
of 2.5A to 3A without runaway saturation. Rod or drum
coreunitsusuallysaturategraduallyanditisacceptableto
exceed manufacturers’ published saturation currents by
20% or so. Second, it should have low DCR, under 0.05Ω
so that copper loss is kept low. Inductance value is not
critical. Generally, for low voltage inputs down to 2V, a
10µHinductorisrecommended(suchasCoilcraftDO3316-
103). For inputs above 4V to 5V use a 22µH unit (such as
CoilcraftDO3316-223). Switchingfrequencycanreachup
to 400kHz so the core material should be able to handle
high frequency without loss. Ferrite or molypermalloy
cores are a better choice than powdered iron. If EMI is a
concernatoroidalinductorissuggested,suchasCoiltronics
CTX20-4.
2A
With the 2V input a value of 3.3µH is acceptable. Since the
inductance is so low, usually a smaller core size can be
used. Efficiency will not be as high as for the continuous
case since peak currents will necessarily be higher.
Table1listsinductorsuppliersalongwithappropriatepart
numbers.
Table 1. Recommended Inductors
VENDOR
Coilcraft
PART NO.
VALUE(
µH)
PHONE NO.
(708) 639-6400
DO3316-103
DO3316-153
DO3316-223
10
15
22
10
20
10
20
10
15
22
Coiltronics
Dale
CTX10-2
CTX20-4
(407) 241-7876
(605) 665-9301
(708) 956-0666
LPT4545-100LA
LPT4545-200LA
CD105-100
CD105-150
CDR125-220
For a boost converter, duty cycle can be calculated by the
following formula:
Sumida
V
IN
DC = 1–
VOUT
Capacitor Selection
AspecialsituationexistswheretheVOUT/VIN differentialis
high, such as a 2V-to-12V converter. The required duty
cycle is higher than the LT1302 can provide, so the
converter must be designed for discontinuous operation.
This means that inductor current goes to zero during the
switch off-time. In the 2V-to-12V case, inductance must
be low enough so that current in the inductor can reach
2A in a single cycle. Inductor value can be defined by:
The output capacitor should have low ESR for proper
performance. A high ESR capacitor can result in “mode-
hopping” between current mode and Burst Mode at high
load currents because the output voltage will increase by
I
SW × ESR when the inductor current is flowing into the
diode. Figure 4 shows output voltage of an LT1302-5
boostconverterwithtwo220µFAVXTPScapacitorsatthe
output. Ripple voltage at a 510mA load is about 30mVP-P
7
LT1302/LT1302-5
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APPLICATIONS INFORMATION
Input Capacitor
and there is no low frequency component. The total ESR
is under 0.03Ω. If a single 100µF aluminum electrolytic
capacitor is used instead, the converter mode-hops be-
tween current mode and Burst Mode due to high ESR,
causing the voltage comparator to trip as shown in Figure
5. The ripple voltage is now over 500mVP-P and contains
a low frequency component. Maximum allowable output
capacitor ESR can be calculated by the following formula:
The input supply should be decoupled with a good quality
electrolytic capacitor close to the LT1302 to provide a
stable input supply. Long leads or traces from power
source to the switcher can have considerable impedance
at the LT1302’s switching frequency. The input capacitor
provides a low impedance at high frequency. A 0.1µF
ceramiccapacitorisrequiredrightattheVIN pin. Whenthe
input voltage can be above 5V, a 10Ω/1µF decoupling
network for VIN is recommended as detailed in Figure 6.
This network is also recommended when driving a trans-
former.
VOS × VOUT
VREF ×1A
ESRMAX
=
where,
VOS = 15mV
V
> 5V
IN
V
REF = 1.24V
10Ω
V
IN
SW
LT1302
+
47µF
TO
100µF
+
1µF
• • •
VOUT
50mV/DIV
PGND
GND
AC COUPLED
1302 F06
510mA
ILOAD
10mA
Figure 6. A 10Ω/1µF Decoupling Network at VIN Is
Recommended When Input Voltage Is Above 5V
500µs/DIV
1302 F04
Figure 4. Low ESR Output Capacitor Results in Stable
Operation. Ripple Voltage is Under 30mVP-P
Table 2 lists capacitor vendors along with device types.
Table 2. Recommended Capacitors
VENDOR
AVX
Sanyo
SERIES
TPS
OS-CON
595D
TYPE
PHONE NO.
Surface Mount
Through Hole
Surface Mount
(803) 448-9411
(619) 661-6835
(603) 224-1961
VOUT
200mV/DIV
AC COUPLED
Sprague
510mA
Diode Selection
ILOAD
10mA
A 2A Schottky diode such as Motorola MBRS130LT3 has
been found to be the best available. Other choices include
1N5821 or MBRS130T3. Do not use “general purpose”
diodes such as 1N4001. They are much too slow for use
in switching regulator applications.
500µs/DIV
1302 F05
Figure 5. Inexpensive Electrolytic Capacitor Has High
ESR, Resulting in Mode-Hop, Ripple Voltage Amplitude Is
Over 500mVP-P and Includes Low Frequency Component
8
LT1302/LT1302-5
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APPLICATIONS INFORMATION
Frequency Compensation
behavior in the 4th graticule is the result of the LT1302’s
BurstModecomparatorturningoffallswitching asoutput
voltage rises above its threshold.
Obtaining proper RC values for the frequency compensa-
tion network is largely an empirical procedure, since
variations in input and output voltage, topology, capacitor
ESR and inductance make a simple formula elusive. As an
example, considerthecaseofa2.5Vto5Vboostconverter
supplying 500mA. To determine optimum compensation,
the circuit is built and a transient load is applied to the
circuit. Figure 7 shows the setup.
In Figure 7c, the 0.1µF capacitor has been replaced by a
0.01µF unit. Undershoot is less but the response is still
underdamped. Figure 7d shows the results of the 0.1µF
capacitor and a 10k resistor in series. Now some amount
of damping is observed, and behavior is more controlled.
Figure 7e details response with a 0.01µF/10k series net-
work. Undershoot is down to around 100mV, or 2%. A
slight underdamping is still noticeable.
In Figure 7a, the VC pin is simply left floating. Although
output voltage is maintained and transient response is
good, switch current rises instantaneously to the internal
current limit upon application of load. This is an undesir-
able situation as it places maximum stress on the switch
and the other power components. Additionally, efficiency
iswelldownfromitsoptimalvalue.Next,a0.1µFcapacitor
is connected with no resistor. Figure 7b details response.
Although the circuit eventually stabilizes, the loop is quite
underdamped. Initial output “sag” exceeds 5%. Aberrant
Finally, a 0.01µF/24k series network results in the re-
sponse shown in Figure 7f. This has optimal damping,
undershoot less than 100mV and settles in less than 1ms.
The VC pin is sensitive to high frequency noise. Some
layouts may inject enough noise to modulate peak switch
current at 1/2 the switching frequency. A small capacitor
connected from VC to ground will eliminate this. Do not
exceed 1/10 of the compensation capacitor value.
V
IN
2.5V
NC
L1
10µH
V
I
T
SHDN
IN
+
C1
330µF
SW
LT1302-5
D1
0.1µF
PGND
GND
SENSE
V
C
10Ω
2W
500Ω
+
+
C2
220µF
C3
220µF
R
PULSE
GENERATOR
C
MTP3055EL
50Ω
C1, C2, C3 = AVX TPS SERIES
D1 = MOTOROLA MBRS130LT3
L1 = COILCRAFT DO3316-103K
1302 F07
Figure 7. Boost Converter with Simulated Load
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
510mA
ILOAD
510mA
ILOAD
10mA
10mA
2ms/DIV
1302 F07b
2ms/DIV
1302 F07a
Figure 7b. 0.1µF from VC to Ground.
Better, but More Improvement Needed
Figure 7a. VC Pin Left Unconnected. Output Shows
Low Frequency Components Under Load
9
LT1302/LT1302-5
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APPLICATIONS INFORMATION
IT Pin
VOUT
100mV/DIV
AC COUPLED
The IT pin is used to disable Burst Mode, forcing the
LT1302 to operate in current mode even at light load. To
disable Burst Mode, 3.3k resistor R1 is connected from IT
to gound. More conservative frequency compensation
must be used when in this mode. A 0.1µF capacitor and
4.7k resistor from VC to ground has been found to be
adequate. Low frequency Burst Mode ripple can be
reduced or eliminated using this technique in many appli-
cations.
510mA
ILOAD
10mA
2ms/DIV
1302 F07c
Figure 7c. 0.01µF from VC to Ground.
Underdamped Response Requires Series R
To illustrate, the transient load response of Figure 8’s
circuit is pictured without and with R1. Figure 8a shows
output voltage and inductor current without the resistor.
Note the 6kHz burst rate when the converter is delivering
25mA. By adding the 3.3k resistor, the low frequency
bursting is eliminated, as shown in Figure 8b. This feature
is useful in systems that contain audio circuitry. At very
light or zero load, switching frequency drops and eventu-
VOUT
100mV/DIV
AC COUPLED
510mA
ILOAD
10mA
2ms/DIV
1302 F07d
Figure 7d. 0.1µF with 10k Series RC.
Classic Overdamped Response
V
IN
2.5V
10µH
V
SENSE
IN
SW
+
C1
330µF
VOUT
100mV/DIV
AC COUPLED
LT1302-5
MBRS130LT3
0.1µF
PGND
GND
V
C
I
T
+
+
220µF
10V
220µF
10V
4.7k
0.1µF
R1
3.3k
510mA
ILOAD
10mA
1302 F08
V
2ms/DIV
1302 F07e
OUT
5V
600mA
Figure 7e. 0.01µF, 10k Series RC Shows Good
Transient Response. Slight Underdamping
Still Noticeable
Figure 8. Addition of R1 Eliminates Low Frequency
Output Ripple in This 2.5V to 5V Boost Converter
VOUT
VOUT
100mV/DIV
100mV/DIV
AC COUPLED
AC COUPLED
INDUCTOR
CURRENT
1A/DIV
510mA
ILOAD
10mA
525mA
ILOAD
25mA
1ms/DIV
1302 F08a
2ms/DIV
1302 F07f
Figure 7f. 0.01µF, 24k Series RC
Results in Optimum Response
Figure 8a. IT Pin Floating. Note 6kHz Burst Rate at
ILOAD = 25mA. 0.1µF/4.7k Compensation Network
Causes 220mV Undershoot
10
LT1302/LT1302-5
U
W U U
APPLICATIONS INFORMATION
ally reaches audio frequencies, but at a much lighter load
than without the IT feature. At some input voltage/load
current combinations, some residual bursting may occur
at frequencies out of the audio band.
The IT pin cannot be used as a soft-start. Large capacitors
connected to the pin will cause erratic operation. If oper-
ating the device in Burst Mode, let the pin float. Keep high
dV/dt signals away from the pin.
Figure8cdetailsefficiencywithandwithouttheadditionof
R1. Burst Mode operation keeps efficiency high at light
load with IT floating. Efficiency falls off at light load with
R1addedbecausetheLT1302cannottransitionintoBurst
Mode.
VOUT
100mV/DIV
AC COUPLED
INDUCTOR
CURRENT
1A/DIV
Layout
525mA
ILOAD
25mA
1ms/DIV
1302 F08b
The high speed, high current switching associated with
the LT1302 mandates careful attention to layout. Follow
thesuggestedcomponentplacementinFigure9forproper
operation. High current functions are separated by the
package from sensitive control functions. Feedback resis-
tors R1 and R2 should be close to the feedback pin (pin4).
Noisecaneasilybecoupledintothispinifcareisnottaken.
A small capacitor (100pF to 200pF) from FB to ground
provides a high frequency bypass. If the LT1302 is oper-
ated off a three-cell or higher input, R3 (2Ω to 10Ω) in
series with VIN is recommended. This isolates the device
from noise spikes on the input supply. Do not put in R3 if
the device must operate from a 2V input, as input current
will cause the voltage at the LT1302’s VIN pin to go below
2V. The 0.1µF ceramic bypass capacitor C3 (use X7R, not
Z5U) should be mounted as close as possible to the
package. When R3 is used, C3 should be a 1µF tantalum
unit. Grounding should be segregated as illustrated. C3’s
ground trace should not carry switch current. Run a
Figure 8b. 3.3k Resistor from IT Pin to Ground Forces
LT1302 into Current Mode Regardless of Load. Audio
Frequency Component Eliminated
90
I
T
FLOATING
80
70
60
50
40
30
3.3k I TO GND
T
1
10
100
1000
OUTPUT CURRENT (mA)
1302 F08c
Figure 8c. 3.3k Resistor for IT to Ground Increases
Efficiency at Moderate Load, Decreases at Light Load
V
IN
R2
5
6
7
8
4
C3
R3
2Ω
L1
3
2
1
R1
200pF
R
SHUTDOWN
LT1302
C
+
C1
C
C
D1
C2
V
OUT
GND (BATTERY AND LOAD RETURN)
1302 F09
Figure 9. Suggested Component Placement for LT1302
11
LT1302/LT1302-5
U
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APPLICATIONS INFORMATION
separate ground trace up under the package as shown.
The battery and load return should go to the power side of
the ground copper.
Table 3. S8 Package, 8-Lead Plastic SO
COPPER AREA
THERMAL RESISTANCE
TOPSIDE*
BACKSIDE BOARD AREA (JUNCTION-TO-AMBIENT)
2500 sq. mm 2500 sq. mm 2500 sq. mm
1000 sq. mm 2500 sq. mm 2500 sq. mm
60°C/W
62°C/W
65°C/W
69°C/W
73°C/W
80°C/W
83°C/W
Thermal Considerations
The LT1302 contains a thermal shutdown feature which
protects against excessive internal (junction) tempera-
ture. If the junction temperature of the device exceeds the
protection threshold, the device will begin cycling be-
tween normal operation and an off state. The cycling is not
harmful to the part. The thermal cycling occurs at a slow
rate, typically10mstoseveralseconds, whichdependson
the power dissipation and the thermal time constants of
the package and heat sinking. Raising the ambient tem-
perature until the device begins thermal shutdown gives a
good indication of how much margin there is in the
thermal design.
225 sq. mm
100 sq. mm
100 sq. mm
100 sq. mm
100 sq. mm
2500 sq. mm 2500 sq. mm
2500 sq. mm 2500 sq. mm
1000 sq. mm 2500 sq. mm
225 sq. mm 2500 sq. mm
100 sq. mm 2500 sq. mm
* Pins 1 and 8 attached to topside copper
N8 Package, 8-Lead DIP:
Thermal Resistance (Junction-to-Ambient) = 100°C/W
Calculating Temperature Rise
Power dissipation internal to the LT1302 in a boost
regulator configuration is approximately equal to:
For surface mount devices heat sinking is accomplished
by using the heat spreading capabilities of the PC board
and its copper traces. Experiments have shown that the
heat spreading copper layer does not need to be electri-
cally connected to the tab of the device. The PCB material
can be very effective at transmitting heat between the pad
area attached to pins 1 and 8 of the device, and a ground
or power plane layer either inside or on the opposite side
of the board. Although the actual thermal resistance of the
PCB material is high, the length/area ratio of the thermal
resistance between the layer is small. Copper board stiff-
eners and plated through holes can also be used to spread
the heat generated by the device.
2
VOUT + VD
IOUT OUT
VOUT + VD
IOUT OUTR
P = I2OUT
R
−
D
V
R
V
VIN −
VIN −
VIN
VIN
IOUT
V
+ VD − VIN
(
)
OUT
+
27
The first term in this equation is due to switch “on-
resistance.” The second term is from the switch driver. R
is switch resistance, typically 0.15Ω. VD is the diode
forward drop.
Table 3 lists thermal resistance for the SO package.
Measured values of thermal resistance for several differ-
ent board sizes and copper areas are listed for each
surface mount package. All measurements were taken in
The temperature rise can be calculated from:
∆T = PD × θJA
still air on 3/32" FR-4 board with 1oz copper. This data can
where:
be used as a rough guideline in estimating thermal resis-
tance. The thermal resistance for each application will be
affectedbythermalinteractionswithothercomponentsas
well as board size and shape.
∆T = Temperature Rise
PD = Device Power Dissipation
θJA = Thermal Resistance (Junction-to-Ambient)
12
LT1302/LT1302-5
U
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APPLICATIONS INFORMATION
As an example, consider a boost converter with the
following specifications:
VIN = 3V
VOUT = 6V
IOUT = 700mA
Total power loss in the LT1302, assuming R = 0.15Ω and
VD = 0.45V, is:
2
0.7 6 + 0.45 − 3
2
) (
(
)(
)
6 + 0.45
0.7× 6 × 0.15
6 + 0.45
0.7× 6 × 0.15
3
P = 700mA 0.15Ω
−
+
(
)
D
27
3 −
3 −
3
= 223mW + 89mW = 312mW
Using the CS8 package with 100 sq. mm topside and
backside heat sinking:
∆T = (312mW)(84°C/W) = 25.9°C rise
With the N8 package:
∆T = 31.2°C
At a 70°C ambient, die temperature would be 101.2°C.
13
LT1302/LT1302-5
U
TYPICAL APPLICATIONS
Single Cell to 5V/150mA Converter
5V/150mA
OUTPUT
L1
3.3µH
D1
R1
301k
220Ω
10Ω
100k
1%
2N3906
1.5V
CELL
(169k FOR 3.3V)
100k
100k
I
V
V
IN
SW
SHDN
FB
L
IN
SET
SW1
LT1073
LT1302
56.2k
1%
A
O
I
T
FB
GND
V
C
GND
SW2
PGND
100pF
20k
4.99k
1%
+
+
C1
47µF
C2
220µF
0.1µF
0.01µF
36.5k
1%
L1 = COILCRAFT DO3316-332
D1 = MOTOROLA MBRS130LT3
C1 = AVX TPSD476M016R0150
C2 = AVX TPSE227M010R0100
COILCRAFT (708) 639-2361
1302 TA03
2V to 12V/120mA Converter
NC
5
L1
3.3µH
6
V
I
T
IN
3
4
7
8
SW
SHDN
SHUTDOWN
C3
0.1µF
LT1302
+
C1
100µF
PGND
GND
1
FB
V
C
2 CELLS
D1
2
R1
100k
1%
R2
866k
1%
R
C
100pF
+
+
20k
C2
C2
33µF
33µF
C
C
0.02µF
OUTPUT
12V
LT1302 • TA04
120mA
C1 = AVX TPSD107M010R0100
C2 = AVX TPSD336M025R0200
D1 = MOTOROLA MBRS130LT3
L1 = COILCRAFT DO3316-332
14
LT1302/LT1302-5
U
TYPICAL APPLICATIONS
3 Cell to 3.3V Buck-Boost Converter with Auxiliary 12V Regulated Output
V
IN
2.5V-8V
7
6
10Ω
C3
+
SHUTDOWN
47µF
T1D
T1E
5
16V
4
SHDN
FB
V
IN
SW
D2
13V
LT1302
100k
1%
0.1µF
I
T
2
V
C
C1
+
GND
PGND
100µF
T1B
16V
IN
12V
OUT
ADJ
9
+
120mA
22µF
25V
D1
330k
1%
169k
1%
24k
LT1121
200pF
SHDN
+
3
1
T1A
10
GND
3.3µF
C2
+
4700pF
330µF
T1C
150k
1%
6.3V
8
1302 TA05
3.3V OUTPUT
400mA
T1 = DALE LPE-6562-A069, 1:3:1:1:1 TURNS RATIO, 10µH PRIMARY. DALE (605) 665-9301
D1, D2 = MOTOROLA MBRS130LT3
C1 = AVX TPSE107016R0100
C2 = AVX TPSE337006R0100
C3 = AVX TPSD476016R0150
2 Li-Ion Cell to 5.8V/600mA DC/DC Converter
C2
220µF
L1
10V
22µH
+
V
IN
4V TO 9V
10Ω
L2
MBRS130LT3
22µH
365k
1%
SW
FB
V
T
V
5.8V
600mA
IN
OUT
I
+
+
C1
100µF
16V
1µF
LT1302
+
C3
220µF
10V
SHDN
GND
V
C
100k
1%
PGND
20k
10nF
SHUTDOWN
V
OUT
1302 TA07
DUTY CYCLE =
L1, L2
=
=
=
COILCRAFT DO3316-223
AVX TPSE107016R0100
AVX TPSE227010R0100
V
+ V
OUT
IN
PEAK SWITCH VOLTAGE = V + V
C1
C2, C3
IN
OUT
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
15
LT1302/LT1302-5
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
N8 Package
8-Lead Plastic DIP
0.400*
(10.160)
MAX
8
7
6
3
5
4
0.255 ± 0.015*
(6.477 ± 0.381)
1
2
0.130 ± 0.005
0.300 – 0.325
0.045 – 0.065
(3.302 ± 0.127)
(1.143 – 1.651)
(7.620 – 8.255)
0.065
(1.651)
TYP
0.009 – 0.015
(0.229 – 0.381)
0.125
0.015
(0.380)
MIN
(3.175)
MIN
+0.025
0.045 ± 0.015
(1.143 ± 0.381)
0.325
–0.015
+0.635
8.255
(
)
–0.381
0.100 ± 0.010
(2.540 ± 0.254)
0.018 ± 0.003
(0.457 ± 0.076)
N8 0694
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTURSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm).
S8 Package
8-Lead Plastic SOIC
0.189 – 0.197*
(4.801 – 5.004)
7
5
8
6
0.150 – 0.157*
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
3
4
2
0.010 – 0.020
(0.254 – 0.508)
× 45°
0.053 – 0.069
(1.346 – 1.752)
0.004 – 0.010
(0.101 – 0.254)
0.008 – 0.010
(0.203 – 0.254)
0°– 8° TYP
0.016 – 0.050
0.406 – 1.270
0.050
(1.270)
BSC
0.014 – 0.019
(0.355 – 0.483)
SO8 0294
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006 INCH (0.15mm).
LT/GP 0295 10K • PRINTED IN USA
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7487
16
●
●
LINEAR TECHNOLOGY CORPORATION 1995
(408) 432-1900 FAX: (408) 434-0507 TELEX: 499-3977
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