LT1306_技术文档

LT1306

Synchronous, Fixed Frequency

Step-Up DC/DC Converter

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FEATURES

DESCRIPTIO

■

Output Disconnected from Input During Shutdown

The LT^®1306 is a fully integrated, fixed frequency syn-

chronous boost converter capable of generating 5V at 1A

from a Li-Ion cell. The device contains both the main

power switch and synchronous rectifier on chip and

automatically disconnects the output from the input in

shutdown, eliminating the need for external load discon-

nect circuitry. Additionally, the output remains regulated

when V_INexceeds V_OUT, allowing difficult step-up/step-

down converter functions to be easily realized using a

single inductor.

■

Output Voltage Remains Regulated

When V_IN> V_OUT

■

Controlled Input Current During Start-Up

■

300kHz Current Mode PWM Operation

■

Can Be Externally Synchronized

■

Internal 2A Switches

■

Operates with V_INas Low as 1.8V

■

Automatic Burst Mode Operation at Light Loads

■

Quiescent Current: 160µA

Shutdown Current: 9µA Typ

■

The internal 300kHz oscillator of the LT1306 can be easily

synchronizedtoanexternalclockfrom425kHzto500kHz.

This allows switching harmonics to be tightly controlled

and eliminates any beat frequencies that may result from

amultifrequencysystem.TheLT1306automaticallyshifts

into power saving Burst Mode^TMoperation at light loads.

At heavy loads the LT1306 operates in fixed frequency

current mode. No-load quiescent current is 160µA and

reduces to 9µA in shutdown mode.

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APPLICATIO S

■

Satellite Phones

■

Portable Instruments

■

Personal Digital Assistants

Palmtop Computers

■

, LTC and LT are registered trademarks of Linear Technology Corporation.

The LT1306 is available in an SO-8 package.

Burst Mode is a trademark of Linear Technology Corporation.

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TYPICAL APPLICATION

D1

Efficiency

90

C1

1µF

L1

10µH

1-CELL

Li-Ion

85

80

75

70

65

60

V

IN

= 4.2V

V

SW CAP

OUT

IN

V

= 3.6V

IN

5V

1A

S/S

R1

V

IN

= 2.6V

LT1306

+

768k

C

IN1

22µF

C

IN2

0.1µF

FB

+

V

GND

C

O1

O2

220µF

1µF

V

= 5V

O

R3

R2

249k

C

: AVX TAJC226M010

IN1

L1 = 10µH

C

P

118k

C

Z

68nF

: AVX TPSE227M010R0100

O1

(FIGURE 1)

68pF

, C : CERAMIC

IN1 O2

C1: AVX TAJA105K020

D1: MMBD914LT1

L1: CTX10-2

1

10

100

1000

LOAD CURRENT (mA)

1306 F01

1306 TA01

Figure 1. Single Li-Ion Cell to 5V Converter

1

LT1306

W W U W

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ABSOLUTE MAXIMUM RATINGS

PACKAGE/ORDER INFORMATION

(Note 1)

V_INVoltage ............................................................. 10V

S/S Voltage ............................................................... 7V

FB Voltage .............................................................. 10V

ORDER PART

NUMBER

TOP VIEW

1

2

3

4

8

7

6

5

V

S/S

C

LT1306ES8

V

_OUTVoltage.......................................................... 5.5V

FB

V

IN

Junction Temperature.......................................... 125°C

Operating Temperature Range (Note 2) .. – 40°C to 85°C

Storage Temperature Range ................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec).................. 300°C

V

CAP

SW

OUT

GND

S8 PART MARKING

1306

S8 PACKAGE

8-LEAD PLASTIC SO

T_JMAX= 125°C, θ_JA= 90°C/ W

Consult factory for Industrial and Military grade parts.

ELECTRICAL CHARACTERISTICS

The ● denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at T_A= 25°C. V_IN= 2.5V, V_S/S= V_IN, V_Copen unless otherwise noted.

PARAMETER

CONDITIONS

MIN

TYP

1.24

0.002

10

MAX

1.26

0.1

25

UNITS

V

Reference Voltage

Measured at the FB Pin

●

1.22

Reference Line Regulation

FB Input Bias Current

1.8V ≤ V ≤ 7V

%/V

nA

IN

V

FB

= V

REF

–1

Error Amplifier Transconductance

Error Amplifier Output Source Current

Error Amplifier Output Sink Current

Error Amplifier Output Clamp Voltage

∆I = ±0.2µA

80

5

150

7.5

220

11

µΩ

V

FB

V

FB

V

FB

= 1V, V = 0.8V

µA

V

C

= 1.5V, V = 0.8V

5

7.5

11

C

= 1V

1.18

1.55

1.28

1.38

1.8

15

V

IN

Undervoltage Lockout Threshold

V

Idle Mode Output Leakage Current

Output Source Current in Shutdown

Switching Frequency

V

= 1.5V, V = 5.5V, V = 1.7V

●

6

µA

FB

OUT

SW

= 0V, V = V = 7V, V = 7.2V, V = 0V

–3

OUT

IN

SW

CAP

S/S

1.8V ≤ V ≤ 7V, 0°C ≤ T ≤ 85°C

260

225

310

305

415

390

kHz

IN

A

1.8V ≤ V ≤ 7V, T = –40°C

IN

A

Maximum Duty Cycle

V

FB

V

FB

= 1V, 0°C ≤ T ≤ 85°C

80

65

90

80

%

A

= 1V, T = –40°C

A

Switch Current Limit

Duty Cycle = 0.1 (Note 3)

Duty Cycle = 0.8 (Note 3)

2.3

2.0

A

Burst Mode Operation Switch Current Limit

250

0.45

0.49

mA

V

Switch V

I

= 2A

0.575

0.675

CESAT

SW

Rectifier V

V

CESAT

Stepdown Mode Rectifier Voltage

V

OUT

V

OUT

= 0V, I = 1A

0.3 + V

1.3

0.7 + V

1.8

V

SW

IN

= 2.2V, I = 1A

SW

Switch and Rectifier Leakage Current

V

OUT

= 0V, V = V = 7V, V = 7.2V, V = 0V

●

0.1

20

µA

IN

SW

CAP

S/S

2

LT1306

ELECTRICAL CHARACTERISTICS

The ● denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at T_A= 25°C. V_IN= 2.5V, V_S/S= V_IN, V_Copen unless otherwise noted.

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

S/S Pin Current

V

S/S

V

S/S

= V

= 0V

6

–3

µA

IN

Shutdown Pin Input High Voltage

Shutdown Pin Input Low Voltage

1.2

V

0.45

50

Shutdown Delay

12

20

µs

kHz

mA

µA

V

Synchronization Frequency Range

Operating Supply Current

425

500

8

4.5

160

9

Quiescent Supply Current

V

= V , V = 1.5V

●

250

16

S/S

IN FB

Shutdown Supply Current

CAP Pin Leakage Current

= 0V

S/S

= V

= 7V, V = 2.5V, I = 0

10

IN

CAP

S/S

SW

Output Boost-to-Stepdown Threshold

Output Stepdown-to-Boost Threshold

V

IN

V

– 0.1

V

IN

Note 1: Absolute Maximum Ratings are those values beyond which the life

to the device may be impaired.

temperature range are assured by design, characterization and correlation

with statistical process controls.

Note 2: The LT1306E is guaranteed to meet performance specifications

from 0°C to 70°C. Specifications over the –40°C to 85°C operating

Note 3: Switch current limit guaranteed by design/correlation to static

tests.

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TYPICAL PERFORMANCE CHARACTERISTICS

Maximum Load Current vs

Input Voltage

Reference Voltage vs

Temperature

S/S Pin Current vs S/S Pin Voltage

1.5

1.0

0.5

1.239

1.238

1.237

1.236

1.235

1.234

1.233

1.232

1.231

5

4

T

= –40°C

A

T

V

= 5V

O

= 25°C

A

V

= 3.3V

O

3

2

T

= 85°C

A

1

0

–1

–2

–3

–4

–5

L = 10µH

= 125°C

T

= 25°C

= 50°C

A

T

J

0

1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0

(V)

0

1

2

3

4

5

–20

0

20

40

60

100

–40

80

V

S/S

(V)

TEMPERATURE (°C)

IN

1306 • G01

1306 • G03

1306 • G02

3

LT1306

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TYPICAL PERFORMANCE CHARACTERISTICS

Shutdown Supply Current vs

Input Voltage

Idle-Mode Supply Current vs

Temperature

S/S Pin Current vs Temperature

40

35

30

25

20

15

10

5

155

150

145

140

135

5.0

2.5

0

V

= 2.5V

T

= 25°C

S/S

A

T

= 85°C

A

T

= –40°C

A

V

= 0V

0

S/S

–2.5

0

4

6

8

10

12

–40 –20

0

20

40

60

80 100

2

– 40

20

40

60

80

100

–20

INPUT VOLTAGE (V)

TEMPERATURE (°C)

1306 • G04

1306 • G06

1306 • G05

Oscillator Frequency Line

Regulation

Frequency vs Temperature

Maximum Duty Ratio

315

310

305

300

295

290

285

280

275

270

265

95

90

85

80

75

70

65

60

320

315

310

305

300

V

= 2.5V

IN

40

TEMPERATURE (°C)

80

100

1

2

3

4

5

10

–40 –20

0

20

60

0

6

7

8

9

40

TEMPERATURE (°C)

100

–40

0

20

60

80

–20

V

(V)

IN

1306 • G09

1306 • G07

1306 • G08

Switch Saturation Voltage

vs Current

Maximum Allowable Rise Time of

Synchronizing Pulse

Current Limit vs Duty Cycle

600

500

400

300

200

100

0

0.7

0.6

0.5

0.4

0.3

0.2

0.1

0

3.0

2.8

T

A

= 25°C

T

= 25°C

A

2.6

T

= 85°C

A

T

= –40°C

A

2.4

2.2

2.0

0.5

1.0

2.0

2.5

1

1.5

2.0

2.5

3.0

3.5

0

1.5

0

10 20 30 40 50 60 70 80 90

DUTY CYCLE (%)

SYNCHRONIZING PULSE AMPLITUDE (V)

SWITCH CURRENT (A)

1306 • G10

1306 • G12

1306 • G11

4

LT1306

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TYPICAL PERFORMANCE CHARACTERISTICS

Continuous-Conduction Mode

Switching Waveforms in Boost

Operation

Rectifier Saturation Voltage

vs Current

Stepdown-Mode Rectifier Voltage

vs Current

0.7

0.6

0.5

0.4

0.3

0.2

0.1

0

1.90

1.85

1.80

1.75

1.70

1.65

1.60

1.55

V

A

= 6V

= 5V

= 25°C

IN

OUT

T

= 85°C

A

T

V_SW

5V/DIV

T

= –40°C

A

I_L

T

= 25°C

A

0.5A/DIV

V_O

0.1V/DIV

AC

0.5

1.0

RECTIFIER CURRENT (A)

2.0

2.5

0

1.5

0.5

1.0

2.0

0

1.5

V

_IN= 4.2V

2µs/DIV

V_O= 5V

RECTIFIER CURRENT (A)

1306 • G13

1306 • G14

Transient Response of the

Converter in Figure 1 with a

50mA to 800mA Load Step

Continuous-Conduction Mode

Switching Waveforms in

Stepdown Mode

Start-Up to Shutdown Transient

Response*

V_S/S

5V/DIV

LOAD

CURRENT

0.5A/DIV

DC

V_SW

5V/DIV

V_SW

5V/DIV

INDUCTOR

CURRENT

1A/DIV

I_L

0.5V/DIV

2A/DIV

V_O

OUTPUT

0.1V/DIV

AC

V_O

5V/DIV

50mV/DIV

AC

2µs/DIV

V_IN= 2.5V

V_IN= 6V

_O= 5V

V_IN= 3.6V

V_O= 5V

1ms/DIV

V

*Notice that the Input Start-Up Current is well Controlled and the

Output Voltage Falls to Zero in Shutdown.

5

LT1306

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PIN FUNCTIONS

CAP (Pin 6): Power Supply to the Synchronous Rectifier

Driver. The bootstrap capacitor and the blocking diode

are tied to this pin. The CAP voltage switches between a

lowlevelofV_IN–V_DtoahighleveldeterminedbytheV_SW

high level.

V_C(Pin 1): Compensation Pin for Error Amplifier. V_Cis

the output of the transconductance error amplifier. Loop

frequency compensation is done by connecting an RC

network from the V_Cpin to ground.

FB (Pin 2): Inverting Input of the Error Amplifier. Connect

the resistor divider tap here. Set output voltage according

to V_OUT= 1.24V (1 + R1/R2).

V_IN(Pin 7): Supply or Battery Input Pin. Must be closely

bypassed to ground plane.

V_OUT(Pin 3): Output of the Switching Regulator and Emit-

ter of the Synchronous Rectifier. Connect appropriate

output capacitor from here to ground. V_OUTmust be kept

below 5.5V.

S/S (Pin 8): Shutdown and Synchronization Pin. Shut-

down is active low with a typical threshold of 0.9V. For

normal operation, the S/S pin is tied to V_IN. To externally

synchronize the switching regulator, drive the S/S pin

with a pulse train.

GND (Pin 4): Ground. Connect to local ground plane.

SW (Pin 5): Switch Pin. The collectors of the grounded

powerswitchandthesynchronousrectifier.KeeptheSW

trace as short as possible to minimize EMI.

W

BLOCK DIAGRA

V

IN

V

C

7

–

+

1

I

> 0

RECT

UVLO

A5

DCM

CONTROL

CAP

6

1.65V

1.24V

+

–

A1

m

X3

–

+

g

X5

X4

IDLE

FB

2

A3

V

B

5

SW

OUT

X1

–

+

3

Q2

S

R

A4

I

RECT

RECTIFIER

Q1

Q

X2

+

–

V

CE2

X4

+

RAMP

COMPENSATION

+

Σ

A2

SENSE

AMP

R

S

–

300kHz OSC

SYNC

CLK

S/S

8

PWM CONTROL

4

1306 F02

GND

SHDN

REF/BIAS

SHUTDOWN

DELAY

Figure 2. LT1306 Block Diagram

6

LT1306

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OPERATIO

The LT1306 is a fixed frequency current mode PWM

regulator with integrated power transistor Q1 and syn-

chronous rectifier Q2.

and switch Q1’s on-time decrease. Hysteretic comparator

A3 determines if V_Cis too low for the LT1306 to operate

efficiently. As V_Cfalls below the trip voltage V_B, the output

of A3 goes high. All circuits except the error amplifier,

comparatorsA3andA5,andtherectifierdrivercontrol X5,

are turned off. After the remaining energy stored in the

inductor is delivered to the output through the synchro-

nous rectifier Q2, the LT1306 stops switching. In this idle

state, the LT1306 draws only 160µA from the input. With

switching stopped and the load being powered by the

output filter capacitor, the output voltage decreases. V_C

then starts to increase. Q1 does not start to switch until V_C

rises above the upper trip point of A3. The LT1306 again

delivers power to the output as a current mode PWM

converter except that the switch current limit is only about

250mA due to the low value of V_C. If the load is still light,

the output voltage will rise and V_Cwill fall, causing the

converter to idle again. Power delivery therefore occurs in

bursts. The on-off cycle frequency, or burst frequency,

depends on the operating conditions, the inductance and

the output filter capacitance. The output voltage ripple in

Burst Mode operation is usually higher than either CCM or

DCMoperation. BurstModeoperationincreaseslightload

efficiency because it delivers more energy to the output

during each clock cycle than is possible with DCM

operation’s extremely low peak switch current. This al-

lows fewer switching cycles per unit time to maintain a

given output. Chip supply current therefore becomes a

small fraction of the total input current.

In the Block Diagram, Figure 2, the PWM control circuit

is enclosed within the dashed line. It consists of the

current sense amplifier (A2), the oscillator, the compen-

sating ramp generator, the PWM comparator (A4), the

logic (X1 and X2), the power transistor driver (X4) and

the main power switch (Q1). Notice that the clock (CLK)

“blanks”Q1conduction.Theinternaloscillatorfrequency

is 300kHz.

Thepulsewidthoftheclockdeterminesthemaximumon

dutyratioofQ1.IntheLT1306thisissetto88%.Q1turns

on at the trailing edge of the clock pulse. To prevent

subharmonic oscillation above 50% duty ratio, a com-

pensatingramp(generatedfromtheoscillatorsawtooth)

is added to the sensed Q1 current. Q1 is turned off when

this sum exceeds the error amplifier A1 output, V_C. Q1’s

absolute current limit is reached when V_C’s upward

excursion is clamped internally at 1.28V.

The error amplifier output, V_C, determines the peak switch

current required to regulate the output voltage. V_Cis a

measure of the output power. At heavy loads, the average

and the peak inductor currents are both high. V_Cmoves to

the upper end of its operating range and the LT1306 oper-

ates in continuous conduction mode (CCM).

As load decreases, the average inductor current de-

creases.InCCM,thepeak-to-peakinductorcurrentripple

to the first order depends only on the inductance, the

inputandtheoutputvoltages.Whentheaverageinductor

current falls below 1/2 of the peak-to-peak inductor

current ripple, the converter enters discontinuous con-

duction mode (DCM). The switching frequency remains

constant except that the inductor current always returns

to zero within each switching cycle.

The synchronous rectifier is represented as NPN transis-

tor, Q2, in the Block Diagram (Figure 2). A rectifier drive

circuit, X5, supplies variable base drive to Q2 and controls

the voltage across the rectifier. The supply voltage, V_CAP

,

for the driver is generated locally with the bootstrap cir-

cuit, D1 and C1 (Figure 1). When Q1 is on, the bootstrap

capacitor C1 is charged from the input to the voltage

V_IN– V_D1(ON)– V_CESAT1. The charging current flows from

the input through D1, C1 and Q1 to ground. After Q1 is

switched off, the node SW goes above V_Oby the rectifier

drop V_CESAT2. D1 becomes back-biased and the CAP volt-

In both CCM and DCM, the output voltage is regulated

with negative feedback. A1 amplifies the error voltage

betweentheinternallygenerated1.24Vreference andthe

attenuated output voltage. The RC network from the V_C

pin to ground provides the loop compensation.

ageispusheduptoV_O+V_CESAT2+V_IN–V_D1(ON)–V_CESAT1

.

C1 supplies the base drive to Q2. The consumed charge is

replenished during the Q1 on interval.

Further reduction in the load moves V_Ctowards the lower

end of its operating range. Both the peak inductor current

7

LT1306

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OPERATIO

In boost operation, X5 drives the rectifier Q2 into satura-

tion. The voltage across the rectifier is V_CESAT. As the

inductor current decreases, Q2’s base drive also de-

creases. X5 ceases supplying base current to Q2 when the

inductor current falls to zero.

A hysteretic comparator in driver X5 controls the mode

of operation. DC transfer characteristics of the compara-

tor are shown in Figure 3 and Figure 4.

A logic low at the S/S pin (Pin 8) initiates shutdown. First,

all circuit blocks in the LT1306 are switched off. The

synchronous rectifier Q2 and its driver are kept on to

allow stored inductive energy to flow to the output. As V_O

drops below V_IN, the voltage across the rectifier Q2

increases so that the inductor voltage reverses. Inductor

current continues to fall to zero. Driver X5 then turns off

and the rectifier, Q2, becomes an open circuit. The

LT1306 dissipates only 9µA in shutdown.

If V_IN> V_O, Q2 will no longer be driven into saturation.

InsteadthevoltageacrossQ2isallowedtoincreasesothat

the inductor voltage reverses polarity as Q1 switches.

Since the inductor voltage is bipolar, volt-second balance

can be maintained regardless of the input voltage. The

LT1306 is therefore capable of operating as a step-down

regulator with the basic boost topology. Input

start-up current is also well controlled since the inductor

currentcannotincreaseduringQ1’soff-timewithnegative

inductor voltage.

The LT1306 is guaranteed to start with a minimum V_INof

1.8V. Comparator A5 senses the input voltage and gen-

erates an undervoltage lockout (UVLO) signal if V_INfalls

below this minimum. In UVLO, V_Cis pulled low and Q1

stops switching. The LT1306 draws 160µA from the

input.

The rectifier voltage drop depends on both the input and

the output voltages. Efficiency in the step-down mode is

less than that of a linear regulator. For sustained step-

down operation, the maximum output current will be

limited by the package thermal characteristics.

MODE

BOOST

STEPDOWN

1306 F03

1306 F04

0

V

IN

– 0.1V

V

IN

V

O

V

+ 0.1V

V

IN

O

Figure 3. DC Transfer Characteristics of the Mode Control

Comparator Plotted with V_Oas an Independent Variable.

V_INis Considered Fixed.

Figure 4. DC Transfer Characteristics of the Mode Control

Comparator Plotted with V_INas an Independent Variable.

V_Ois Considered Fixed.

8

LT1306

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APPLICATIONS INFORMATION

The inductor should be able to handle the full load peak

inductor current without saturation. The peak inductor

current can be as high as 2A. This places a lower limit on

the core size of the inductor. Powder iron cores have

unacceptable core losses and are not suitable for high

efficiency applications. Most ferrite core materials have

manageable core losses and are recommended. Inductor

DC winding resistance (DCR) also needs to be considered

for efficiency. Usually there are trade-offs between core

loss, DCR, saturation current, cost and size.

Output Voltage Setting

The output voltage of the LT1306 is set with a resistive

divider, R1andR2(Figure1andFigure5), fromtheoutput

to ground. The divider tap is tied to the FB pin. Current

through R2 should be significantly higher than the FB pin

input bias current (≤25nA). With R2 = 249k, the input bias

current of the error amplifier is 0.5% of the current in R1.

V

O

R1

R2

R1

R2

V

= 1.24V 1 +

O

(

)

For EMI sensitive applications, one may want to use

magneticallyshieldedortoroidalinductorstocontainfield

radiation. Table 1 lists a number of inductors suitable for

LT1306 applications.

FB PIN

V

O

– 1

R1 = R2

(

)

1.24

1306 F05

Table 1. Inductors Suitable for Use with the LT1306

Figure 5. Feedback Resistive Divider

PART

NO.

VALUE MAX DCR

CORE

TYPE

HEIGHT

(mm)

VENDOR

(µH)

5.0

10

(Ω)

BH Electronics 511-0033

0.023

0.09

Toroid

Open

Toroid

Open

4.8

3.0

5.2

6.0

5.0

3.4

Synchronization and Shutdown

Coilcraft

DO3308-103

DO3316-472

DO3316-103

DO3316-153

CTX5-2

The S/S pin (Pin 8) can be used to synchronize the

oscillator or disconnect the load from the input. The S/S

pin is tied to the input (V_IN> 1.8V) for normal operation.

The oscillator in the LT1306 can be externally synchro-

nized by driving the S/S pin with a pulse train (See the

graph “Maximum Allowable Rise Time of Synchronizing

Pulse” in the Typical Performance Characteristics). The

synchronization is positive edge triggered. The recom-

mended frequency of the external clock ranges from

425kHz to 500kHz. If synchronization results in switching

jitter, reducing the rising edge dv/dt of the external clock

pulse usually cures the problem.

4.7

10

0.018

0.029

0.046

0.021

0.032

0.034

0.072

15

Coiltronics

5

CTX10-2

10

Murata

Sumida

LQN6C4R7

CDRH73-100

4.7

10

Magnetic

Shielding

CD43-4R7

4.7

0.109

Open

3.2

Capacitors

Shutdown will be activated if the S/S pin voltage stays

below the shutdown threshold (0.45V) for more than

50µs. This shutdown delay is reset whenever the S/S pin

goes above the shutdown threshold.

The output filter capacitor is usually chosen based on its

equivalent series resistance (ESR) and the acceptable

change in output voltage as a result of load transients. The

output voltage ripple at the switching frequency can be

estimated by considering the peak inductor current and

the capacitor ESR.

Inductor

The value of the energy storage inductor L1 (Figure 1) is

usually selected so that the peak-to-peak ripple current is

less than 40% of the average inductor current. For 1- or

2-cell alkaline or single Li-Ion to 5V applications, 10µH to

20µH is recommended for the LT1306 running at 300kHz.

A 5µH to 10µH inductor can be used if the LT1306 is

externally synchronized at 500kHz.

I_OV_O

(

)(

)

I_PEAK≈I_IN≈

V

IN

ESR I

V

O

(

)( )( )

O

output ripple (ESR)(I_PEAK) =

V

IN

9

LT1306

U

W U U

APPLICATIONS INFORMATION

Since a boost converter produces high output current

ripple, one also needs to consider the maximum ripple

current rating of the output capacitor. Capacitor reliability

will be affected if the ripple current exceeds the maximum

allowable ratings. This maximum rating is usually

specified as the RMS ripple current. In the LT1306 the

RMS output capacitor ripple current is:

switch and can cause the current limit comparator to trip

erratically.ForboostapplicationswhereV_INisafewtenths

ofavoltbelowV_O, a1µFor2.2µFtantalumcapacitor(such

as AVX TAJ series) can be used for C1. The ESR of the

tantalumcapacitorlimitsthechargingcurrent. Alowvalue

resistor (2Ω to 5Ω) can also be added in series with C1 for

furtherlimitingthechargingcurrentalthoughthistendsto

lower the converter efficiency slightly.

V – V

O

IN

I

Frequency Compensation

O

V

IN

Current mode switching regulators have two feedback

loops. The inner current feedback loop controls the

inductor current in response to the outer loop. The outer

or overall feedback loop tightly regulates the output

voltage. The high frequency gain asymptote of the inner

current loop rolls off at –20dB/decade and crosses the

unitygainaxisatafrequencyω_cbetween1/6to2/3ofthe

switching frequency. The current loop is stable and is

widebandcomparedtotheoverallvoltagefeedbackloop.

The low frequency current loop gain is not high (usually

between unity and 10) but it increases the low frequency

impedance of the inductor as seen by the output filter

capacitor. (In a boost regulator, the inductor is con-

nected to the output during the switch off-time.) Current

mode control introduces an effective series resistance

(>>DCR) to the inductor that damps the LC tank re-

sponse.Thecomplexhigh-QpolesoftheLCfilterarenow

separated, resulting in a dominant pole determined by

the filter capacitance and the load resistance and a

second high frequency pole.

For 2-cell to 5V applications, 220µF low ESR solid tanta-

lum capacitors (AVX TPS series or Sprague 593D series)

work well. To reduce output voltage ripple due to heavy

load transients or Burst Mode operation, higher capaci-

tance may be used. For through-hole applications, Sanyo

OS-CON capacitors are also good choices.

In a boost regulator, the input capacitor ripple current is

much lower. Maximum ripple current rating and input

voltageripplesarenotusuallyofconcern.A22µFtantalum

capacitor soldered near the input pin is generally an

adequate bypass.

Bootstrap Supply

Diode D1 and capacitor C1 generate a pulsating supply

voltage,V_CAP,whichishigherthantheoutput.Therectifier

drive circuit runs off this supply. During rectifier on-time,

the rectifier base current drains C1. Q2 base current and

the maximum allowable V_CAPripple voltage determine the

size of C1. A 1µF capacitor is sufficient to keep V_CAPripple

below0.3V.Fora2-cellinput(V_IN>1.8V)overanextended

temperature range, a BAT54 Schottky diode may be used

forD1.TheuseofaSchottkydiodeincreasesthebootstrap

voltageandtheoperatingheadroomfortherectifierdriver,

X5. Diodes like a 1N4148 or 1N914 work well for 2-cell

inputs over the 0°C to 70°C commercial temperature

range.

For a boost regulator the control to output transfer func-

tion can be shown to have a dominant pole at the load

corner frequency

1

ω_P=

R_L

2

C

( )

O

The charge drawn from C1 during the rectifier on-time has

to be replenished during the switch on-interval. As duty

cycle decreases, the amplitude of the C1 charging current

can increase dramatically especially when delivering high

power to the load. This charging current flows through the

and a moving right-half plane (RHP) zero with a minimum

value of

2

R 1–D

(

)

L

MAX

ω_Z=

L

10

LT1306

U

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APPLICATIONS INFORMATION

where

The low frequency zero 1/R3C_Zof the compensation

network is placed at ω_P/2.

Output Voltage

R_L= MaximumLoad =

2

C_Z=

MaximumDCLoadCurrent

R3ω_P

D_MAX= MaximumConverter Duty Cycle

The capacitor C_Pensures adequate gain margin beyond

the RHP zero. The high frequency pole 1/R3C_Pof the

amplifier frequency response is placed beyond ω_Z.

V_O– V_IN(MIN)+ 0.5

=

V_O+ 0.1

There is also a second pole at the current loop crossover

frequency ω_C(Figure 6). ω_Zis much lower in frequency

thanω_C.Theloopiscompensatedbyadjustingthemidband

gainwithresistorR3(Figure7)sothattheoverallloopgain

crosses 0dB before the minimum frequency RHP zero

(i.e., corresponding to the highest duty ratio). The value of

R3 can be estimated with the fromula:

1

C_P=

3ω_ZR3

Higher output filter capacitance rolls off the gain response

from a lower corner frequency so higher midband gain is

required in the compensation network to make the overall

loop gain cross 0dB just below ω_Z.

390V_O(1–D_MAX)C_OR_L

Layout Consideration

R3 =

L

To minimize EMI and high frequency resonances, it is

essential to keep the SW and the CAP trace leads as short

as possible. The input and the output bypass capacitors

C_INand C_OUTshould be placed close to the IC package and

soldered to the ground plane. A ground plane under the

switching regulator is highly recommended. Figure 8

shows a suggested component placement and PC board

layout.

Duetothelowtransconductanceoftheerroramplifier, the

gain setting resistor R3 is AC-coupled with capacitor C_Z.

ThispreventsR3frominducinganoffsettotheinputofthe

error amplifier. It also creates a pole at DC and a low

frequency zero.

The amplitude response of the error amplifier with the

compensation network shown is:

1+ S •R3 •C_Z

ˆ

(

)

V_C

R2

R1+R2

= g_m

ˆ

V_O

S •C_Z1+ S •R3 •C_P

(

)

]

[

C_Z>> C_P

11

LT1306

U

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APPLICATIONS INFORMATION

GAIN

(dB)

(g )(R3)(R2)

R1 + R2

m

MIDBAND GAIN =

ˆ

V

C

ˆ

V

2

O

R (1 – D

L

)

MAX

RHP ZERO =

L

AMPLITUDE RESPONSE

OF THE ERROR AMPLIFIER

AMOUNT OF

MIDBAND GAIN

NEEDED

CURRENT LOOP

CROSSOVER

FREQUENCY

ω

C

ω

Z

ω

P

0

1

(R3)(C )

Z

R

(C )

L

O

(

)

2

OVERALL LOOP GAIN

AFTER COMPENSTION

LOOP GAIN

CROSSOVER

1

ω

FREQUENCY ≈

Z

3

AMPLITUDE RESPONSE OF

CONTROL-TO-OUTPUT

TRANSFER FUNCTION BEFORE

COMPENSATION

ˆ

V

O

ˆ

V

C

1306 F06

ˆ

V_C

V_O

Figure 6. Gain Asymptotes of the Control-to-Output

and Error Amplifier

Transfer Function

ˆ

V_C

V_O

SW

L

V

IN

Q2

V

O

PWM CONTROL

LOGIC

Q1

g

R1

RECTIFIER

I

O

–

FB

m

+

1.24V

R2

R

C

O

L

LT1306

GND

V

C

R3

C

P

C

Z

1306 F07

Figure 7. Current Mode Boost Converter Overall-Loop Compensation

12

LT1306

U

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APPLICATIONS INFORMATION

GROUND PLANE

C

Z

R3

S/S

V

IN

C

P

V

C

R2

LT1306

1

8

7

6

5

D1

R1

2

3

4

C

IN2

IN1

C1

+

V

OUT

C

O2

O1

VIAS

L1

GND

1306 F08

Figure 8. Recommended Component Placement for LT1306.

Notice That the Input and the Output Capacitors Are Grounded

at the Same Point. A Ground Plane Under the DC/DC Converter

Is Highly Recommended. Use Multiple Vias to Tie Pin 4 Copper

to the Ground Plane

13

LT1306

U

TYPICAL APPLICATIO S

2-Cell NiMH to 3.3V Output

D1

C1

1µF

L1

4.7µH

+

2V

TO 3V

V

SW

CAP

OUT

IN

2V/500kHz

3.3V

1A

S/S

R1

412k

LT1306

+

C

IN2

22µF

IN1

FB

0.1µF

+

V

C

GND

CERAMIC

C

O

220µF

R3

95k

R2

C

P

249k

C

: AVX TAJC226M010

IN1

O1

39pF

: AVX TPSE227M010R0100

C

Z

C1: AVX TAJA105K020

D1: CMDSH-3

L1: LQN6C4R7

5.6nF

1306 F09

Efficiency

90

85

80

75

70

65

60

V

= 3.3V

O

L1 = 4.7µH

V

= 3V

IN

V

= 2.5V

IN

V

IN

= 1.8V

1

10

100

1000 2000

LOAD CURRENT (mA)

1306 F09a

14

LT1306

U

PACKAGE DESCRIPTION

Dimensions in inches (millimeters) unless otherwise noted.

S8 Package

8-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.189 – 0.197*

(4.801 – 5.004)

7

5

8

6

0.150 – 0.157**

(3.810 – 3.988)

0.228 – 0.244

(5.791 – 6.197)

1

3

4

2

0.010 – 0.020

(0.254 – 0.508)

× 45°

0.053 – 0.069

(1.346 – 1.752)

0.004 – 0.010

(0.101 – 0.254)

0.008 – 0.010

(0.203 – 0.254)

0°– 8° TYP

0.016 – 0.050

(0.406 – 1.270)

0.050

(1.270)

BSC

0.014 – 0.019

(0.355 – 0.483)

TYP

*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD

FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

SO8 1298

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tation that the interconnection ofits circuits as described herein willnotinfringe on existing patentrights.

15

LT1306

TYPICAL APPLICATIO S

U

4-Cell NiMH to 5V Output

Efficiency

D1

90

85

80

75

70

65

60

V

= 5V

O

V

= 4.8V

= 3.6V

IN

L1 = 10µH

C1

L1

1µF

10µH

+

3.6V

TO 6.5V

IN

V

SW

CAP

OUT

IN

5V

1A

S/S

R1

LT1306

V

= 6V

IN

+

768k

C

IN1

22µF

C

IN2

FB

0.1µF

C

+

V

C

GND

O2

CERAMIC

C

O1

1µF

220µF

CERAMIC

R3

75k

R2

249k

C

P

C

: AVX TAJC226M010

IN1

22pF

1

10

100

1000 2000

: AVX TPSE227M010R0100

O1

C

Z

LOAD CURRENT (mA)

C1: AVX TAJA105K020

D1: MMBD914LT1

L1: CTX10-3

15nF

1306 F10a

1306 F09

Transient Response with Step Input (4V to 6V)

V_IN

5V/DIV

V_SW

5V/DIV

I_L

500mA/DIV

V_O

0.1V/DIV

AC

0.5ms/DIV

RELATED PARTS

PART NUMBER

LT1302

DESCRIPTION

COMMENTS

5V/600mA from 2V, 2A Internal Switch, 200µA I

High Output Current Micropower DC/DC Converter

2-Cell Micropower DC/DC Converter

Q

LT1304

5V/200mA, Low-Battery Detector Active in Shutdown

3.3V at 75mA from One Cell, MSOP Package

5V at 1A from Single Li-Ion Cell

LT1307/LT1307B

LT1308A/LT1308B

LT1316

Single Cell, Micropower, 600kHz PWM DC/DC Converters

High Output Current Micropower DC/DC Converter

Burst ModeOperation DC/DC with Programmable Current Limit

Micropower, 600kHz PWM DC/DC Converters

Single-Cell Micropower DC/DC Converter

1.5V Minimum, Precise Control of Peak Current Limit

LT1317/LT1317B

LT1610

100µA I , Operate with V as Low as 1.5V

Q IN

3V at 30mA from 1V, 1.7MHz Fixed Frequency

LT1613

1.4MHz Switching Regulator in 5-Lead SOT-23

Micropower Step-Up DC/DC in 5-Lead SOT-23

High Efficiency N-channel Switching Regulator Controller

5V at 200mA from 4.4V Input, Tiny SOT-23 package

LT1615

20µA I , 36V/350mA Internal Switch, V as Low as 1.2V

Q IN

LTC1624

V

= 1.19V to 30V in Stepdown; V = 3.5V to 36V

OUT IN

SO-8 Package

LT1949

600kHz, 1A Switch PWM DC/DC Converter

1.1A, 0.5Ω/30V Internal Switch, V as Low as 1.5V

IN

1306f LT/TP 0400 4K • PRINTED IN USA

LinearTechnology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

16

●

(408)432-1900 FAX:(408)434-0507 www.linear-tech.com

LINEAR TECHNOLOGY CORPORATION 1999


型号：	LT1306
厂家：	Linear
描述：	Synchronous, Fixed Frequency Step-Up DC/DC Converter 同步，固定频率升压型DC / DC转换器转换器
文件：	总16页 (文件大小：217K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

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