LT1933HDCB#TRM [Linear]
IC 1.05 A SWITCHING REGULATOR, 600 kHz SWITCHING FREQ-MAX, PDSO6, 2 X 3 MM, 0.75 MM HEIGHT, PLASTIC, MO-229, DFN-6, Switching Regulator or Controller;
| 型号： | LT1933HDCB#TRM |
| 厂家： | 
Linear |
| 描述： | IC 1.05 A SWITCHING REGULATOR, 600 kHz SWITCHING FREQ-MAX, PDSO6, 2 X 3 MM, 0.75 MM HEIGHT, PLASTIC, MO-229, DFN-6, Switching Regulator or Controller 稳压器 开关 |
| 文件： | 总16页 (文件大小：629K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LT1933
600mA, 500kHz Step-Down
Switching Regulator
in SOT-23
U
FEATURES
DESCRIPTIO
The LT®1933 is a current mode PWM step-down DC/DC
converter with an internal 0.75A power switch, packaged
in a tiny 6-lead SOT-23. The wide input range of 3.6V to
36V makes the LT1933 suitable for regulating power from
a wide variety of sources, including unregulated wall
transformers, 24V industrial supplies and automotive
batteries. Its high operating frequency allows the use of
tiny, low cost inductors and ceramic capacitors, resulting
in low, predictable output ripple.
■
Wide Input Range: 3.6V to 36V
■
5V at 600mA from 16V to 36V Input
■
3.3V at 600mA from 12V to 36V Input
■
5V at 500mA from 6.3V to 36V Input
■
3.3V at 500mA from 4.5V to 36V Input
■
Fixed Frequency 500kHz Operation
■
Uses Tiny Capacitors and Inductors
■
Soft-Start
Internally Compensated
■
■
Low Shutdown Current: <2µA
Output Adjustable Down to 1.25V
Low Profile (1mm) SOT-23 (ThinSOT™) Package
Cycle-by-cycle current limit provides protection against
shorted outputs, and soft-start eliminates input current
surge during start up. The low current (<2µA) shutdown
provides output disconnect, enabling easy power man-
agement in battery-powered systems.
■
■
U
APPLICATIO S
, LTC and LT are registered trademarks of Linear Technology Corporation.
ThinSOT is a trademark of Linear Technology Corporation.
■
Automotive Battery Regulation
■
Industrial Control Supplies
■
Wall Transformer Regulation
■
Distributed Supply Regulation
■
Battery-Powered Equipment
U
TYPICAL APPLICATIO
3.3V Step-Down Converter
Efficiency
95
1N4148
V
IN
= 12V
V
IN
V
IN
BOOST
4.5V TO 36V
90
85
80
75
70
65
LT1933
0.1µF
V
= 5V
OUT
22µH
OFF ON
SHDN
GND
SW
FB
V
OUT
3.3V/500mA
V
= 3.3V
OUT
MBRM140
16.5k
2.2µF
22µF
10k
1933 TA01a
100
200
LOAD CURRENT (mA)
500
600
0
300
400
1933 TA01b
1933f
1
LT1933
W W U W
U W
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ABSOLUTE AXI U RATI GS
PACKAGE/ORDER I FOR ATIO
(Note 1)
Input Voltage (VIN) ....................................–0.4V to 36V
BOOST Pin Voltage .................................................. 43V
BOOST Pin Above SW Pin ....................................... 20V
SHDN Pin ..................................................–0.4V to 36V
FB Voltage ...................................................–0.4V to 6V
Operating Temperature Range (Note 2)
LT1933E ................................................. –40°C to 85°C
LT1933I ................................................ –40°C to 125°C
Maximum Junction Temperature .......................... 125°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
LT1933ES6
LT1933IS6
BOOST 1
GND 2
FB 3
6 SW
5 V
IN
4 SHDN
S6 PART MARKING
S6 PACKAGE
6-LEAD PLASTIC TSOT-23
LTAGN
LTAGP
TJMAX = 125°C, θJA = 165°C/ W,
θJC = 102°C/ W
Consult factory for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C.
VIN = 12V, VBOOST = 17V, unless otherwise noted. (Note 2)
PARAMETER
CONDITIONS
MIN
TYP
3.35
1.245
40
MAX
3.6
UNITS
V
Undervoltage Lockout
Feedback Voltage
●
●
1.225
1.265
120
2.5
V
FB Pin Bias Current
Quiescent Current
V
= Measured V + 10mV (Note 4)
nA
mA
µA
%/V
kHz
kHz
%
FB
REF
Not Switching
1.6
Quiescent Current in Shutdown
Reference Line Regulation
Switching Frequency
V
V
V
V
= 0V
0.01
0.01
500
55
2
SHDN
= 5V to 36V
= 1.1V
IN
FB
FB
400
600
= 0V
Maximum Duty Cycle
Switch Current Limit
●
88
94
(Note 3)
0.75
1.05
370
A
Switch V
I
= 400mA
500
2
mV
µA
V
CESAT
SW
Switch Leakage Current
Minimum Boost Voltage Above Switch
BOOST Pin Current
I
I
= 400mA
= 400mA
1.9
18
2.3
25
SW
SW
mA
V
SHDN Input Voltage High
SHDN Input Voltage Low
SHDN Bias Current
2.3
0.3
V
V
V
= 2.3V (Note 5)
= 0V
34
0.01
50
0.1
µA
µA
SHDN
SHDN
with statistical process controls. The LT1933I specifications are
Note 1: Absolute Maximum Ratings are those values beyond which the life
guaranteed over the –40°C to 125°C temperature range.
of the device may be impaired.
Note 3: Current limit guaranteed by design and/or correlation to static test.
Slope compensation reduces current limit at higher duty cycle.
Note 4: Current flows out of pin.
Note 5: Current flows into pin.
Note 2: The LT1933E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
1933f
2
LT1933
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency, VOUT = 5V
Efficiency, VOUT = 3.3V
Switch Current Limit
1200
1000
800
600
400
200
0
100
100
T
= 25°C
OUT
T
= 25°C
T = 25°C
A
A
A
V
= 5V
V
= 3.3V
OUT
TYPICAL
90
90
V
IN
= 12V
V
= 5V
IN
MINIMUM
V
= 12V
V
IN
= 24V
IN
80
80
V
= 24V
IN
70
60
70
D1 = MBRM140
L1 = Toko D53LCB 33µH
D1 = MBRM140
L1 = Toko D53LCB 22µH
60
100
200
LOAD CURRENT (mA)
500
600
100
200
LOAD CURRENT (mA)
500
600
0
300
400
0
300
400
0
20
40
60
80
100
DUTY CYCLE (%)
1933 G01
1933 G02
1933 G03
Maximum Load Current
Maximum Load Current
Switch Voltage Drop
800
700
600
500
400
800
700
600
500
400
600
500
400
300
200
100
0
T
= 25°C
OUT
T = 25°C
A
A
V
= 5V
V
= 3.3V
OUT
T
= 25°C
A
L = 22µH
L = 33µH
T
= 85°C
A
T
= –40°C
A
L = 15µH
L = 22µH
0
5
10
15
20
25
30
0
5
10
15
20
25
30
0
0.1
0.2
0.3
0.4
0.5
0.6
SWITCH CURRENT (A)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
1933 G04
1933 G05
1933 G06
Feedback Voltage
Undervoltage Lockout
Switching Frequency
1.260
1.255
1.250
1.245
1.240
1.235
1.230
3.8
3.6
3.4
3.2
3.0
600
550
500
450
400
–50 –25
0
25
50
75 100 125
–50 –25
0
25
50
75 100 125
–50 –25
0
25
50
75 100 125
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
1933 G07
1933 G08
1933 G09
1933f
3
LT1933
TYPICAL PERFOR A CE CHARACTERISTICS
U W
Frequency Foldback
Soft-Start
SHDN Pin Current
700
600
500
400
300
200
100
0
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
200
150
100
50
T
= 25°C
T
= 25°C
T = 25°C
A
A
A
DC = 30%
0
0.0
0.5
1.0
1.5
0
1
2
3
4
0
4
8
12
16
FB PIN VOLTAGE (V)
SHDN PIN VOLTAGE (V)
SHDN PIN VOLTAGE (V)
1933 G10
1933 G11
1933 G12
Typical Minimum Input Voltage
Typical Minimum Input Voltage
Switch Current Limit
8
7
6
5
4
6.0
5.5
5.0
4.5
4.0
3.5
3.0
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
V
A
L = 33µH
= 5V
V
= 3.3V
OUT
OUT
T
= 25°C
T = 25°C
A
L = 22µH
TO START
TO START
TO RUN
TO RUN
1
10
100
1
10
100
–50 –25
0
25
50
75 100 125
LOAD CURRENT (mA)
LOAD CURRENT (mA)
TEMPERATURE (°C)
1933 G13
1933 G14
1933 G15
Operating Waveforms,
Discontinuous Mode
Operating Waveforms
VSW 10V/DIV
VSW 10V/DIV
L 200mA/DIV
I
I
L 200mA/DIV
VOUT 10mV/DIV
VOUT 10mV/DIV
1933 G16
1933 G16
VIN = 12V, VOUT = 3.3V, IOUT = 400mA,
VIN = 12V, VOUT = 3.3V, IOUT = 20mA,
L = 22µH, COUT = 22µF
L = 22µH, COUT = 22µF
1933f
4
LT1933
U
U
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PI FU CTIO S
BOOST (Pin 1): The BOOST pin is used to provide a drive
voltage, higher than the input voltage, to the internal
bipolar NPN power switch.
SHDN (Pin 4): The SHDN pin is used to put the LT1933 in
shutdown mode. Tie to ground to shut down the LT1933.
Tie to 2.3V or more for normal operation. If the shutdown
feature is not used, tie this pin to the VIN pin. SHDN also
provides a soft-start function; see the Applications Infor-
mation section.
GND(Pin2):TietheGNDpintoalocalgroundplanebelow
the LT1933 and the circuit components. Return the feed-
back divider to this pin.
VIN (Pin 5): The VIN pin supplies current to the LT1933’s
internal regulator and to the internal power switch. This
pin must be locally bypassed.
FB (Pin 3): The LT1933 regulates its feedback pin to
1.245V. Connect the feedback resistor divider tap to this
pin. Set the output voltage according to VOUT = 1.245V
(1 + R1/R2). A good value for R2 is 10k.
SW (Pin 6): The SW pin is the output of the internal power
switch. Connect this pin to the inductor, catch diode and
boost capacitor.
W
BLOCK DIAGRA
V
IN
V
IN
5
C2
INT REG
AND
UVLO
D2
BOOST
1
Σ
ON OFF
SLOPE
COMP
R
S
Q
Q
R3
SHDN
C3
4
DRIVER
Q1
C4
L1
SW
OSC
V
6
OUT
C1
D1
FREQUENCY
FOLDBACK
V
C
g
m
1.245V
GND
FB
2
3
R2
R1
1933 BD
1933f
5
LT1933
OPERATIO
U
(Refer to Block Diagram)
The LT1933 is a constant frequency, current mode step
down regulator. A 500kHz oscillator enables an RS flip-
flop, turning on the internal 750mA power switch Q1. An
amplifier and comparator monitor the current flowing
between the VIN and SW pins, turning the switch off when
this current reaches a level determined by the voltage at
VC.Anerroramplifiermeasurestheoutputvoltagethrough
anexternalresistordividertiedtotheFBpinandservosthe
VC node. If the error amplifier’s output increases, more
current is delivered to the output; if it decreases, less
currentisdelivered.Anactiveclamp(notshown)ontheVC
node provides current limit. The VC node is also clamped
to the voltage on the SHDN pin; soft-start is implemented
by generating a voltage ramp at the SHDN pin using an
external resistor and capacitor.
An internal regulator provides power to the control cir-
cuitry. This regulator includes an undervoltage lockout to
preventswitchingwhenVIN islessthan~3.35V.TheSHDN
pin is used to place the LT1933 in shutdown, disconnect-
ing the output and reducing the input current to less than
2µA.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate the
internal bipolar NPN power switch for efficient operation.
The oscillator reduces the LT1933’s operating frequency
when the voltage at the FB pin is low. This frequency
foldbackhelpstocontroltheoutputcurrentduringstartup
and overload.
U
W U U
APPLICATIO S I FOR ATIO
FB Resistor Network
voltage of:
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1%
resistors according to:
VIN(MIN) = (VOUT + VD)/DCMAX – VD + VSW
with DCMAX = 0.88
The maximum input voltage is determined by the absolute
maximum ratings of the VIN and BOOST pins and by the
minimum duty cycle DCMIN = 0.08 (corresponding to a
minimum on time of 130ns):
R1 = R2(VOUT/1.245 – 1)
R2 should be 20k or less to avoid bias current errors.
Reference designators refer to the Block Diagram.
VIN(MAX) = (VOUT + VD)/DCMIN – VD + VSW
Input Voltage Range
Note that this is a restriction on the operating input
voltage; the circuit will tolerate transient inputs up to the
absolute maximum ratings of the VIN and BOOST pins.
The input voltage range for LT1933 applications depends
on the output voltage and on the absolute maximum
ratings of the VIN and BOOST pins.
The minimum input voltage is determined by either the
LT1933’s minimum operating voltage of ~3.35V, or by its
maximum duty cycle. The duty cycle is the fraction of time
thattheinternalswitchisonandisdeterminedbytheinput
and output voltages:
Inductor Selection and Maximum Output Current
A good first choice for the inductor value is:
L = 5 (VOUT + VD)
whereVD isthevoltagedropofthecatchdiode(~0.4V)and
L is in µH. With this value the maximum load current will
be above 500mA. The inductor’s RMS current rating must
be greater than your maximum load current and its satu-
ration current should be about 30% higher. For robust
operation in fault conditions the saturation current should
DC = (VOUT + VD)/(VIN – VSW + VD)
where VD is the forward voltage drop of the catch diode
(~0.4V) and VSW is the voltage drop of the internal switch
(~0.4V at maximum load). This leads to a minimum input
1933f
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LT1933
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APPLICATIO S I FOR ATIO
be ~1A. To keep efficiency high, the series resistance
(DCR) should be less than 0.2Ω. Table 1 lists several
vendors and types that are suitable.
for 0.5A forward current and a maximum reverse voltage
of 40V. The MBRM140 provides better efficiency, and will
handle extended overload conditions.
Of course, such a simple design guide will not always
result in the optimum inductor for your application. A
larger value provides a slightly higher maximum load
current, and will reduce the output voltage ripple. If your
loadislowerthan500mA, thenyoucandecreasethevalue
oftheinductorandoperatewithhigherripplecurrent. This
allowsyoutouseaphysicallysmallerinductor,oronewith
a lower DCR resulting in higher efficiency. There are
several graphs in the Typical Performance Characteristics
section of this data sheet that show the maximum load
current as a function of input voltage and inductor value
for several popular output voltages. Low inductance may
result in discontinuous mode operation, which is OK, but
further reduces maximum load current. For details of
maximum output current and discontinuous mode opera-
tion, see Linear Technology Application Note 44. Finally,
for duty cycles greater than 50% (VOUT/VIN < 0.5), there is
a minimum inductance required to avoid subharmonic
oscillations. Choosing L greater than 3(VOUT + VD) µH
prevents subharmonic oscillations at all duty cycles.
Input Capacitor
Bypass the input of the LT1933 circuit with a 2.2µF or
higher value ceramic capacitor of X7R or X5R type. Y5V
types have poor performance over temperature and ap-
plied voltage, and should not be used. A 2.2µF ceramic is
adequate to bypass the LT1933 and will easily handle the
ripplecurrent.However,iftheinputpowersourcehashigh
impedance, or there is significant inductance due to long
wires or cables, additional bulk capacitance may be nec-
essary. This can be provided with a low performance
electrolytic capacitor.
Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage ripple
at the LT1933 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
A 2.2µF capacitor is capable of this task, but only if it is
placed close to the LT1933 and the catch diode; see the
PCB Layout section. A second precaution regarding the
ceramic input capacitor concerns the maximum input
voltage rating of the LT1933. A ceramic input capacitor
combined with trace or cable inductance forms a high
quality (under damped) tank circuit. If the LT1933 circuit
is plugged into a live supply, the input voltage can ring to
twice its nominal value, possibly exceeding the LT1933’s
Catch Diode
A0.5Aor1ASchottkydiodeisrecommendedforthecatch
diode, D1. The diode must have a reverse voltage rating
equal to or greater than the maximum input voltage. The
ON Semiconductor MBR0540 is a good choice; it is rated
Table 1. Inductor Vendors
Vendor
URL
Part Series
DO1608C
MSS5131
MSS6122
CR43
Inductance Range (µH)
10 to 22
Size (mm)
Coilcraft
www.coi1craft.com
2.9 × 4.5 × 6.6
3.1 × 5.1 × 5.1
2.2 × 6.1 × 6.1
3.5 × 4.3 × 4.8
3.0 × 5.0 × 5.0
3.0 × 5.7 × 5.7
2.0 × 5.0 × 5.0
3.0 × 5.0 × 5.0
2.8 × 4.8 × 4.8
2.9 × 4.5 × 6.6
3.2 × 4.0 × 4.5
10 to 22
10 to 33
Sumida
www.sumida.com
10 to 22
CDRH4D28
CDRH5D28
D52LC
10 to 33
22 to 47
Toko
www.toko.com
10 to 22
D53LC
22 to 47
Würth Elektronik
www.we-online.com
WE-TPC MH
WE-PD4 S
WE-PD2 S
10 to 22
10 to 22
10 to 47
1933f
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LT1933
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APPLICATIO S I FOR ATIO
voltage rating. This situation is easily avoided; see the Hot
Plugging Safely section.
may be required to get the full benefit (see the Compensa-
tion section).
High performance electrolytic capacitors can be used for
theoutputcapacitor.LowESRisimportant,sochooseone
that is intended for use in switching regulators. The ESR
should be specified by the supplier, and should be 0.1Ω or
less. Such a capacitor will be larger than a ceramic
capacitor and will have a larger capacitance, because the
capacitor must be large to achieve low ESR. Table 2 lists
several capacitor vendors.
Output Capacitor
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated by
the LT1933 to produce the DC output. In this role it
determines the output ripple, and low impedance at the
switching frequency is important. The second function is
to store energy in order to satisfy transient loads and
stabilize the LT1933’s control loop.
Figure 1 shows the transient response of the LT1933 with
several output capacitor choices. The output is 3.3V. The
load current is stepped from 100mA to 400mA and back
to 100mA, and the oscilloscope traces show the output
voltage. The upper photo shows the recommended value.
The second photo shows the improved response (less
voltage drop) resulting from a larger output capacitor and
a phase lead capacitor. The last photo shows the response
to a high performance electrolytic capacitor. Transient
performance is improved due to the large output capaci-
tance, but output ripple (as shown by the broad trace) has
increased because of the higher ESR of this capacitor.
Ceramic capacitors have very low equivalent series resis-
tance (ESR) and provide the best ripple performance. A
good value is
C
OUT = 60/VOUT
where COUT is in µF. Use X5R or X7R types, and keep in
mind that a ceramic capacitor biased with VOUT will have
less than its nominal capacitance. This choice will provide
low output ripple and good transient response. Transient
performance can be improved with a high value capacitor,
but a phase lead capacitor across the feedback resistor R1
Table 2. Capacitor Vendors
Vendor
Phone
URL
Part Series
Comments
Panasonic
(714) 373-7366
www.panasonic.com
Ceramic,
Polymer,
Tantalum
EEF Series
Kemet
Sanyo
(864) 963-6300
(408) 749-9714
www.kemet.com
Ceramic,
Tantalum
T494, T495
POSCAP
www.sanyovideo.com Ceramic,
Polymer,
Tantalum
Murata
AVX
(404) 436-1300
www.murata.com
www.avxcorp.com
Ceramic
Ceramic,
Tantalum
TPS Series
Taiyo Yuden (864) 963-6300
www.taiyo-yuden.com Ceramic
1933f
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LT1933
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APPLICATIO S I FOR ATIO
V
VOUT
50mV/DIV
OUT
16.5k
FB
10k
22µF
IOUT
200mA/DIV
1933 F01a
VOUT
50mV/DIV
V
OUT
470pF
16.5k
FB
22µF
2x
10k
IOUT
200mA/DIV
1933 F01b
VOUT
50mV/DIV
V
OUT
16.5k
FB
+
100µF
10k
IOUT
200mA/DIV
SANYO
4TPB100M
1933 F01c
Figure 1. Transient Load Response of the LT1933 with Different
Output Capacitors as the Load Current is Stepped from 100mA to
400mA. VIN = 12V, VOUT = 3.3V, L = 22µH.
1933f
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LT1933
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APPLICATIO S I FOR ATIO
BOOST Pin Considerations
energy stored in the inductor, the circuit will rely on some
minimum load current to get the boost circuit running
properly. This minimum load will depend on input and
output voltages, and on the arrangement of the boost
circuit. The minimum load generally goes to zero once the
circuit has started. Figure 3 shows a plot of minimum load
to start and to run as a function of input voltage. In many
cases the discharged output capacitor will present a load
to the switcher which will allow it to start. The plots show
theworst-casesituationwhereVIN isrampingveryslowly.
For lower start-up voltage, the boost diode can be tied to
VIN; however, this restricts the input range to one-half of
the absolute maximum rating of the BOOST pin.
Capacitor C3 and diode D2 are used to generate a boost
voltage that is higher than the input voltage. In most cases
a 0.1µF capacitor and fast switching diode (such as the
1N4148 or 1N914) will work well. Figure 2 shows two
ways to arrange the boost circuit. The BOOST pin must be
at least 2.3V above the SW pin for best efficiency. For
outputs of 3V and above, the standard circuit (Figure 2a)
is best. For outputs between 2.5V and 3V, use a 0.47µF
capacitor and a small Schottky diode (such as the BAT-
54). For lower output voltages the boost diode can be tied
to the input (Figure 2b). The circuit in Figure 2a is more
efficient because the BOOST pin current comes from a
lower voltage source. You must also be sure that the
maximumvoltageratingoftheBOOSTpinisnotexceeded.
Minimum Input Voltage VOUT = 3.3V
6.0
The minimum operating voltage of an LT1933 application
is limited by the undervoltage lockout (~3.35V) and by the
maximumdutycycleasoutlinedabove.Forproperstartup,
the minimum input voltage is also limited by the boost
circuit.Iftheinputvoltageisrampedslowly,ortheLT1933
is turned on with its SHDN pin when the output is already
in regulation, then the boost capacitor may not be fully
charged. Because the boost capacitor is charged with the
V
A
L = 22µH
= 3.3V
OUT
T
= 25°C
5.5
5.0
4.5
4.0
3.5
3.0
TO START
TO RUN
D2
1
10
100
LOAD CURRENT (mA)
1933 F03a
C3
BOOST
LT1933
Minimum Input Voltage VOUT = 5V
V
IN
V
OUT
V
SW
IN
8
7
6
5
4
V
A
L = 33µH
= 5V
OUT
= 25°C
GND
T
1933 F02a
V
– V ≅ V
SW OUT
BOOST
TO START
TO RUN
MAX V
≅ V + V
IN OUT
BOOST
(2a)
D2
C3
BOOST
LT1933
V
IN
V
V
SW
OUT
IN
GND
1
10
100
1933 F02b
LOAD CURRENT (mA)
V
– V ≅ V
BOOST
BOOST
SW
IN
IN
1933 F03b
MAX V
≅ 2V
(2b)
Figure 3. The Minimum Input Voltage Depends
on Output Voltage, Load Current and Boost Circuit
Figure 2. Two Circuits for Generating the Boost Voltage
1933f
10
LT1933
U
W U U
APPLICATIO S I FOR ATIO
At light loads, the inductor current becomes discontinu-
ous and the effective duty cycle can be very high. This
reduces the minimum input voltage to approximately
300mV above VOUT. At higher load currents, the inductor
current is continuous and the duty cycle is limited by the
maximum duty cycle of the LT1933, requiring a higher
input voltage to maintain regulation.
Shorted and Reversed Input Protection
If the inductor is chosen so that it won’t saturate exces-
sively, an LT1933 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT1933 is absent. This may occur in battery charging
applications or in battery backup systems where a battery
or some other supply is diode OR-ed with the LT1933’s
output. If the VIN pin is allowed to float and the SHDN pin
is held high (either by a logic signal or because it is tied to
VIN), then the LT1933’s internal circuitry will pull its
quiescent current through its SW pin. This is fine if your
system can tolerate a few mA in this state. If you ground
the SHDN pin, the SW pin current will drop to essentially
zero. However, if the VIN pin is grounded while the output
is held high, then parasitic diodes inside the LT1933 can
Soft-Start
The SHDN pin can be used to soft-start the LT1933,
reducing the maximum input current during start up. The
SHDN pin is driven through an external RC filter to create
a voltage ramp at this pin. Figure 4 shows the start up
waveforms with and without the soft-start circuit. By
choosing a large RC time constant, the peak start up
current can be reduced to the current that is required to
regulate the output, with no overshoot. Choose the value
of the resistor so that it can supply 60µA when the SHDN
pin reaches 2.3V.
RUN
5V/DIV
RUN
SHDN
GND
IIN
100mA/DIV
1933 F04a
VOUT
5V/DIV
50µs/DIV
RUN
5V/DIV
RUN
15k
SHDN
GND
0.1µF
IIN
100mA/DIV
VOUT
5V/DIV
1933 F04b
0.5ms/DIV
Figure 4. To Soft-Start the LT1933, Add a Resistor and Capacitor to the SHDN Pin.
VIN = 12V, VOUT = 3.3V, COUT = 22µF, RLOAD = 10Ω.
1933f
11
LT1933
U
W U U
APPLICATIO S I FOR ATIO
pulllargecurrentsfromtheoutputthroughtheSWpinand
the VIN pin. Figure 5 shows a circuit that will run only when
the input voltage is present and that protects against a
shorted or reversed input.
Hot Plugging Safely
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT1933 circuits. However, these ca-
pacitors can cause problems if the LT1933 is plugged into
a live supply (see Linear Technology Application Note 88
for a complete discussion). The low loss ceramic capaci-
tor combined with stray inductance in series with the
powersourceformsanunderdampedtankcircuit, andthe
voltage at the VIN pin of the LT1933 can ring to twice the
nominal input voltage, possibly exceeding the LT1933’s
rating and damaging the part. If the input supply is poorly
controlled or the user will be plugging the LT1933 into an
energizedsupply, theinputnetworkshouldbedesignedto
prevent this overshoot.
D4
5
4
1
6
V
V
BOOST
IN
IN
LT1933
V
SHDN
SW
OUT
GND
2
FB
3
BACKUP
D4: MBR0540
1933 F05
Figure 5. Diode D4 Prevents a Shorted Input from Discharging
a Backup Battery Tied to the Output; It Also Protects the Circuit
from a Reversed Input. The LT1933 Runs Only When the Input
is Present
Figure6showsthewaveformsthatresultwhenanLT1933
circuit is connected to a 24V supply through six feet of 24-
gauge twisted pair. The first plot is the response with a
CLOSING SWITCH
SIMULATES HOT PLUG
I
IN
V
IN
DANGER!
LT1933
2.2µF
V
IN
20V/DIV
RINGING V MAY EXCEED
IN
ABSOLUTE MAXIMUM
RATING OF THE LT1933
+
I
IN
5A/DIV
LOW
STRAY
IMPEDANCE
ENERGIZED
24V SUPPLY
INDUCTANCE
DUE TO 6 FEET
(2 METERS) OF
TWISTED PAIR
20µs/DIV
(6a)
V
LT1933
2.2µF
IN
20V/DIV
+
+
+
10µF
35V
AI.EI.
I
IN
5A/DIV
(6b)
20µs/DIV
1Ω
V
LT1933
2.2µF
IN
20V/DIV
0.1µF
I
IN
5A/DIV
1933 F06
20µs/DIV
(6c)
Figure 6. A Well Chosen Input Network Prevents Input Voltage Overshoot and
Ensures Reliable Operation When the LT1933 is Connected to a Live Supply
1933f
12
LT1933
U
W U U
APPLICATIO S I FOR ATIO
2.2µF ceramic capacitor at the input. The input voltage
rings as high as 35V and the input current peaks at 20A.
One method of damping the tank circuit is to add another
capacitor with a series resistor to the circuit. In Figure 6b
an aluminum electrolytic capacitor has been added. This
capacitor’s high equivalent series resistance damps the
circuit and eliminates the voltage overshoot. The extra
capacitor improves low frequency ripple filtering and can
slightly improve the efficiency of the circuit, though it is
likely to be the largest component in the circuit. An
alternative solution is shown in Figure 6c. A 1Ω resistor is
added in series with the input to eliminate the voltage
overshoot (it also reduces the peak input current). A 0.1µF
capacitor improves high frequency filtering. This solution
is smaller and less expensive than the electrolytic capaci-
tor. For high input voltages its impact on efficiency is
minor, reducing efficiency less than one half percent for a
5V output at full load operating from 24V.
that the capacitor on the VC node (CC) integrates the error
amplifier output current, resulting in two poles in the loop.
RC provides a zero. With the recommended output capaci-
tor, the loop crossover occurs above the RCCC zero. This
simple model works well as long as the value of the
inductor is not too high and the loop crossover frequency
is much lower than the switching frequency. With a larger
ceramiccapacitor(verylowESR),crossovermaybelower
and a phase lead capacitor (CPL) across the feedback
divider may improve the phase margin and transient
response. Large electrolytic capacitors may have an ESR
large enough to create an additional zero, and the phase
lead may not be necessary.
If the output capacitor is different than the recommended
capacitor, stability should be checked across all operating
conditions, including load current, input voltage and tem-
perature. The LT1375 data sheet contains a more thor-
oughdiscussionofloopcompensationanddescribeshow
to test the stability using a transient load.
Frequency Compensation
The LT1933 uses current mode control to regulate the
output. This simplifies loop compensation. In particular,
the LT1933 does not require the ESR of the output
capacitor for stability allowing the use of ceramic capaci-
tors to achieve low output ripple and small circuit size.
PCB Layout
For proper operation and minimum EMI, care must be
taken during printed circuit board layout. Figure 8 shows
the recommended component placement with trace,
ground plane and via locations. Note that large, switched
currents flow in the LT1933’s VIN and SW pins, the catch
diode (D1) and the input capacitor (C2). The loop formed
by these components should be as small as possible and
tied to system ground in only one place. These compo-
nents, along with the inductor and output capacitor,
Figure 7 shows an equivalent circuit for the LT1933
control loop. The error amp is a transconductance ampli-
fier with finite output impedance. The power section,
consistingofthemodulator, powerswitchandinductor, is
modeled as a transconductance amplifier generating an
output current proportional to the voltage at the VC node.
Note that the output capacitor integrates this current, and
CURRENT MODE
POWER STAGE
LT1933
–
+
0.7V
SW
g
OUT
m
C
PL
R1
1.1mho
C
SHUTDOWN
–
FB
g
=
V
IN
V
m
150µmhos
ESR
+
C1
V
OUT
1.245V
R
C
C1
ERROR
AMPLIFIER
+
100k
SYSTEM
GROUND
C2
D1
C1
C
C
500k
80pF
R2
GND
1933 F08
VIAS TO LOCAL GROUND PLANE
OUTLINE OF LOCAL GROUND PLANE
1933 F07
Figure 7. Model for Loop Response
Figure 8. A Good PCB Layout Ensures Proper, Low EMI Operation
1933f
13
LT1933
shouldbeplacedonthesamesideofthecircuitboard, and
their connections should be made on that layer. Place a
local, unbroken ground plane below these components,
andtiethisgroundplanetosystemgroundatonelocation,
ideally at the ground terminal of the output capacitor C1.
The SW and BOOST nodes should be as small as possible.
Finally, keep the FB node small so that the ground pin and
groundtraceswillshielditfromtheSWandBOOSTnodes.
Include two vias near the GND pin of the LT1933 to help
remove heat from the LT1933 to the ground plane.
estimated by calculating the total power loss from an
efficiency measurement and subtracting the catch diode
loss. The resulting temperature rise at full load is nearly
independentofinputvoltage. Thermalresistancedepends
on the layout of the circuit board, but a value of 125°C/W
is typical.
Die temperature rise was measured on a two-layer, five by
five cm circuit board in still air. The LT1933 producing 5V
at 500mA showed a temperature rise of 28°C, allowing it
to deliver full load to 97°C ambient. Above this tempera-
ture the load current should be reduced. For 3.3V at
500mA the temperature rise is 24°C.
High Temperature Considerations
The die temperature of the LT1933 must be lower than the
maximum rating of 125°C. This is generally not a concern
unless the ambient temperature is above 85°C. For higher
temperatures, care should be taken in the layout of the
circuit to ensure good heat sinking of the LT1933. The
maximum load current should be derated as the ambient
temperature approaches 125°C.
Other Linear Technology Publications
Application notes AN19, AN35 and AN44 contain more
detailed descriptions and design information for Buck
regulators and other switching regulators. The LT1376
data sheet has a more extensive discussion of output
ripple, loop compensation and stability testing. Design
Note DN100 shows how to generate a bipolar output
supply using a Buck regulator.
ThedietemperatureiscalculatedbymultiplyingtheLT1933
power dissipation by the thermal resistance from junction
to ambient. Power dissipation within the LT1933 can be
U
TYPICAL APPLICATIO S
1.8V Step-Down Converter
5V Step-Down Converter
D2
D2
5
4
1
V
IN
V
IN
BOOST
6.3V TO 36V
C3
0.1µF
L1
LT1933
33µH
5
4
1
6
6
V
IN
V
BOOST
SW
OFF ON
SHDN
SW
IN
3.6V TO 20V
V
OUT
5V/500mA
C3
0.1µF
L1
LT1933
GND
2
FB
3
R1
30.1k
D1
10µH
OFF ON
SHDN
V
OUT
1.8V/500mA
C2
C1
R2
10k
GND
2
FB
3
R1
4.42k
D1
2.2µF
22µF
6.3V
C2
2.2µF
C1
1933 TA02c
R2
10k
22µF
2x
1933 TA02a
3.3V Step-Down Converter
12V Step-Down Converter
D2
D3, 6V
D2
5
4
1
5
4
1
V
V
IN
IN
V
IN
BOOST
V
IN
BOOST
4.5V TO 36V
14.5V TO 36V
C3
0.1µF
C3
0.1µF
L1
L1
47µH
LT1933
LT1933
22µH
6
6
OFF ON
SHDN
SW
OFF ON
SHDN
SW
V
V
OUT
OUT
3.3V/500mA
12V/450mA
GND
2
FB
3
GND
2
FB
3
R1
16.5k
D1
R1
86.6k
D1
C2
2.2µF
C1
22µF
6.3V
C2
2.2µF
C1
10µF
R2
10k
R2
10k
1933 TA02b
1933 TA02d
1933f
14
LT1933
U
PACKAGE DESCRIPTION
S6 Package
6-Lead Plastic SOT-23
(Reference LTC DWG # 05-08-1634)
2.80 – 3.10
(NOTE 4)
0.62
MAX
0.95
REF
1.22 REF
1.50 – 1.75
(NOTE 4)
2.60 – 3.00
1.4 MIN
3.85 MAX 2.62 REF
PIN ONE ID
0.25 – 0.50
TYP 6 PLCS
NOTE 3
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.95 BSC
0.90 – 1.30
0.20 BSC
DATUM ‘A’
0.90 – 1.45
0.35 – 0.55 REF
1.90 BSC
0.09 – 0.15
0.09 – 0.20
(NOTE 3)
NOTE 3
NOTE:
S6 SOT-23 0502
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
ATTENTION: ORIGINAL SOT23-6L PACKAGE.
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. PACKAGE EIAJ REFERENCE IS SC-74A (EIAJ)
MOST SOT23-6L PRODUCTS CONVERTED TO THIN SOT23
PACKAGE, DRAWING # 05-08-1636 AFTER APPROXIMATELY
APRIL 2001 SHIP DATE
1933f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
15
LT1933
TYPICAL APPLICATIO S
U
2.5V Step-Down Converter
D2
5
4
1
V
IN
V
BOOST
IN
3.6V TO 36V
C3
L1
15µH
LT1933
0.47µF
6
OFF ON
SHDN
SW
V
OUT
2.5V/500mA
GND
2
FB
3
R1
10.5k
D1
C2
2.2µF
C1
22µF
R2
10k
1933 TA03
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V : 3.6V to 25V, V
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Dual 600mA I , 1.5MHz, Synchronous Step-Down
V : 2.5V to 5.5V, V
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V : 2.3V to 5.5V, V
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60V, 2.75A I , 200kHz/500kHz, High Efficiency Step-Down V : 5.5V to 60V, V
= 1.2V, I = 2.5mA, I = 30µA,
Q SD
OUT
IN
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TSSOP16E Package
Burst Mode is a registered trademark of Linear Technology Corporation.
1933f
LT/TP 0704 1K • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
16
●
●
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©LINEAR TECHNOLOGY CORPORATION 2004
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