LT3475EFE-1#TRPBF [Linear]
LT3475/LT3475-1 - Dual Step-Down 1.5A LED Driver; Package: TSSOP; Pins: 20; Temperature Range: -40°C to 85°C;
| 型号： | LT3475EFE-1#TRPBF |
| 厂家： | 
Linear |
| 描述： | LT3475/LT3475-1 - Dual Step-Down 1.5A LED Driver; Package: TSSOP; Pins: 20; Temperature Range: -40°C to 85°C 驱动器 |
| 文件： | 总20页 (文件大小：233K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LT3475/LT3475-1
Dual Step-Down
1.5A LED Driver
U
DESCRIPTIO
FEATURES
True Color PWMTM Delivers Constant Color with
The LT®3475/LT3475-1 are dual step-down DC/DC
converters designed to operate as a constant-current
source. An internal sense resistor monitors the output
current allowing accurate current regulation ideal for
driving high current LEDs. The high side current sense al-
lowsgroundedcathodeLEDoperation.Highoutputcurrent
accuracy is maintained over a wide current range, from
50mA to 1.5A, allowing a wide dimming range. Unique
PWM circuitry allows a dimming range of 3000:1, avoid-
ing the color shift normally associated with LED current
dimming.
■
3000:1 Dimming Range
■
Wide Input Range: 4V to 36V Operating, 40V
Maximum
■
Accurate and Adjustable Control of LED Current
from 50mA to 1.5A
■
High Side Current Sense Allows Grounded Cathode
LED Operation
■
Open LED (LT3475) and Short Circuit Protection
■
LT3475-1 Drives LED Strings Up to 25V
■
Accurate and Adjustable 200kHz to 2MHz
Switching Frequency
The high switching frequency offers several advantages,
permitting the use of small inductors and ceramic capaci-
tors. Small inductors combined with the 20 lead TSSOP
surface mount package save space and cost versus
alternative solutions. The constant switching frequency
combined with low-impedance ceramic capacitors result
in low, predictable output ripple.
■
Anti-Phase Switching Reduces Ripple
■
Uses Small Inductors and Ceramic Capacitors
■
Available in the Compact 20-Lead TSSOP Thermally
Enhanced Surface Mount Package
U
APPLICATIO S
■
Automotive and Avionic Lighting
Withitswideinputrangeof4Vto36V,theLT3475/LT3475-1
regulate a broad array of power sources. A current mode
PWM architecture provides fast transient response and
cycle-by-cycle current limiting. Frequency foldback and
thermal shutdown provide additional protection.
■
Architectural Detail Lighting
■
Display Backlighting
Constant-Current Sources
■
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners. Patents Pending.
U
TYPICAL APPLICATIO
Dual Step-Down 1.5A LED Driver
Efficiency
V
IN
5V TO 36V
4.7μF
95
V
SHDN
BOOST2
IN
V
IN
= 12V
TWO SERIES CONNECTED
WHITE 1.5A LEDS
BOOST1
90
85
80
0.22μF
0.22μF
LT3475
10μH
10μH
SW1
SW2
SINGLE WHITE 1.5A LED
OUT1
LED1
OUT2
LED2
75
70
*DIMMING
CONTROL
DIMMING*
CONTROL
PWM1
PWM2
V
C1
V
C2
65
60
55
REF
R
T
2.2μF
0.1μF
2.2μF
0.1μF
1.5A LED
CURRENT
1.5A LED
CURRENT
V
ADJ1
V
ADJ2
24.3k
GND
0.5
1
0
1.5
LED CURRENT (A)
3475 TA01
*SEE APPLICATIONS SECTION FOR DETAILS
f
= 600kHz
SW
3475 TA01b
3475fb
1
LT3475/LT3475-1
W W
U W
ABSOLUTE AXI U RATI GS
PIN CONFIGURATION
(Note 1)
TOP VIEW
V Pin .........................................................(-0.3V), 40V
IN
OUT1
LED1
1
2
3
4
5
6
7
8
9
20
19
18
17
16
15
14
13
12
11
PWM1
BOOST Pin Voltage ...................................................60V
BOOST Above SW Pin...............................................30V
OUT, LED, Pins (LT3475)...........................................15V
OUT, LED Pins (LT3475-1).........................................25V
PWM Pin...................................................................15V
V
ADJ1
BOOST1
SW1
V
C1
REF
V
SHDN
GND
IN
21
V
IN
SW2
BOOST2
LED2
R
T
V
C
Pin ......................................................................6V
T
V
ADJ
C2
V
V , R , REF Pins..........................................................3V
ADJ2
OUT2 10
PWM2
SHDN Pin...................................................................V
IN
FE PACKAGE
Maximum Junction Temperature (Note 2)............. 125°C
20-LEAD PLASTIC TSSOP
Operating Temperature Range (Note 3)
T
= 125°C, θ = 30°C/W, θ = 8°C/W
JA JC
EXPOSED PAD (PIN 21) IS GROUND AND MUST
BE ELECTRICALLY CONNECTED TO THE PCB.
JMAX
LT3475E/LT3475E-1............................. –40°C to 85°C
LT3475I/LT3475I-1............................. –40°C to 125°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature Range (Soldering, 10 sec) ....... 300°C
ORDER INFORMATION
LEAD FREE FINISH
LT3475EFE#PBF
LT3475IFE#PBF
TAPE AND REEL
PART MARKING*
LT3475EFE
PACKAGE DESCRIPTION
TEMPERATURE RANGE
–40°C to 85°C
LT3475EFE#TRPBF
LT3475IFE#TRPBF
LT3475EFE-1#TRPBF
LT3475IFE-1#TRPBF
20-Lead Plastic TSSOP
20-Lead Plastic TSSOP
20-Lead Plastic TSSOP
20-Lead Plastic TSSOP
LT3475IFE
–40°C to 125°C
–40°C to 85°C
LT3475EFE-1#PBF
LT3475IFE-1#PBF
LT3475FE-1
LT3475FE-1
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3)
PARAMETER
CONDITIONS
MIN
TYP
3.7
6
MAX
UNITS
V
●
Minimum Input Voltage
Input Quiescent Current
Shutdown Current
4
8
2
Not Switching
mA
μA
SHDN = 0.3V, VBOOST = VOUT = 0V
0.01
3475fb
2
LT3475/LT3475-1
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3)
PARAMETER
CONDITIONS
Tied to V • 2/3
MIN
TYP
MAX
UNITS
LED Pin Current
V
ADJ
0.97
0.94
0.336
0.325
0.31
1.00
1.03
1.04
0.364
0.375
0.385
A
A
A
A
A
REF
●
V
Tied to V • 7/30
0.350
ADJ
REF
LT3475E/LT3475E-1 0°C to 85°C
●
●
REF Voltage
1.22
1.25
0.05
0.0002
40
1.27
V
%/V
%/μA
nA
Reference Voltage Line Regulation
Reference Voltage Load Regulation
4V < V < 40V
IN
0 < I < 500μA
REF
●
●
●
V
ADJ
Pin Bias Current (Note 4)
400
640
Switching Frequency
Maximum Duty Cycle
R = 24.3k
T
530
90
600
kHz
R = 24.3k
95
80
98
%
%
%
T
R = 4.32k
T
R = 100k
T
Switching Phase
R = 24.3k
150
180
80
210
Deg
kHz
V
T
Foldback Frequency
SHDN Threshold (to Switch)
SHDN Pin Current (Note 5)
PWM Threshold
R = 24.3k, V
T
= 0V
OUT
2.5
7
2.6
9
2.74
11
V
SHDN
=
2.6V
μA
V
0.3
0.8
0.8
50
1.2
V Switching Threshold
C
V
V Source Current
C
V = 1V
C
μA
μA
V/A
mA/μA
A/V
V
V Sink Current
C
V = 1V
C
50
LED to V Transresistance
500
1
C
LED to V Current Gain
C
V to Switch Current Gain
C
2.6
1.8
10
V Clamp Voltage
C
●
●
V Pin Current in PWM Mode
C
V = 1V, V = 0.3V
C PWM
400
14.5
50
nA
V
OUT Pin Clamp Voltage (LT3475)
OUT Pin Current in PWM Mode
Switch Current Limit (Note 6)
13.5
2.3
14
V
OUT
= 4V, V
= 0.3V
PWM
25
μA
A
2.7
350
25
3.2
500
40
Switch V
I
I
=1.5A
=1.5A
mV
mA
μA
V
CESAT
SW
BOOST Pin Current
SW
Switch Leakage Current
Minimum Boost Voltage Above SW
0.1
1.8
10
2.5
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
Note 3: The LT3475E and LT3475E-1 are guaranteed to meet performance
specifications from 0°C to 85°C. Specifications over the –40°C to 85°C
operating temperature range are assured by design, characterization and
correlation with statistical process controls. The LT3475I and LT3475I-1
are guaranteed to meet performance specifications over the –40°C to
125°C operating temperature range.
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
Note 4: Current flows out of pin.
Note 5: Current flows into pin.
Note 6: Current limit is guaranteed by design and/or correlation to static
test. Slope compensation reduces current limit at higher duty cycles.
3475fb
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LT3475/LT3475-1
U W
TYPICAL PERFOR A CE CHARACTERISTICS
LED Current vs VADJ
LED Current vs Temperature
Switch On Voltage
1.50
1.25
1.2
1.0
0.8
0.6
600
500
400
300
200
100
0
T
= 25°C
T
= 25°C
A
A
V
ADJ
= V
• 2/3
REF
1.00
0.75
V
ADJ
= V
• 7/30
REF
0.50
0.25
0
0.4
0.2
0
1.5
SWITCH CURRENT (A)
50
TEMPERATURE (˚C)
100 125
0
0.5
1.0
2.0
0
0.25
0.5
V
0.75
(V)
1
1.25
–50 –25
0
25
75
ADJ
3475 G01
3475 G02
3475 G03
Switch Current Limit
vs Duty Cycle
Switch Current Limit vs
Temperature
Current Limit vs Output Voltage
3.0
3.5
3.0
T
= 25°C
A
TYPICAL
3.0
2.5
2.5
2.0
1.5
1.0
0.5
0
2.5
2.0
1.5
1.0
0.5
2.0
1.5
MINIMUM
1.0
0.5
0
T
= 25°C
A
0
50
TEMPERATURE (°C)
100 125
1.5
0
20
40
60
80
100
–50 –25
0
25
75
0
0.5 1.0
2.0 2.5 3.0 3.5 4.0
V (V)
OUT
DUTY CYCLE (%)
3475 G04
3475 G05
3475 G06
Oscillator Frequency
vs Temperature
Oscillator Frequency Foldback
Oscillator Frequency vs RT
700
650
600
550
700
600
T
= 25°C
= 24.3kΩ
R
= 24.3kΩ
T
= 25°C
A
T
T
A
R
1000
500
400
300
200
100
0
500
450
400
10
50
TEMPERATURE (˚C)
100 125
0.5
1.0
V
1.5
(V)
2.5
1
10
100
–50 –25
0
25
75
0
2.0
R
(kΩ)
T
OUT
3475 G09
3475 G07
3475 G08
3475fb
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LT3475/LT3475-1
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Open-Circuit Output Voltage and
Input Current
Boost Pin Current
Quiescent Current
7
6
35
30
50
45
40
14
T
= 25°C
T
= 25°C
T
= 25°C
A
A
A
INPUT CURRENT
LT3475-1
LT3475
12
10
8
5
25
35
30
25
20
15
10
5
4
3
2
1
20
15
10
5
LT3475-1
6
OUTPUT VOLTAGE
LT3475
4
2
0
0
0
0
10
20
(V)
40
0
30
0.5
1.0
2.0
20
(V)
0
1.5
0
10
30
40
V
SWITCH CURRENT (A)
V
IN
IN
3475 G11
3475 G10
3475 G12
Minimum Input Voltage, Single
1.5A White LED
Minimum Input Voltage, Two Series
Connected 1.5A White LEDs
Reference Voltage
1.28
1.27
1.26
1.25
10
9
6
5
4
3
2
1
0
T
= 25°C
T
= 25°C
A
A
TO START
TO RUN
8
TO START
LED VOLTAGE
LED VOLTAGE
7
TO RUN
1.24
1.23
1.22
6
5
50
100 125
0
0.5
1
1.5
–50 –25
0
25
75
0
0.5
1
1.5
LED CURRENT (A)
TEMPERATURE (˚C)
LED CURRENT (A)
3475 G13
3475 G15
3475 G14
U
U
U
PI FU CTIO S
OUT1, OUT2 (Pins 1, 10): The OUT pin is the input to the
current sense resistor. Connect this pin to the inductor
and the output capacitor.
BOOST1, BOOST2 (Pins 3, 8): The BOOST pin is used to
provide a drive voltage, higher than the input voltage, to
the internal bipolar NPN power switch.
LED1, LED2 (Pins 2, 9): The LED pin is the output of
the current sense resistor. Connect the anode of the LED
here.
GND (Pins 15, Exposed Pad Pin 21): Ground. Tie the GND
pin and the exposed pad directly to the ground plane. The
exposed pad metal of the package provides both electrical
contact to ground and good thermal contact to the printed
circuit board. The exposed pad must be soldered to the
circuitboardforproperoperation.Usealargegroundplane
and thermal vias to optimize thermal performance.
V (Pins 5, 6): The V pins supply current to the internal
IN
IN
circuitry and to the internal power switches and must be
locally bypassed.
SW1, SW2 (Pins 4, 7): The SW pin is the output of the
internal power switch. Connect this pin to the inductor,
switching diode and boost capacitor.
3475fb
5
LT3475/LT3475-1
U
U
U
PI FU CTIO S
R (Pin 14): The R pin is used to set the internal
V , V (Pins 18, 13): The V pin is the output of the
C1 C2 C
T
T
oscillator frequency. Tie a 24.3k resistor from R to GND
internal error amp. The voltage on this pin controls the
peak switch current. Use this pin to compensate the
control loop.
T
for a 600kHz switching frequency.
SHDN (Pin 16): The SHDN pin is used to shut down the
switching regulator and the internal bias circuits. The
2.6V switching threshold can function as an accurate
undervoltage lockout. Pull below 0.3V to shut down the
LT3475/LT3475-1. Pull above 2.6V to enable the LT3475/
V
, V
(Pins 19, 12): The V
pin is the input to
ADJ1 ADJ2
ADJ
the internal voltage-to-current amplifier. Connect the V
ADJ
pin to the REF pin for a 1.5A output current. For lower
output currents, program the V pin using the following
ADJ
LT3475-1. Tie to V if the SHDN function is unused.
formula: I
= 1.5A • V /1.25V.
IN
LED ADJ
REF (Pin 17): The REF pin is the buffered output of the
PWM1, PWM2 (Pins 20, 11): The PWM pin controls the
internal reference. Either tie the REF pin to the V
pin
connection of the V pin to the internal circuitry. When
ADJ
C
for a 1.5A output current, or use a resistor divider to
the PWM pin is low, the V pin is disconnected from the
C
generate a lower voltage at the V
unconnected if unused.
pin. Leave this pin
internal circuitry and draws minimal current. If the PWM
ADJ
feature is unused, leave this pin unconnected.
BLOCK DIAGRAM
V
IN
R
T
C
IN
V
SHDN
R
V
IN
IN
T
INT REG
MASTER
OSC
AND
UVLO
D1
D2
BOOST2
BOOST1
C1
C2
SLOPE COMP
SLOPE COMP
C1
C2
∑
∑
Q
R
S
R
S
Q
Q
MOSC 1
MOSC 2
Q
Q1
Q2
SLAVE
OSC
SLAVE
OSC
SW1
SW2
L1
L2
DRIVER
DRIVER
D3
D4
FREQUENCY
FOLDBACK
FREQUENCY
FOLDBACK
OUT1
LED1
OUT2
LED2
–
+
–
+
C
OUT1
C
OUT2
0.067Ω
gm1
100Ω
100Ω
0.067Ω
gm2
2V
2V
D
D
LED 2
LED1
1.25V
PWM 1
PWM2
Q3
Q4
V
C1
V
C2
1.25k
1.25k
C
C1
C
C2
V
V
REF
ADJ2
GND
EXPOSED
PAD
ADJ1
3475 BD
3475fb
6
LT3475/LT3475-1
OPERATION
The LT3475 is a dual constant frequency, current mode
regulator with internal power switches capable of gen-
erating constant 1.5A outputs. Operation can be best
understood by referring to the Block Diagram.
programming of LED pin currents of less than 1.5A. LED
pin current can also be programmed by tying the V pin
ADJ
directly to a voltage source.
An LED can be dimmed with pulse width modulation
using the PWM pin and an external NFET. If the PWM
pin is unconnected or is pulled high, the part operates
If the SHDN pin is tied to ground, the LT3475 is shut
down and draws minimal current from the input source
nominally. If the PWM pin is pulled low, the V pin is dis-
tied to V . If the SHDN pin exceeds 1V, the internal bias
C
IN
connected from the internal circuitry and draws minimal
current from the compensation capacitor. Circuitry draw-
ing current from the OUT pin is also disabled. This way,
circuits turn on, including the internal regulator, reference
and oscillator. The switching regulators will only begin to
operate when the SHDN pin exceeds 2.6V.
the V pin and the output capacitor store the state of
C
Theswitcherisacurrentmoderegulator.Insteadofdirectly
modulatingthedutycycleofthepowerswitch,thefeedback
loop controls the peak current in the switch during each
cycle. Compared to voltage mode control, current mode
control improves loop dynamics and provides cycle-by-
cycle current limit.
the LED pin current until the PWM is pulled high again.
This leads to a highly linear relationship between pulse
width and output light, allowing for a large and accurate
dimming range.
TheR pinallowsprogrammingoftheswitchingfrequency.
T
Forapplicationsrequiringthesmallestexternalcomponents
possible, a fast switching frequency can be used. If low
dropout or very high input voltages are required, a slower
switching frequency can be programmed.
A pulse from the oscillator sets the RS flip-flop and turns
on the internal NPN bipolar power switch. Current in the
switch and the external inductor begins to increase. When
this current exceeds a level determined by the voltage at
During startup V
will be at a low voltage. The NPN,
V , current comparator C1 resets the flip-flop, turning
OUT
C
Q3, can only operate correctly with sufficient voltage
off the switch. The current in the inductor flows through
the external Schottky diode and begins to decrease. The
cycle begins again at the next pulse from the oscillator.
of ≈1.7V at V , A comparator senses V
and forces
OUT
OUT
the V pin high until V
rises above 2V, and Q3 is op-
OUT
C
erating correctly.
In this way, the voltage on the V pin controls the current
C
through the inductor to the output. The internal error
amplifier regulates the output current by continually
The switching regulator performs frequency foldback
during overload conditions. An amplifier senses when
OUT
adjusting the V pin voltage. The threshold for switching
C
V
is less than 2V and begins decreasing the oscillator
on the V pin is 0.8V, and an active clamp of 1.8V limits
C
frequencydownfromfullfrequencyto15%ofthenominal
frequency when V = 0V. The OUT pin is less than 2V
the output current.
OUT
during startup, short circuit, and overload conditions.
Frequency foldback helps limit switch current under these
conditions.
The voltage on the V
pin sets the current through the
ADJ
LED pin. The NPN, Q3, pulls a current proportional to the
voltage on the V pin through the 100Ω resistor. The gm
ADJ
amplifier servos the V pin to set the current through the
The switch driver operates either from V or from the
C
IN
0.067Ω resistor and the LED pin. When the voltage drop
across the 0.067Ω resistor is equal to the voltage drop
across the 100Ω resistor, the servo loop is balanced.
BOOST pin. An external capacitor and Schottky diode
are used to generate a voltage at the BOOST pin that
is higher than the input supply. This allows the driver
to saturate the internal bipolar NPN power switch for
efficient operation.
Tying the REF pin to the V pin sets the LED pin current
to 1.5A. Tying a resistor divider to the REF pin allows the
ADJ
3475fb
7
LT3475/LT3475-1
APPLICATIONS INFORMATION
Open Circuit Protection
OUT
10k
22V
TheLT3475hasinternalopen-circuitprotection. IftheLED
is absent or is open circuit, the LT3475 clamps the voltage
on the LED pin at 14V. The switching regulator then oper-
ates at a very low frequency to limit the input current. The
LT3475-1 has no internal open circuit protection. With the
LT3475-1, be careful not to violate the ABSMAX voltage of
V
C
100k
3475 F01
th BOOST pin; if V > 25V, external open circuit protection
IN
Figure 1. External Overvoltage Protection
Circuitry for the LT3475-1
circuitry (as shown in Figure 1) may be necessary.The
output voltage during an open LED condition is shown in
the Typical Performance Characteristics section.
LT3475
V
IN
V
IN
2.6V
Undervoltage Lockout
R1
R2
V
C
SHDN
Undervoltagelockout(UVLO)istypicallyusedinsituations
wheretheinputsupplyiscurrentlimited,orhashighsource
resistance. A switching regulator draws constant power
from the source, so the source current increases as the
source voltage drops. This looks like a negative resistance
loadtothesourceandcancausethesourcetocurrentlimit
or latch low under low source voltage conditions. UVLO
prevents the regulator from operating at source voltages
where these problems might occur.
9μA
C1
GND
3475 F02
Figure 2. Undervoltage Lockout
Keep the connections from the resistors to the SHDN pin
short and make sure the coupling to the SW and BOOST
pins is minimized. If high resistance values are used, the
SHDN pin should be bypassed with a 1nF capacitor to
prevent coupling problems from switching nodes.
An internal comparator will force the part into shut-
down when V falls below 3.7V. If an adjustable UVLO
IN
threshold is required, the SHDN pin can be used. The
threshold voltage of the SHDN pin comparator is 2.6V. An
internal resistor pulls 9μA to ground from the SHDN pin
at the UVLO threshold.
Setting the Switching Frequency
The LT3475 uses a constant frequency architecture that
can be programmed over a 200kHz to 2MHz range with a
single external timing resistor from the R pin to ground.
Choose resistors according to the following formula:
T
A graph for selecting the value of R for a given operating
T
2.6V
R2 =
frequency is shown in the Typical Applications section.
VTH – 2.6V
– 9μA
Table 1. Switching Frequencies
SWITCHING FREQUENCY (MHz)
R1
R (kΩ)
T
V
= UVLO Threshold
TH
2
4.32
6.81
9.09
11.8
16.9
24.3
40.2
57.6
100
1.5
1.2
1
Example: Switching should not start until the input is
above 8V.
V
TH
= 8V
0.8
0.6
0.4
0.3
0.2
R1=100k
2.6V
8V – 2.6V
100k
R2 =
= 57.6k
– 9μA
3475fb
8
LT3475/LT3475-1
APPLICATIONS INFORMATION
Table 1 shows suggested R selections for a variety of
The maximum operating voltage is determined by the
T
switching frequencies.
absolute maximum ratings of the V and BOOST pins,
IN
and by the minimum duty cycle.
Operating Frequency Selection
V
OUT + V
DCMIN
F
V
=
– V + VSW
F
IN MAX
(
)
The choice of operating frequency is determined by
several factors. There is a tradeoff between efficiency and
component size. A higher switching frequency allows the
use of smaller inductors at the cost of increased switching
losses and decreased efficiency.
with DC
where t
= t
• f
MIN
ON(MIN)
is equal to 140ns and f is the switching
ON(MIN)
frequency.
Anotherconsiderationisthemaximumdutycycle.Incertain
applications, the converter needs to operate at a high duty
cycle in order to work at the lowest input voltage possible.
The LT3475 has a fixed oscillator off time and a variable
on time. As a result, the maximum duty cycle increases
as the switching frequency is decreased.
Example: f = 750kHz, V
= 3.4V
OUT
DCMIN =140ns •750kHz = 0.105
3.4V + 0.4V
V
=
– 0.4V + 0.4V = 36V
IN MAX
(
)
0.105
The minimum duty cycle depends on the switching fre-
quency. Running at a lower switching frequency might
allow a higher maximum operating voltage. Note that
this is a restriction on the operating input voltage; the
circuit will tolerate transient inputs up to the Absolute
Input Voltage Range
Theminimumoperatingvoltageisdeterminedeitherbythe
LT3475’s undervoltage lockout of 4V, or by its maximum
duty cycle. The duty cycle is the fraction of time that the
internal switch is on and is determined by the input and
output voltages:
Maximum Ratings of the V and BOOST pins. The input
IN
voltage should be limited to the V operating range (36V)
IN
during overload conditions (short circuit or start up).
V
+ VF
(
)
OUT
DC =
V – V + VF
(
)
IN
SW
Minimum On Time
The LT3475 will regulate the output current at input volt-
where V is the forward voltage drop of the catch diode
F
ages greater than V
. For example, an application
IN(MAX)
(~0.4V) and V is the voltage drop of the internal switch
SW
with an output voltage of 3V and switching frequency of
(~0.4V at maximum load). This leads to a minimum input
voltage of:
1.2MHz has a V
of 20V, as shown in Figure 3. Figure
IN(MAX)
4 shows operation at 35V. Output ripple and peak inductor
V
OUT + V
F
V
=
– V + VSW
F
IN MIN
(
)
DCMAX
with DC
where t
= 1–t
• f
V
OUT
MAX
OFF(MIN)
500mV/DIV
(AC COUPLED)
is equal to 167ns and f is the switching
0FF(MIN)
I
L
frequency.
1A/DIV
Example: f = 600kHz, V
= 4V
OUT
V
SW
20V/DIV
DCMAX =1−167ns •600kHz = 0.90
4V + 0.4V
3475 F03
V
=
– 0.4V + 0.4V = 4.9V
IN MIN
(
)
0.9
Figure 3. Operation at VIN(MAX) = 20V.
VOUT = 3V and fSW = 1.2MHHz
3475fb
9
LT3475/LT3475-1
APPLICATIONS INFORMATION
current have significantly increased. Exceeding V
IN(MAX)
Table 2. Inductors
is safe if the external components have adequate ratings
to handle the peak conditions and if the peak inductor
current does not exceed 3.2A. A saturating inductor may
further reduce performance.
VALUE
I
DCR
( )
HEIGHT
(mm)
RMS
PART NUMBER
Sumida
(μH)
(A)
CR43-3R3
3.3
4.7
3.3
3.3
4.7
5.0
5.6
10
1.44
1.15
1.10
1.57
1.32
2.20
2.0
0.086
0.109
0.063
0.049
0.072
0.032
0.036
0.048
0.076
0.072
0.130
0.050
3.5
3.5
1.8
3.0
3.0
2.8
2.8
3.0
3.0
3.4
3.4
4.0
CR43-4R7
CDRH4D16-3R3
CDRH4D28-3R3
CDRH4D28-4R7
CDRH6D26-5R0
CDRH6D26-5R6
CDRH5D28-100
CDRH5D28-150
CDRH73-100
CDRH73-150
CDRH104R-150
Coilcraft
V
OUT
500mV/DIV
(AC COUPLED)
I
L
1A/DIV
1.30
1.10
1.68
1.33
3.1
V
SW
20V/DIV
15
10
3475 F04
15
Figure 4. Operation above VIN(MAX). Output
Ripple and Peak Inductor Current Increases
15
DO1606T-332
DO1606T-472
DO1608C-332
DO1608C-472
MOS6020-332
MOS6020-472
DO3316P-103
DO3316P-153
3.3
4.7
3.3
4.7
3.3
10
1.30
1.10
2.00
1.50
1.80
1.50
3.9
0.100
0.120
0.080
0.090
0.046
0.050
0.038
0.046
2.0
2.0
2.9
2.9
2.0
2.0
5.2
5.2
Inductor Selection and Maximum Output Current
A good first choice for the inductor value is:
1.2MHz
L = (VOUT + VF )•
f
where V is the voltage drop of the catch diode (~0.4V),
F
10
f is the switching frequency and L is in μH. With this value
the maximum load current will be above 1.6A at all duty
cycles. The inductor’s RMS current rating must be greater
than the maximum load current and its saturation current
should be at least 30% higher. For highest efficiency,
the series resistance (DCR) should be less than 0.15Ω.
Table 2 lists several vendors and types that are suitable.
For robust operation at full load and high input voltages
15
3.1
The optimum inductor for a given application may differ
from the one indicated by this simple design guide. A larger
valueinductorprovidesahighermaximumloadcurrent, and
reduces the output voltage ripple. If your load is lower than
themaximumloadcurrent,thenyoucanrelaxthevalueofthe
inductor and operate with higher ripple current. This allows
you to use a physically smaller inductor, or one with a lower
DCRresultinginhigherefficiency.Inaddition,lowinductance
may result in discontinuous mode operation, which further
reduces maximum load current. For details of maximum
outputcurrentanddiscontinuousmodeoperation,seeLinear
Technology’s Application Note 44. Finally, for duty cycles
(V > 30V), use an inductor with a saturation current
IN
higher than 3.2A.
greater than 50% (V /V > 0.5), a minimum inductance
is required to avoid sub-harmonic oscillations:
OUT IN
800kHz
LMIN = (VOUT + VF )•
f
3475fb
10
LT3475/LT3475-1
APPLICATIONS INFORMATION
Thecurrentintheinductorisatrianglewavewithanaverage
value equal to the load current. The peak switch current
is equal to the output current plus half the peak-to-peak
inductor ripple current. The LT3475 limits its switch cur-
rentinordertoprotectitselfandthesystemfromoverload
faults. Therefore, the maximum output current that the
LT3475 will deliver depends on the switch current limit,
the inductor value, and the input and output voltages.
Input Capacitor Selection
Bypass the input of the LT3475 circuit with a 4.7μF or
higher ceramic capacitor of X7R or X5R type. A lower
value or a less expensive Y5V type will work if there is
additional bypassing provided by bulk electrolytic capaci-
tors or if the input source impedance is low. The following
paragraphs describe the input capacitor considerations in
more detail.
When the switch is off, the potential across the inductor
is the output voltage plus the catch diode drop. This gives
the peak-to-peak ripple current in the inductor
Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input ca-
pacitor is required to reduce the resulting voltage ripple at
the LT3475 input and to force this switching current into a
tight local loop, minimizing EMI. The input capacitor must
have low impedance at the switching frequency to do this
effectively, and it must have an adequate ripple current rat-
ing. With two switchers operating at the same frequency
but with different phases and duty cycles, calculating the
input capacitor RMS current is not simple. However, a
conservativevalueistheRMSinputcurrentforthechannel
1– DC V
)(
+ VF
(
)
OUT
ΔIL =
L • f
(
)
where f is the switching frequency of the LT3475 and L
is the value of the inductor. The peak inductor and switch
current is
ΔIL
2
ISW PK = IL PK = IOUT
+
(
)
(
)
that is delivering most power (V
• I ):
OUT OUT
To maintain output regulation, this peak current must be
less than the LT3475’s switch current limit I . I is at
LIM LIM
VOUT(V – VOUT
)
IOUT
2
least2.3Aatlowdutycyclesanddecreaseslinearlyto1.8A
at DC = 0.9. The maximum output current is a function of
the chosen inductor value:
IN
CINRMS = IOUT
•
<
V
IN
ΔIL
2
and is largest when V = 2V
(50% duty cycle). As the
IN
OUT
IOUT MAX = ILIM
–
(
)
second, lower power channel draws input current, the
input capacitor’s RMS current actually decreases as the
out-of-phase current cancels the current drawn by the
higher power channel. Considering that the maximum
load current from a single channel is ~1.5A, RMS ripple
current will always be less than 0.75A.
ΔIL
2
= 2.3A• 1–0.25•DC –
(
)
Choosing an inductor value so that the ripple current is
smallwillallowamaximumoutputcurrentneartheswitch
current limit.
The high frequency of the LT3475 reduces the energy
storage requirements of the input capacitor, so that the
capacitance required is less than 10μF. The combination
of small size and low impedance (low equivalent series
resistance or ESR) of ceramic capacitors makes them the
preferred choice. The low ESR results in very low voltage
ripple. Ceramic capacitors can handle larger magnitudes
of ripple current than other capacitor types of the same
value. Use X5R and X7R types.
One approach to choosing the inductor is to start with the
simple rule given above, look at the available inductors,
and choose one to meet cost or space goals. Then use
these equations to check that the LT3475 will be able to
deliver the required output current. Note again that these
equations assume that the inductor current is continu-
ous. Discontinuous operation occurs when I
is less
OUT
than ΔI /2.
L
3475fb
11
LT3475/LT3475-1
APPLICATIONS INFORMATION
An alternative to a high value ceramic capacitor is a
lower value ceramic along with a larger electrolytic
capacitor.Theelectrolyticcapacitorlikelyneedstobegreater
than 10μF in order to meet the ESR and ripple current
requirements. The input capacitor is likely to see high
surge currents when the input source is applied. Tanta-
lum capacitors can fail due to an over-surge of current.
Only use tantalum capacitors with the appropriate surge
current rating. The manufacturer may also recommend
operation below the rated voltage of the capacitor.
RMS current rating of the output capacitor is usually not
of concern. It can be estimated with the formula:
IC(RMS) = ΔIL / 12
The low ESR and small size of ceramic capacitors make
them the preferred type for LT3475 applications. Not all
ceramic capacitors are the same, however. Many of the
higher value capacitors use poor dielectrics with high
temperature and voltage coefficients. In particular Y5V
and Z5U types lose a large fraction of their capacitance
with applied voltage and at temperature extremes.
Because loop stability and transient response depend on
A final caution is in order regarding the use of ceramic
capacitors at the input. A ceramic input capacitor can
combine with stray inductance to form a resonant tank
circuit. If power is applied quickly (for example by plug-
ging the circuit into a live power source) this tank can ring,
doubling the input voltage and damaging the LT3475. The
solution is to either clamp the input voltage or dampen the
tank circuit by adding a lossy capacitor in parallel with the
ceramic capacitor. For details, see Application Note 88.
the value of C , this loss may be unacceptable. Use X7R
OUT
and X5R types. Table 3 lists several capacitor vendors.
Table 3. Low ESR Surface Mount Capacitors.
VENDOR
Taiyo-Yuden
AVX
TYPE
SERIES
X5R, X7R
X5R, X7R
X5R, X7R
Ceramic
Ceramic
Ceramic
TDK
Output Capacitor Selection
For most LEDs, a 2.2μF, 6.3V ceramic capacitor (X5R or
X7R) at the output results in very low output voltage ripple
and good transient response. Other types and values will
also work. The following discusses tradeoffs in output
ripple and transient performance.
Diode Selection
The catch diode (D3 from the Block Diagram) conducts
current only during switch off time. Average forward cur-
rent in normal operation can be calculated from:
The output capacitor filters the inductor current to
generate an output with low voltage ripple. It also stores
energy in order to satisfy transient loads and stabilizes the
LT3475’s control loop. Because the LT3475 operates at a
high frequency, minimal output capacitance is necessary.
In addition, the control loop operates well with or without
the presence of output capacitor series resistance (ESR).
Ceramic capacitors, which achieve very low output ripple
and small circuit size, are therefore an option.
I
= I
(V – V )/V
D(AVG)
OUT IN OUT IN
The only reason to consider a diode with a larger current
rating than necessary for nominal operation is for the
worst-case condition of shorted output. The diode cur-
rent will then increase to one half the typical peak switch
current limit.
Peak reverse voltage is equal to the regulator input
voltage. Use a diode with a reverse voltage rating greater
than the input voltage. Table 4 lists several Schottky
diodes and their manufacturers.
You can estimate output ripple with the following
equation:
Diode reverse leakage can discharge the output capacitor
during LED off times while PWM dimming. If operating at
high ambient temperatures, use a low leakage Schottky
for the widest PWM dimming range.
V
= ΔI / (8 • f • C ) for ceramic capacitors
L OUT
RIPPLE
where ΔI is the peak-to-peak ripple current in the
L
inductor. The RMS content of this ripple is very low so the
3475fb
12
LT3475/LT3475-1
APPLICATIONS INFORMATION
Table 4. Schottky Diodes
pin for full efficiency. For outputs of 3.3V and higher, the
standard circuit (Figure 5a) is best. For outputs between
2.8V and 3.3V, use a small Schottky diode (such as the
BAT-54). Forloweroutputvoltages, theboostdiodecanbe
tiedtotheinput(Figure5b).ThecircuitinFigure5aismore
efficient because the BOOST pin current comes from a
lower voltage source. The anode of the boost diode can
be tied to another source that is at least 3V. For example, if
youaregeneratinga3.3Voutput, andthe3.3Voutputison
whenever the LED is on, the BOOST pin can be
connected to the 3.3V output. For LT3475-1 applications
with higher output voltages, an additional Zener diode
may be necessary (Figure 5d) to maintain pin voltage
below the absolute maximum. In any case, be sure that
the maximum voltage at the BOOST pin is both less than
60V and the voltage difference between the BOOST and
SW pins is less than 30V.
V at 1A
(mV)
V
I
(A)
V at 2A
F
R
AVE
F
(V)
(A)
(mV)
On Semiconductor
MBR0540
MBRM120E
MBRM140
Diodes Inc
B120
40
20
40
0.5
1
620
530
550
1
20
30
40
40
40
1
1
500
500
530
510
B130
B140HB
1
DFLS140
1.1
2
B240
500
International Rectifier
10BQ030
30
1
420
BOOST Pin Considerations
The minimum operating voltage of an LT3475 application
is limited by the undervoltage lockout (~3.7V) and by the
maximum duty cycle. The boost circuit also limits the
minimum input voltage for proper start up. If the input
voltage ramps slowly, or the LT3475 turns on when the
output is already in regulation, the boost capacitor may
not be fully charged. Because the boost capacitor charges
The capacitor and diode tied to the BOOST pin gener-
ate a voltage that is higher than the input voltage. In
most cases, a 0.22μF capacitor and fast switching diode
(such as the CMDSH-3 or MMSD914LT1) will work well.
Figure 5 shows three ways to arrange the boost circuit.
The BOOST pin must be more than 2.5V above the SW
D2
D2
C3
C3
BOOST
LT3475
BOOST
LT3475
V
IN
V
V
V
V
SW
V
SW
OUT
IN
OUT
IN
IN
GND
GND
V
– V ≅ V
V
– V ≅ V
SW IN
BOOST
MAX V
SW
OUT
BOOST
MAX V ≅ 2V
BOOST IN
≅ V + V
BOOST
IN
OUT
(5a)
(5b)
D2
D2
V
> 3V
IN2
BOOST
LT3475
BOOST
LT3475
C3
C3
V
V
V
V
OUT
V
SW
V
SW
IN
OUT
IN
IN
IN
GND
GND
3475 F05
3475 F05
V
– V ≅ V
V
– V – V
SW Z
BOOST
MAX V
SW
IN2
BOOST
MAX V
≅ V + V
≅ V + V
– V
OUT Z
BOOST
IN2
IN
BOOST
IN
MINIMUM VALUE FOR V = 3V
IN2
(5c)
(5d)
Figure 5. Generating the Boost Voltage
3475fb
13
LT3475/LT3475-1
APPLICATIONS INFORMATION
with the energy stored in the inductor, the circuit will rely
on some minimum load current to get the boost circuit
running properly. This minimum load will depend on input
and output voltages, and on the arrangement of the boost
circuit. The minimum load current generally goes to zero
once the circuit has started. The typical performance char-
acteristics section shows a plot of minimum load to start
and to run as a function of input voltage. Even without an
output load current, in many cases the discharged output
capacitor will present a load to the switcher that will allow
the voltage on the V
pin by tying a low on resistance
ADJ
FET to the resistor divider string. This allows the se-
lection of two different LED currents. For reliable op-
eration program an LED current of no less than 50mA.
The maximum current dimming ratio (I
) can be
MAX
RATIO
calculated from the maximum LED current (I
) and the
minimum LED current (I ) as follows:
MIN
I
/I
= I
MAX MIN RATIO
Another dimming control circuit (Figure 8) uses the PWM
pin and an external NFET tied to the cathode of the LED.
An external PWM signal is applied to the PWM pin and the
gate of the NFET (For PWM dimming ratios of 20 to 1 or
less, theNFETcanbeomitted). TheaverageLEDcurrentis
proportionaltothedutycycleofthePWMsignal.Whenthe
PWM signal goes low, the NFET turns off, turning off the
LED and leaving the output capacitor charged. The PWM
it to start. The plots show the worst case, where V is
ramping very slowly.
IN
Programming LED Current
The LED current can be set by adjusting the voltage on the
V
pin. For a 1.5A LED current, either tie V to REF or
ADJ
ADJ
to a 1.25V source. For lower output currents, program the
V
ADJ
using the following formula: I = 1.5A • V /1.25V.
pin is pulled low as well, which disconnects the V pin,
LED
ADJ
C
Voltages less than 1.25V can be generated with a voltage
divider from the REF pin, as shown in Figure 6. In order
to have accurate LED current, precision resistors are
preferred (1% or better is recommended). Note that the
storingthevoltageinthecapacitortiedthere.UsetheC-RC
string shown in Figure 8 and Figure 9 tied to the V pin for
C
proper operation during startup. When the PWM pin goes
high again, the LED current returns rapidly to its previous
onstatesincethecompensationandoutputcapacitorsare
at the correct voltage. This fast settling time allows the
V
pin sources a small amount of bias current, so use
ADJ
the following formula to choose resistors:
VADJ
1.25V – VADJ
R2 =
REF
+ 50nA
R1
R1
LT3475
V
ADJ
To minimize the error from variations in V pin current,
GND
ADJ
R2
3475 F07
use resistors with a parallel resistance of less than 4k. Use
resistorstringswithahighenoughseriesresistancesoasnot
to exceed the 500μA current compliance of the REF pin.
DIM
Dimming Control
Figure 7. Dimming with a MOSFET and Resistor Divider
There are several different types of dimming control
circuits. One dimming control circuit (Figure 7) changes
PWM
100Hz TO
10kHz
PWM
LED
V
C
10k
0.1μF
LT3475
GND
REF
3.3nF
R1
R2
LT3475
GND
3475 F08
V
ADJ
3475 F06
Figure 8. Dimming Using PWM Signal
Figure 6. Setting VADJ with a Resistor Divider
3475fb
14
LT3475/LT3475-1
APPLICATIONS INFORMATION
LT3475 to maintain diode current regulation with PWM
pulse widths as short as 7.5 switching cycles (12.5μs for
Layout Hints
As with all switching regulators, careful attention must
be paid to the PCB layout and component placement. To
maximize efficiency, switch rise and fall times are made
as short as possible. To prevent electromagnetic interfer-
ence (EMI) problems, proper layout of the high frequency
switching path is essential. The voltage signal of the SW
and BOOST pins have sharp rise and fall edges. Minimize
the area of all traces connected to the BOOST and SW
pins and always use a ground plane under the switching
regulator to minimize interplane coupling. In addition, the
f
= 600kHz). Maximum PWM period is determined by
SW
the system and is unlikely to be longer than 12ms. Using
PWM periods shorter than 100μs is not recommended.
The maximum PWM dimming ratio (PWM
) can be
RATIO
calculated from the maximum PWM period (t
) and
MAX
minimum PWM pulse width (t ) as follows:
MIN
t
/t
= PWM
MAX MIN RATIO
Total dimming ratio (DIM
) is the product of the PWM
RATIO
dimming ratio and the current dimming ratio.
ground connection for frequency setting resistor R and
capacitors at V , V pins (refer to the Block Diagram)
T
Example:
C1 C2
should be tied directly to the GND pin and not shared
with the power ground path, ensuring a clean, noise-free
connection.
I
t
I
= 1A, I
= 3.3μs (f = 1.4MHz)
= 0.1A, t
= 9.9ms
MAX
MAX
MIN
MIN
SW
= 1A/0.1A =10:1
RATIO
PWM
= 9.9ms/3.3μs = 3000:1
PWM1
SHDN
PWM2
RATIO
DIM
= 10 • 3000 = 30000:1
RATIO
To achieve the maximum PWM dimming ratio, use the
circuit shown in Figure 9. This allows PWM pulse widths
as short as 4.5 switching cycles (7.5μs for f = 600kHz).
SW
Note that if you use the circuit in Figure 9, the rising edge
of the two PWM signals must align within 100ns.
V
IN
220pF
R
V
C
T
10k
0.1μF
LT3475
GND
1M
3.3nF
R
T
PWM1
3475 F09
3475 F10
VIA TO LOCAL GND PLANE
Figure 9. Extending the PWM Dimming Range
Figure 10. Recommended Component Placement
3475fb
15
LT3475/LT3475-1
TYPICAL APPLICATIONS
Dual Step-Down 1A LED Driver
V
IN
5V TO 36V
C1
4.7μF
50V
D3
V
SHDN
BOOST2
D4
IN
BOOST1
C4
0.22μF
6.3V
C3
0.22μF
6.3V
L2
10μH
L1
10μH
LT3475
SW1
SW2
D1
D2
C5
2.2μF
6.3V
C2
2.2μF
6.3V
OUT1
LED1
OUT2
LED2
LED 1
LED 2
C7
V
V
C2
C1
C6
0.1μF
REF
R
0.1μF
T
R2
1k
V
ADJ1
V
ADJ2
GND
R3
2k
R1
24.3k
3475 TA02
C1 TO C5: X5R OR X7R
D1, D2: DFLS140
f
= 600kHz
SW
D3, D4: MBR0540
LED CURRENT = 1A
Dual Step-Down 1.5A LED Driver with 1200 : 1 True Color PWM Dimming
V
IN
6V TO 36V
C1
4.7μF
50V
D3
V
SHDN
BOOST2
D4
IN
BOOST1
C2
0.22μF
6.3V
C3
0.22μF
6.3V
L2
10μH
L1
10μH
LT3475
SW1
SW2
C4
C5
2.2μF
6.3V
2.2μF
6.3V
D1
D2
OUT1
LED1
PWM1
OUT2
LED2
1.5A LED
CURRENT
LED 1
LED 2
1.5A LED
CURRENT
PWM2
V
V
C2
C1
REF
R3
10k
C6
3.3nF
C7
3.3nF
R4
10k
R
T
V
V
ADJ2
ADJ1
M1
M2
GND
C8
0.1μF
C9
0.1μF
C8
220p
1M
R2
R1
24.3k
M3
3475 TA03
f
= 600kHz
SW
PWM1
D1, D2: B260
D3, D4: MBR0540
C1 TO C5: X5R OR X7R
M1, M2: Si2302ADS
M3: 2n7002L
PWM2
3475fb
16
LT3475/LT3475-1
TYPICAL APPLICATIONS
Step-Down 3A LED Driver
V
IN
5V TO 36V
C1
4.7μF
50V
D3
V
SHDN
BOOST2
D4
IN
BOOST1
C2
0.22μF
6.3V
C3
0.22μF
6.3V
L2
10μH
L1
10μH
LT3475
SW1
SW2
C5
2.2μF
6.3V
D1
D2
OUT1
OUT2
LED1
LED2
C4
2.2μF
6.3V
V
C1
V
C2
C6
0.1μF
C7
0.1μF
R
T
REF
V
V
ADJ2
ADJ1
R1
24.3k
3A LED
CURRENT
LED 1
GND
D1, D2: B240A
f
= 600kHz
3475 TA04
SW
D3, D4: MBR0540
C1 TO C5: X5R OR X7R
Dual Step-Down LED Driver with Series Connected LEDs
V
IN
10V TO 36V
C1
4.7μF
50V
D3
V
IN
SHDN
BOOST2
D4
BOOST1
C2
0.22μF
10V
C3
0.22μF
10V
L2
15μH
L1
15μH
LT3475
SW1
SW2
D1
D2
OUT1
LED1
OUT2
LED2
C4
2.2μF
10V
C5
2.2μF
10V
LED 1
LED 2
V
C1
V
C2
C6
0.1μF
C7
0.1μF
1.5A LED
CURRENT
1.5A LED
CURRENT
R
REF
T
V
ADJ1
V
ADJ2
R1
24.3k
LED 3
LED 4
GND
D1, D2: B240A
f
= 600kHz
3475 TA05
SW
D3, D4: MMSD4148T1
C1 TO C5: X5R OR X7R
3475fb
17
LT3475/LT3475-1
TYPICAL APPLICATIONS
Dual Step-Down 1.5A Red LED Driver
V
IN
5V TO 28V
C1
4.7μF
35V
D3
D4
V
SHDN
BOOST2
IN
BOOST1
C2
0.22μF
35V
C3
0.22μF
35V
L2
10μH
L1
10μH
LT3475
SW1
SW2
D1
D2
OUT1
LED1
OUT2
LED2
C4
2.2μF
6.3V
C5
2.2μF
6.3V
V
C1
V
C2
C6
0.1μF
C7
0.1μF
R
T
REF
V
V
ADJ2
ADJ1
R1
24.3k
1.5A LED
CURRENT
1.5A LED
CURRENT
LED 1
LED 2
GND
D1, D2: B240A
f
= 600kHz
3475 TA06
SW
D3, D4: MMSD4148T1
C1 TO C5: X5R OR X7R
3475fb
18
LT3475/LT3475-1
PACKAGE DESCRIPTION
FE Package
20-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663)
Exposed Pad Variation CB
6.40 – 6.60*
3.86
(.152)
(.252 – .260)
3.86
(.152)
20 1918 17 16 15 14 1312 11
6.60 ±0.10
2.74
(.108)
4.50 ±0.10
6.40
(.252)
BSC
2.74
(.108)
SEE NOTE 4
0.45 ±0.05
1.05 ±0.10
0.65 BSC
5
7
8
1
2
3
4
6
9 10
RECOMMENDED SOLDER PAD LAYOUT
1.20
(.047)
MAX
4.30 – 4.50*
(.169 – .177)
0.25
REF
0° – 8°
0.65
(.0256)
BSC
0.09 – 0.20
(.0035 – .0079)
0.50 – 0.75
(.020 – .030)
0.05 – 0.15
(.002 – .006)
FE20 (CB) TSSOP 0204
0.195 – 0.30
(.0077 – .0118)
TYP
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS 4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
MILLIMETERS
(INCHES)
2. DIMENSIONS ARE IN
3. DRAWING NOT TO SCALE
3475fb
InformationfurnishedbyLinearTechnologyCorporationisbelievedtobeaccurateandreliable.However,
no responsibility is assumed for its use. Linear Technology Corporation makes no representation that
the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LT3475/LT3475-1
TYPICAL APPLICATION
Dual Step-Down 1.5A LED Driver with Four Series Connected LED Output
V
IN
21V TO 36V
C1
4.7μF
50V
V
SHDN
BOOST2
IN
D3
D4
D1
D2
BOOST1
C2
0.22μF
16V
C3
0.22μF
16V
L1
L2
LT3475-1
33μH
33μH
SW1
SW2
R1
1k
R2
1k
D5
D6
OUT1
LED1
OUT2
LED2
12V TO 18V LED VOLTAGE
12V TO 18V LED VOLTAGE
R5
10k
R4
10k
V
V
C2
C1
D8
D7
Q1
C5
2.2μF
25V
C4
2.2μF
25V
C7
C6
REF
R
T
0.1μF
0.1μF
V
V
ADJ2
ADJ1
R3
24.3k
R6
100k
R7
GND
Q2
1.5A LED
CURRENT*
1.5A LED
CURRENT*
100k
f
= 600kHz
SW
3475 TA08
D1, D4: 7.5V ZENER DIODE
D2, D3: MMSD4148
D5, D6: B240A
D7, D8: 22V ZENER DIODE
R1, R2: USE 0.5W RESISTOR OF TWO 2k 0.25W RESISTORS IN PARALLEL
Q1, Q2: MMBT3904
C1 TO C5: X5R or X7R
*DERATE LED CURRENT AT ELEVATED AMBIENT TEMPERATURES TO MAINTAIN LT3475-1 JUNCTION TEMPERATURE BELOW 125 °C.
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
= 1.6V, V
MS10 Package
LT1618
Constant-Current, 1.4MHz, 1.5A Boost
Converter
V
= 18V, V = 35V, Analog/PWM, I < 1μA,
OUT(MAX) SD
IN(MIN)
IN(MAX)
LT3466
LT3474
Dual Full Function Step-Up LED Driver
Drivers Up to 20 LEDs, V : 2.7V to 24V, V
= 40V, DFN, TSSOP16E Packages
OUT(MAX)
IN
36V, 1A (I ), 2MHz Step-Down
V
= 4V, V
= 36V, 400:1 True Color PWM, I < 1μA,
IN(MAX) SD
LED
IN(MIN)
LED Driver
TSSOP16E Package
LT3477
LT3479
LT3846
42V, 3A, 3.5MHz Boost, Buck-Boost,
Buck LED Driver
V
= 2.5V, V
= 25V, V
= 40V, Analog/PWM, I < 1μA,
SD
IN(MIN)
IN(MAX)
OUT(MAX)
OUT(MAX)
QFN, TSSOP20E Packages
3A, Full-Featured DC/DC Converter with
Soft-Start and Inrush Current Protection
V
= 2.5V, V = 24V, V
= 40V, Analog/PWM, I < 1μA,
SD
IN(MIN)
IN(MAX)
DFN, TSSOP Packages
Dual 1.3A, 2MHz, LED Driver
V : 2.5V to 24V, V
DFN, TSSOP16E Packages
= 36V, 1000:1 True Color PWMTM Dimmin,
OUT(MAX)
IN
3475fb
LT 1007 REV B • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
20
●
●
© LINEAR TECHNOLOGY CORPORATION 2006
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
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