LT3756HMSE-2#PBF [Linear]
LT3756/LT3756-1/LT3756-2 - 100VIN, 100VOUT LED Controller; Package: MSOP; Pins: 16; Temperature Range: -40°C to 125°C;
| 型号： | LT3756HMSE-2#PBF |
| 厂家： | 
Linear |
| 描述： | LT3756/LT3756-1/LT3756-2 - 100VIN, 100VOUT LED Controller; Package: MSOP; Pins: 16; Temperature Range: -40°C to 125°C 驱动 光电二极管 接口集成电路 |
| 文件： | 总24页 (文件大小：309K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LT3756/LT3756-1/LT3756-2
100V , 100V
IN
OUT
LED Controller
FeaTures
DescripTion
n
3000:1 True Color PWM™ Dimming
The LT®3756, LT3756-1 and LT3756-2 are DC/DC control-
lers designed to operate as a constant-current source for
driving high current LEDs. They drive a low side external
N-channelpowerMOSFETfromaninternalregulated7.15V
supply. The fixed frequency, current-mode architecture
results in stable operation over a wide range of supply
and output voltages. A ground referenced voltage FB pin
serves as the input for several LED protection features,
and also makes it possible for the converter to operate
as a constant-voltage source. A frequency adjust pin
allows the user to program the frequency from 100kHz
to 1MHz to optimize efficiency, performance or external
component size.
n
Wide Input Voltage Range: 6V to 100V
n
Output Voltage Up to 100V
n
Constant-Current and Constant-Voltage Regulation
n
100mV High Side Current Sense
n
Drives LEDs in Boost, Buck Mode, Buck-Boost Mode,
SEPIC or Flyback Topology
n
Adjustable Frequency: 100kHz to 1MHz
n
Open LED Protection
n
Programmable Undervoltage Lockout with Hysteresis
n
Improved Open LED Status Pin (LT3756-2)
n
Frequency Synchronization (LT3756-1)
n
PWM Disconnect Switch Driver
n
CTRL Pin Provides Analog Dimming
The LT3756/LT3756-1/LT3756-2 sense output current at
the high side of the LED string. High side current sensing
is the most flexible scheme for driving LEDs, allowing
boost, buck mode or buck-boost mode configuration.
The PWM input provides LED dimming ratios of up to
3000:1, and the CTRL input provides additional analog
dimming capability.
n
Low Shutdown Current: <1µA
Programmable Soft-Start
n
nꢀ
Thermally Enhanced 16-Lead QFN (3mm × 3mm)
and MSOP Packages
applicaTions
n
High Power LED Applications
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear
Technology Corporation. True Color PWM is a trademark of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
n
Current Limited Constant Voltage Applications
n
Protected by U.S. Patents, including 7199560 and 7321203.
Battery Charging
Typical applicaTion
94% Efficient 30W White LED Headlamp Driver
Efficiency vs VIN
V
IN
8V TO 60V
100
22µH
(100V TRANSIENT)
4.7µF
1M
4.7µF
1M
V
IN
SHDN/UVLO
96
FB
V
ISP
185k
REF
14k
332k
LT3756-2
92
0.27Ω
370mA
CTRL
ISN
INTV
CC
40.2k
88
GATE
100k
SENSE
OPENLED
PWM
SS
30W
0.018Ω
84
LED
STRING
RT
PWMOUT
V
C
GND INTV
0.01µF
80
CC
28.7k
375kHz
1%
0
20
60
80
40
(V)
V
INTV
10k
IN
CC
375612 TA01b
0.001µF
4.7µF
375612 TA01a
375612fb
ꢀ
LT3756/LT3756-1/LT3756-2
absoluTe MaxiMuM raTings (Note 1)
V ..........................................................................100V
RT ............................................................................1.5V
SENSE......................................................................0.5V
Operating Junction Temperature Range (Notes 2, 3)
LT3756E, LT3756I ..............................–40°C to 125°C
LT3756H ............................................ –40°C to 150°C
Storage Temperature Range...................–65°C to 150°C
Lead Temperature (Soldering, 10 sec)
IN
SHDN/UVLO............................................................100V
ISP, ISN ...................................................................100V
INTV ...................................................... V + 0.3V, 8V
CC
IN
GATE, PWMOUT (Note 4)..........................INTV + 0.3V
CC
CTRL, PWM, OPENLED.............................................12V
VC, V , SS, FB .........................................................3V
REF
SYNC ..........................................................................8V
MSE.................................................................. 300°C
pin conFiguraTion
TOP VIEW
16 15 14 13
TOP VIEW
V
1
2
3
4
12 FB
1
2
3
4
5
6
7
8
REF
PWMOUT
FB
16 GATE
15 SENSE
PWM
SYNC OR OPENLED
SS
11 PWMOUT
17
ISN
14 V
IN
GND
ISP
13 INTV
GATE
17
GND
10
9
CC
VC
12 SHDN/UVLO
11 RT
SENSE
CTRL
V
10 SS
REF
5
6
7
8
PWM
9
SYNC OR OPENLED
MSE PACKAGE
16-LEAD PLASTIC MSOP
T
JMAX
= 125°C (E-,I-GRADES), T
= 150°C (H-GRADE), θ = 43°C/W, θ = 4°C/W
JMAX
JA
JC
EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB
UD PACKAGE
16-LEAD (3mm s 3mm) PLASTIC QFN
= 125°C, θ = 68°C/W, θ = 4.2°C/W
T
JMAX
JA
JC
EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB
orDer inForMaTion
LEAD FREE FINISH
LT3756EMSE#PBF
LT3756IMSE#PBF
LT3756EMSE-1#PBF
LT3756IMSE-1#PBF
LT3756EMSE-2#PBF
LT3756IMSE-2#PBF
LT3756HMSE-2#PBF
LT3756EUD#PBF
TAPE AND REEL
PART MARKING*
3756
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3756EMSE#TRPBF
LT3756IMSE#TRPBF
LT3756EMSE-1#TRPBF
LT3756IMSE-1#TRPBF
LT3756EMSE-2#TRPBF
LT3756IMSE-2#TRPBF
LT3756HMSE-2#TRPBF
LT3756EUD#TRPBF
LT3756IUD#TRPBF
16-Lead Plastic MSOP
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
–40°C to 150°C
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
3756
16-Lead Plastic MSOP
37561
37561
37562
37562
37562
LDMQ
LDMQ
LDMR
LDMR
LFKB
16-Lead Plastic MSOP
16-Lead Plastic MSOP
16-Lead Plastic MSOP
16-Lead Plastic MSOP
16-Lead Plastic MSOP
16-Lead (3mm × 3mm) Plastic QFN
16-Lead (3mm × 3mm) Plastic QFN
16-Lead (3mm × 3mm) Plastic QFN
16-Lead (3mm × 3mm) Plastic QFN
16-Lead (3mm × 3mm) Plastic QFN
16-Lead (3mm × 3mm) Plastic QFN
LT3756IUD#PBF
LT3756EUD-1#PBF
LT3756IUD-1#PBF
LT3756EUD-2#PBF
LT3756IUD-2#PBF
LT3756EUD-1#TRPBF
LT3756IUD-1#TRPBF
LT3756EUD-2#TRPBF
LT3756IUD-2#TRPBF
LFKB
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
This product is only offered in trays. For more information go to: http://www.linear.com/packaging/
375612fb
ꢁ
LT3756/LT3756-1/LT3756-2
elecTrical characTerisTics The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, SHDN/UVLO = 24V, CTRL = 2V, PWM = 5V, unless otherwise
noted.
PARAMETER
CONDITIONS
Tied to INTV
CC
MIN
TYP
MAX
UNITS
l
V
IN
IN
Minimum Operating Voltage
V
IN
6
V
V
Shutdown I
SHDN/UVLO = 0V, PWM = 0V
SHDN/UVLO = 1.15V, PWM = 0V
0.1
1
5
µA
µA
Q
V
V
V
Operating I (Not Switching)
PWM = 0V
1.4
2.00
0.006
108
40
1.7
mA
V
IN
Q
l
l
Voltage
100µA ≤ I
≤ 0µA
1.965
98
2.045
REF
REF
VREF
Line Regulation
6V ≤ V ≤ 100V
%/V
mV
µA
IN
SENSE Current Limit Threshold
118
13
SENSE Input Bias Current
Current Out of Pin
Current Out of Pin
SS Pull-Up Current
8
10
µA
Error Amplifier
l
l
l
l
ISP/ISN Full-Scale Current Sense Threshold
ISP/ISN Current Sense Threshold at CTRL = 0V
CTRL Pin Range for Current Sense Threshold Adjustment
CTRL Input Bias Current
FB = 0V, ISP = 48V
96
–12
0
100
103
–7
mV
mV
V
CTRL = 0V, FB = 0V, ISP = 48V
–9.5
1.1
100
100
200
3
Current Out of Pin
ISN = 0V
50
nA
V
LED Current Sense Amplifier Input Common Mode Range (V
ISP/ISN Short-Circuit Threshold
)
2.9
115
0
ISN
150
mV
V
ISP/ISN Short-Circuit Fault Sensing Common Mode Range (V
ISP/ISN Input Bias Current (Combined)
)
ISN
PWM = 5V (Active), ISP = ISN = 48V
PWM = 0V (Standby), ISP = ISN = 48V
55
0
µA
µA
0.1
LED Current Sense Amplifier g
VC Output Impedance
V
= 100mV
120
µS
kΩ
nA
m
(ISP – ISN)
1V < VC < 2V
PWM = 0V
15000
VC Standby Input Bias Current
–20
20
lꢀ
FB Regulation Voltage (V
)
FB
1.220
1.232
1.250
1.250
1.270
1.265
V
V
ISP = ISN
FB Amplifier g
FB = V , ISP = ISN
480
40
µS
nA
V
m
FB
FB Pin Input Bias Current
FB Open LED Threshold
Current Out of Pin
100
OPENLED Falling
(LT3756 and LT3756-2)
V
–
V
–
V
–
FB
FB
FB
65mV
50mV
40mV
FB Overvoltage Threshold
PWMOUT Falling
V
+
V
+
V +
FB
75mV
V
FB
FB
50mV
60mV
4
V/V
VC Current Mode Gain – (∆V /∆V
)
SENSE
VC
Oscillator
l
Switching Frequency
R = 100k
T
90
925
100
1000
125
1050
kHz
kHz
T
R = 10k
Minimum Off-Time
170
ns
Linear Regulator
INTV Regulation Voltage
7
7.15
1
7.3
V
V
CC
Dropout (V – INTV
)
I
= –10mA, V = 7V
INTVCC IN
IN
CC
INTV Undervoltage Lockout
3.9
14
4.1
17
8
4.3
23
12
V
CC
INTV Current Limit
mA
µA
CC
INTV Current in Shutdown
SHDN/UVLO = 0V, INTV = 7V
CC
CC
375612fb
ꢂ
LT3756/LT3756-1/LT3756-2
elecTrical characTerisTics The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, SHDN/UVLO = 24V, CTRL = 2V, PWM = 5V, unless otherwise
noted.
PARAMETER
CONDITIONS
MIN
1.5
45
TYP
MAX
UNITS
Logic Inputs/Outputs
PWM Input High Voltage
PWM Input Low Voltage
PWM Pin Resistance to GND
l
l
V
V
0.4
50
60
0
kΩ
mV
V
PWMOUT Output Low (V
)
OL
PWMOUT Output High (V
)
OH
INTV
–
CC
0.05
lꢀ
l
SHDN/UVLO Threshold Voltage Falling
E-, I-Grades
H Grade
1.185
1.175
1.220
20
1.245
1.245
V
V
SHDN/UVLO Rising Hysteresis
SHDN/UVLO Input Low Voltage
SHDN/UVLO Pin Bias Current Low
SHDN/UVLO Pin Bias Current High
mV
V
I
Drops Below 1µA
0.4
2.5
VIN
SHDN/UVLO = 1.15V
SHDN/UVLO = 1.30V
1.7
1.5
2.05
10
µA
nA
mV
100
200
OPENLED Output Low (V
)
OL
I
= 0.5mA
OPENLED
(LT3756 and LT3756-2)
SYNC Pin Resistance to GND
SYNC Input High
SYNC Input Low
LT3756-1 Only
30
kΩ
V
LT3756-1 Only
LT3756-1 Only
0.4
V
Gate Driver
t GATE Driver Output Rise Time
C = 3300pF
35
35
ns
ns
V
r
L
t GATE Driver Output Fall Time
f
C = 3300pF
L
GATE Output Low (V
)
OL
0.05
GATE Output High (V
)
OH
INTV
–
V
CC
0.05
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 3: The LT3756 includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed the maximum operating junction temperature
when overtemperature protection is active. Continuous operating above
the specified maximum operating junction temperature may impair device
reliability.
Note 2: The LT3756E, LT3756E-1 and LT3756E-2 are guaranteed to meet
performance specifications from 0°C to 125°C junction temperature.
Specifications over the –40°C to 125°C operating junction temperature
range are assured by design, characterization and correlation with
statistical process controls. The LT3756I, LT3756I-1 and LT3756I-2 are
guaranteed to meet performance specifications over the –40°C to 125°C
operating junction temperature range. The LT3756H-2 is guaranteed to
meet performance specifications over the full –40°C to 150°C operating
junction temperature range. High junction temperatures degrade operating
lifetimes. Operating lifetime is derated at junction temperatures greater
than 125°C.
Note 4: GATE and PWMOUT pins are driven either to GND or INTV by
CC
internal switches. Do not connect these pins externally to a power supply.
375612fb
ꢃ
LT3756/LT3756-1/LT3756-2
Typical perForMance characTerisTics TA = 25°C, unless otherwise noted.
V(ISP – ISN) Threshold
vs Temperature
V(ISP – ISN) Threshold vs VCTRL
V(ISP – ISN) Threshold vs VISP
103
102
101
100
99
120
100
80
103
102
101
100
99
V
= 2V
V
CTRL
= 2V
CTRL
60
40
20
98
98
0
97
–50 –25
–20
97
0
0.5
1
1.5
2
0
20
40
60
80
100
0
25 50 75 100 125 150
TEMPERATURE (°C)
375612 G03
V
(V)
ISP VOLTAGE (V)
CTRL
375612 G01
375612 G02
FB Regulation Voltage
vs Temperature
VREF Voltage vs Temperature
VREF Voltage vs VIN
1.28
1.27
1.26
1.25
1.24
1.23
1.22
1.21
1.20
2.04
2.03
2.02
2.01
2.00
1.99
1.98
1.97
1.96
2.04
2.03
2.02
2.01
2.00
1.99
1.98
1.97
1.96
0
20
40
60
80
100
–50
0
25 50 75 100 125 150
TEMPERATURE (°C)
375612 G04
–25
–50
0
25 50 75
125 150
–25
100
V
(V)
TEMPERATURE (°C)
IN
375612 G06
375612 G05
SHDN/UVLO Hysteresis Current
vs Temperature
Switching Frequency
vs Temperature
Switching Frequency vs RT
500
450
400
350
300
10000
1000
100
2.4
2.2
2.0
1.8
1.6
R
T
= 26.7k
10
10
100
50
125 150
50
TEMPERATURE (°C)
–50
0
25
75 100
–50
0
25
75 100
125 150
–25
–25
R
(k)
TEMPERATURE (°C)
T
375612 G07
375612 G08
375612 G09
375612fb
ꢄ
LT3756/LT3756-1/LT3756-2
Typical perForMance characTerisTics TA = 25°C, unless otherwise noted.
SENSE Current Limit Threshold
vs Temperature
SHDN/UVLO Threshold
Quiescent Current vs VIN
vs Temperature
110
105
100
95
1.28
1.26
1.24
1.22
1.20
1.18
2.0
1.5
1.0
0.5
0
PWM = 0V
SHDN/UVLO RISING
SHDN/UVLO FALLING
90
0
20
40
60
80
100
–50
0
25 50 75 100 125 150
TEMPERATURE (°C)
375612 G12
–25
–50
0
25 50 75 100 125 150
TEMPERATURE (°C)
375612 G11
–25
V
(V)
IN
375612 G10
INTVCC Current Limit
vs Temperature
INTVCC Voltage vs VIN
INTVCC Voltage vs Temperature
20
18
16
14
12
10
8
6
4
2
0
7.4
7.3
7.2
7.1
7.0
0
20
40
60
80
100
–50
0
25 50 75 100 125 150
TEMPERATURE (°C)
–25
50
TEMPERATURE (°C)
–50
0
25
75 100
125 150
–25
V
(V)
IN
375612 G14
375612 G13
375612 G15
SENSE Current Limit Threshold
vs Duty Cycle
Gate Rise/Fall Time
vs Capacitance
V(ISP-ISN) Threshold vs FB Voltage
125
100
75
50
25
0
100
80
60
40
20
0
110
105
100
95
V
CTRL
= 2V
10% TO 90%
GATE RISE
TIME
GATE
FALL TIME
90
0
25
50
75
100
1.2
1.22
1.24
1.26
1.28
0
2
4
6
8
10
DUTY CYCLE (%)
FB VOLTAGE (V)
CAPACITANCE (nF)
375612 G16
375612 G17
375612 G18
375612fb
ꢅ
LT3756/LT3756-1/LT3756-2
Typical perForMance characTerisTics TA = 25°C, unless otherwise noted.
ISP/ISN Input Bias Current
vs CTRL Voltage
INTVCC Dropout Voltage
vs Current, Temperature
0
–0.5
–1.0
–1.5
–2.0
–2.5
40
30
20
10
0
ISP
ISN
–45°C
25°C
125°C
150°C
V
= 7V
IN
0
3
6
9
12
15
0
0.5
1
1.5
2
LDO CURRENT (mA)
CTRL (V)
375612 G20
375612 G19
pin FuncTions (MSOP/QFN)
PWMOUT (Pin1/Pin 11):BufferedVersion ofPWM Signal
for Driving LED Load Disconnect NMOS or Level Shift.
This pin also serves in a protection function for the FB
overvoltagecondition—willtoggleiftheFBinputisgreater
ISN (Pin 3/Pin 13): Connection Point for the Negative
TerminaloftheCurrentFeedbackResistor. IfISNisgreater
than 2.9V, the LED current can be programmed by I
=
LED
–100mV)/
100mV/R whenV
>1.2VorI =(V
LED
CTRL
LED
CTRL
(10 • R ) when V
≤ 1V. Input bias current is typi-
than the FB regulation voltage (V ) plus 60mV (typical).
LED
CTRL
FB
cally 25µA. Below 3V, ISN is an input to the short-circuit
protection feature that forces GATE to 0V if ISP exceeds
ISN by more than 150mV (typ).
The PWMOUT pin is driven from INTV . Use of a FET with
CC
gate cut-off voltage higher than 1V is recommended.
FB (Pin 2/Pin 12): Voltage Loop Feedback Pin. FB is
intendedforconstant-voltageregulationorforLEDprotec-
tion/open LED detection. The internal transconductance
amplifierwithoutputVCwillregulateFBto1.25V(nominal)
through the DC/DC converter. If the FB input is regulating
the loop, the OPENLED pull-down is asserted. This ac-
tion may signal an open LED fault. If FB is driven above
the FB threshold (by an external power supply spike, for
example),theOPENLEDpull-downwillbede-assertedand
the PWMOUT pin will be driven low to protect the LEDs
from an overcurrent event. Do not leave the FB pin open.
If not used, connect to GND.
ISP (Pin 4/Pin 14): Connection Point for the Positive
Terminal of the Current Feedback Resistor. Input bias
current is dependent upon CTRL pin voltage as shown
in the TPC. ISP is an input to the short-circuit protection
feature when ISN is less than 3V.
VC (Pin 5/Pin 15): Transconductance Error Amplifier
Output Pin Used to Stabilize the Voltage Loop with an RC
Network. This pin is high impedance when PWM is low, a
feature that stores the demand current state variable for
thenextPWMhightransition.Connectacapacitorbetween
this pin and GND; a resistor in series with the capacitor is
recommended for fast transient response.
375612fb
ꢆ
LT3756/LT3756-1/LT3756-2
pin FuncTions
CTRL(Pin6/Pin16):CurrentSenseThresholdAdjustment
by an undervoltage condition (detected by SHDN/UVLO
pin) or thermal limit.
Pin. Regulating threshold V
is 1/10th V
plus
(ISP – ISN)
CTRL
> 1.2V the current
an offset for 0V < V
< 1V. For V
CTRL
CTRL
RT (Pin 11/Pin 5): Switching Frequency Adjustment Pin.
Set the frequency using a resistor to GND (for resistor
values, see the Typical Performance curve or Table 1).
Do not leave the RT pin open.
sensethresholdisconstantatthefull-scalevalueof100mV.
For 1V < V < 1.2V, the dependence of current sense
CTRL
threshold upon V
transitions from a linear function
CTRL
to a constant value, reaching 98% of full-scale value by
SHDN/UVLO (Pin 12/Pin 6): Shutdown and Undervoltage
Detect Pin. An accurate 1.22V falling threshold with ex-
ternally programmable hysteresis detects when power is
OK to enable switching. Rising hysteresis is generated
by the external resistor divider and an accurate internal
2.1µA pull-down current. Above the threshold (but below
6V), SHDN/UVLO input bias current is sub-µA. Below the
falling threshold, a 2.1µA pull-down current is enabled so
the user can define the hysteresis with the external resis-
tor selection. An undervoltage condition resets soft-start.
V
= 1.1V. Do not leave this pin open.
CTRL
V
(Pin7/Pin1):VoltageReferenceOutputPin,Typically
REF
2V.ThispindrivesaresistordividerfortheCTRLpin,either
foranalogdimmingorfortemperaturelimit/compensation
of LED load. Can supply up to 100μA.
PWM (Pin 8/Pin 2): A signal low turns off switcher, idles
oscillator and disconnects VC pin from all internal loads.
PWMOUT pin follows PWM pin. PWM has an internal
pull-down resistor. If not used, connect to INTV .
CC
Tie to 0.4V, or less, to disable the device and reduce V
quiescent current below 1µA.
IN
OPENLED (Pin 9/Pin 3, LT3756 and LT3756-2): An open-
collector pull-down on OPENLED asserts if the FB input
is greater than the FB regulation threshold minus 50mV
(typical). To function, the pin requires an external pull-up
current less than 1mA. When the PWM input is low and
the DC/DC converter is idle, the OPENLED condition is
latched to the last valid state when the PWM input was
high. When PWM input goes high again, the OPENLED
pin will be updated. This pin may be used to report an
open LED fault.
INTV (Pin 13/Pin 7): Regulated Supply for Internal
CC
Loads, GATE Driver and PWMOUT Driver. Supplied from
V
and regulates to 7.15V (typical). INTV must be
IN
CC
bypassed with a 4.7µF capacitor placed close to the pin.
Connect INTV directly to V if V is always less than
CC
IN
IN
or equal to 8V.
V
(Pin 14/Pin 8): Input Supply Pin. Must be locally
IN
bypassed with a 0.22µF (or larger) capacitor placed close
SYNC (Pin 9/Pin 3, LT3756-1 Only): The SYNC pin is used
to the IC.
to synchronize the internal oscillator to an external logic
SENSE (Pin 15/Pin 9): The current sense input for the
level signal. The R resistor should be chosen to program
T
control loop. Kelvin connect this pin to the positive ter-
aninternalswitchingfrequency20%slowerthantheSYNC
pulse frequency. Gate turn-on occurs a fixed delay after
the rising edge of SYNC. For best PWM performance, the
PWM rising edge should occur at least 200ns before the
SYNCrisingedge.Usea50%dutycyclewaveformtodrive
this pin. This pin replaces OPENLED on LT3756-1 option
parts. If not used, tie this pin to GND.
minal of the switch current sense resistor, R , in the
SENSE
source of the NFET. The negative terminal of the current
sense resistor should be connected to the GND plane
close to the IC.
GATE (Pin 16/Pin 10): N-channel FET Gate Driver Output.
Switches between INTV and GND. Driven to GND during
CC
shutdown, fault or idle states.
SS (Pin 10/Pin 4): Soft-Start Pin. This pin modulates
oscillator frequency and compensation pin voltage (VC)
clamp.Thesoft-startintervalissetwithanexternalcapaci-
tor. The pin has a 10µA (typical) pull-up current source
to an internal 2.5V rail. The soft-start pin is reset to GND
GND (Pin 17/Pin 17): Ground. This pin also serves as
current sense input for control loop, sensing negative
terminal of current sense resistor. Solder the exposed pad
directly to ground plane.
375612fb
ꢇ
LT3756/LT3756-1/LT3756-2
block DiagraM
SHDN/UVLO
–
+
A6
FB
VC
PWMOUT PWM
1.25V
V
IN
–
+
SHDN
1.22V
2.1µA
1.3V
LDO
–
+
OVFB
COMPARATOR
A8
7.15V
INTV
CC
A5
+
–
10µA AT
FB = 1.25V
g
m
1.25V
SHORT-CIRCUIT
DETECT
SCILMB
10µA
+
SCILMB
A10
GATE
SENSE
GND
+
+
–
150mV
–
R
Q
–
g
A2
m
DRIVER
S
EAMP
ISN
ISP
PWM
COMPARATOR
+
–
10µA AT
5k
A1
I
+
–
SENSE
A1 = A1
+
–
CTRL
A4
BUFFER
CTRL
+
–
1.1V
A3
Q2
+
RAMP
GENERATOR
VC
SSCLAMP
10µA
50k
100kHz TO 1MHz
OSCILLATOR
OPENLED
FAULT
–
+
140µA
LOGIC
1.25V
+
+
–
1.2V
FB
(LT3756
AND
LT3756-2)
V
REF
–
+
T
FREQ
PROG
LIM
165°C
170k
A7
2V
SS
RT
SYNC (LT3756-1 ONLY)
375612 BD
375612fb
ꢈ
LT3756/LT3756-1/LT3756-2
operaTion
TheLT3756isaconstant-frequency,currentmodecontrol-
ler with a low side NMOS gate driver. The GATE pin and
PWMOUT pin drivers, and other chip loads, are powered
difference between ISP and ISN is monitored to determine
if the output is in a short-circuit condition. If the difference
between ISP and ISN is greater than 150mV (typical), the
SR latch will be reset regardless of the PWM comparator.
These functions are intended to protect the power switch,
as well as various external components in the power path
of the DC/DC converter.
from INTV , which is an internally regulated supply. In
CC
the discussion that follows, it will be helpful to refer to
the Block Diagram of the IC. In normal operation, with the
PWM pin low, the GATE and PWMOUT pins are driven to
GND, the VC pin is high impedance to store the previous
switching state on the external compensation capacitor,
and the ISP and ISN pin bias currents are reduced to
leakage levels. When the PWM pin transitions high, the
PWMOUT pin transitions high after a short delay. At the
same time, the internal oscillator wakes up and gener-
ates a pulse to set the PWM latch, turning on the external
power MOSFET switch (GATE goes high). A voltage input
proportional to the switch current, sensed by an external
current sense resistor between the SENSE and GND input
pins, is added to a stabilizing slope compensation ramp
and the resulting “switch current sense” signal is fed into
the positive terminal of the PWM comparator. The current
in the external inductor increases steadily during the time
the switch is on. When the switch current sense voltage
exceeds the output of the error amplifier, labeled “VC”,
the latch is reset and the switch is turned off. During the
switch off phase, the inductor current decreases. At the
completion of each oscillator cycle, internal signals such
asslopecompensationreturntotheirstartingpointsanda
new cycle begins with the set pulse from the oscillator.
In voltage feedback mode, the operation is similar to that
described above, except the voltage at the VC pin is set
by the amplified difference of the internal reference of
1.25V (nominal) and the FB pin. If FB is lower than the
reference voltage, the switch current will increase; if FB
is higher than the reference voltage, the switch demand
current will decrease. The LED current sense feedback
interacts with the FB voltage feedback so that FB will not
exceed the internal reference and the voltage between ISP
and ISN will not exceed the threshold set by the CTRL pin.
For accurate current or voltage regulation, it is necessary
to be sure that under normal operating conditions, the
appropriate loop is dominant. To deactivate the voltage
loop entirely, FB can be connected to GND. To deactivate
the LED current loop entirely, the ISP and ISN should be
tied together and the CTRL input tied to V
.
REF
Two LED specific functions featured on the LT3756 are
controlled by the voltage feedback pin. First, when the
FB pin exceeds a voltage 50mV lower (–4%) than the FB
regulation voltage, the pull-down driver on the OPENLED
pin is activated (LT3756 and LT3756-2 only). This function
provides a status indicator that the load may be discon-
nected and the constant-voltage feedback loop is taking
controloftheswitchingregulator.WhentheFBpinexceeds
theFBregulationvoltageby60mV(5%typical), thePWM-
OUT pin is driven low, ignoring the state of the PWM input.
In the case where the PWMOUT pin drives a disconnect
NFET, this action isolates the LED load from GND, prevent-
ing excessive current from damaging the LEDs. If the FB
input exceeds both the open LED and the overvoltage
thresholds, then an externally driven overvoltage event
has caused the FB pin to be too high and the OPENLED
pull-down willbede-asserted. The LT3756-2 willre-assert
the OPENLED signal when FB falls below the overvoltage
threshold and remains above the open LED threshold. The
LT3756 is prevented from re-asserting OPENLED until FB
drops below both thresholds.
Through this repetitive action, the PWM control algorithm
establishes a switch duty cycle to regulate a current or
voltage in the load. The VC signal is integrated over many
switching cycles and is an amplified version of the differ-
ence between the LED current sense voltage, measured
between ISP and ISN, and the target difference voltage
set by the CTRL pin. In this manner, the error amplifier
sets the correct peak switch current level to keep the
LED current in regulation. If the error amplifier output
increases, more current is demanded in the switch; if it
decreases, less current is demanded. The switch current
is monitored during the on-phase and the voltage across
the SENSE pin is not allowed to exceed the current limit
threshold of 108mV (typical). If the SENSE pin exceeds
the current limit threshold, the SR latch is reset regard-
less of the output state of the PWM comparator. Likewise,
at an ISP/ISN common mode voltage less than 3V, the
375612fb
ꢀ0
LT3756/LT3756-1/LT3756-2
applicaTions inForMaTion
The following equations should be used to determine the
values of the resistors:
INTV Regulator Bypassing and Operation
CC
The INTV pin requires a capacitor for stable operation
CC
R1+R2
and to store the charge for the large GATE switching cur-
rents. Choose a 10V rated low ESR, X7R or X5R ceramic
capacitor for best performance. A 4.7µF capacitor will be
adequate for many applications. Place the capacitor close
VIN,FALLING = 1.22 •
R2
V
= 2.1µA •R1+ V
IN,FALLING
IN,RISING
to the IC to minimize the trace length to the INTV pin
and also to the IC ground.
CC
V
IN
LT3756
An internal current limit on the INTV output protects
R1
R2
CC
the LT3756 from excessive on-chip power dissipation.
The minimum value of this current should be considered
when choosing the switching NMOS and the operating
frequency.
SHDN/UVLO
375612 F01
Figure 1. Resistor Connection to Set VIN
Undervoltage Shutdown Threshold
I
can be calculated from the following equation:
INTVCC
I
= Q • f
G OSC
INTVCC
LED Current Programming
Careful choice of a lower Q FET will allow higher switch-
G
The LED current is programmed by placing an appropriate
ingfrequencies, leadingtosmallermagnetics. TheINTV
CC
value current sense resistor, R , in series with the LED
LED
pin has its own undervoltage disable (UVLO) set to 4.1V
(typical)toprotecttheexternalFETsfromexcessivepower
dissipation caused by not being fully enhanced. If the
string. The voltage drop across R
is (Kelvin) sensed
LED
by the ISP and ISN pins. Typically, sensing of the current
should be done at the top of the LED string. If this option
is not available, then the current may be sensed at the
bottom of the string, but take caution that the minimum
ISN value does not fall below 3V, which is the lower limit of
the LED current regulation function. The CTRL pin should
be tied to a voltage higher than 1.1V to get the full-scale
100mV (typical) threshold across the sense resistor. The
CTRL pin can also be used to dim the LED current to zero,
although relative accuracy decreases with the decreasing
voltage sense threshold. When the CTRL pin voltage is
less than 1.0V, the LED current is:
INTV pin drops below the UVLO threshold, the GATE
CC
and PWMOUT pins will be forced to 0V and the soft-start
pin will be reset.
Iftheinputvoltage, V , willnotexceed8V, thentheINTV
IN
CC
pin could be connected to the input supply. Be aware that
a small current (less than 12μA) will load the INTV in
CC
shutdown.IfV isnormallyabove,butoccasionallydrops
IN
below the INTV regulation voltage, then the minimum
CC
operating V will be close to 7V. This value is determined
IN
by the dropout voltage of the linear regulator and the 4.5V
(4.1V typical) INTV undervoltage lockout threshold
VCTRL − 100mV
CC
ILED
=
mentioned above.
RLED • 10
Programming the Turn-On and Turn-Off Thresholds
WhentheCTRLpinvoltageisbetween1Vand1.2VtheLED
current varies with CTRL, but departs from the equation
abovebyanincreasingamountasCTRLvoltageincreases.
Ultimately, above CTRL = 1.2V the LED current no longer
with the SHDN/UVLO Pin
ThefallingUVLOvaluecanbeaccuratelysetbytheresistor
divider. A small 2.1µA pull-down current is active when
SHDN/UVLO is below the threshold. The purpose of this
currentistoallowtheusertoprogramtherisinghysteresis.
varies with CTRL. At CTRL = 1.1V, the actual value of I
LED
is ~98% of the equation’s estimate.
375612fb
ꢀꢀ
LT3756/LT3756-1/LT3756-2
applicaTions inForMaTion
When V
lated to:
is higher than 1.2V, the LED current is regu-
operationwillnotexceed1.1V. ForanLEDdriverofbuckor
a buck-boost configuration, the output voltage is typically
level-shifted to a signal with respect to GND as illustrated
in Figure 3. The output can be expressed as:
CTRL
100mV
RLED
ILED
=
R3
VOUT = VBE + 1.25 •
R4
The LED current programming feature can increase total
dimming range by a factor of 10. The CTRL pin should
not be left open (tie to V
can also be used in conjunction with a thermistor to
provide overtemperature protection for the LED load, or
with a resistor divider to V to reduce output power and
switching current when V is low. The presence of a time
if not used). The CTRL pin
REF
+
R3
R
SEN(EXT)
C
V
OUT
OUT
LED
ARRAY
100k
IN
IN
–
LT3756
FB
varying differential voltage signal (ripple) across ISP and
ISN at the switching frequency is expected. The amplitude
of this signal is increased by high LED load current, low
switching frequency and/or a smaller value output filter
capacitor. Some level of ripple signal is acceptable: the
compensation capacitor on the VC pin filters the signal so
the average difference between ISP and ISN is regulated
to the user-programmed value. Ripple voltage amplitude
(peak-to-peak) in excess of 20mV should not cause mis-
operation, but may lead to noticeable offset between the
average value and the user-programmed value.
R4
375612 F03
Figure 3. Feedback Resistor Connection for
Buck Mode or Buck-Boost Mode LED Driver
ISP/ISN Short-Circuit Protection Feature (for SEPIC)
The ISP and ISN pins have a protection feature indepen-
dent of the LED current sense feature that operates at
ISN below 3V. The purpose of this feature is to provide
continuous current sensing when ISN is below the LED
current sense common mode range (during start-up or
an output short-circuit fault) to prevent the development
of excessive switching currents that could damage the
power components in a SEPIC converter. The action
threshold (150mV, typ) is above the default LED current
sense threshold, so that no interference will occur over
the ISN voltage range where these two functions overlap.
This feature acts in the same manner as SENSE current
limit—it prevents GATE from going high (switch turn-on)
untiltheISP/ISNdifferencefallsbelowthethreshold. Ifthe
load has appreciable series inductance, use of a Schottky
clamp from GND to ISN is recommended for the SEPIC
to prevent excessive current flowing from the ISN pin in
a fault.
Programming Output Voltage (Constant-Voltage
Regulation) or Open LED/Overvoltage Threshold
For a boost or SEPIC application, the output voltage can
be set by selecting the values of R3 and R4 (see Figure 2)
according to the following equation:
R3+R4
VOUT = 1.25 •
R4
ForaboosttypeLEDdriver,settheresistorfromtheoutput
to the FB pin such that the expected V during normal
FB
V
IN
LT3756
R3
R4
FB
Dimming Control
375612 F02
There are two methods to control the current source for
dimming using the LT3756. One method uses the CTRL
pin to adjust the current regulated in the LEDs. A second
Figure 2. Feedback Resistor Connection
for Boost or SEPIC LED Drivers
375612fb
ꢀꢁ
LT3756/LT3756-1/LT3756-2
applicaTions inForMaTion
Duty Cycle Considerations
method uses the PWM pin to modulate the current source
between zero and full current to achieve a precisely pro-
grammed average current. To make PWM dimming more
accurate, the switch demand current is stored on the VC
node during the quiescent phase when PWM is low. This
featureminimizesrecoverytimewhenthePWMsignalgoes
high. To further improve the recovery time, a disconnect
switch may be used in the LED current path to prevent the
ISP node from discharging during the PWM signal low
phase. The minimum PWM on or off time will depend on
thechoiceofoperatingfrequencyandexternalcomponent
selection. With operation in discontinuous conduction
mode (DCM), regulated current pulses as short as 1µs are
achievable. But, the best overall combination of PWM and
analog dimming (with CTRL) is available if the minimum
PWM pulse is at least six switching cycles.
Switching duty cycle is a key variable defining converter
operation, therefore, its limits must be considered when
programming the switching frequency for a particular
application. The fixed minimum on-time and minimum
off-time (see Figure 4) and the switching frequency define
the minimum and maximum duty cycle of the switch,
respectively. The following equations express the mini-
mum/maximum duty cycle:
Min Duty Cycle = (minimum on-time) • switching fre-
quency
Max Duty Cycle = 1 – (minimum off-time) • switching
frequency
When calculating the operating limits, the typical values
for on/off-time in the data sheet should be increased by
at least 60ns to allow margin for PWM control latitude,
GATE rise/fall times and SW node rise/fall times.
Programming the Switching Frequency
The RT frequency adjust pin allows the user to program
the switching frequency from 100kHz to 1MHz to optimize
efficiency/performanceorexternalcomponentsize.Higher
frequency operation yields smaller component size but
increases switching losses and gate driving current, and
maynotallowsufficientlyhighorlowdutycycleoperation.
Lowerfrequencyoperationgivesbetterperformanceatthe
cost of larger external component size. For an appropriate
300
C
= 3300pF
GATE
250
200
150
100
50
MINIMUM ON-TIME
MINIMUM OFF-TIME
R resistor value see Table 1. An external resistor from the
T
RT pin to GND is required—do not leave this pin open.
0
–50
Table 1. Switching Frequency vs RT Value
0
25 50 75 100 125 150
TEMPERATURE (°C)
–25
f
(kHz)
R (kΩ)
T
375612 F04
OSC
1000
10.0
11.8
13.0
15.4
17.8
21.0
26.7
35.7
53.6
100
Figure 4. Typical Minimum On and Off
Pulse Width vs Temperature
900
800
700
600
500
400
300
200
100
Thermal Considerations
The LT3756 series is rated to a maximum input voltage
of 100V. Careful attention must be paid to the internal
power dissipation of the IC at higher input voltages to
ensure that a junction temperature of 125°C (150°C for
H-grade) is not exceeded. This junction limit is especially
375612fb
ꢀꢂ
LT3756/LT3756-1/LT3756-2
applicaTions inForMaTion
Driving SYNC with a 50% duty cycle waveform is always
agoodchoice, otherwise, maintainthedutycyclebetween
20%and60%. WhenusingbothPWMandSYNCfeatures,
the PWM signal rising edge should occur at least 200ns
important when operating at high ambient temperatures.
ThemajorityofthepowerdissipationintheICcomesfrom
the supply current needed to drive the gate capacitance of
the external power MOSFET. This gate drive current can
be calculated as:
before the SYNC rising edge (V ) for optimal PWM
IH
performance. If the SYNC pin is not used, it should be
connected to GND.
I
= f • Q
SW G
GATE
A low Q power MOSFET should always be used when op-
G
eratingathighinputvoltages,andtheswitchingfrequency
should also be chosen carefully to ensure that the IC does
not exceed a safe junction temperature. The internal junc-
tion temperature of the IC can be estimated by:
Open LED Detection (LT3756 and LT3756-2)
TheLT3756andLT3756-2provideanopen-collectorstatus
pin, OPENLED, that pulls low when the FB pin is within
~50mV of its 1.25V regulated voltage. If the open LED
clamp voltage is programmed correctly using the FB pin,
then the FB pin should never exceed 1.1V when LEDs are
connected, therefore, the only way for the FB pin to be
within 50mV of the regulation voltage is for an open LED
event to have occurred. The key difference between the
LT3756 and LT3756-2 is the behavior of the OPENLED pin
whentheFBpincrossesandre-crossestheFBovervoltage
thresholdat1.31V(typ). TheLT3756-2asserts/de-asserts
OPENLED freely when crossing the 1.31V threshold.
The LT3756, by comparison, de-asserts OPENLED when
FB exceeds 1.31V and is prevented from re-asserting
OPENLED until the FB pin falls below the 1.2V (typ) open
LED threshold and clears the fault. The LT3756-2 has the
more general purpose behavior and is recommended for
applications using OPENLED.
T = T + [V (I + f • Q ) • θ ]
J
A
IN
Q
SW
G
JA
where T is the ambient temperature, I is the quiescent
A
Q
current of the part (maximum 1.5mA) and θ is the
JA
package thermal impedance (68°C/W for the 3mm × 3mm
QFN package). For example, an application with T
A(MAX)
= 85°C, V
= 60V, f = 400kHz, and having a FET
IN(MAX)
SW
with Q = 20nC, the maximum IC junction temperature
G
will be approximately:
T = 85°C + [60V (1.5mA + 400kHz • 20nC) • 68°C/W]
J
= 124°C
The exposed pad on the bottom of the package must be
soldered to a ground plane. This ground should then be
connectedtoaninternalcoppergroundplanewiththermal
vias placed directly under the package to spread out the
heat dissipated by the IC.
Input Capacitor Selection
If LT3756 junction temperature reaches 165°C, the GATE
and PWMOUT pins will be driven to GND and the soft-
start (SS) pin will be discharged to GND. Switching will
be enabled after device temperature is reduced 10°C. This
functionisintendedtoprotectthedeviceduringmomentary
thermal overload conditions.
Theinputcapacitorsuppliesthetransientinputcurrentfor
the power inductor of the converter and must be placed
andsizedaccordingtothetransientcurrentrequirements.
Theswitchingfrequency,outputcurrentandtolerableinput
voltage ripple are key inputs to estimating the capacitor
value. An X7R type ceramic capacitor is usually the best
choice since it has the least variation with temperature
and DC bias. Typically, boost and SEPIC converters re-
quire a lower value capacitor than a buck mode converter.
Assuming that a 100mV input voltage ripple is acceptable,
the required capacitor value for a boost converter can be
estimated as follows:
Frequency Synchronization (LT3756-1 Only)
TheLT3756-1switchingfrequencycanbesynchronizedto
anexternalclockusingtheSYNCpin.Forproperoperation,
theR resistorshouldbechosenforaswitchingfrequency
T
20% lower than the external clock frequency. The SYNC
pin is disabled during the soft-start period.
VOUT
1µF
A •µs
CIN(µF)=ILED (A)•
• tSW(µs)•
Observation of the following guidelines about the SYNC
waveform will ensure proper operation of this feature.
V
IN
375612fb
ꢀꢃ
LT3756/LT3756-1/LT3756-2
applicaTions inForMaTion
Therefore, a 4.7µF capacitor is an appropriate selection
for a 400kHz boost regulator with 12V input, 48V output
and 1A load.
operating frequencies will require proportionately higher
capacitor values.
Soft-Start Capacitor Selection
WiththesameV voltagerippleof100mV,theinputcapaci-
IN
For many applications, it is important to minimize the
inrush current at start-up. The built-in soft-start circuit
significantly reduces the start-up current spike and output
voltageovershoot. Thesoft-startintervalissetbythesoft-
start capacitor selection according to the equation:
tor for a buck converter can be estimated as follows:
4.7µF
CIN(µF)=ILED (A)• tSW(µs)•
A •µs
A 10µF input capacitor is an appropriate selection for a
400kHz buck mode converter with a 1A load.
2V
10µA
TSS = CSS
•
In the buck mode configuration, the input capacitor has
large pulsed currents due to the current returned through
the Schottky diode when the switch is off. In this buck
convertercaseitisimportanttoplacethecapacitorasclose
as possible to the Schottky diode and to the GND return
of the switch (i.e., the sense resistor). It is also important
to consider the ripple current rating of the capacitor. For
best reliability, this capacitor should have low ESR and
ESL and have an adequate ripple current rating. The RMS
input current for a buck mode LED driver is:
A typical value for the soft-start capacitor is 0.01µF. The
soft-start pin reduces the oscillator frequency and the
maximum current in the switch. The soft-start capacitor
is discharged when SHDN/UVLO falls below its threshold,
during an overtemperature event or during an INTV
CC
undervoltage event. During start-up with SHDN/UVLO,
charging of the soft-start capacitor is enabled after the
first PWM high period.
Power MOSFET Selection
IIN(RMS) = ILED
•
1– D •D
(
)
Forapplicationsoperatingathighinputoroutputvoltages,
the power NMOS FET switch is typically chosen for drain
where D is the switch duty cycle.
voltage V rating and low gate charge Q . Consideration
DS
G
Table 2. Recommended Ceramic Capacitor Manufacturers
MANUFACTURER
TDK
of switch on-resistance, R
, is usually secondary be-
DS(ON)
WEB
cause switching losses dominate power loss. The INTV
CC
www.tdk.com
www.kemet.com
www.murata.com
www.t-yuden.com
regulator on the LT3756 has a fixed current limit to protect
the IC from excessive power dissipation at high V , so the
Kemet
IN
Murata
FET should be chosen so that the product of Q at 7V and
G
Taiyo Yuden
switching frequency does not exceed the INTV current
CC
limit. For driving LEDs be careful to choose a switch with
Output Capacitor Selection
a V rating that exceeds the threshold set by the FB pin
DS
in case of an open-load fault. Several MOSFET vendors
are listed in Table 3. The MOSFETs used in the application
circuits in this data sheet have been found to work well
with the LT3756. Consult factory applications for other
recommended MOSFETs.
The selection of the output capacitor depends on the load
and converter configuration, i.e., step-up or step-down
and the operating frequency. For LED applications, the
equivalent resistance of the LED is typically low and the
output filter capacitor should be sized to attenuate the
current ripple. Use of an X7R type ceramic capacitor is
recommended.
Table 3. MOSFET Manufacturers
VENDOR
WEB
To achieve the same LED ripple current, the required filter
capacitor is larger in the boost and buck-boost mode ap-
plications than that in the buck mode applications. Lower
Vishay Siliconix
Fairchild
www.vishay.com
www.fairchildsemi.com
www.irf.com
International Rectifier
375612fb
ꢀꢄ
LT3756/LT3756-1/LT3756-2
applicaTions inForMaTion
Schottky Rectifier Selection
These equations provide an estimate of the sense resistor
value based on reasonable assumptions about induc-
tor current ripple during steady state switching. Lower
values of sense resistor may be required in applications
where inductor ripple current is higher. Examples include
applications with current limited operation at high duty
cycle, and those with discontinuous conduction mode
(DCM) switching. It is always prudent to verify the peak
inductor current in the application to ensure the sense
resistor selection provides margin to the SENSE current
limit threshold.
The power Schottky diode conducts current during the
interval when the switch is turned off. Select a diode rated
forthemaximumSWvoltage. IfusingthePWMfeaturefor
dimming, it is important to consider diode leakage, which
increaseswiththetemperature,fromtheoutputduringthe
PWM low interval. Therefore, choose the Schottky diode
with sufficiently low leakage current. Table 4 has some
recommended component vendors.
Table 4. Schottky Rectifier Manufacturers
VENDOR
WEB
Inductor Selection
On Semiconductor
Diodes, Inc.
www.onsemi.com
www.diodes.com
www.centralsemi.com
TheinductorusedwiththeLT3756shouldhaveasaturation
current rating appropriate to the maximum switch current
Central Semiconductor
selectedwiththeR
resistor.Chooseaninductorvalue
SENSE
Sense Resistor Selection
based on operating frequency, input and output voltage to
provide a current mode ramp on SENSE during the switch
on-time of approximately 20mV magnitude. The following
equations are useful to estimate the inductor value for
continuous conduction mode operation:
The resistor, R
, between the source of the exter-
SENSE
nal NMOS FET and GND should be selected to provide
adequate switch current to drive the application without
exceeding the 108mV (typical) current limit threshold on
the SENSE pin of LT3756. For buck mode applications,
select a resistor that gives a switch current at least 30%
greater than the required LED current. For buck mode,
select a resistor according to:
RSENSE • VLED V – V
(
)
IN
LED
LBUCK
=
V • 0.02V • f
IN
OSC
RSENSE • VLED • V
IN
LBUCK-BOOST
=
0.07V
V
LED + V • 0.02V • f
IN
OSC
(
)
RSENSE,BUCK
≤
ILED
RSENSE • V
VLED – VIN
(
)
IN
LBOOST
=
For buck-boost, select a resistor according to:
VLED • 0.02V • fOSC
V • 0.07V
IN
RSENSE,BUCK-BOOST
≤
Table 5 provides some recommended inductor vendors.
V + V
I
IN
LED LED
Table 5. Inductor Manufacturers
For boost, select a resistor according to:
VENDOR
WEB
Sumida
www.sumida.com
www.we-online.com
www.cooperet.com
www.vishay.com
www.coilcraft.com
V • 0.07V
IN
RSENSE,BOOST
≤
Würth Elektronik
Coiltronics
Vishay
V
LED
•ILED
The placement of R
should be close to the source of
SENSE
Coilcraft
the NMOS FET and GND of the LT3756. The SENSE input
to LT3756 should be a Kelvin connection to the positive
terminal of R
.
SENSE
375612fb
ꢀꢅ
LT3756/LT3756-1/LT3756-2
applicaTions inForMaTion
Loop Compensation
isimportanttominimizetheareaofthehighdV/dtswitching
node between the inductor, switch drain and anode of the
Schottky rectifier. Use a ground plane under the switching
node to eliminate interplane coupling to sensitive signals.
The lengths of the high dI/dt traces: 1) from the switch
node through the switch and sense resistor to GND, and
2) from the switch node through the Schottky rectifier and
filter capacitor to GND should be minimized. The ground
points of these two switching current traces should come
toacommonpointthenconnecttothegroundplaneunder
the LT3756. Likewise, the ground terminal of the bypass
TheLT3756usesaninternaltransconductanceerrorampli-
fier whose VC output compensates the control loop. The
external inductor, output capacitor and the compensation
resistor and capacitor determine the loop stability.
The inductor and output capacitor are chosen based on
performance, size and cost. The compensation resistor
and capacitor at VC are selected to optimize control loop
response and stability. For typical LED applications, a
2.2nF compensation capacitor at VC is adequate, and
a series resistor should always be used to increase the
slew rate on the VC pin to maintain tighter regulation of
LED current during fast transients on the input supply to
the converter.
capacitor for the INTV regulator should be placed near
CC
the GND of the switching path. Typically, this requirement
will result in the external switch being closest to the IC,
along with the INTV bypass capacitor. The ground for
CC
the compensation network and other DC control signals
should be star connected to the underside of the IC. Do
not extensively route high impedance signals such as FB
and VC, as they may pick up switching noise. In particular,
avoid routing FB and PWMOUT in parallel for more than a
fewmillimetersontheboard.Likewise,minimizeresistance
in series with the SENSE input to avoid changes (most
likely reduction) to the switch current limit threshold.
Board Layout
The high speed operation of the LT3756 demands careful
attention to board layout and component placement. The
exposed pad of the package is the only GND terminal of
the IC and is also important for thermal management of
the IC. It is crucial to achieve a good electrical and thermal
contact between the exposed pad and the ground plane of
theboard.Toreduceelectromagneticinterference(EMI),it
375612fb
ꢀꢆ
LT3756/LT3756-1/LT3756-2
applicaTions inForMaTion
VIAS TO GROUND PLANE
C
SS
R
T
4
9
3
2
1
C
C
R2
R1
5
6
7
8
16
15
14
13
R
C
V
OUT
VIA
x
x
CV
CC
10 11 12
L1
R3
R4
5
6
7
8
4
1
–
3
2
1
M2
3
LED
M1
2
R
SENSE
C
OUT
C
OUT
D1
C
IN
+
LED
R
LED
V
IN
GND
375612 F05
COMPONENT DESIGNATIONS REFER TO “30W WHITE LED HEADLAMP DRIVER WITH THERMAL DERATING” SCHEMATIC
Figure 5. Boost Converter Suggested Layout
375612fb
ꢀꢇ
LT3756/LT3756-1/LT3756-2
Typical applicaTions
30W White LED Headlamp Driver with Thermal Derating
D1
L1, 22µH
V
8V TO 60V
IN
(100V TRANSIENT)
C
R1
1M
R3
1M
C
IN
OUT
V
IN
4.7µF
4.7µF
SHDN/UVLO
FB
R2
185k
R4
V
ISP
REF
14k
16.9k
R
LT3756-2
LED
370mA
0.27Ω
CTRL
ISN
GATE
100k
NTC
RT1
INTV
CC
30W LED STRING
M1
100k
SENSE
OPENLED
R
SENSE
0.018Ω
PWM
SS
PWMOUT
GND INTV
C
SS
V
C
RT
CC
0.01µF
R
T
R
C
28.7k
C
VCC
10k
375kHz
4.7µF
C
C
M2
0.001µF
375612 TA02a
M1: VISHAY SILICONIX Si7454DP
D1: DIODES INC PDS5100
SEE SUGGESTED LAYOUT, FIGURE 5
L1: COILTRONICS DR127-220
RT1: MURATA NCP18WM104J
M2: VISHAY SILICONIX Si2328DS
V
(ISP – ISN) Threshold vs Temperature
for NTC Resistor Divider
120
100
80
60
40
20
0
25
45
65
85
105
125
TEMPERATURE (°C)
375512 TA02b
375612fb
ꢀꢈ
LT3756/LT3756-1/LT3756-2
Typical applicaTions
Buck-Boost Mode LED Driver
L1
68µH
Efficiency vs VIN
D1
V
IN
V
OUT
9V TO
65V
100
90
80
70
60
50
C1
1µF
C3
4.7µF
V
IN
4.7µF 1M
100V
1M
SHDN/UVLO
FB
V
IN
V
ISP
185k
REF
13k
LT3756-2
1Ω
CTRL
INTV
ISN
CC
M1
GATE
100k
24V TO 32V
LED STRING
100mA
SENSE
OPENLED
PWM
SS
0.068Ω
RT
PWMOUT
0.1µF
V
C
GND INTV
1.5k
0
40
(V)
60
80
CC
20
35.7k
V
IN
C2
2.2µF
10V
M2
300kHz
39k
375612 TA03b
V
IN
4700pF
Q1
L1: COILCRAFT MSS1038-683
D1: ON SEMICONDUCTOR MBRS3100T3
M1: VISHAY SILICONIX Si2328DS
M2: ZETEX ZXM6IP03F
1k
375612 TA03a
Q1: ZETEX FMMT493
90% Efficient, 20W SEPIC LED Driver
C4
1µF
L1A
33µH
Efficiency vs VIN
D1
V
IN
100
96
92
88
84
80
8V TO
80V
C3
1:1
C1
4.7µF
100V
10µF
s2
1M
V
IN
511k
35V
SHDN/UVLO
FB
L1B
V
25k
185k
REF
CTRL
ISP
INTV
LT3756-2
CC
0.1Ω
1A
100k
ISN
M1
OPENLED
PWM
SS
RT
V
GATE
20W
LED
STRING
SENSE
PWMOUT
0.033Ω
0.01µF
0
40
(V)
60
80
20
GND INTV
C
CC
V
IN
28.7k
400kHz
C2
4.7µF
10V
375612 TA04b
30k
0.001µF
M2
375612 TA04a
L1: COILCRAFT MSD1278T-333
M1: VISHAY SILICONIX Si7430DP
D1: ON SEMICONDUCTOR MBRS3200T
M2: ZETEX ZXM61N03F
375612fb
ꢁ0
LT3756/LT3756-1/LT3756-2
package DescripTion
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev A)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 p 0.102
(.112 p .004)
2.845 p 0.102
(.112 p .004)
0.889 p 0.127
(.035 p .005)
1
8
0.35
REF
5.23
(.206)
MIN
1.651 p 0.102
(.065 p .004)
1.651 p 0.102
(.065 p .004)
3.20 – 3.45
(.126 – .136)
0.12 REF
DETAIL “B”
CORNER TAIL IS PART OF
THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
DETAIL “B”
16
9
0.305 p 0.038
0.50
(.0197)
BSC
NO MEASUREMENT PURPOSE
4.039 p 0.102
(.159 p .004)
(NOTE 3)
(.0120 p .0015)
TYP
0.280 p 0.076
(.011 p .003)
RECOMMENDED SOLDER PAD LAYOUT
16151413121110
9
REF
DETAIL “A”
0o – 6o TYP
0.254
(.010)
3.00 p 0.102
(.118 p .004)
(NOTE 4)
4.90 p 0.152
(.193 p .006)
GAUGE PLANE
0.53 p 0.152
(.021 p .006)
1 2 3 4 5 6 7 8
DETAIL “A”
0.86
(.034)
REF
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
0.1016 p 0.0508
(.004 p .002)
MSOP (MSE16) 0608 REV A
0.50
(.0197)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
375612fb
ꢁꢀ
LT3756/LT3756-1/LT3756-2
package DescripTion
UD Package
16-Lead Plastic QFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1691)
0.70 ±0.05
3.50 ± 0.05
2.10 ± 0.05
1.45 ± 0.05
(4 SIDES)
PACKAGE OUTLINE
0.25 ±0.05
0.50 BSC
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
BOTTOM VIEW—EXPOSED PAD
PIN 1 NOTCH R = 0.20 TYP
OR 0.25 × 45° CHAMFER
R = 0.115
TYP
0.75 ± 0.05
3.00 ± 0.10
(4 SIDES)
15 16
PIN 1
TOP MARK
(NOTE 6)
0.40 ± 0.10
1
2
1.45 ± 0.10
(4-SIDES)
(UD16) QFN 0904
0.200 REF
0.25 ± 0.05
0.00 – 0.05
0.50 BSC
NOTE:
1. DRAWING CONFORMS TO JEDEC PACKAGE OUTLINE MO-220 VARIATION (WEED-2)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
375612fb
ꢁꢁ
LT3756/LT3756-1/LT3756-2
revision hisTory (Revision history begins at Rev B)
REV
DATE
DESCRIPTION
PAGE NUMBER
B
03/10 Revised Entire Data Sheet to Include H-Grade
1-24
375612fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
ꢁꢂ
LT3756/LT3756-1/LT3756-2
Typical applicaTion
Buck Mode 1A LED Driver with High Dimming Ratio and Open LED Reporting
Efficiency vs VIN
100
96
92
88
84
80
V
IN
C3
24V TO
80V
4.7µF
s5
C1
1µF
s2
1M
200k
200k
200k
V
ISP
IN
SHDN/UVLO
25V
0.1Ω
1A
1.5k
61.9k
Q2
V
ISN
FB
REF
CTRL
PWM
20k
M2
Q1
1k
PWMOUT
5 WHITE LEDs
20W
LT3756-2
INTV
CC
100k
L1
33µH
20
50
(V)
60
70
80
30
40
D1
V
OPENLED
IN
V
IN
375612 TA05b
C4
4.7µF
M1
SS
GATE
PWM Dimming Waveforms
RT
VC
SENSE
0.1µF
GND INTV
CC
0.033Ω
28.7k
375kHz
C2
4.7µF
47k
V
PWM
M1: VISHAY SILICONIX Si3430DV
D1: DIODES INC B1100/B
L1: WÜRTH 74456133
M2: VISHAY SILICONIX Si5435BDC
Q1: ZETEX FMMT493
Q2: ZETEX FMMT593
0.001µF
V
SW
50V/DIV
375612 TA05a
1A
I
LED
0A
375612 TA05c
10µs/DIV
relaTeD parTs
PART NUMBER
DESCRIPTION
36V, 1A (I ), 2MHz, Step-Down LED Driver
COMMENTS
LT3474
V : 4V to 36V, V
= 13.5V, True Color PWM Dimming = 400:1,
LED
IN
SD
OUT(MAX)
I
< 1µA, TSSOP16E Package
LT3475
Dual 1.5A (I ), 36V, 2MHz Step-Down LED Driver
V : 4V to 36V, V
SD
= 13.5V, True Color PWM Dimming = 3000:1,
LED
IN
OUT(MAX)
I
< 1µA, TSSOP20E Package
LT3476
Quad Output 1.5A, 36V, 2MHz High Current LED Driver V : 2.8V to 16V, V
= 36V, True Color PWM Dimming = 1000:1,
OUT(MAX)
IN
with 1000:1 Dimming
I
< 10µA, 5mm × 7mm QFN Package
SD
LT3477
3A, 42V, 3MHz Boost, Buck-Boost, Buck LED Driver
V : 2.5V to 25V, V
= 40V, Dimming = Analog/PWM, I < 1µA,
OUT(MAX) SD
IN
QFN and TSSOP20E Packages
LT3478/LT3478-1
LT3486
4.5A, 42V, 2.5MHz High Current LED Driver with
3000:1 Dimming
V : 2.8V to 36V, V
SD
= 42V, True Color PWM Dimming = 3000:1,
IN
OUT(MAX)
I
< 3µA, TSSOP16E Package
Dual 1.3A, 2MHz High Current LED Driver
Triple 0.75A, 2.1MHz, 45V LED Driver
1.5A, 2.5MHz, 45V LED Driver
V : 2.5V to 24V, V
= 36V, True Color PWM Dimming = 1000:1,
IN
SD
OUT(MAX)
I
< 1µA, 5mm × 3mm DFN and TSSOP16E Packages
LT3496
V : 3V to 30V, V
= 45V, Dimming = 3000:1, I < 1µA,
IN
OUT(MAX) SD
4mm × 5mm QFN and TSSOP16E Packages
LT3517
V : 3V to 30V, V = 45V, Dimming = 3000:1, I < 1µA,
IN
OUT(MAX)
SD
4mm × 4mm QFN and TSSOP16E Packages
LT3518
2.3A, 2.5MHz, 45V LED Driver
V : 3V to 30V, V = 45V, Dimming = 3000:1, I < 1µA,
IN
OUT(MAX)
SD
4mm × 4mm QFN and TSSOP16E Packages
LT3755/LT3755-1/
LT3755-2
40V , 75V , Full Featured LED Controller
IN OUT
V : 4.5V to 40V, V = 75V, True Color PWM Dimming = 3000:1,
SD
IN
OUT(MAX)
I
< 1µA, 3mm × 3mm QFN-16 and MS16E Packages
LTC®3783
High Current LED Controller
V : 3V to 36V, V
SD
= Ext FET, True Color PWM Dimming = 3000:1,
IN
OUT(MAX)
I
< 20µA, 5mm × 4mm QFN10 and TSSOP16E Packages
375612fb
LT 0310 REV B • PRINTED IN USA
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
ꢁꢃ
●
●
LINEAR TECHNOLOGY CORPORATION 2008
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
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