LTC1435AI [Linear]
High Efficiency Low Noise Synchronous Step-Down Switching Regulator; 高效率，低噪声同步降压型开关稳压器
| 型号： | LTC1435AI |
| 厂家： | 
Linear |
| 描述： | High Efficiency Low Noise Synchronous Step-Down Switching Regulator |
| 文件： | 总20页 (文件大小：430K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC1435A
High Efficiency Low Noise
Synchronous Step-Down
Switching Regulator
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FEATURES
DESCRIPTION
The LTC®1435A is a synchronous step-down switching
regulator controller that drives external N-channel power
MOSFETs using a fixed frequency architecture. A wide
dutycyclerangeof5%to99%allowshighVIN tolowVOUT
DC/DC conversion, as well as low dropout operation that
extendsoperatingtimeinbattery-operatedsystems.Burst
ModeTM operation provides high efficiency at low load
currents.
■
Dual N-Channel MOSFET Synchronous Drive
■
Programmable Fixed Frequency
■
Wide VIN Range: 3.5V to 36V Operation
Low Minimum On-Time (≤300ns) for High
■
Frequency, Low Duty Cycle Applications
Very Low Dropout Operation: 99% Duty Cycle
Low Standby Current
■
■
■
■
■
■
■
■
Secondary Feedback Control
Programmable Soft Start
Remote Output Voltage Sense
Logic Controlled Micropower Shutdown: IQ < 25µA
Foldback Current Limiting (Optional)
Current Mode Operation for Excellent Line and Load
Transient Response
The operating frequency is set by an external capacitor
allowing maximum flexibility in optimizing efficiency. A
secondary winding feedback control pin, SFB, guarantees
regulation regardless of load on the main output by
forcing continuous operation. Burst Mode operation is
inhibited when the SFB pin is pulled low, which reduces
noise and RF interference.
■
■
Output Voltages from 1.19V to 9V
Available in 16-LUead Narrow SO and SSOP Packages
Soft start is provided by an external capacitor that can be
used to properly sequence supplies. The operating cur-
rent level is user-programmable via an external current
sense resistor. Wide input supply range allows operation
from 3.5V to 30V (36V maximum).
APPLICATIONS
■
Notebook and Palmtop Computers, PDAs
Cellular Telephones and Wireless Modems
Portable Instruments
Battery-Operated Devices
DC Power Distribution Systems
■
■
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
■
■
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TYPICAL APPLICATION
V
IN
4.5V TO 22V
C
V
IN
OSC
C
IN
+
C
OSC
43pF
22µF
35V
× 2
M1
Si4412DY
RUN/SS
TG
C
R
SS
0.1µF
SENSE
0.033Ω
V
OUT
1.6V/3A
I
SW
TH
L1
C
D
C
B
LTC1435A
INTV
4.7µH
330pF
CMDSH-3
R1
C
C
R
B
OUT
C
35.7k
CC
+
0.1µF
100µF
6.3V
× 2
10k
SGND
BOOST
R2
102k
+
D1
MBRS140T3
100pF
4.7µF
M2
Si4412DY
V
BG
OSENSE
PGND
SENSE
–
+
SENSE
1000pF
1435A F01
Figure 1. High Efficiency Step-Down Converter
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LTC1435A
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
Input Supply Voltage (VIN).........................36V to –0.3V
Topside Driver Supply Voltage (BOOST)....42V to –0.3V
Switch Voltage (SW)............................. VIN + 5V to –5V
EXTVCC Voltage ........................................ 10V to –0.3V
SENSE+, SENSE– Voltages...... INTVCC + 0.3V to –0.3V
ITH, VOSENSE Voltages .............................. 2.7V to –0.3V
SFB, Run/SS Voltages .............................. 10V to –0.3V
Peak Driver Output Current < 10µs (TG, BG) ............. 2A
INTVCC Output Current ........................................ 50mA
Operating Ambient Temperature Range
LTC1435AC ............................................ 0°C to 70°C
LTC1435AI ......................................... –40°C to 85°C
Junction Temperature (Note 1)............................. 125°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
TOP VIEW
ORDER PART
NUMBER
C
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
TG
OSC
RUN/SS
BOOST
SW
LTC1435ACG
LTC1435ACS
LTC1435AIG
LTC1435AIS
I
TH
SFB
V
IN
SGND
INTV
BG
CC
V
OSENSE
–
SENSE
PGND
EXTV
+
SENSE
CC
G PACKAGE
S PACKAGE
16-LEAD PLASTIC SSOP 16-LEAD PLASTIC SO
TJMAX = 125°C, θJA = 130°C/ W (G)
TJMAX = 125°C, θJA = 110°C/ W (S)
Consult factory for Military grade parts.
TA = 25°C, VIN = 15V, VRUN/SS = 5V unless otherwise noted.
ELECTRICAL CHARACTERISTICS
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Main Control Loop
I
V
Feedback Current
(Note 2)
(Note 2)
10
50
nA
V
IN OSENSE
V
Feedback Voltage
●
1.178 1.19
1.202
OSENSE
∆V
∆V
Reference Voltage Line Regulation
Output Voltage Load Regulation
V
= 3.6V to 20V (Note 2)
0.002 0.01
%/V
LINEREG
IN
I
I
Sinking 5µA (Note 2)
Sourcing 5µA
●
●
0.5
0.8
–0.8
%
%
LOADREG
TH
TH
–0.5
V
Secondary Feedback Threshold
Secondary Feedback Current
Output Overvoltage Lockout
V
V
Ramping Negative
= 1.5V
●
1.16
1.24
1.19
–1
1.22
–2
V
µA
V
SFB
SFB
SFB
I
SFB
V
1.28
1.32
OVL
I
Input DC Supply Current
Normal Mode
EXTV = 5V (Note 3)
CC
Q
3.6V < V < 30V
280
16
1.3
3
150
250
µA
µA
V
µA
mV
ns
IN
Shutdown
V
= 0V, 3.6V < V < 15V
25
2
RUN/SS
IN
V
Run Pin Threshold
●
0.8
1.5
130
RUN/SS
I
Soft Start Current Source
Maximum Current Sense Threshold
Minimum On-Time
V
V
= 0V
= 0V, 5V
4.5
180
300
RUN/SS
RUN/SS
OSENSE
∆V
SENSE(MAX)
–
t
Tested with Square Wave, SENSE = 1.6V,
∆V = 20mV (Note 5
ON(MIN)
)
SENSE
TG Transition Time
Rise Time
Fall Time
TG t
TG t
C
C
= 3000pF
= 3000pF
50
50
150
150
ns
ns
r
f
LOAD
LOAD
BG Transition Time
Rise Time
Fall Time
BG t
BG t
C
C
= 3000pF
= 3000pF
50
40
150
150
ns
ns
r
f
LOAD
LOAD
Internal V Regulator
CC
V
V
V
V
Internal V Voltage
6V < V < 30V, V = 4V
EXTVCC
●
●
4.8
4.5
5.0
–0.2
130
4.7
5.2
–1
230
V
%
mV
V
INTVCC
CC
IN
INT
EXT
INTV Load Regulation
I
I
I
= 15mA, V
= 15mA, V
= 15mA, V
= 4V
= 5V
LDO
LDO
CC
INTVCC
INTVCC
INTVCC
EXTVCC
EXTVCC
EXTVCC
EXTV Voltage Drop
CC
EXTV Switchover Voltage
Ramping Positive
EXTVCC
CC
2
LTC1435A
ELECTRICAL CHARACTERISTICS TA = 25°C, VIN = 15V, VRUN/SS = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Oscillator
f
Oscillator Frequency
C
OSC
= 100pF (Note 4)
112
125
138
kHz
OSC
The
temperature range.
LTC1435ACG/LTC1435ACS: 0°C ≤ T ≤ 70°C
●
denotes specifications which apply over the full operating
Note 3: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
A
Note 4: Oscillator frequency is tested by measuring the C
charge and
OSC
LTC1435AIG/LTC1435AIS: –40°C ≤ T ≤ 85°C
A
discharge currents and applying the formula:
Note 1: T is calculated from the ambient temperature T and power
8.4(108)
J
A
–1
1
1
+
f
(kHz) =
OSC
dissipation P according to the following formula:
D
(
C
) (
I
)
(pF) + 11
I
OSC
CHG DIS
LTC1435ACG/LTC1435AIG: T = T + (P )(130°C/W)
J
A
D
Note 5: The minimum on-time test condition corresponds to an inductor
(see Minimum On-Time
Considerations in the Applications Information section).
LTC1435ACS/LTC1435AIS: T = T + (P )(110°C/W)
J
A
D
peak-to-peak ripple current ≥40% of I
MAX
Note 2: The LTC1435A is tested in a feedback loop which servos V
OSENSE
to the balance point for the error amplifier (V = 1.19V).
ITH
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TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Input Voltage
VOUT = 3.3V
Efficiency vs Input Voltage
VOUT = 5V
Efficiency vs Load Current
100
95
90
85
80
75
70
65
60
55
50
100
95
90
85
80
75
70
100
95
90
85
80
75
70
V
V
= 10V
IN
V
= 3.3V
V
= 5V
OUT
OUT
= 5V
OUT
R
= 0.033Ω
SENSE
I
= 1A
LOAD
I
= 1A
LOAD
I
= 100mA
LOAD
CONTINUOUS
MODE
Burst Mode
OPERATION
I
= 100mA
LOAD
0
10
15
20
25
30
0.001
0.01
0.1
1
10
0
10
15
20
25
30
5
5
INPUT VOLTAGE (V)
LOAD CURRENT (A)
INPUT VOLTAGE (V)
1435A G02
1435A G03
1435A G01
VIN – VOUT Dropout Voltage
vs Load Current
Load Regulation
VITH Pin Voltage vs Output Current
0
–0.25
–0.50
–0.75
–1.00
–1.25
–1.50
3.0
2.5
0.5
0.4
0.3
0.2
0.1
R
= 0.033Ω
R
OUT
= 0.033Ω
SENSE
SENSE
V
DROP OF 5%
2.0
1.5
1.0
0.5
0
Burst Mode
OPERATION
CONTINUOUS
MODE
0
0
1.0
1.5
2.0
2.5
3.0
0.5
0
10 20 30 40 50 60 70 80 90 100
OUTPUT CURRENT (%)
0
0.5
1.0
1.5
2.0
2.5
3.0
LOAD CURRENT (A)
LOAD CURRENT (A)
1435A G05
1435A G06
1435A G04
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LTC1435A
TYPICAL PERFORMANCE CHARACTERISTICS
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EXTVCC Switch Drop
vs INTVCC Load Current
Input Supply and Shutdown
Current vs Input Voltage
INTVCC Regulation
vs INTVCC Load Current
2.5
2.0
1.5
1.0
100
80
200
180
160
140
120
100
80
0.5
0.3
V
= 0V
EXTVCC
70°C
V
= 5V
OUT
25°C
60
70°C
25°C
EXTV = V
CC
OUT
0
–55°C
V
OUT
= 3.3V
40
EXTV = OPEN
CC
60
–0.3
–0.5
40
0.5
0
20
0
20
SHUTDOWN
10
INPUT VOLTAGE (V)
0
0
15
20
25
30
0
2
4
6
12 14 16 18 20
5
10
INTV LOAD CURRENT (mA)
8
10
0
15
20
5
INTV LOAD CURRENT (mA)
CC
CC
1435A G07
1435A G09
1435A G08
Normalized Oscillator Frequency
vs Temperature
RUN/SS Pin Current
vs Temperature
SFB Pin Current vs Temperature
10
5
0
–0.25
–0.50
–0.75
4
3
2
1
f
O
–1.00
–1.25
–1.50
–5
–10
0
60
TEMPERATURE (°C)
110 135
–40 –15 10
35
60
85 110 135
60
TEMPERATURE (°C)
110 135
–40 –15
10
35
85
–40 –15
10
35
85
TEMPERATURE (°C)
1435A G10
1435A G11
1435A G12
Maximum Current Sense
Threshold Voltage vs Temperature
Transient Response
Transient Response
154
152
150
148
VOUT
50mV/DIV
VOUT
50mV/DIV
ILOAD = 1A to 3A
1435A G15
I
LOAD = 50mA to 1A
1435A G14
146
–40 –15 10
35
60
85 110 135
TEMPERATURE (°C)
1435A G13
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LTC1435A
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TYPICAL PERFORMANCE CHARACTERISTICS
Soft Start: Load Current vs Time
Burst Mode Operation
VOUT
20mV/DIV
RUN/SS
5V/DIV
INDUCTOR
CURRENT
1A/DIV
VITH
200mV/DIV
1435A G17
ILOAD = 50mA
1435A G16
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PIN FUNCTIONS
ever EXTVCC is higher than 4.7V. See EXTVCC connection
in Applications Information section. Do notexceed10V on
this pin. Connect to VOUT if VOUT ≥ 5V.
COSC (Pin 1): External capacitor COSC from this pin to
ground sets the operating frequency.
RUN/SS (Pin 2): Combination of Soft Start and Run
Control Inputs. A capacitor to ground at this pin sets the
ramp timeto fullcurrentoutput. The timeis approximately
0.5s/µF. Forcing this pin below 1.3V causes the device to
be shut down. In shutdown all functions are disabled.
PGND (Pin 10): Driver Power Ground. Connects to source
of bottom N-channel MOSFET and the (–) terminal of CIN.
BG (Pin 11): High Current Gate Drive for Bottom
N-Channel MOSFET. Voltage swing at this pin is from
ground to INTVCC.
ITH (Pin 3): Error Amplifier Compensation Point. The
current comparator threshold increases with this control
voltage. Nominal voltage range for this pin is 0V to 2.5V.
INTVCC (Pin 12): Output of the Internal 5V Regulator and
EXTVCC Switch. The driver and control circuits are pow-
ered from this voltage. Must be closely decoupled to power
ground with a minimum of 2.2µF tantalum or electrolytic
capacitor.
SFB (Pin 4): Secondary Winding Feedback Input. Nor-
mally connected to a feedback resistive divider from the
secondary winding. This pin should be tied to: ground to
force continuous operation; INTVCC in applications that
don’tuseasecondarywinding;andaresistivedividerfrom
the output in applications using a secondary winding.
VIN (Pin 13): Main Supply Pin. Must be closely decoupled
to the IC’s signal ground pin.
SW (Pin 14): Switch Node Connection to Inductor. Volt-
age swing at this pin is from a Schottky diode (external)
voltage drop below ground to VIN.
SGND (Pin 5): Small-Signal Ground. Must be routed
separately from other grounds to the (–) terminal of COUT
.
VOSENSE (Pin 6): Receives the feedback voltage from an
BOOST (Pin 15): Supply to Topside Floating Driver. The
bootstrap capacitor is returned to this pin. Voltage swing
at this pin is from INTVCC to VIN + INTVCC.
external resistive divider across the output.
SENSE– (Pin 7): The (–) Input to the Current Comparator.
SENSE+ (Pin 8): The (+) Input to the Current Comparator.
Built-in offsets between SENSE– and SENSE+ pins in
conjunction with RSENSE set the current trip thresholds.
TG (Pin 16): High Current Gate Drive for Top N-Channel
MOSFET. This is the output of a floating driver with a
voltage swing equal to INTVCC superimposed on the
switch node voltage SW.
EXTVCC (Pin 9): Input to the Internal Switch Connected to
INTVCC. This switch closes and supplies VCC power when-
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LTC1435A
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FUNCTIONAL DIAGRA
V
IN
+
C
IN
C
OSC
1
C
OSC
SFB
13
V
IN
SGND 5
4
INTV
CC
1.19V
REF
D
B
1µA
BOOST
15
–
C
B
1.19V
+
TG
16
SHUTDOWN
OSC
DROP
OUT
DET
OV
+
–
S
R
Q
SWITCH
LOGIC
1.28V
0.6V
+
–
SW
14
V
OSENSE
6
V
SEC
I
2
V
FB
–
+
–
+
–
+
I
1
D1
EA
R2
4k
1.19V
Ω
g
m
= 1m
180k
+
V
IN
INTV
CC
C
SEC
+
–
INTV
CC
12
+
SHUTDOWN
5V
LDO
REG
3µA
R1
RUN
SOFT
START
4.8V
+
–
BG
11
30k
8k
V
OUT
6V
+
C
OUT
PGND
10
R
C
+
–
2
8
7
9
EXTV
CC
RUN/SS
3
SENSE
SENSE
I
TH
C
SS
C
C
D *
FB
R
SENSE
1435A • FD
* FOLDBACK CURRENT LIMITING OPTION
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(Refer to Functional Diagram)
OPERATION
Main Control Loop
erence,whichinturncausestheITHvoltagetoincreaseuntil
theaverageinductorcurrentmatchesthenewloadcurrent.
WhilethetopMOSFETisoff,thebottomMOSFETisturned
on until either the inductor current starts to reverse, as
indicated by current comparator I2, or the beginning of the
next cycle.
The LTC1435A uses a constant frequency, current mode
step-down architecture. During normal operation, the top
MOSFET is turned on each cycle when the oscillator sets
the RS latch, and turned off when the main current com-
parator I1 resets the RS latch. The peak inductor current at
which I1 resets the RS latch is controlled by the voltage on
the ITH pin , which is the output of error amplifier EA. The
VOSENSEpin,describedinthePinFunctionssection,allows
EA to receive an output feedback voltage VFB from an ex-
ternal resistive divider. When the load current increases,
it causes a slight decrease in VFB relative to the 1.19V ref-
The top MOSFET driver is biased from floating bootstrap
capacitor CB, which normally is recharged during each off
cycle. However, when VIN decreases to a voltage close to
VOUT, the loop may enter dropout and attempt to turn on
thetopMOSFETcontinuously.Thedropoutdetectorcounts
thenumberofoscillatorcyclesthatthetopMOSFETremains
6
LTC1435A
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(Refer to Functional Diagram)
OPERATION
on and periodically forces a brief off period to allow CB to either of which causes drive to be returned to the TG pin
recharge. on the next cycle.
The main control loop is shut down by pulling the RUN/SS Twoconditionscanforcecontinuoussynchronousopera-
pin low. Releasing RUN/SS allows an internal 3µA current tion, even when the load current would otherwise dictate
sourcetochargesoftstartcapacitorCSS.WhenCSS reaches low current operation. One is when the common mode
1.3V, the main control loop is enabled with the ITH voltage voltage of the SENSE+ and SENSE– pins is below 1.4V and
clamped at approximately 30% of its maximum value. As the other is when the SFB pin is below 1.19V. The latter
C
SS continuestocharge, ITH isgraduallyreleasedallowing conditionisusedtoassistinsecondarywindingregulation
normal operation to resume.
as described in the Applications Information section.
Comparator OV guards against transient overshoots
> 7.5% by turning off the top MOSFET and keeping it off
until the fault is removed.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
oftheotherLTC1435AcircuitryisderivedfromtheINTVCC
pin. The bottom MOSFET driver supply pin is internally
connectedtoINTVCC intheLTC1435A. WhentheEXTVCC
pin is left open, an internal 5V low dropout regulator
suppliesINTVCC power.IfEXTVCC istakenabove4.8V,the
5V regulator is turned off and an internal switch is turned
on to connect EXTVCC to INTVCC. This allows the INTVCC
powertobederivedfromahighefficiencyexternalsource
such as the output of the regulator itself or a secondary
winding, as described in the Applications Information
section.
Low Current Operation
TheLTC1435AiscapableofBurstModeoperationinwhich
theexternalMOSFETsoperateintermittentlybasedonload
demand. The transition to low current operation begins
when comparator I2 detects current reversal and turns off
thebottomMOSFET. IfthevoltageacrossRSENSE doesnot
exceed the hysteresis of I2 (approximately 20mV) for one
fullcycle,thenonfollowingcyclesthetopandbottomdrives
are disabled. This continues until an inductor current peak
exceeds 20mV/RSENSE or the ITH voltage exceeds 0.6V,
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APPLICATIONS INFORMATION
Allowing a margin for variations in the LTC1435A and
external component values yields:
The basic LTC1435A application circuit is shown in Figure
1, High Efficiency Step-Down Converter. External compo-
nentselectionisdrivenbytheloadrequirementandbegins
with the selection of RSENSE. Once RSENSE is known, COSC
andLcanbechosen.Next,thepowerMOSFETsandD1are
selected. Finally, CIN and COUT are selected. The circuit
shown in Figure 1 can be configured for operation up to an
input voltage of 28V (limited by the external MOSFETs).
100mV
R
=
SENSE
I
MAX
The LTC1435A works well with RSENSE values ≥ 0.005Ω.
COSC Selection for Operating Frequency
TheLTC1435Ausesaconstantfrequencyarchitecturewith
the frequency determined by an external oscillator capaci-
tor COSC. Each time the topside MOSFET turns on, the
voltage COSC is reset to ground. During the on-time, COSC
is charged by a fixed current. When the voltage on the ca-
pacitorreaches1.19V,COSC isresettoground.Theprocess
then repeats.
RSENSE Selection for Output Current
RSENSE ischosenbasedontherequiredoutputcurrent.The
LTC1435A current comparator has a maximum threshold
of 150mV/RSENSE and an input common mode range of
SGND to INTVCC. The current comparator threshold sets
the peak of the inductor current, yielding a maximum av-
erage output current IMAX equal to the peak value less half
the peak-to-peak ripple current ∆IL.
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LTC1435A
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APPLICATIONS INFORMATION
greater core losses. A reasonable starting point for setting
ripplecurrentis∆IL =0.4(IMAX).Remember,themaximum
∆IL occurs at the maximum input voltage.
The value of COSC is calculated from the desired operating
frequency:
4
1.37(10 )
Frequency (kHz)
Theinductorvaluealsohasaneffectonlowcurrentopera-
tion. The transition to low current operation begins when
theinductorcurrentreacheszerowhilethebottomMOSFET
is on. Lower inductor values (higher ∆IL) will cause this to
occur at higher load currents, which can cause a dip in
efficiency in the upper range of low current operation. In
Burst Mode operation, lower inductance values will cause
the burst frequency to decrease.
C
(pF) =
– 11
OSC
A graph for selecting COSC vs frequency is given in Figure
2. As the operating frequency is increased the gate charge
losses will be higher, reducing efficiency (see Efficiency
Considerations). The maximum recommended switching
frequency is 400kHz.
300
250
200
150
100
50
The Figure 3 graph gives a range of recommended induc-
tor values vs operating frequency and VOUT
.
60
V
OUT
V
OUT
V
OUT
= 5.0V
= 3.3V
≤ 2.5V
50
40
30
20
10
0
0
0
100
200
300
400
500
OPERATING FREQUENCY (kHz)
1435A F02
Figure 2. Timing Capacitor Value
0
100
150
200
250
300
50
OPERATING FREQUENCY (kHz)
1435A F03
Inductor Value Calculation
Figure 3. Recommended Inductor Values
The operating frequency and inductor selection are inter-
related in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because of
MOSFETgatechargelosses.Inadditiontothisbasictrade-
off, the effect of inductor value on ripple current and low
current operation must also be considered.
For low duty cycle, high frequency applications where the
required minimum on-time,
VOUT
tON(MIN)
=
,
V
f
(
IN(MAX))( )
is less than 350ns, there may be further restrictions on the
inductance to ensure proper operation. See Minimum On-
Time Considerations section for more details.
Theinductorvaluehasadirecteffectonripplecurrent.The
inductor ripple current ∆IL decreases with higher induc-
tance or frequency and increases with higher VIN or VOUT
Inductor Core Selection
:
Once the value for L is known, the type of inductor must be
selected.Highefficiencyconvertersgenerallycannotafford
the core loss found in low cost powdered iron cores, forc-
ingtheuseofmoreexpensiveferrite,molypermalloyorKool
Mµ® cores. Actual core loss is independent of core size for
Kool Mµ is a registered trademark of Magnetics, Inc.
1
V
OUT
∆I =
V
1–
L
OUT
f L
( )( )
V
IN
Accepting larger values of∆IL allows the use of low induc-
tances, but results in higher output voltage ripple and
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a fixed inductor value, but it is very dependent on induc-
tance selected. As inductance increases, core losses go
down. Unfortunately, increased inductance requires more
turns of wire and therefore copper losses will increase.
V
V
OUT
Main Switch Duty Cycle =
IN
V − V
(
)
IN
OUT
Synchronous Switch Duty Cycle =
V
IN
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can
concentrate on copper loss and preventing saturation.
Ferritecorematerialsaturates“hard,”whichmeansthatin-
ductance collapses abruptly when the peak design current
is exceeded. This results in an abrupt increase in inductor
ripplecurrentandconsequentoutputvoltageripple.Donot
allow the core to saturate!
The MOSFET power dissipations at maximum output cur-
rent are given by:
V
V
2
OUT
P
=
I
(
1+δ R
+
) (
)
MAIN
MAX
DS ON
(
)
IN
1.85
k V
I
(
C
f
(
)
)(
)( )
IN
MAX
RSS
Molypermalloy (from Magnetics, Inc.) is a very good, low
losscorematerialfortoroids,butitismoreexpensivethan
ferrite.Areasonablecompromisefromthesamemanufac-
turerisKoolMµ.Toroidsareveryspaceefficient,especially
when you can use several layers of wire. Because they
generally lack a bobbin, mounting is more difficult. How-
ever, designs for surface mount are available which do not
increase the height significantly.
V − V
2
IN
OUT
P
=
I
(
1+δ R
) (
)
SYNC
MAX
DS ON
(
)
V
IN
where δ is the temperature dependency of RDS(ON) and k
is a constant inversely related to the gate drive current.
Both MOSFETs have I2R losses while the topside
N-channel equation includes an additional term for tran-
sition losses, which are highest at high input voltages.
For VIN < 20V the high current efficiency generally im-
proves with larger MOSFETs, while for VIN > 20V the
transition losses rapidly increase to the point that the use
of a higher RDS(ON) device with lower CRSS actual pro-
videshigherefficiency.ThesynchronousMOSFETlosses
are greatest at high input voltage or during a short circuit
when the duty cycle in this switch is nearly 100%. Refer
totheFoldbackCurrentLimitingsectionforfurtherappli-
cations information.
Power MOSFET and D1 Selection
TwoexternalpowerMOSFETsmustbeselectedforusewith
the LTC1435A: an N-channel MOSFET for the top (main)
switchandanN-channelMOSFETforthebottom(synchro-
nous) switch.
The peak-to-peak gate drive levels are set by the INTVCC
voltage. This voltage is typically 5V during start-up (see
EXTVCC PinConnection).Consequently,logiclevelthresh-
oldMOSFETsmustbeusedinmostLTC1435Aapplications.
The only exception is applications in which EXTVCC is
poweredfromanexternalsupplygreaterthan8V(mustbe
less than 10V), in which standard threshold MOSFETs
(VGS(TH)<4V)maybeused.PaycloseattentiontotheBVDSS
specificationfortheMOSFETsaswell;manyofthelogiclevel
MOSFETs are limited to 30V or less.
Theterm(1+δ)isgenerallygivenforaMOSFETintheform
of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltageMOSFETs.CRSS isusuallyspecifiedintheMOSFET
characteristics. The constant k = 2.5 can be used to esti-
mate the contributions of the two terms in the main switch
dissipation equation.
SelectioncriteriaforthepowerMOSFETsincludethe“ON”
resistance RDS(ON), reverse transfer capacitance CRSS, in-
put voltage and maximum output current. When the
LTC1435Aisoperatingincontinuousmodethedutycycles
for the top and bottom MOSFETs are given by:
The Schottky diode D1 shown in Figure 1 conducts during
the dead-time between the conduction of the two large
power MOSFETs. This prevents the body diode of the bot-
tomMOSFETfromturningonandstoringchargeduringthe
dead-time, which could cost as much as 1% in efficiency.
A 1A Schottky is generally a good size for 3A regulators.
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CIN and COUT Selection
Insurfacemountapplicationsmultiplecapacitorsmayhave
to be paralleled to meet the ESR or RMS current handling
requirementsoftheapplication.Aluminumelectrolyticand
drytantalumcapacitorsarebothavailableinsurfacemount
configurations. Inthecaseoftantalum, itiscriticalthatthe
capacitors are surge tested for use in switching power
supplies. An excellent choice is the AVX TPS series of
surface mount tantalum, available in case heights ranging
from 2mm to 4mm. Other capacitor types include Sanyo
OS-CON, Nichicon PL series and Sprague 593D and 595D
series. Consultthemanufacturerforotherspecificrecom-
mendations.
In continuous mode, the source current of the top
N-channel MOSFET is a square wave of duty cycle VOUT
/
VIN. To prevent large voltage transients, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS capacitor current is given by:
1/2
]
V
V − V
OUT
(
)
OUT IN
[
C required I
≈I
IN
RMS MAX
V
IN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is com-
monlyusedfordesignbecauseevensignificantdeviations
donotoffermuchrelief.Notethatcapacitormanufacturer’s
ripple current ratings are often based on only 2000 hours
of life. This makes it advisable to further derate the capaci-
tor or to choose a capacitor rated at a higher temperature
thanrequired. Severalcapacitorsmayalsobeparalleledto
meet size or height requirements in the design. Always
consult the manufacturer if there is any question.
INTVCC Regulator
An internal P-channel low dropout regulator produces the
5V supply that powers the drivers and internal circuitry
within the LTC1435A. The INTVCC pin can supply up to
15mA and must be bypassed to ground with a minimum
of2.2µFtantalumorlowESRelectrolytic. Goodbypassing
isnecessarytosupplythehightransientcurrentsrequired
by the MOSFET gate drivers.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR require-
ment is satisfied the capacitance is adequate for filtering.
The output ripple (∆VOUT) is approximated by:
High input voltage applications, in which large MOSFETs
are being driven at high frequencies, may cause the maxi-
mum junction temperature rating for the LTC1435A to be
exceeded. The IC supply current is dominated by the gate
charge supply current when not using an output derived
EXTVCC source. The gate charge is dependent on operat-
ingfrequencyasdiscussedintheEfficiencyConsiderations
section.Thejunctiontemperaturecanbeestimatedbyusing
the equations given in Note 1 of the Electrical Character-
istics. For example, the LTC1435A is limited to less than
17mA from a 30V supply:
1
∆V
≈ ∆I ESR +
L
OUT
4fC
OUT
where f = operating frequency, COUT = output capacitance
and ∆IL= ripple current in the inductor. The output ripple
is highest at maximum input voltage since ∆IL increases
withinputvoltage.With∆IL =0.4IOUT(MAX)theoutputripple
will be less than 100mV at max VIN assuming:
TJ = 70°C + (17mA)(30V)(100°C/W) = 126°C
COUT required ESR < 2RSENSE
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked when
operating in continuous mode at maximum VIN.
Manufacturers such as Nichicon, United Chemicon and
Sanyoshouldbeconsideredforhighperformancethrough-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR(size)
product of any aluminum electrolytic at a somewhat
higher price. Once the ESR requirement for COUT has been
met, the RMS current rating generally far exceeds the
IRIPPLE(P-P) requirement.
EXTVCC Connection
The LTC1435A contains an internal P-channel MOSFET
switchconnectedbetweentheEXTVCCandINTVCCpins.The
switchclosesandsuppliestheINTVCC powerwheneverthe
EXTVCC pinisabove4.8V,andremainscloseduntilEXTVCC
drops below 4.5V. This allows the MOSFET driver and
10
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controlpowertobederivedfromtheoutputduringnormal
operation (4.8V < VOUT < 9V) and from the internal regu-
lator when the output is out of regulation (start-up, short
circuit). Do not apply greater than 10V to the EXTVCC pin
and ensure that EXTVCC < VIN.
+
V
IN
C
IN
1N4148
V
SEC
V
IN
+
L1
1:N
1µF
N-CH
N-CH
TG
OPTIONAL
EXT V
R
CC
SENSE
EXTV
CC
CONNECTION
Significant efficiency gains can be realized by powering
INTVCC fromtheoutput,sincetheVINcurrentresultingfrom
the driver and control currents will be scaled by a factor of
DutyCycle/Efficiency.For5Vregulatorsthissupplymeans
connecting the EXTVCC pin directly to VOUT. However, for
3.3Vandotherlowervoltageregulators,additionalcircuitry
is required to derive INTVCC power from the output.
V
OUT
5V ≤ V
≤ 9V
SEC
+
R6
R5
LTC1435A
C
OUT
SW
SFB
BG
SGND
PGND
1435A F04a
Figure 4a. Secondary Output Loop and EXTVCC Connection
Thefollowinglistsummarizesthefourpossibleconnections
for EXTVCC:
+
V
+
IN
1µF
C
IN
1. EXTVCC left open (or grounded). This will cause INTVCC
to be powered from the internal 5V regulator resulting
in an efficiency penalty of up to 10% at high input volt-
ages.
0.22µF
BAT85
BAT85
BAT85
L1
V
IN
N-CH
N-CH
TG
VN2222LL
R
EXTV
CC
SENSE
2. EXTVCC connected directly to VOUT. This is the normal
connection for a 5V regulator and provides the highest
efficiency.
V
OUT
LTC1435A
+
SW
C
OUT
BG
1435A F04b
3. EXTVCC connectedtoanoutput-derivedboostnetwork.
For 3.3V and other low voltage regulators, efficiency
gains can still be realized by connecting EXTVCC to an
output-derived voltage which has been boosted to
greater than 4.8V. This can be done with either the in-
ductive boost winding as shown in Figure 4a or the
capacitivechargepumpshowninFigure4b.Thecharge
pump has the advantage of simple magnetics.
PGND
Figure 4b. Capacitive Charge Pump for EXTVCC
topside MOSFET is to be turned on, the driver places the
CB voltage across the gate source of the MOSFET. This en-
hances the MOSFET and turns on the topside switch. The
switchnodevoltageSWrisestoVIN andtheBoostpinrises
to VIN + INTVCC. The value of the boost capacitor CB needs
to be 100 times greater than the total input capacitance of
the topside MOSFET. In most applications 0.1µF is ad-
equate.ThereversebreakdownonDB mustbegreaterthan
VIN(MAX).
4. EXTVCC connected to an external supply. If an external
supplyisavailableinthe5Vto10Vrange(EXTVCC≤V IN),
it may be used to power EXTVCC providing it is compat-
ible with the MOSFET gate drive requirements. When
driving standard threshold MOSFETs, the external sup-
ply must always be present during operation to prevent
MOSFET failure due to insufficient gate drive.
Output Voltage Programming
The output voltage is set by a resistive divider according
to the following formula:
Topside MOSFET Driver Supply (CB, DB)
An external bootstrap capacitor CB connected to the Boost
pinsuppliesthegatedrivevoltageforthetopsideMOSFET.
CapacitorCB intheFunctionalDiagramischargedthrough
diode DB from INTVCC when the SW pin is low. When the
R2
VOUT = 1.19V 1+
,VOUT ≥ 1.19V
R1
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The external resistive divider is connected to the output as
shown in Figure 5 allowing remote voltage sensing.
Foldback Current Limiting
AsdescribedinPowerMOSFETandD1Selection,theworst-
case dissipation for either MOSFET occurs with a short-
circuitedoutput,whenthesynchronousMOSFETconducts
the current limit value almost continuously. In most appli-
cations this will not cause excessive heating, even for
extended fault intervals. However, when heat sinking is at
a premium or higher RDS(ON) MOSFETs are being used,
foldback current limiting should be added to reduce the
current in proportion to the severity of the fault.
1.19V ≤ V
≤ 9V
OUT
R2
V
OSENSE
100pF
LTC1435A
SGND
R1
1435A F05
Figure 5. Setting the LTC1435A Output Voltage
Run/Soft Start Function
Foldback current limiting is implemented by adding diode
D
FB between the output and the ITH pin as shown in the
TheRUN/SSpinisadualpurposepinthatprovidesthesoft
startfunctionandameanstoshutdowntheLTC1435A.Soft
startreducessurgecurrentsfromVIN bygraduallyincreas-
ingtheinternalcurrentlimit.Powersupplysequencingcan
also be accomplished using this pin.
Functional Diagram. In a hard short (VOUT = 0V) the cur-
rentwillbereducedtoapproximately25%ofthemaximum
output current. This technique may be used for all applica-
tions with regulated output voltages of 1.8V or greater.
An internal 3µA current source charges up an external
capacitor CSS. When the voltage on RUN/SS reaches 1.3V
theLTC1435Abeginsoperating.AsthevoltageonRUN/SS
continues to ramp from 1.3V to 2.4V, the internal current
limit is also ramped at a proportional linear rate. The cur-
rent limit begins at approximately 50mV/RSENSE (at VRUN/
SS=1.3V)andendsat150mV/RSENSE(VRUN/SS>2.7V).The
output current thus ramps up slowly, charging the output
capacitor. If RUN/SS has been pulled all the way to ground
thereisadelaybeforestartingofapproximately500ms/µF,
followed by an additional 500ms/µF to reach full current.
SFB Pin Operation
When the SFB pin drops below its ground referenced 1.19V
threshold,continuousmodeoperationisforced.Incontinu-
ous mode, the large N-channel main and synchronous
switchesareusedregardlessoftheloadonthemainoutput.
In addition to providing a logic input to force continuous
synchronous operation, the SFB pin provides a means to
regulateaflybackwindingoutput.Continuoussynchronous
operation allows power to be drawn from the auxiliary
windingswithoutregardtotheprimaryoutputload.TheSFB
pin provides a way to force continuous synchronous op-
eration as needed by the flyback winding.
tDELAY = 5(105)CSS Seconds
PullingtheRUN/SSpinbelow1.3VputstheLTC1435Ainto
alowquiescentcurrentshutdown(IQ <25µA).Thispincan
be driven directly from logic as shown in Figure 6. Diode
D1inFigure6reducesthestartdelaybutallowsCSS toramp
up slowly for the soft start function; this diode and CSS can
be deleted if soft start is not needed. The RUN/SS pin has
an internal 6V Zener clamp (See Functional Diagram).
Thesecondaryoutputvoltageissetbytheturnsratioofthe
transformerinconjunctionwithapairofexternalresistors
returnedtotheSFBpinasshowninFigure4a. Thesecond-
ary regulated voltage, VSEC, in Figure 4a is given by:
R6
V
≈ N +1 V
> 1.19 1+
(
)
SEC
OUT
R5
where N is the turns ratio of the transformer and VOUT is
3.3V OR 5V
RUN/SS
RUN/SS
D1
the main output voltage sensed by VOSENSE
.
C
SS
C
SS
Minimum On-Time Considerations
1435 F06
Minimumon-time,tON(MIN),isthesmallestamountoftime
that the LTC1435A is capable of turning the top MOSFET
Figure 6. RUN/SS Pin Interfacing
12
LTC1435A
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on and off again. It is determined by internal timing delays
and the gate charge required to turn on the top MOSFET.
Low duty cycle applications may approach this minimum
on-time limit. If the duty cycle falls below what can be
accommodated by the minimum on-time, the LTC1435A
will begin to skip cycles. The output voltage will continue
toberegulated,buttheripplecurrentandripplevoltagewill
increase. Therefore this limit should be avoided.
Because of the sensitivity of the LTC1435A current com-
paratorwhenoperatingclosetotheminimumon-timelimit,
it is important to prevent stray magnetic flux generated by
the inductor from inducing noise on the current sense re-
sistor,whichmayoccurwhenaxialtypecoresareused.By
orienting the sense resistor on the radial axis of the induc-
tor (see Figure 8), this noise will be minimized.
INDUCTOR
The minimum on-time for the LTC1435A in a properly
configured application is less than 300ns but increases at
low ripple current amplitudes (see Figure 7). If an appli-
cationisexpectedtooperateclosetotheminimumon-time
limit, an inductor value must be chosen that is low enough
toprovidesufficientrippleamplitudetomeettheminimum
on-time requirement. To determine the proper value, use
the following procedure:
L
1435A F08
Figure 8. Allowable Inductor/RSENSE Layout Orientations
Efficiency Considerations
The efficiency of a switching regulator is equal to the out-
putpowerdividedbytheinputpowertimes100%.Itisoften
useful to analyze individual losses to determine what is
limitingtheefficiencyandwhichchangewouldproducethe
most improvement. Efficiency can be expressed as:
1. Calculate on-time at maximum supply, tON(MIN)
(1/f)(VOUT/VIN(MAX)).
=
2. Use Figure 7 to obtain the peak-to-peak inductor ripple
currentasapercentageofIMAX necessarytoachievethe
calculated tON(MIN)
3. Ripple amplitude ∆IL(MIN) = (% from Figure 7)(IMAX
where IMAX = 0.1/RSENSE
.
Efficiency = 100% – (L1 + L2 + L3 + ...)
)
whereL1, L2, etc. aretheindividuallossesasapercentage
of input power.
.
V
IN(MAX) – VOUT
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
lossesinLTC1435Acircuits.LTC1435AVINcurrent,INTVCC
current,I2Rlosses,andtopsideMOSFETtransitionlosses.
tON(MIN)
4. LMAX
=
∆IL(MIN)
ChooseaninductorlessthanorequaltothecalculatedLMAX
to ensure proper operation.
1. The VIN current is the DC supply current given in the
electricalcharacteristicswhichexcludesMOSFETdriver
and control currents. VIN current results in a small
(< 1%) loss which increases with VIN.
400
350
2. INTVCC current is the sum of the MOSFET driver and
controlcurrents.TheMOSFETdrivercurrentresultsfrom
switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched from low to high
to low again, a packet of charge dQ moves from INTVCC
toground.TheresultingdQ/dtisacurrentoutofINTVCC
that is typically much larger than the control circuit cur-
rent. In continuous mode, IGATECHG = f(QT + QB), where
QT and QB are the gate charges of the topside and bot-
tom side MOSFETs.
RECOMMENDED
REGION FOR MIN
300
ON-TIME AND
MAX EFFICIENCY
250
200
0
10
20
30
40
50
60
70
INDUCTOR RIPPLE CURRENT (% OF I
)
MAX
1435A F07
Figure 7. Minimum On-Time vs Inductor Ripple Current
13
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theFigure1circuitwillprovideadequatecompensationfor
most applications.
BypoweringEXTVCCfromanoutput-derivedsource,the
additional VIN current resulting from the driver and
control currents will be scaled by a factor of
Duty Cycle/Efficiency. For example, in a 20V to 5V ap-
plication, 10mA of INTVCC current results in approxi-
mately3mAofVIN current.Thisreducesthemidcurrent
loss from 10% or more (if the driver was powered di-
rectly from VIN) to only a few percent.
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
dischargedbypasscapacitorsareeffectivelyputinparallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
the load rise time is limited to approximately (25)(CLOAD).
Thus a 10µF capacitor would require a 250µs rise time,
limiting the charging current to about 200mA.
3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L and
RSENSE, but is “chopped” between the topside main
MOSFET and the synchronous MOSFET. If the two
MOSFETs have approximately the same RDS(ON), then
the resistance of one MOSFET can simply be summed
with the resistances of L and RSENSE to obtain I2R
losses. For example, if each RDS(ON) = 0.05Ω,
RL = 0.15Ω, and RSENSE = 0.05Ω, then the total resis-
tanceis0.25Ω.Thisresultsinlossesrangingfrom3%
to 10% as the output current increases from 0.5A to
2A.I2Rlossescausetheefficiencytodropathighoutput
currents.
Automotive Considerations:
Plugging into the Cigarette Lighter
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserveorevenrechargebatterypacksduringoperation.
But before you connect, be advised: you are plugging into
the supply from hell. The main battery line in an automo-
bileisthesourceofanumberofnastypotentialtransients,
including load dump, reverse battery and double battery.
4. Transition losses apply only to the topside MOSFET(s),
andonlywhenoperatingathighinputvoltages(typically
20Vorgreater).Transitionlossescanbeestimatedfrom:
Load dump is the result of a loose battery cable. When the
cablebreaksconnection,thefieldcollapseinthealternator
cancauseapositivespikeashighas60Vwhichtakesseveral
hundredmillisecondstodecay.Reversebatteryisjustwhat
it says, while double battery is a consequence of tow truck
operatorsfindingthata24Vjumpstartcrankscoldengines
faster than 12V.
Transition Loss = 2.5 (VIN)1.85(IMAX)(CRSS)(f)
Other losses, including CIN and COUT ESR dissipative
losses, Schottky conduction losses during dead-time,
andinductorcorelosses,generallyaccountforlessthan
2% total additional loss.
ThenetworkshowninFigure9isthemoststraightforward
approach to protect a DC/DC converter from the ravages
of an automotive battery line. The series diode prevents
current from flowing during reverse battery, while the
transientsuppressorclampstheinputvoltageduringload
dump. Note that the transient suppressor should not
Checking Transient Response
The regulator loop response can be checked by looking at
theloadtransientresponse.Switchingregulatorstakesev-
eral cycles to respond to a step in DC (resistive) load cur-
rent. When a load step occurs, VOUT immediately shifts by
an amount equal to (∆ILOAD)(ESR), where ESR is the ef-
fective series resistance of COUT. ∆ILOAD also begins to
chargeordischargeCOUTwhichgeneratesafeedbackerror
signal. The regulator loop then acts to return VOUT to its
steady-state value. During this recovery time VOUT can be
monitored for overshoot or ringing, which would indicate
astabilityproblem.TheITH externalcomponentsshownin
12V
50A I RATING
PK
V
IN
LTC1435A
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
1435A F09
Figure 9. Automotive Application Protection
14
LTC1435A
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conduct during double battery operation, but must still
clamptheinputvoltagebelowbreakdownoftheconverter.
Although the LTC1435A has a maximum input voltage of
36V, most applications will be limited to 30V by the
2
1.6V
22V
PMAIN
=
3 1+ 0.005 50°C − 25°C 0.042Ω
( )
(
)(
) (
]
)
[
1.85
+ 2.5 22V
3A 100pF 250kHz =88mW
(
)
(
)(
)(
)
MOSFET BVDSS
.
The most stringent requirement for the synchronous
N-channel MOSFET occurs when VOUT = 0 (i.e. short cir-
cuit). In this case the worst-case dissipation rises to:
Design Example
As a design example, assume VIN = 12V(nominal), VIN =
22V(max), VOUT = 1.6V, IMAX = 3A and f = 250kHz, RSENSE
and COSC can immediately be calculated:
2
P
= I
(
1+δ R
DS ON
(
)
SYNC
SC AVG
)
(
)
(
)
With the 0.033Ω sense resistor ISC(AVG) = 4A will result,
increasingtheSi4412DYdissipationto950mWatadietem-
perature of 105°C.
RSENSE = 100mV/3A = 0.033Ω
COSC = 1.37(104)/250 – 11 = 43pF
Referring to Figure 3, a 4.7µH inductor falls within the rec-
ommended range. To check the actual value of the ripple
current the following equation is used:
CIN is chosen for an RMS current rating of at least 1.5A at
temperature. COUT is chosen with an ESR of 0.03Ω for low
outputripple. Theoutputrippleincontinuousmodewillbe
highest at the maximum input voltage. The output voltage
ripple due to ESR is approximately:
V
f L
( )( )
V
OUT
OUT
∆I =
1–
L
V
IN
VORIPPLE = RESR(∆IL) = 0.03Ω(1.3A) = 39mVP-P
The highest value of the ripple current occurs at the maxi-
mum input voltage:
PC Board Layout Checklist
1.6V
1.6V
22V
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1435A. These items are also illustrated graphically in
thelayoutdiagramofFigure10.Checkthefollowinginyour
layout:
∆IL =
1–
= 1.3A
250kHz 4.7µH
(
)
The lowest duty cycle also occurs at maximum input volt-
age. The on-time during this condition should be checked
to make sure it doesn’t violate the LTC1435A’s minimum
on-timeandcausecycleskippingtooccur.Therequiredon-
time at VIN(MAX) is:
1. Are the signal and power grounds segregated? The
LTC1435Asignalgroundpinmustreturntothe(–)plate
ofCOUT.Thepowergroundconnectstothesourceofthe
bottom N-channel MOSFET, anode of the Schottky di-
ode,and(–)plateofCIN,whichshouldhaveasshortlead
lengths as possible.
VOUT
1.6V
tON(MIN)
=
=
= 291ns
V
f
22V 250kHz
(
IN(MAX))( )
(
)(
)
The∆IL waspreviouslycalculatedtobe1.3A,whichis43%
of IMAX. From Figure 7, the LTC1435A minimum on-time
at 43% ripple is about 235ns. Therefore, the minimum on-
time is sufficient and no cycle skipping will occur.
2. Does the VOSENSE pin connect directly to the feedback
resistors? The resistive divider R1, R2 must be con-
nectedbetweenthe(+)plateofCOUT andsignalground.
The 100pF capacitor should be as close as possible to
the LTC1435A.
3. AretheSENSE– andSENSE+ leadsroutedtogetherwith
minimumPCtracespacing?Thefiltercapacitorbetween
SENSE+ and SENSE– should be as close as possible to
the LTC1435A.
ThepowerdissipationonthetopsideMOSFETcanbeeasily
estimated. Choosing a Siliconix Si4412DY results in:
RDS(ON) = 0.042Ω, CRSS = 100pF. At maximum input volt-
age with T(estimated) = 50°C:
15
LTC1435A
U
W U U
APPLICATIONS INFORMATION
4. Does the (+) plate of CIN connect to the drain of the
topsideMOSFET(s)ascloselyaspossible?Thiscapaci-
tor provides the AC current to the MOSFET(s).
6. KeeptheswitchingnodeSWawayfromsensitivesmall-
signal nodes. Ideally the switch node should be placed
at the furthest point from the LTC1435A.
5. Is the INTVCC decoupling capacitor connected closely
betweenINTVCC andthepowergroundpin?Thiscapaci-
tor carries the MOSFET driver peak currents.
7. SGND should be exclusively used for grounding exter-
nal components on COSC, ITH, VOSENSE and SFB pins.
8. If operating close to the minimum on-time limit, is the
sense resistor oriented on the radial axis of the induc-
tor? See Figure 8.
+
C
M1
OSC
1
2
16
15
C
IN
C
TG
OSC
C
SS
RUN/SS
BOOST
C
V
IN
C1
R
C
3
14
I
SW
TH
C
B
C
4
5
13
12
C2
D
D1
LTC1435A
SFB
V
0.1µF
B
IN
SGND
INTV
CC
–
100pF
+
M2
6
7
11
10
4.7µF
BG
V
OSENSE
–
SENSE
PGND
1000pF
8
9
+
SENSE
EXTV
CC
L1
–
R1
C
OUT
+
V
OUT
R
SENSE
R2
BOLD LINES INDICATE
HIGH CURRENT PATHS
+
1435A F10
Figure 10. LTC1435A Layout Diagram
Intel Mobile CPU VID Power Converter
U
TYPICAL APPLICATIONS
V
IN
4.5V TO 22V
4.7Ω
1
13
16
C
V
IN
OSC
C
IN
+
C
OSC
43pF
2
10µF
30V
× 2
M1
Si4410
RUN/SS
TG
C
R
C
F
0.1µF
SS
0.1µF
SENSE
0.015Ω
V
OUT
3
14
I
1.3V TO 2.0V
7A
SW
TH
L1
3.3µH
C
C
D
B
LTC1435A
INTV
1000pF
6
CMDSH-3
12
15
3
5
C
R
C2
220pF
C
10k
V
CC
SENSE
CC
0.22µF
LTC1706-19
FB
5
6
C
OUT
+
SGND
BOOST
820µF
4V
+
50pF
4.7µF
11
10
D1
M2
Si4410
× 2
V
BG
OSENSE
VID
0 1 2 3 GND
MBRS140T3
PGND
7 8 1 2
4
–
+
SENSE
SENSE
FROM µP
7
8
1000pF
1435A TA07
16
LTC1435A
U
TYPICAL APPLICATIONS
Dual Output 5V and Synchronous 12V Application
V
IN
5.4V TO 28V
0.01µF
C
C
OSC
IN
+
68pF
1
22µF
35V
× 2
IRLL014
16
15
14
13
12
11
10
9
M1
C
TG
BOOST
SW
OSC
Si4412DY
4.7k
C
SS
0.1µF
2
3
4
5
6
7
8
RUN/SS
R
C
C
C1
470pF
10k
I
TH
T1
C
C2
51pF
C
SEC
+
10µH
3.3µF
SFB
V
IN
1:1.42
35V
LTC1435A
R
SENSE
0.1µF
CMDSH-3
0.033Ω
V
OUT
5V/3.5A
SGND
INTV
CC
+
100pF
R1
35.7k
1%
4.7µF
V
BG
OSENSE
M2
Si4412DY
MBRS140T3
C
OUT
+
100µF
–
SENSE
SENSE
PGND
10V
× 2
R2
20k
1%
1000pF
+
EXTV
CC
100Ω
100Ω
SGND
V
OUT2
12V
1435A TA04
11.3k
1%
100k
1%
T1: DALE LPE6562-A236
120mA
3.3V/4.5A Converter with Foldback Current Limiting
V
IN
4.5V TO 28V
C
C
OSC
68pF
IN
+
22µF
35V
× 2
1
16
15
14
13
12
11
10
9
M1
C
TG
BOOST
SW
OSC
Si4410DY
C
SS
0.1µF
2
3
4
5
6
7
8
RUN/SS
R
C
C
C1
10k
330pF
I
TH
I
TH
PIN 3
C
C2
51pF
IN4148
SFB
V
INTV
IN
CC
L1
10µH
LTC1435A
R
0.1µF
CMDSH-3
SENSE
0.025Ω
V
OUT
3.3V/4.5A
SGND
INTV
CC
+
100pF
R1
4.7µF
35.7k
1%
V
BG
OSENSE
M2
Si4410DY
MBRS140T3
C
OUT
+
100µF
–
SENSE
SENSE
PGND
10V
× 2
100pF
R2
20k
1%
1000pF
OPTIONAL:
CONNECT TO 5V
+
EXTV
CC
SGND
(PIN 5)
1435A TA01
17
LTC1435A
TYPICAL APPLICATIONS
U
Constant-Current/Constant-Voltage High Efficiency Battery Charger
E1
V
IN
+
C1*
22µF
35V
+
C2*
22µF
35V
C4
R7
0.1µF
C11
C5
0.1µF
1.5M
E3
GND
E3
56pF
LTC1435A
1
2
3
4
5
6
7
8
16
Q1
SHDN
C
OSC
TG
C12
0.1µF
Si4412DY
C13
0.033µF
15
14
13
12
11
10
9
R5
1k
L1
27µH
R1
0.025Ω
RUN/SS BOOST
D1
E6
I
TH
SW
BATT
C6
0.33µF
C14
1000pF
+
C3
22µF
35V
SFB
V
D2
IN
E7
GND
SGND INTV
CC
C9
100pF
Q2
Si4412DY
V
BG
OSENSE
–
SENSE
PGND
C15
0.1µF
+
SENSE EXTV
CC
LT1620
C8
C10
100pF
100pF
1
8
7
6
5
C7
SENSE
AVG
4.7µF
+
2
3
4
16V
I
PROG
OUT
R2
1M
0.1%
GND
NIN
V
CC
PIN
R3
105k
0.1%
R4
76.8k
0.1%
C16
0.33µF
C18
0.1µF
JP1A
JP1B
R6
10k
1%
C17
0.01µF
1435A TA06
E5
GND
E4
PROG
*CONSULT CAPACITOR MANUFACTURER FOR RECOMMENDED
ESR RATING FOR CONTINUOUS 4A OPERATION
I
R
PROG
Current Programming Equation
(I
PROG
)(R6) – 0.04
10(R1)
I
=
BATT
Efficiency
100
95
V
= 24V
IN
V
= 16V
= 12V
BATT
V
BATT
90
V
= 6V
BATT
85
80
75
0
1
2
3
4
5
BATTERY CHARGE CURRENT (A)
1435A TA05
18
LTC1435A
U
TYPICAL APPLICATIONS
Dual Output 5V and 12V Application
V
IN
5.4V TO 28V
C
C
IN
OSC
68pF
+
22µF
35V
× 2
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
M1
C
TG
BOOST
SW
OSC
IRF7403
C
SS
0.1µF
RUN/SS
MBRS1100T3
R
C
C
C1
10k
510pF
I
TH
24V
C
T1
C2
C
SEC
+
51pF
10µH
3.3µF
SFB
V
IN
1:2.2
25V
LTC1435A
0.1µF
CMDSH-3
V
OUT
5V/3.5A
SGND
INTV
CC
R
SENSE
0.033Ω
+
100pF
R1
4.7µF
35.7k
1%
V
BG
OSENSE
M2
IRF7403
C
MBRS140T3
OUT
+
100µF
–
10V
× 2
SENSE
SENSE
PGND
R2
20k
1%
1000pF
+
EXTV
CC
100Ω
100Ω
SGND
10k
90.9k
V
OUT2
1435A TA02
12V
T1: DALE LPE6562-A092
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
G Package
16-Lead Plastic SSOP (0.209)
0.239 – 0.249*
(LTC DWG # 05-08-1640)
(6.07 – 6.33)
0.068 – 0.078
(1.73 – 1.99)
0.205 – 0.212**
(5.20 – 5.38)
16 15 14 13 12 11 10
9
0° – 8°
0.301 – 0.311
(7.65 – 7.90)
0.0256
(0.65)
BSC
0.005 – 0.009
(0.13 – 0.22)
0.022 – 0.037
(0.55 – 0.95)
0.002 – 0.008
(0.05 – 0.21)
0.010 – 0.015
(0.25 – 0.38)
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
G16 SSOP 1197
5
7
8
1
2
3
4
6
S Package
16-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.386 – 0.394*
(9.804 – 10.008)
0.010 – 0.020
(0.254 – 0.508)
16
15
14
13
12
11
10
9
× 45°
0.053 – 0.069
(1.346 – 1.752)
0.004 – 0.010
(0.101 – 0.254)
0.008 – 0.010
(0.203 – 0.254)
0° – 8° TYP
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
0.050
(1.270)
TYP
0.014 – 0.019
(0.355 – 0.483)
0.016 – 0.050
0.406 – 1.270
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
S16 0695
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
1
2
3
4
5
6
7
8
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
19
LTC1435A
TYPICAL APPLICATION
U
Low Dropout 2.9V/3A Converter
V
IN
3.5V TO 20V
C
OSC
68pF
C
IN
+
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
M1
22µF
35V
× 2
C
TG
BOOST
SW
OSC
1/2 Si9802DY
C
SS
0.1µF
RUN/SS
R
C
C
C1
10k
330pF
I
TH
C
C2
51pF
SFB
V
INTV
IN
CC
L1
10µH
LTC1435A
R
0.1µF
CMDSH-3
SENSE
0.033Ω
V
OUT
2.9V/3A
SGND
INTV
CC
+
100pF
R1
4.7µF
35.7k
1%
V
BG
OSENSE
M2
100pF
MBRS140T3
C
OUT
1/2 Si9802DY
+
100µF
–
SENSE
SENSE
PGND
10V
× 2
R2
24.9k
1%
1000pF
OPTIONAL:
CONNECT TO 5V
+
EXTV
CC
SGND
1435A TA03
L1: SUMIDA CDRH125-10
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1142HV/LTC1142
LTC1148HV/LTC1148
Dual High Efficiency Synchronous Step-Down Switching Regulators Dual Synchronous, V ≤ 20V
IN
High Efficiency Sychronous Step-Down Switching
Regulator Controllers
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IN
LTC1159
LT®1375/LT1376
High Efficiency Synchronous Step-Down Switching Regulator
1.5A, 500kHz Step-Down Switching Regulators
Synchronous, V ≤ 40V, For Logic Threshold FETs
IN
High Frequency, Small Inductor, High Efficiency
Switchers, 1.5A Switch
LTC1430
High Power Step-Down Switching Regulator Controller
High Efficiency 5V to 3.3V Conversion at Up to 15A
Full-Featured Single Controller
LTC1436A/LTC1436A-PLL/ High Efficiency Low Noise Synchronous Step-Down
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LT1510
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5V Standby in Shutdown
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Dual High Efficiency, Low Noise, Synchronous Step-Down
Switching Regulator
LTC1539
Dual High Efficiency, Low Noise, Synchronous Step-Down
Switching Regulator
5V Standby in Shutdown
LTC1706-19
VID Voltage Programmer
Creates a Programmable 1.3V to 2V Supply for Intel
Mobile Pentium® II Processor When Used with the
LTC1435A
Pentium is a registered trademark of Intel Corporation.
1435af LT/GP 0798 4K • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1998
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
20
●
●
(408)432-1900 FAX:(408)434-0507 www.linear-tech.com
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