LTC1474IS8 [Linear]
Low Quiescent Current High Efficiency Step-Down Converters; 低静态电流高效率降压型转换器
| 型号： | LTC1474IS8 |
| 厂家： | 
Linear |
| 描述： | Low Quiescent Current High Efficiency Step-Down Converters |
| 文件： | 总20页 (文件大小：353K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC1474/LTC1475
Low Quiescent Current
High Efficiency Step-Down
Converters
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FEATURES
DESCRIPTION
The LTC®1474/LTC1475 series are high efficiency step-
down converters with internal P-channel MOSFET power
switches that draw only 10µA typical DC supply current at
no load while maintaining output voltage. The LTC1474
uses logic-controlled shutdown while the LTC1475 fea-
tures pushbutton on/off.
■
High Efficiency: Over 92% Possible
Very Low Standby Current: 10µA Typ
■
■
■
■
■
■
■
Available in Space Saving 8-Lead MSOP Package
Internal 1.4Ω Power Switch (VIN = 10V)
Wide VIN Range: 3V to 18V Operation
Very Low Dropout Operation: 100% Duty Cycle
Low-Battery Detector Functional During Shutdown
Programmable Current Limit with Optional
Current Sense Resistor (10mA to 400mA Typ)
Short-Circuit Protection
The low supply current coupled with Burst ModeTM opera-
tion enables the LTC1474/LTC1475 to maintain high effi-
ciency over a wide range of loads. These features, along
with their capability of 100% duty cycle for low dropout
andwideinputsupplyrange, maketheLTC1474/LTC1475
idealformoderatecurrent(upto300mA)battery-powered
applications.
■
■
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■
Few External Components Required
Active Low Micropower Shutdown: IQ = 6µA Typ
Pushbutton On/Off (LTC1475 Only)
3.3V, 5V and Adjustable Output Versions
The peak switch current is user-programmable with an
optionalsenseresistor(defaultsto325mAminimumifnot
used) providing a simple means for optimizing the design
for lower current applications. The peak current control
also provides short-circuit protection and excellent start-
upbehavior.Alow-batterydetectorthatremainsfunctional
in shutdown is provided .
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APPLICATIONS
■
Cellular Telephones and Wireless Modems
4mA to 20mA Current Loop Step-Down Converter
Portable Instruments
Battery-Operated Digital Devices
Battery Chargers
Inverting Converters
Intrinsic Safety Applications
■
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■
The LTC1474/LTC1475 series availability in 8-lead MSOP
and SO packages and need for few additional components
provide for a minimum area solution.
■
■
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
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TYPICAL APPLICATION
LTC1474 Efficiency
100
V
IN
LOW BATTERY OUT
4V TO 18V
V
IN
= 5V
+
90
80
70
60
50
10µF
25V
V
IN
= 10V
0.1µF
7
V
IN
1
2
6
3
V
OUT
SENSE
V
V
IN
= 15V
FB
3.3V AT 250mA
LTC1474-3.3
+
L1
100µH
LBI
LBO
SW
LOW BATTERY IN
RUN SHDN
100µF
6.3V
100k
8
5
RUN
GND
4
D1
MBR0530
L = 100µH
V
R
= 3.3V
OUT
SENSE
1474/75 F01
= 0Ω
L1 = SUMIDA CDRH74-101
0.03
0.3
3
30
300
Figure 1. High Efficiency Step-Down Converter
LOAD CURRENT (mA)
1474/75 TA01
1
LTC1474/LTC1475
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ABSOLUTE MAXIMUM RATINGS
Input Supply Voltage (VIN).........................–0.3V to 20V
Switch Current (SW, SENSE).............................. 750mA
Switch Voltage (SW).............. (VIN –20V) to (VIN +0.3V)
VFB (Adjustable Versions) ..........................–0.3V to 12V
Operating Ambient Temperature Range
Commercial ............................................ 0°C to 70°C
Industrial ............................................ –40°C to 85°C
Junction Temperature (Note 1)............................ 125°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
V
OUT (Fixed Versions)................................ –0.3V to 20V
LBI, LBO ....................................................–0.3V to 20V
RUN, SENSE ..................................–0.3V to (VIN +0.3V)
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PACKAGE/ORDER INFORMATION
TOP VIEW
TOP VIEW
TOP VIEW
TOP VIEW
V
/V
RUN
1
2
3
4
8
7
6
5
V
/V
ON
1
2
3
4
8
7
6
5
OUT FB
OUT FB
V
/V
LBO
LBI
1
2
3
4
8 RUN
V
/V
1
2
3
4
8 ON
OUT FB
OUT FB
LBO
LBI
V
LBO
LBI/OFF
GND
V
IN
7 V
IN
LBO
LBI/OFF
GND
7 V
IN
IN
6 SENSE
5 SW
6 SENSE
5 SW
SENSE
SW
SENSE
SW
GND
GND
MS8 PACKAGE
8-LEAD PLASTIC MSOP
MS8 PACKAGE
8-LEAD PLASTIC MSOP
S8 PACKAGE
S8 PACKAGE
8-LEAD PLASTIC SO
8-LEAD PLASTIC SO
TJMAX = 125°C, θJA = 150°C/ W
TJMAX = 125°C, θJA = 150°C/ W
TJMAX = 125°C, θJA = 110°C/ W
TJMAX = 125°C, θJA = 110°C/ W
ORDER PART NUMBER
ORDER PART NUMBER
ORDER PART NUMBER
ORDER PART NUMBER
LTC1474CMS8
LTC1474CMS8-3.3
LTC1474CMS8-5
LTC1475CMS8
LTC1475CMS8-3.3
LTC1475CMS8-5
LTC1474CS8
LTC1475CS8
LTC1474IS8
LTC1475IS8
LTC1474CS8-3.3
LTC1474CS8-5
LTC1474IS8-3.3
LTC1474IS8-5
LTC1475CS8-3.3
LTC1475CS8-5
MS8 PART MARKING
MS8 PART MARKING
S8 PART MARKING
S8 PART MARKING
LTBW
LTCR
LTCS
LTBK
LTCP
LTCQ
1475
1474
1475I
14753
14755
1474I
14743
14745
14743I
14745I
Consult factory for Military grade parts.
2
LTC1474/LTC1475
TA = 25°C, VIN = 10V, VRUN = open, RSENSE = 0, unless otherwise noted.
ELECTRICAL CHARACTERISTICS
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX UNITS
V
Feedback Voltage
LTC1474/LTC1475
I
= 50mA
●
1.205
1.230 1.255
V
FB
LOAD
V
Regulated Output Voltage
LTC1474-3.3/LTC1475-3.3
LTC1474-5/LTC1475-5
I
= 50mA
OUT
LOAD
●
●
3.234
4.900
3.300 3.366
5.000 5.100
V
V
I
I
Feedback Current
●
0
30
nA
FB
LTC1474/LTC1475 Only
No Load Supply Current (Note 3)
Output Voltage Line Regulation
Output Voltage Load Regulation
Output Ripple
I
= 0 (Figure 1 Circuit)
10
5
µA
mV
mV
SUPPLY
LOAD
∆V
V
= 7V to 12V, I = 50mA
LOAD
20
15
OUT
IN
I
I
= 0mA to 50mA
= 10mA
2
LOAD
LOAD
50
mV
P-P
I
Input DC Supply Current (Note 2)
Active Mode (Switch On)
Sleep Mode (Note 3)
Shutdown
(Exclusive of Driver Gate Charge Current)
Q
V
V
V
= 3V to 18V
= 3V to 18V
= 3V to 18V, V
100
9
6
175
15
12
µA
µA
µA
IN
IN
IN
= 0V
RUN
R
ON
Switch Resistance
I
= 100mA
1.4
1.6
Ω
SW
I
Current Comp Max Current Trip Threshold
R
SENSE
R
SENSE
= 0Ω
= 1.1Ω
325
70
400
76
mA
mA
PEAK
85
V
V
Current Comp Sense Voltage Trip Threshold
Voltage Comparator Hysteresis
Switch Off-Time
●
●
90
100
5
110
mV
mV
SENSE
HYST
OFF
t
V
V
at Regulated Value
= 0V
3.5
4.75
65
6.0
µs
µs
OUT
OUT
V
V
V
Low Battery Comparator Threshold
Run/ON Pin Threshold
1.16
0.4
1.23
0.7
0.7
0.70
0.8
0.015
0
1.27
1.0
V
V
LBI, TRIP
RUN
OFF Pin Threshold (LTC1475 Only)
Sink Current into Pin 2
0.4
1.0
V
LBI, OFF
I
I
I
I
I
V
V
V
V
V
= 0V, V = 0.4V
LBO
0.45
0.4
mA
µA
µA
µA
µA
LBO, SINK
RUN, SOURCE
SW, LEAK
LBI, LEAK
LBO, LEAK
LBI
Source Current from Pin 8
Switch Leakage Current
= 0V
1.2
1
RUN
= 18V, V = 0V, V = 0V
RUN
IN
SW
Leakage Current into Pin 3
Leakage Current into Pin 2
= 18V, V = 18V
0.1
0.5
LBI
LBI
IN
= 2V, V
= 5V
0
LBO
The
temperature range.
Note 1: T is calculated from the ambient temperature T and power
●
denotes specifications which apply over the full operating
Note 2: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 3: No load supply current consists of sleep mode DC current (9µA
typical) plus a small switching component (about 1µA for Figure 1 circuit)
J
A
dissipation P according to the following formulas:
D
necessary to overcome Schottky diode and feedback resistor leakage.
LTC1474CS8/LTC1475CS8: T = T + (P • 110°C/W)
J
A
D
LTC1474CMS8/LTC1475CMS8: T = T + (P • 150°C/W)
J
A
D
3
LTC1474/LTC1475
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TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Input Voltage
Line Regulation
Load Regulation
100
95
90
85
80
75
70
40
30
40
30
FIGURE 1 CIRCUIT
FIGURE 1 CIRCUIT
FIGURE 1 CIRCUIT
LOAD
L: CDRH73-101
I
= 100mA
20
V
= 15V
= 10V
IN
R
= 0Ω
20
I
= 25mA
SENSE
LOAD
10
I
= 200mA
LOAD
10
V
IN
0
R
= 0.33Ω
SENSE
0
V
= 5V
IN
–10
–20
–30
I
= 1mA
LOAD
–10
–20
0
150
200
250
300
0
4
8
12
16
0
4
8
12
16
50
100
LOAD CURRENT (mA)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
1474/75 G01
1474/75 G02
1474/75 G03
Switch Resistance vs
Input Voltage
Current Trip Threshold vs
Temperature
Supply Current in Shutdown
500
400
300
200
100
0
10.0
7.5
5.0
2.5
0
5
V
= 10V
IN
R
= 0Ω
SENSE
4
3
2
1
0
T = 70°C
T = 25°C
R
= 1.1Ω
SENSE
40
TEMPERATURE (°C)
0
20
60
80
10
0
5
10
15
20
0
5
15
20
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
1474/75 G04
1474/75 G05
1474/75 G06
Switch Leakage Current vs
Temperature
Off-Time vs Output Voltage
VIN DC Supply Current
80
60
40
20
0
120
1.0
0.8
0.6
0.4
0.2
0
V
= 10V
IN
V
IN
= 18V
ACTIVE MODE
100
80
60
40
20
0
SLEEP MODE
40
0
20
60
80
100
0
20
40
60
80
100
0
4
8
12
16
20
% OF REGULATED OUTPUT VOLTAGE (%)
INPUT VOLTAGE (V)
TEMPERATURE (°C)
1474/75 G07
1474/75 G08
1474/75 G09
4
LTC1474/LTC1475
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PIN FUNCTIONS
VOUT/VFB (Pin 1): Feedback of Output Voltage. In the fixed
versions, an internal resistive divider divides the output
voltage down for comparison to the internal 1.23V refer-
ence. In the adjustable versions, this divider must be
implemented externally.
SW (Pin 5): Drain of Internal PMOS Power Switch. Cath-
ode of Schottky diode must be closely connected to this
pin.
SENSE(Pin6):CurrentSenseInputforMonitoringSwitch
Current and Source of Internal PMOS Power Switch.
Maximum switch current is programmed with a resistor
between SENSE and VIN pins.
LBO (Pin 2): Open Drain Output of the Low Battery
Comparator. This pin will sink current when Pin 3 is below
1.23V.
VIN (Pin 7): Main Supply Pin.
LBI/OFF (Pin 3): Input to Low Battery Comparator. This
input is compared to the internal 1.23V reference. For the
LTC1475, a momentary ground on this pin puts regulator
in shutdown mode.
RUN/ON (Pin 8): On LTC1474, voltage level on this pin
controls shutdown/run mode (ground = shutdown, open/
high = run). On LTC1475, a momentary ground on this pin
puts regulator in run mode. A 100k series resistor must be
used between Pin 8 and the switch or control voltage.
GND (Pin 4): Ground Pin.
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FUNCTIONAL DIAGRA
LBI/OFF
100mV
V
IN
1µA
V
–
+
7
IN
C
+
R
ON
ON
SENSE
(OPTIONAL)
×
V
CC
–
+
6
V
SENSE
5Ω
LTC1474: RUN
LTC1475: ON
8
2
4.75µs
20×
1×
1-SHOT
TRIGGER
STRETCH
OUT
WAKEUP
SW
5
LBO
V
OUT
+
+
LB
3M
(5V VERSION)
1.68M
1.23V
V /V
OUT FB
–
(3.3V VERSION)
READY
1
1.23V
REFERENCE
1M
LTC1474: LBI
LTC1475: LBI/OFF
GND
4
3
OUTPUT DIVIDER IS
IMPLEMENTED EXTERNALLY IN
ADJUSTABLE VERSIONS
CONNECTION NOT PRESENT IN LTC1474 SERIES
CONNECTION PRESENT IN LTC1474 SERIES ONLY
×
1474/75 FD
5
LTC1474/LTC1475
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(Refer to Functional Diagram)
OPERATIO
The LTC1474/LTC1475 are step-down converters with
internal power switches that use Burst Mode operation to
keep the output capacitor charged to the proper output
voltage while minimizing the quiescent current. Burst
Mode operation functions by using short “burst” cycles to
ramp the inductor current through the internal power
switch and external Schottky diode, followed by a sleep
cycle where the power switch is off and the load current is
supplied by the output capacitor. During sleep mode, the
LTC1474/LTC1475 draw only 9µA typical supply current.
At light loads, the burst cycles are a small percentage of
thetotalcycletime;thustheaveragesupplycurrentisvery
low, greatly enhancing efficiency.
Peak Inductor Current Programming
The current comparator provides a means for program-
mingthemaximuminductor/switchcurrentforeachswitch
cycle. The 1X sense MOSFET, a portion of the main power
MOSFET, is used to divert a sample (about 5%) of the
switch current through the internal 5Ω sense resistor. The
current comparator monitors the voltage drop across the
series combination of the internal and external sense
resistorsandtripswhenthevoltageexceeds100mV. Ifthe
external sense resistor is not used (Pins 6 and 7 shorted),
thecurrentthresholddefaultstothe400mAmaximumdue
to the internal sense resistor.
Off-Time
Burst Mode Operation
The off-time duration is 4.75µs when the feedback voltage
is close to the reference; however, as the feedback voltage
drops, the off-time lengthens and reaches a maximum
valueofabout65µswhenthisvoltageiszero.Thisensures
that the inductor current has enough time to decay when
the reverse voltage across the inductor is low such as
during short circuit.
At the beginning of the burst cycle, the switch is turned on
andtheinductorcurrentrampsup.Atthistime,theinternal
currentcomparatorisalsoturnedontomonitortheswitch
current by measuring the voltage across the internal and
optional external current sense resistors. When this volt-
age reaches 100mV, the current comparator trips and
pulses the 1-shot timer to start a 4.75µs off-time during
which the switch is turned off and the inductor current
ramps down. At the end of the off-time, if the output
voltage is less than the voltage comparator threshold, the
switchisturnedbackonandanothercyclecommences.To
minimize supply current, the current comparator is turned
on only during the switch-on period when it is needed to
monitorswitchcurrent.Likewise,the1-shottimerwillonly
be on during the 4.75µs off-time period.
Shutdown Mode
Both LTC1474 and LTC1475 provide a shutdown mode
that turns off the power switch and all circuitry except for
the low battery comparator and 1.23V reference, further
reducing DC supply current to 6µA typical. The LTC1474’s
run/shutdown mode is controlled by a voltage level at the
RUN pin (ground = shutdown, open/high = run). The
LTC1475’s run/shutdown mode, on the other hand, is
controlledbyaninternalS/Rflip-floptoprovidepushbutton
on/offcontrol. Theflip-flopisset(runmode)byamomen-
tary ground at the ON pin and reset (shutdown mode) by
a momentary ground at the LBI/OFF pin.
The average inductor current during a burst cycle will
normally be greater than the load current, and thus the
output voltage will slowly increase until the internal volt-
age comparator trips. At this time, the LTC1474/LTC1475
go into sleep mode, during which the power switch is off
and only the minimum required circuitry is left on: the
voltage comparator, reference and low battery compara-
tor. During sleep mode, with the output capacitor supply-
ing the load current, the VFB voltage will slowly decrease
until it reaches the lower threshold of the voltage com-
parator (about 5mV below the upper threshold). The
voltagecomparatorthentripsagain,signalingtheLTC1474/
LTC1475 to turn on the circuitry necessary to begin a new
burst cycle.
Low Battery Comparator
The low battery comparator compares the voltage on the
LBI pin to the internal reference and has an open drain
N-channel MOSFET at its output. If LBI is above the
reference, the output FET is off and the LBO output is high
impedance. If LBI is below the reference, the output FET is
on and sinks current. The comparator is still active in
shutdown.
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LTC1474/LTC1475
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APPLICATIONS INFORMATION
The basic LTC1474/LTC1475 application circuit is shown ments. Lower peak currents have the advantage of lower
inFigure1,ahighefficiencystep-downconverter.External output ripple (∆VOUT = IPEAK • ESR), lower noise, and less
componentselectionisdrivenbytheloadrequirementand stress on alkaline batteries and other circuit components.
begins with the selection of RSENSE. Once RSENSE is Also, lower peak currents allow the use of inductors with
known, L can be chosen. Finally D1, CIN and COUT are smaller physical size.
selected.
Peak currents as low as 10mA can be programmed with
the appropriate sense resistor. Increasing RSENSE above
RSENSE Selection
10Ω, however, gives no further reduction of IPEAK
.
The current sense resistor (RSENSE) allows the user to
program the maximum inductor/switch current to opti-
mizetheinductorsizeforthemaximumload.TheLTC1474/
LTC1475currentcomparatorhasamaximumthresholdof
100mV/(RSENSE + 0.25). The maximum average output
current IMAX is equal to this peak value less half the peak-
to-peak ripple current ∆IL.
For RSENSE values less than 1Ω, it is recommended that
the user parallel standard resistors (available in values ≥
1Ω) instead of using a special low valued shunt resistor.
Although a single resisor could be used with the desired
value, these low valued shunt resistor types are much
more expensive and are currently not available in case
sizes smaller than 1206. Three or four 0603 size standard
resistors require about the same area as one 1206 size
current shunt resistor at a fraction of the cost.
Allowing a margin for variations in the LTC1474/LTC1475
and external components, the required RSENSE can be
calculated from Figure 2 and the following equation:
At higher supply voltages and lower inductances, the peak
currents may be slightly higher due to current comparator
overshoot and can be predicted from the second term in
the following equation:
RSENSE = (0.067/IMAX) – 0.25
(1)
for 10mA < IMAX < 200mA.
5
−7
2.5 10
V − V
IN OUT
(
)
(
)
0.1
0.25 +R
4
(2)
I
=
+
PEAK
FOR LOWEST NOISE
L
SENSE
3
FOR BEST EFFICIENCY
Note that RSENSE only sets the maximum inductor current
peak. At lower dI/dt (lower input voltages and higher
inductances), theobservedpeakcurrentatloadslessthan
IMAX may be less than this calculated peak value due to the
voltage comparator tripping before the current ramps up
high enough to trip the current comparator. This effect
improves efficiency at lower loads by keeping the I2R
losses down (see Efficiency Considerations section).
2
1
0
0
100
150
200
250
300
50
MAXIMUM OUTPUT CURRENT (mA)
1474/75 F02
Figure 2. RSENSE Selection
Inductor Value Selection
For IMAX above 200mA, RSENSE is set to zero by shorting
Pins 6 and 7 to provide the maximum peak current of
400mA (limited by the fixed internal sense resistor). This
400mA default peak current can be used for lower IMAX if
desired to eliminate the need for the sense resistor and
associated decoupling capacitor. However, for optimal
performance,thepeakinductorcurrentshouldbesettono
morethanwhatisneededtomeettheloadcurrentrequire-
Once RSENSE and IPEAK are known, the inductor value can
be determined. The minimum inductance recommended
as a function of IPEAK and IMAX can be calculated from:
0.75 V
+ V t
(
)
OUT
D OFF
L
≥
MIN
(3)
I
−I
PEAK MAX
where tOFF = 4.75µs.
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LTC1474/LTC1475
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APPLICATIONS INFORMATION
section, increased inductance requires more turns of wire
and therefore copper losses will increase.
If the LMIN calculated is not practical, a larger IPEAK should
be used. Although the above equation provides the mini-
mum, betterperformance(efficiency, line/loadregulation,
noise) is usually gained with higher values. At higher
inductances, peak current and frequency decrease (im-
proving efficiency) and inductor ripple current decreases
(improving noise and line/load regulation). For a given
inductor type, however, as inductance is increased, DC
resistance (DCR) increases, increasing copper losses,
and current rating decreases, both effects placing an
upper limit on the inductance. The recommended range of
inductances for small surface mount inductors as a func-
tion of peak current is shown in Figure 3. The values in this
range are a good compromise between the trade-offs
discussedabove.Ifspaceisnotapremium,inductorswith
larger cores can be used, which extends the recom-
mended range of Figure 3 to larger values.
Ferrite and Kool Mµdesigns have very low core loss and
are preferred at high switching frequencies, so design
goals can concentrate on copper loss and preventing
saturation. Ferrite core material saturates “hard,” which
means that inductance collapses abruptly when the peak
design current is exceeded. This results in an abrupt
increase in inductor current above IPEAK and consequent
increase in voltage ripple. Do not allow the core to satu-
rate! Coiltronics, Coilcraft, Dale and Sumida make high
performance inductors in small surface mount packages
with low loss ferrite and Kool Mµ cores and work well in
LTC1474/LTC1475 regulators.
Catch Diode Selection
The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
diode conducts most of the time. As VIN approaches VOUT
the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short-circuited. Under this condition, the diode must
safely handle IPEAK at close to 100% duty cycle.
1000
500
To maximize both low and high current efficiency, a fast
switching diode with low forward drop and low reverse
leakage should be used. Low reverse leakage current is
critical to maximize low current efficiency since the leak-
agecanpotentiallyapproachthemagnitudeoftheLTC1474/
LTC1475 supply current. Low forward drop is critical for
high current efficiency since loss is proportional to for-
warddrop. Theseareconflictingparameters(seeTable1),
but a good compromise is the MBR0530 0.5A Schottky
diode specified in the application circuits.
100
50
10
100
1000
PEAK INDUCTOR CURRENT (mA)
1474/75 F03
Figure 3. Recommended Inductor Values
Inductor Core Selection
Once the value of L is known, the type of inductor must be
selected. High efficiency converters generally cannot
affordthecorelossfoundinlowcostpowderedironcores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of core
size for a fixed inductor value, but is very dependent on
inductanceselected. Asinductanceincreases, corelosses
go down. Unfortunately, as discussed in the previous
Table 1. Effect of Catch Diode on Performance
FORWARD
DROP
NO LOAD
SUPPLY CURRENT EFFICIENCY*
DIODE (D1) LEAKAGE
BAS85
200nA
1µA
0.6V
0.4V
0.3V
9.7µA
10µA
16µA
77.9%
83.3%
84.6%
MBR0530
MBRS130
20µA
*Figure 1 circuit with V = 15V, I
= 0.1A
IN
OUT
Kool Mµ is a registered trademark of Magnetics, Inc.
8
LTC1474/LTC1475
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APPLICATIONS INFORMATION
CIN and COUT Selection
negligible ESR. AVX and Marcon are good sources for
these capacitors.
At higher load currents, when the inductor current is
continuous, the source current of the P-channel MOSFET
is a square wave of duty cycle VOUT/VIN. To prevent large
voltage transients, a low ESR input capacitor sized for the
maximum RMS current must be used. The maximum
capacitor current is given by:
In surface mount applications multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
surfacemountconfigurations. Inthecaseoftantalum, itis
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS series of surface mount tantalums, available in case
heights ranging from 2mm to 4mm. Other capacitor types
include SANYO OS-CON, Nichicon PL series and Sprague
595D series. Consult the manufacturer for other specific
recommendations.
1/2
]
I
V
V − V
(
)
MAX OUT IN
OUT
[
C Required I
=
IN
RMS
V
IN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is com-
monlyusedfordesignbecauseevensignificantdeviations
donotoffermuchrelief.Notethatcapacitormanufacturer’s
ripplecurrentratingsareoftenbasedon2000hoursoflife.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Do not underspecify this component. An addi-
tional 0.1µF ceramic capacitor is also required on VIN for
high frequency decoupling.
To avoid overheating, the output capacitor must be sized
to handle the ripple current generated by the inductor. The
worst-case ripple current in the output capacitor is given
by:
IRMS = IPEAK/2
Once the ESR requirement for COUT has been met, the
RMS current rating generally far exceeds the IRIPPLE(P-P)
requirement.
The selection of COUT is driven by the required effective
series resistance (ESR) to meet the output voltage ripple
andlineregulationrequirements.Theoutputvoltageripple
during a burst cycle is dominated by the output capacitor
ESR and can be estimated from the following relation:
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
oftenusefultoanalyzeindividuallossestodeterminewhat
is limiting efficiency and which change would produce the
most improvement. Efficiency can be expressed as:
25mV < ∆VOUT, RIPPLE = ∆IL • ESR
where ∆IL ≤ IPEAK and the lower limit of 25mV is due to the
voltage comparator hysteresis. Line regulation can also
vary with COUT ESR in applications with a large input
voltage range and high peak currents.
Efficiency = 100% – (L1 + L2 + L3 + ...)
whereL1, L2, etc. aretheindividuallossesasapercentage
of input power.
ESR is a direct function of the volume of the capacitor.
ManufacturerssuchasNichicon,AVXandSpragueshould
be considered for high performance capacitors. The
OS-CONsemiconductordielectriccapacitoravailablefrom
SANYO has the lowest ESR for its size at a somewhat
higher price. Typically, once the ESR requirement is satis-
fied, the capacitance is adequate for filtering. For lower
current applications with peak currents less than 50mA,
10µF ceramic capacitors provide adequate filtering and
are a good choice due to their small size and almost
Although all dissipative elements in the circuit produce
losses, threemainsourcesusuallyaccountformostofthe
losses in LTC1474/LTC1475 circuits: VIN current, I2R
losses and catch diode losses.
1. The VIN current is due to two components: the DC bias
current and the internal P-channel switch gate charge
current. The DC bias current is 9µA at no load and
increases proportionally with load up to a constant
100µA during continuous mode. This bias current is so
9
LTC1474/LTC1475
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APPLICATIONS INFORMATION
small that this loss is negligible at loads above a
milliamp but at no load accounts for nearly all of the
loss. The second component, the gate charge current,
results from switching the gate capacitance of the
internalP-channelswitch.Eachtimethegateisswitched
from high to low to high again, a packet of charge dQ
moves from VIN to ground. The resulting dQ/dt is the
currentoutofVIN whichistypicallymuchlargerthanthe
DC bias current. In continuous mode, IGATECHG = fQP
where QP is the gate charge of the internal switch. Both
the DC bias and gate charge losses are proportional to
VIN and thus their effects will be more pronounced at
higher supply voltages.
To minimize no-load supply current, resistor values in the
megohm range should be used. The increase in supply
current due to the feedback resistors can be calculated
from:
V
V
OUT
V
IN
OUT
∆I
=
VIN
R1+R2
A 10pF feedforward capacitor across R2 is necessary due
to the high impedances to prevent stray pickup and
improve stability.
V
OUT
2. I2R losses are predicted from the internal switch,
inductor and current sense resistor. At low supply
voltages where the switch on-resistance is higher and
the switch is on for longer periods due to higher duty
cycle, theswitchlosseswilldominate. Keepingthepeak
currents low with the appropriate RSENSE and with
larger inductance helps minimize these switch losses.
Athighersupplyvoltages, theselossesareproportional
to load and result in the flat efficiency curves seen in
Figure 1.
R2
10pF
1
LTC1474
LTC1475
V
FB
R1
GND
4
1474/75 F04
Figure 4. LTC1474/LTC1475 Adjustable Configuration
Low Battery Comparator
The LTC1474/LTC1475 have an on-chip low battery com-
parator that can be used to sense a low battery condition
when implemented as shown in Figure 5. The resistive
divider R3/R4 sets the comparator trip point as follows:
3. The catch diode loss is due to the VDID loss as the diode
conducts current during the off-time and is more pro-
nounced at high supply voltage where the on-time is
short. This loss is proportional to the forward drop.
However, as discussed in the Catch Diode section,
diodes with lower forward drops often have higher
leakage current, so although efficiency is improved, the
no load supply current will increase.
R4
R3
V
= 1.23 1+
TRIP
The divided down voltage at the LBI pin is compared to the
internal 1.23V reference. When VLBI < 1.23V, the LBO
output sinks current. The low battery comparator is active
all the time, even during shutdown mode.
Adjustable Applications
Foradjustableversions,theoutputvoltageisprogrammed
with an external divider from VOUT to VFB (Pin 1) as shown
in Figure 4. The regulated voltage is determined by:
V
IN
LTC1474/LTC1475
R4
LBI
LBO
R2
R1
–
V
=1.23 1+
OUT
(4)
R3
+
1.23V
REFERENCE
1474/75 F05
Figure 5. Low Battery Comparator
10
LTC1474/LTC1475
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APPLICATIONS INFORMATION
LTC1475 Pushbutton On/Off and
the depressed switch state is detected by the microcon-
trollerthroughitsinput. Themicrocontrollerthenpullsthe
LBI/OFF pin low with the connection to one of its ouputs.
With the LBI/OFF pin low, the LTC1475 powers down
turningthemicrocontrolleroff. NotethatsincetheI/Opins
of most microcontrollers have a reversed bias diode
between input and supply, a blocking diode with less than
1µA leakage is necessary to prevent the powered down
microcontroller from pulling down on the ON pin.
Microprocessor Interface
TheLTC1475providespushbuttoncontrolofpoweron/off
for use with handheld products. A momentary ground on
the ON pin sets an internal S/R latch to run mode while a
momentary ground on the LBI/OFF pin resets the latch to
shutdown mode. See Figure 6 for a comparsion of on/off
operation of the LTC1474 and LTC1475 and Figure 7 for a
comparison of the circuit implementations.
Figure19intheTypicalApplicationssectionshowshowto
use the low battery comparator to provide a low battery
lockout on the “ON” switch. The LBO output disconnects
the pushbutton from the ON pin when the comparator has
tripped, preventing the LTC1475 from attempting to start
up again until VIN is increased.
In the LTC1475, the LBI/OFF pin has a dual function as
both the shutdown control pin and the low battery com-
parator input. Since the “OFF” pushbutton is normally
open, it does not affect the normal operation of the low
battery comparator. In the unpressed state, the LBI/OFF
input is the divided down input voltage from the resistive
divider to the internal low battery comparator and will
normally be above or just below the trip threshold of
1.23V. When shutdown mode is desired, the LBI/OFF pin
is pulled below the 0.7V threshold to invoke shutdown.
100k
100k
ON
LTC1475
LBI/OFF
RUN
LTC1474
V
IN
ON
RUN
RUN
OFF
LTC1474
MODE
RUN
SHUTDOWN
RUN
1474/75 F07
ON OVERRIDES LBI/OFF
WHILE ON IS LOW
Figure 7. Simplified Implementation of
LTC1474 and LTC1475 On/Off
ON
Absolute Maximum Ratings and Latchup Prevention
LBI/OFF
LTC1475
TheabsolutemaximumratingsspecifythatSW(Pin5)can
never exceed VIN (Pin 7) by more than 0.3V. Normally this
situation should never occur. It could, however, if the
output is held up while the supply is pulled down. A
condition where this could potentially occur is when a
battery is supplying power to an LTC1474 or LTC1475
regulator and also to one or more loads in parallel with the
the regulator’s VIN. If the battery is disconnected while the
LTC1474 or LTC1475 regulator is supplying a light load
and one of the parallel circuits is a heavy load, the input
capacitor of the LTC1474 or LTC1475 regulator could be
pulled down faster than the output capacitor, causing the
absolute maximum ratings to be exceeded. The result is
often a latchup which can be destructive if VIN is reapplied.
Battery disconnect is possible as a result of mechanical
stress, bad battery contacts or use of a lithium-ion battery
RUN
RUN
MODE
SHUTDOWN
1474/75 F06
Figure 6. Comparison of LTC1474 and LTC1475
Run/Shutdown Operation
The ON pin has precedence over the LBI/OFF pin. As seen
in Figure 6, if both pins are grounded simultaneously, run
mode wins.
Figure 18 in the Typical Applications section shows an
example for the use of the LTC1475 to control on/off of a
microcontroller with a single pushbutton. With both the
microcontroller and LTC1475 off, depressing the
pushbuttongroundstheLTC1475ONpinandstartsupthe
LTC1475 regulator which then powers up the microcon-
troller. When the pushbutton is depressed a second time,
11
LTC1474/LTC1475
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APPLICATIONS INFORMATION
with a built-in internal disconnect. The user needs to
assess his/her application to determine whether this situ-
ationcouldoccur.Ifso,additionalprotectionisnecessary.
where P is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to the
ambient temperature.
Prevention against latchup can be accomplished by sim-
ply connecting a Schottky diode across the SW and VIN
pins as shown in Figure 8. The diode will normally be
reverse biased unless VIN is pulled below VOUT at which
time the diode will clamp the (VOUT – VIN) potential to less
than the 0.6V required for latchup. Note that a low leakage
Schottky should be used to minimize the effect on no-load
supplycurrent.SchottkydiodessuchasMBR0530,BAS85
and BAT84 work well. Another more serious effect of the
protection diode leakage is that at no load with nothing to
provide a sink for this leakage current, the output voltage
can potentially float above the maximum allowable toler-
ance. To prevent this from occuring, a resistor must be
connected between VOUT and ground with a value low
enough to sink the maximum possible leakage current.
The junction temperature is given by:
TJ = TA + TR
As an example consider the LTC1474/LTC1475 in dropout
at an input voltage of 3.5V, a load current of 300mA, and
an ambient temperature of 70°C. From the typical perfor-
mancegraphofswitchresistance,theon-resistanceofthe
P-channel switch at 70°C is 3.5Ω. Therefore, power dissi-
pated by the part is:
P = I2 • RDS(ON) = 0.315W
For the MSOP package, the θJA is 150°C/W. Thus the
junction temperature of the regulator is:
TJ = 70°C + (0.315)(150) = 117°C
whichisnearthemaximumjunctiontemperatureof125oC.
Note that at higher supply voltages, the junction tempera-
ture is lower due to reduced switch resistance.
LATCHUP
PROTECTION
SCHOTTKY
PC Board Layout Checklist
V
V
SW
LTC1474
OUT
IN
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC1474/LTC1475. These items are also illustrated
graphically in the layout diagram of Figure 9. Check the
following in your layout:
+
LTC1475
1474/75 F08
Figure 8. Preventing Absolute Maximum
Ratings from Being Exceeded
1. Is the Schottky diode cathode closely connected to SW
(Pin 5)?
Thermal Considerations
In the majority of the applications, the LTC1474/LTC1475
do not dissipate much heat due to their high efficiency.
However, in applications where the switching regulator is
running at high ambient temperature with low supply
voltage and high duty cycles, such as dropout with the
switch on continuously, the user will need to do some
thermal analysis. The goal of the thermal analysis is to
determine whether the power dissipated by the regulator
exceeds the maximum junction temperature of the part.
The temperature rise is given by:
2. Is the 0.1µF input decoupling capacitor closely con-
nected between VIN (Pin 7) and ground (Pin 4)? This
capacitor carries the high frequency peak currents.
3. When using adjustable version, is the resistive divider
closely connected to the (+) and (–) plates of COUT with
a 10pF capacitor connected across R2?
4. Is the 1000pF decoupling capacitor for the current
sense resistor connected as close as possible to Pins 6
and 7? If no current sense resistor is used, Pins 6 and
7 should be shorted.
TR = P • θJA
12
LTC1474/LTC1475
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APPLICATIONS INFORMATION
OUTPUT DIVIDER REQUIRED WITH
ADJUSTABLE VERSION ONLY
10pF
LTC1474
100k
8
7
1
2
3
4
V
OUT
V
FB
RUN
R2
R1
L
V
IN
LBO
LBI
1000pF
R
SENSE
+
6
5
SENSE
SW
C
OUT
GND
1474/75 F09
D1
0.1µF
C
IN
+
V
IN
BOLD LINES INDICATE HIGH PATH CURRENTS
Figure 9. LTC1474/LTC1475 Layout Diagram (See Board Layout Checklist)
150mA. The minimum inductance is, therefore, from the
5. Are the signal and power grounds segregated? The
equation (3) and assuming VD = 0.4V,
signal ground consists of the (–) plate of COUT, Pin 4 of
the LTC1474/LTC1475 and the resistive divider. The
power ground consists of the Schottky diode anode,
the (–) plate of CIN and the 0.1µF decoupling capacitor.
0.75 3.3 + 0.4 4.75µs
(
)(
)
L
=
= 264µH
MIN
0.15 − 0.1
6. Is a 100k resistor connected in series between RUN
(Pin 8) and the RUN control voltage? The resistor
should be as close as possible to Pin 8.
From Figure 3, an inductance of 270µH is chosen from the
recommended region. The CDRH73-271 or CD54-271 is a
good choice for space limited applications.
Design Example (Refer to RSENSE and Inductor
Selection)
For the feedback resistors, choose R1 = 1M to minimize
supply current. R2 can then be calculated from the equa-
tion (4) to be:
As a design example, assume VIN = 10V, VOUT = 3V, and
a maximum average output current IMAX = 100mA. With
this information, we can easily calculate all the important
components:
V
OUT
R2 =
−1 • R1= 1.43M
1.23
From the equation (1),
For the catch diode, the MBR0530 will work well in this
application.
RSENSE = (0.067/0.1) – 0.25 = 0.42Ω
Using the standard resistors (1Ω, 1Ω and 2Ω) in parallel
provides 0.4Ω without having to use a more expensive
low value current shunt type resistor (see RSENSE Selec-
tion section).
Fortheinputandoutputcapacitors, AVX4.7µFand100µF,
respectively, low ESR TPS series work well and meet the
RMS current requirement of 100mA/2 = 50mA. They are
available in small “C” case sizes with 0.15Ω ESR. The
0.15Ω output capacitor ESR will result in 25mV of output
voltage ripple.
With RSENSE = 0.4Ω, the peak inductor current IPEAK is
calculated from (2), neglecting the second term, to be
Figure 10 shows the complete circuit for this example.
13
LTC1474/LTC1475
TYPICAL APPLICATIONS
U
10pF
V
IN
3.5V TO 18V
+
†
4.7µF
35V
0.1µF
1000pF
7
1Ω**
1Ω**
2Ω**
1.43M
1%
V
V
IN
OUT
1
2
6
3
SENSE
LBI
V
3V
FB
100mA
1M
1%
L*
270µH
+
LTC1474
LBO
SW
††
100µF
6.3V
*
SUMIDA CDRH73-271
100k
8
5
** 3 PARALLEL STANDARD RESISTORS
PROVIDE LEAST EXPENSIVE SOLUTION
RUN
RUN
GND
4
D1
MBR0530
(SEE R SELECTION SECTION)
SENSE
AVX TPSC475M035
AVX TPSC107M006
†
††
1474/75 F10
Figure 10. High Efficiency 3V/100mA Regulator (Design Example)
+
IN
4mA TO 20mA
††
1000pF
6
D2
1µF
× 3
2Ω
7
12V
7.5M
1M
V
IN
V
OUT
1
2
SENSE
V
OUT
3.3V
10mA
LTC1474-3.3
3
†
L*
330µH
LBI
LBO
10µF**
100k
TO A/D
MBR0530
8
5
RUN
RUN
SW
GND
4
D1
MBR0530
–
IN
4mA TO 20mA
*
COILCRAFT DO1608-334
** MARCON THCS50E1E106Z,
1474/75 F11
AVX 1206ZG106Z
†
† †
OPTIONAL RESISTOR FOR SENSING LOOP CURRENT BY A/D CONVERTER
MOTOROLA MMBZ5242BL
Figure 11. High Efficiency 3.3V/10mA Output from 4mA to 20mA Loop
14
LTC1474/LTC1475
U
TYPICAL APPLICATIONS
MBR0530
V
OUT
–12V
70mA
††
22µF
10pF
+
+
V
25V
IN
3.5V TO 6V
+
0.1µF
22µF**
4.7M
1%
7
16V
V
OUT
1
6
3
V
IN
12V
SENSE
LBI
V
FB
70mA
536k
1%
††
2
5
22µF
25V
L*
200µH
LTC1474
LBO
100k
+
8
RUN
RUN
SW
V
IN
(V)
3.5
4
I
LOAD(MAX)
GND
4
†
30mA
50mA
70mA
90mA
10µF
25V
L*
200µH
D1
MBR0530
*
COILTRONICS CTX200-4
** AVX TPSC226M016
†
5
AVX TPSC106M025
AVX TPSD226M025
††
6
1474/75 F12
Figure 12. 5V to ±12V Regulator
V
IN
3.5V TO 12V
+
0.1µF
10µF**
25V
7
V
OUT
1
6
V
IN
5V
200mA AT V = 10V
SENSE
V
OUT
IN
+
LTC1474-5
†
2
5
33µF
10V
3
8
L*
LBI
LBO
SW
10µF**
100µH
25V
+
100k
RUN
RUN
GND
4
V
IN
(V)
I
LOAD(MAX)
3.5
4
70mA
95mA
L*
100µH
D1
MBR0530
*
COILTRONICS CTX100-4
** AVX TPSC106MO25
5
125mA
180mA
200mA
225mA
†
AVX TPSC336M010
8
10
12
1474/75 F13
Figure 13. 5V Buck-Boost Converter
15
LTC1474/LTC1475
TYPICAL APPLICATIONS
U
V
IN
3.5V TO 12V
+
0.1µF
10µF**
25V
7
††
TP0610
10M
ON/OFF
1
2
6
3
V
IN
SENSE
V
OUT
LTC1474-5
V
IN
(V)
3.5
5
I
L*
100µH
LOAD(MAX)
LBI
LBO
SW
+
†
100mA
140mA
190mA
240mA
1474/75 F14
33µF
10V
8
5
RUN
GND
4
8
D1
12
MBR0530
V
OUT
–5V
140mA AT V = 5V
*
SUMIDA CDRH74-101
IN
** AVX TPSC106M025
†
AVX TPSC336M010
††
RUN: ON/OFF = 0, SHUTDOWN: 0N/OFF = V
IN
Figure 14. Positive-to-Negative (–5V) Converter
V
IN
10pF
8V TO 18V
+
0.1µF
4.7µF**
7
MBR0530
35V
V
4.69M
OUT
1
2
6
3
V
IN
4-NiCd
SENSE
LBI
V
FB
200mA
+
†
47µF
1M
L*
100µH
LTC1474
LBO
SW
16V
100k
5
8
CHARGER
ON/OFF
RUN
GND
4
D1
MBR0530
*
SUMIDA CDRH73-101
** AVX TPSC475M035
1474/75 F15
†
AVX TPSD476M016
Figure 15. 4-NiCd Battery Charger
16
LTC1474/LTC1475
U
TYPICAL APPLICATIONS
V
IN
4V TO 18V
+
0.1µF
†
4.7µF
35V
2.2M
7
V
OUT
1
2
6
3
V
IN
3.3V
SENSE
V
OUT
250mA
+
††
LTC1474-3.3
100µF
6.3V
L*
100µH
LBI
LBO
100k
1M
5
8
RUN
RUN
SW
GND
4
D1
MBR0530
*
SUMIDA CDRH73-101
AVX TPSC475M035
†
1474/75 F16
††
AVX TPSC107M006
Figure 16. High Efficiency 3.3V Regulator with Low Battery Lockout
V
IN
5.7V TO 18V
+
0.1µF
4.7µF**
7
35V
V
3.65M
OUT
1
2
6
3
V
IN
5V
SENSE
V
OUT
250mA
+
LTC1475-5
†
33µF
10V
L*
100µH
LBO
SW
LBI/OFF
ON
100k
5
8
OFF
GND
4
1M
D1
ON
MBR0530
*
SUMIDA CDRH73-101
** AVX TPSC475M035
†
AVX TPSC336M010
1474/75 F17
Figure 17. Pushbutton On/Off 5V/250mA Regulator
V
IN
4V TO 18V
V
CC
+
0.1µF
4.7µF**
35V
7
6
MMBD914LT1
V
100k
OUT
SENSE
V
1
8
2
IN
3.3V
V
OUT
ON
250mA
+
†
LTC1475-3.3
100µF
6.3V
ON/OFF
0.1µF
L*
100µH
LBO
2.2M
5
3
SW
LBI/OFF
GND
4
D1
1M
MBR0530
µC
*
SUMIDA CDRH73-101
1474/75 F18
** AVX TPSC475M035
†
AVX TPSC107M006
Figure 18. LTC1475 Regulator with 1-Button Toggle On/Off
17
LTC1474/LTC1475
U
Dimensions in inches (millimeters) unless otherwise noted.
PACKAGE DESCRIPTION
MS8 Package
8-Lead Plastic MSOP
(LTC DWG # 05-08-1660)
0.118 ± 0.004*
(3.00 ± 0.102)
8
7
6
5
0.118 ± 0.004**
(3.00 ± 0.102)
0.192 ± 0.004
(4.88 ± 0.10)
1
2
3
4
0.040 ± 0.006
(1.02 ± 0.15)
0.034 ± 0.004
(0.86 ± 0.102)
0.007
(0.18)
0° – 6° TYP
SEATING
PLANE
0.012
(0.30)
REF
0.021 ± 0.006
(0.53 ± 0.015)
0.006 ± 0.004
(0.15 ± 0.102)
MSOP (MS8) 1197
0.0256
(0.65)
TYP
*
DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH,
PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
18
LTC1474/LTC1475
U
Dimensions in inches (millimeters) unless otherwise noted.
PACKAGE DESCRIPTION
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
7
5
8
6
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
0.053 – 0.069
3
4
2
0.010 – 0.020
(0.254 – 0.508)
× 45°
(1.346 – 1.752)
0.004 – 0.010
(0.101 – 0.254)
0.008 – 0.010
(0.203 – 0.254)
0°– 8° TYP
0.016 – 0.050
0.406 – 1.270
0.050
(1.270)
TYP
0.014 – 0.019
(0.355 – 0.483)
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
SO8 0996
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
19
LTC1474/LTC1475
TYPICAL APPLICATION
U
10pF
V
IN
3.5V to 18V
+
0.1µF
4.7µF**
1.02M
1%
7
35V
V
1.8M
OUT
1
2
6
8
V
IN
2.5V
SENSE
ON
V
FB
1M
250mA
+
†
1M
1%
100k
100µF
6.3V
MMBT2N2222LT1
L*
100µH
LBO
SW
LTC1475
5
3
LBI/OFF
GND
4
1M
D1
MBR0530
ON
OFF
*
SUMIDA CDRH73-101
** AVX TPSC475M035
†
AVX TPSC107M006
1474/75 F19
Figure 19. Pushbutton On/Off with Low Battery Lockout
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
I = 80µA Max
LTC1096/LTC1098
Micropower Sampling 8-Bit Serial I/O A/D Converter
150mA Low Dropout Regulator
Q
LT1121/LT1121-3.3/LT1121-5
Linear Regulator, I = 30µA
Q
LTC1174/LTC1174-3.3/LTC1174-5 High Efficiency Step-Down and Inverting DC/DC Converters
Selectable I
= 300mA or 600mA
PEAK
LTC1265
1.2A High Efficiency Step-Down DC/DC Converter
1.5A 500kHz Step-Down Switching Regulators
Burst Mode Operation, Internal MOSFET
LT1375/LT1376
500kHz, Small Inductor, High
Efficiency Switchers, 1.5A Switch
LTC1440/LTC1441/LTC1442
LT1495/LT1496
Ultralow Power Comparator with Reference
1.5µA Precision Rail-to-Rail Op Amps
300mA Low Dropout Regulator
I = 2.8µA Max
Q
I = 1.5µA Max
Q
LT1521/LT1521-3/LT1521-3.3/
LT1521-5
Linear Regulator, I = 12µA
Q
LTC1574/LTC1574-3.3/LTC1574-5 High Efficiency Step-Down DC/DC Converters with Internal Schottky Diode LTC1174 with Internal Schottky Diode
LT1634-1.25
Micropower Precision Shunt Reference
I
= 10µA
Q(MIN)
14745fa LT/TP 0398 4K REV A • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1997
Linear Technology Corporation
●
1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900
20
●
●
FAX: (408) 434-0507 TELEX: 499-3977 www.linear-tech.com
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