LTC1624CS8_技术文档

LTC1624

High Efficiency SO-8

N-Channel Switching

Regulator Controller

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FEATURES

DESCRIPTION

The LTC^®1624 is a current mode switching regulator

controllerthat drivesanexternalN-channelpowerMOSFET

using a fixed frequency architecture. It can be operated in

all standard switching configurations including boost,

step-down, inverting and SEPIC. Burst Mode^TMoperation

provides high efficiency at low load currents. A maximum

highdutycyclelimitof95%provideslowdropoutoperation

whichextendsoperatingtimeinbattery-operatedsystems.

■

N-Channel MOSFET Drive

■

Implements Boost, Step-Down, SEPIC

and Inverting Regulators

■

Wide V_INRange: 3.5V to 36V Operation

■

Wide V_OUTRange: 1.19V to 30V in Step-Down

Configuration

■

±

1% 1.19V Reference

■

Low Dropout Operation: 95% Duty Cycle

200kHz Fixed Frequency

Theoperatingfrequencyisinternallysetto200kHz,allowing

smallinductorvaluesandminimizingPCboardspace.The

operatingcurrentlevelisuser-programmableviaanexternal

current sense resistor. Wide input supply range allows

operation from 3.5V to 36V (absolute maximum).

Low Standby Current

Very High Efficiency

Remote Output Voltage Sense

Logic-Controlled Micropower Shutdown

Internal Diode for Bootstrapped Gate Drive

Current Mode Operation for Excellent Line and

Load Transient Response

A multifunction pin (I_TH/RUN) allows external

compensation for optimum load step response plus

shutdown. Soft start can also be implemented with the

I_TH/RUN pin to properly sequence supplies.

■

Available in an 8-Lead SO Package

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APPLICATIONS

■

Notebook and Palmtop Computers, PDAs

Cellular Telephones and Wireless Modems

Battery-Operated Digital Devices

DC Power Distribution Systems

Battery Chargers

■

, LTC and LT are registered trademarks of Linear Technology Corporation.

Burst Mode is a trademark of Linear Technology Corporation.

■

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TYPICAL APPLICATION

V

IN

4.8V TO 28V

1000pF

–

V

SENSE

IN

C

IN

+

R

22µF

35V

× 2

SENSE

0.05Ω

BOOST

I

/RUN

TH

C

M1

Si4412DY

LTC1624

470pF

TG

V

FB

R

C

L1

10µH

C

B

100pF

6.8k

V

0.1µF

OUT

3.3V

2A

SW

GND

D1

R2

35.7k

MBRS340T3

C

OUT

100µF

10V

R1

20k

× 2

1624 F01

Figure 1. High Efficiency Step-Down Converter

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LTC1624

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ABSOLUTE MAXIMUM RATINGS

PACKAGE/ORDER INFORMATION

Input Supply Voltage (V_IN).........................36V to –0.3V

Topside Driver Supply Voltage (BOOST)....42V to –0.3V

Switch Voltage (SW)..................................36V to –0.6V

Differential Boost Voltage

ORDER PART

TOP VIEW

NUMBER

–

SENSE

1

2

3

4

8

7

6

5

V

IN

I

TH

/RUN

BOOST

TG

LTC1624CS8

LTC1624IS8

(BOOST to SW) ....................................7.8V to –0.3V

V

FB

SENSE^–Voltage

GND

SW

V_IN< 15V.................................. (V_IN+ 0.3V) to –0.3V

V_IN≥ 15V .......................... (V_IN+0.3V) to (V_IN–15V)

I_TH/RUN, V_FBVoltages ............................ 2.7V to – 0.3V

Peak Driver Output Current < 10µs (TG) .................... 2A

Operating Temperature Range

S8 PACKAGE

8-LEAD PLASTIC SO

S8 PART MARKING

1624

1624I

T_JMAX= 125°C, θ_JA= 110°C/ W

LTC1624CS ............................................ 0°C to 70°C

LTC1624IS......................................... –40°C to 85°C

Junction Temperature (Note 1)............................. 125°C

Storage Temperature Range ................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec).................. 300°C

Consult factory for Military grade parts.

T_A= 25°C, V_IN= 15V, unless otherwise noted.

ELECTRICAL CHARACTERISTICS

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

Main Control Loop

I

V

Feedback Current

(Note 2)

10

50

nA

V

IN FB

V

Feedback Voltage

●

1.1781

1.19

0.002

1.2019

0.01

FB

∆V

Reference Voltage Line Regulation

Output Voltage Load Regulation

V

IN

= 3.6V to 20V (Note 2)

%/V

LINE REG

(Note 2)

LOAD REG

I

Sinking 5µA

Sourcing 5µA

●

0.5

–0.5

0.8

–0.8

%

TH

V

Output Overvoltage Lockout

1.24

0.6

1.28

1.32

V

OVL

I

Input DC Supply Current

Normal Mode

(Note 3)

Q

550

16

900

30

µA

Shutdown

V

= 0V

ITH/RUN

V

Run Threshold

0.8

V

ITH/RUN

I

Run Current Source

Run Pullup Current

V

= 0.3V

= 1V

–0.8

–50

–2.5

–160

–5.0

–350

µA

ITH/RUN

∆V

Maximum Current Sense Threshold

V

FB

= 1.0V

145

160

185

mV

SENSE(MAX)

TG Transition Time

Rise Time

TG t

C

LOAD

C

LOAD

= 3000pF

50

150

ns

r

f

Fall Time

f

Oscillator Frequency

Boost Voltage

●

175

4.8

200

5.15

3

225

5.5

5

kHz

V

OSC

V

SW = 0V, I

= 5mA, V = 8V

BOOST

IN

∆V

Boost Load Regulation

= 2mA to 20mA

%

BOOST

The

●

denotes specifications which apply over the full operating

T = T + (P • 110°C/W)

J A D

temperature range.

Note 2: The LTC1624 is tested in a feedback loop which servos V to

FB

LTC1624CS: 0°C ≤ T ≤ 70°C

the midpoint for the error amplifier (V = 1.8V).

A

ITH

LTC1624IS: –40°C ≤ T ≤ 85°C

A

Note 3: Dynamic supply current is higher due to the gate charge being

Note 1: T is calculated from the ambient temperature T and power

delivered at the switching frequency. See Applications Information.

J

A

dissipation P according to the following formula:

D

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LTC1624

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TYPICAL PERFORMANCE CHARACTERISTICS

Efficiency vs Input Voltage

_OUT= 3.3V

Efficiency vs Load Current

V_OUT= 5V

Efficiency vs Load Current

_OUT= 3.3V

V

100

95

90

85

80

75

70

100

95

90

85

80

75

70

100

95

90

85

80

75

70

V

= 3.3V

SENSE

OUT

V

R

= 3.3V

SENSE

OUT

V

= 5V

OUT

= 10V

R

= 0.033Ω

V

IN

V

= 5V

IN

R

= 0.033Ω

SENSE

V

= 10V

IN

I

= 1A

LOAD

I

= 0.1A

LOAD

20

30

0

5

10

15

25

2.5

25

0.001

0.01

0.1

1

10

0.001

0.01

0.1

1

10

LOAD CURRENT (A)

INPUT VOLTAGE (V)

LOAD CURRENT (A)

1624 G07

1624 G08

1624 G09

Efficiency vs Input Voltage

V_OUT= 5V

V_IN– V_OUTDropout Voltage

vs Load Current

Input Supply Current vs

Input Voltage

100

95

90

85

80

75

70

0.7

0.6

0.5

0.4

0.3

0.2

0.1

0

700

600

500

400

300

200

100

0

V

= 5V

OUT

SENSE

R

V

= 0.033Ω

DROP OF 5%

V

= 1.21V

SENSE

OUT

FB

R

= 0.033Ω

SLEEP MODE

I

= 1A

LOAD

I

= 0.1A

LOAD

SHUTDOWN

20

INPUT VOLTAGE (V)

30

15

20

INPUT VOLTAGE (V)

25

30

35

0

5

10

15

25

0

5

10

0

1.0

1.5

2.0

3.0

0.5

LOAD CURRENT (A)

1624 G10

1624 G11

1624 G05

Boost Load Regulation

Boost Voltage vs Temperature

Boost Line Regulation

6.0

5.5

5.0

4.5

4.0

6

5

4

3

2

1

0

6

5

4

3

2

1

0

I

= 1mA

LOAD

V

= 15V

= 5V

IN

V

IN

I

V

= 1mA

BOOST

V

= 0V

SW

60

TEMPERATURE (°C)

110 135

–40 –15

10

35

85

20

INPUT VOLTAGE (V)

30

35

0

5

10

15

25

20

BOOST LOAD CURRENT (mA)

30

0

5

10

15

1624 G15

1624 G04

1624 G06

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LTC1624

TYPICAL PERFORMANCE CHARACTERISTICS

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I_TH/RUN Pin Source Current vs

Temperature

V_ITHvs Output Current

I_ITHvs V_ITH

300

250

200

150

100

50

5

4

3

2

1

0

200

2.4

I

/RUN = 1V

TH

150

50

I

TH

/RUN = 0V

ACTIVE

MODE

1.2

0.8

ACTIVE

MODE

SHUTDOWN

3

0

I

OUT(MAX)

60

TEMPERATURE (°C)

110 135

–40 –15

10

35

85

0

0.8

1.2

(V)

2.4

I

V

OUT

ITH

1624 G14

(a)

(b)

1624 G01

1624 G02

Operating Frequency vs

Temperature

Maximum Current Sense

Threshold vs Temperature

Frequency vs Feedback Voltage

250

200

150

100

50

250

200

150

100

50

170

168

166

164

162

160

158

156

154

152

150

V

OUT

IN REGULATION

V

= 0V

FB

0

0.25

0.50

0.75

1.00

1.25

–40

10

35

60

85 110 135

–40

10

35

60

85 110 135

–15

TEMPERATURE (°C)

FEEDBACK VOLTAGE

1624 G03

1448 G12

1448 G13

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PIN FUNCTIONS

SENSE^–(Pin 1): Connects to the (–) input for the current

comparator. Built-in offsets between the SENSE^–and V_IN

pinsinconjunctionwithR_SENSEsetthecurrenttripthresh-

olds. Do not pull this pin more than 15V below V_INor more

than 0.3V below ground.

shutdownallfunctionsaredisabledandTGpinisheldlow.

V_FB(Pin 3): Receives the feedback voltage from an exter-

nal resistive divider across the output.

GND (Pin 4): Ground. Connect to the (–) terminal of C_OUT

,

the Schottky diode and the (–) terminal of C_IN.

I_TH/RUN (Pin 2): Combination of Error Amplifier Compen-

sation Point and Run Control Inputs. The current com-

parator threshold increases with this control voltage.

Nominalvoltagerangeforthispinis1.19Vto2.4V.Forcing

this pin below 0.8V causes the device to be shut down. In

SW (Pin 5): Switch Node Connection to Inductor. In step-

down applications the voltage swing at this pin is from a

Schottky diode (external) voltage drop below ground to

V_IN.

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LTC1624

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PIN FUNCTIONS

swing at this pin is from INTV_CCto V_IN+ INTV_CCin step-

down applications. In non step-down topologies the volt-

age at this pin is constant and equal to INTV_CCif SW = 0V.

TG (Pin 6): High Current Gate Drive for Top N-Channel

MOSFET. This is the output of a floating driver with a

voltage swing equal to INTV_CCsuperimposed on the

switch node voltage SW.

V_IN(Pin 8): Main Supply Pin and the (+) Input to the

CurrentComparator.Mustbecloselydecoupledtoground.

BOOST (Pin 7): Supply to Topside Floating Driver. The

bootstrap capacitor C_Bis returned to this pin. Voltage

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(Refer to Functional Diagram)

OPERATIO

Main Control Loop

implemented by ramping the voltage on the I_TH/RUN pin

from 1.19V to its 2.4V maximum (see Applications Infor-

mation section).

The LTC1624 uses a constant frequency, current mode

architecture. During normal operation, the top MOSFET is

turned on each cycle when the oscillator sets the RS latch

andturnedoffwhenthemaincurrentcomparatorI₁resets

the RS latch. The peak inductor current at which I₁resets

the RS latch is controlled by the voltage on the I_TH/RUN

pin, which is the output of error amplifier EA. The V_FBpin,

described in the pin functions, allows EA to receive an

output feedback voltage from an external resistive divider.

When the load current increases, it causes a slight

decrease in V_FBrelative to the 1.19V reference, which in

turn causes the I_TH/RUN voltage to increase until the

average inductor current matches the new load current.

While the top MOSFET is off, the internal bottom MOSFET

isturnedonforapproximately300nsto400nstorecharge

the bootstrap capacitor C_B.

Comparator OV guards against transient output over-

shoots >7.5% by turning off the top MOSFET and keeping

it off until the fault is removed.

Low Current Operation

The LTC1624 is capable of Burst Mode operation in which

the external MOSFET operates intermittently based on

load demand. The transition to low current operation

begins when comparator B detects when the I_TH/RUN

voltage is below 1.5V. If the voltage across R_SENSEdoes

not exceed the offset of I₂(approximately 20mV) for one

full cycle, then on following cycles the top and internal

bottom drives are disabled. This continues until the I_TH

voltageexceeds1.5V,whichcausesdrivetobereturnedto

the TG pin on the next cycle.

The top MOSFET driver is biased from the floating boot-

strap capacitor C_Bthat is recharged during each off cycle.

The dropout detector counts the number of oscillator

cycles that the top MOSFET remains on and periodically

forces a brief off period to allow C_Bto recharge.

INTV_CCPower/Boost Supply

Power for the top and internal bottom MOSFET drivers is

derived from V_IN. An internal regulator supplies INTV_CC

power. To power the top driver in step-down applications

an internal high voltage diode recharges the bootstrap

capacitorC_BduringeachoffcyclefromtheINTV_CCsupply.

A small internal N-channel MOSFET pulls the switch node

(SW) to ground each cycle after the top MOSFET has

turned off ensuring the bootstrap capacitor is kept fully

charged.

ThemaincontrolloopisshutdownbypullingtheI_TH/RUN

pin below its 1.19V clamp voltage. Releasing I_TH/RUN

allows an internal 2.5µA current source to charge com-

pensation capacitor C_C. When the I_TH/RUN pin voltage

reaches0.8VthemaincontrolloopisenabledwiththeI_TH/

RUN voltage pulled up by the error amp. Soft start can be

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LTC1624

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FUNCTIONAL DIAGRA

(Shown in a step-down application)

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LTC1624

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APPLICATIONS INFORMATION

The LTC1624 can be used in a wide variety of switching

regulator applications, the most common being the step-

down converter. Other switching regulator architectures

includestep-up,SEPICandpositive-to-negativeconverters.

Accepting larger values of ∆I_Lallows the use of low

inductances, but results in higher output voltage ripple

and greater core losses. A reasonable starting point for

setting ripple current is ∆I_L= 0.4(I_MAX). Remember, the

maximum ∆I_Loccurs at the maximum input voltage.

ThebasicLTC1624step-downapplicationcircuitisshown

in Figure 1 on the first page. External component selection

is driven by the load requirement and begins with the

selection of R_SENSE. Once R_SENSEis known, the inductor

can be chosen. Next, the power MOSFET and D1 are

selected. Finally, C_INand C_OUTare selected. The circuit

shown in Figure 1 can be configured for operation up to an

input voltage of 28V (limited by the external MOSFETs).

The inductor value also has an effect on low current

operation. Lower inductor values (higher ∆I_L) will cause

Burst Mode operation to begin at higher load currents,

which can cause a dip in efficiency in the upper range of

low current operation. In Burst Mode operation lower

inductance values will cause the burst frequency to

decrease. In general, inductor values from 5µH to 68µH

are typical depending on the maximum input voltage and

output current. See also Modifying Burst Mode Operation

section.

Step-Down Converter: R_SENSESelection for

Output Current

R_SENSEis chosen based on the required output current.

The LTC1624 current comparator has a maximum thresh-

old of 160mV/R_SENSE. The current comparator threshold

sets the peak of the inductor current, yielding a maximum

average output current I_MAXequal to the peak value less

half the peak-to-peak ripple current, ∆I_L.

Step-Down Converter: Inductor Core Selection

Once the value for L is known, the type of inductor must be

selected. High efficiency converters generally cannot

affordthecorelossfoundinlowcostpowderedironcores,

forcing the use of more expensive ferrite, molypermalloy

orKoolMµ^®cores. Actualcorelossisindependentofcore

size for a fixed inductor value, but it is very dependent on

inductanceselected. Asinductanceincreases,corelosses

go down. Unfortunately, increased inductance requires

more turns of wire and, therefore, copper losses will

increase.

Allowing a margin for variations in the LTC1624 and

external component values yields:

100mV

R

=

SENSE

I

MAX

The LTC1624 works well with values of R_SENSEfrom

0.005Ω to 0.5Ω.

Ferrite designs have very low core loss and are preferred

at high switching frequencies, so design goals can con-

centrate on copper loss and preventing saturation. Ferrite

core material saturates “hard,” which means that induc-

tance collapses abruptly when the peak design current is

exceeded. This results in an abrupt increase in inductor

ripple current and consequent output voltage ripple. Do

not allow the core to saturate!

Step-Down Converter: Inductor Value Calculation

With the operating frequency fixed at 200kHz smaller

inductor values are favored. Operating at higher frequen-

cies generally results in lower efficiency because of

MOSFET gate charge losses. In addition to this basic

trade-off, the effect of inductor value on ripple current and

low current operation must also be considered.

Molypermalloy (from Magnetics, Inc.) is a very good, low

losscorematerialfortoroids,butitismoreexpensivethan

ferrite. A reasonable compromise from the same manu-

facturer is Kool Mµ. Toroids are very space efficient,

especially when you can use several layers of wire.

Because they generally lack a bobbin, mounting is more

difficult. However, designs for surface mount that do not

increase the height significantly are available.

Theinductorvaluehasadirecteffectonripplecurrent.The

inductor ripple current ∆I_Ldecreases with higher induc-

tance and increases with higher V_INor V_OUT

:

V − V

V

+ V

IN

OUT OUT D

∆I =

L

V + V

f L

( )( )

IN

D

where V_Dis the output Schottky diode forward drop.

Kool Mu is a registered trademark of Magnetics, Inc.

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LTC1624

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APPLICATIONS INFORMATION

Step-Down Converter: Power MOSFET Selection

characteristics. The constant k = 2.5 can be used to

estimate the contributions of the two terms in the P_MAIN

dissipation equation.

One external N-channel power MOSFET must be selected

for use with the LTC1624 for the top (main) switch.

Step-Down Converter: Output Diode Selection (D1)

The peak-to-peak gate drive levels are set by the INTV_CC

voltage. This voltage is typically 5V. Consequently, logic

level threshold MOSFETs must be used in most LTC1624

applications. If low input voltage operation is expected

(V_IN< 5V) sublogic level threshold MOSFETs should be

used. PaycloseattentiontotheBV_DSSspecificationforthe

MOSFETs as well; many of the logic level MOSFETs are

limited to 30V or less.

The Schottky diode D1 shown in Figure 1 conducts during

theoff-time. Itisimportanttoadequatelyspecifythediode

peak current and average power dissipation so as not to

exceed the diode ratings.

The most stressful condition for the output diode is under

short circuit (V_OUT= 0V). Under this condition, the diode

must safely handle I_SC(PK)at close to 100% duty cycle.

Under normal load conditions, the average current con-

ducted by the diode is simply:

Selection criteria for the power MOSFET include the “ON”

resistance R_DS(ON), reverse transfer capacitance C_RSS

,

input voltage and maximum output current. When the

LTC1624 is operating in continuous mode the duty cycle

for the top MOSFET is given by:

V

OUT

−

IN

I

=I

DIODE AVG

LOAD AVG

(

)

(

)

V + V

IN

D

V

+ V

D

OUT

Remember to keep lead lengths short and observe proper

grounding (see Board Layout Checklist) to avoid ringing

and increased dissipation.

Main Switch Duty Cycle =

V + V

IN

D

The MOSFET power dissipation at maximum output

current is given by:

The forward voltage drop allowable in the diode is calcu-

lated from the maximum short-circuit current as:

2

) (

V

+ V

D

OUT

P

V + V

IN D

P

=

I

1+ δ R

+

D

(

)

MAIN

MAX

DS ON

(

)

V ≈

V + V

D

IN

I

V

IN

SC AVG

(

)

1.85

)

k V

(

I

C

f

(

)(

)( )

IN

MAX RSS

where P_Dis the allowable diode power dissipation and will

be determined by efficiency and/or thermal requirements

(see Efficiency Considerations).

where δ is the temperature dependency of R_DS(ON)and k

is a constant inversely related to the gate drive current.

MOSFETs have I²R losses, plus the P_MAINequation

includes an additional term for transition losses that are

highest at high output voltages. For V_IN< 20V the high

currentefficiencygenerallyimproveswithlargerMOSFETs,

whileforV_IN>20Vthetransitionlossesrapidlyincreaseto

the point that the use of a higher R_DS(ON)device with lower

C_RSSactual provides higher efficiency. The diode losses

are greatest at high input voltage or during a short circuit

when the diode duty cycle is nearly 100%.

Step-Down Converter: C_INand C_OUTSelection

In continuous mode the source current of the top

N-channel MOSFET is a square wave of approximate duty

cycle V_OUT/V_IN. To prevent large voltage transients, a low

ESR input capacitor sized for the maximum RMS current

must be used. The maximum RMS capacitor current is

given by:

1/2

]

V

V − V

OUT

(

)

OUT IN

[

C Required I

≈I

Theterm(1+δ)isgenerallygivenforaMOSFETintheform

of a normalized R_DS(ON)vs Temperature curve, but

δ = 0.005/°C can be used as an approximation for low

voltageMOSFETs.C_RSSisusuallyspecifiedintheMOSFET

IN

RMS MAX

V

IN

This formula has a maximum at V_IN= 2V_OUT, where

I_RMS= I_OUT/2. This simple worst-case condition is com-

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LTC1624

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APPLICATIONS INFORMATION

monlyusedfordesignbecauseevensignificantdeviations

donotoffermuchrelief.Notethatcapacitormanufacturer’s

ripple current ratings are often based on only 2000 hours

of life. This makes it advisable to further derate the

capacitor, or to choose a capacitor rated at a higher

temperaturethanrequired.Severalcapacitorsmayalsobe

paralleled to meet size or height requirements in the

design. Always consult the manufacturer if there is any

question.

ratings that are ideal for input capacitor applications.

Consult the manufacturer for other specific recommend-

ations.

INTV_CCRegulator

An internal regulator produces the 5V supply that powers

the drivers and internal circuitry within the LTC1624.

Good V_INbypassing is necessary to supply the high

transient currents required by the MOSFET gate drivers.

The selection of C_OUTis driven by the required effective

series resistance (ESR). Typically, once the ESR require-

ment is satisfied the capacitance is adequate for filtering.

The output ripple (∆V_OUT) is determined by:

High input voltage applications in which large MOSFETs

are being driven at high frequencies may cause the maxi-

mum junction temperature rating for the LTC1624 to be

exceeded. The supply current is dominated by the gate

charge supply current as discussed in the Efficiency

Considerations section. The junction temperature can be

estimated by using the equations given in Note 1 of the

Electrical Characteristics table. For example, the LTC1624

is limited to less than 17mA from a 30V supply:

1

∆V

≈ ∆I ESR +

L

OUT

4fC

OUT

where f = operating frequency, C_OUT= output capacitance

and ∆I_L= ripple current in the inductor. The output ripple

is highest at maximum input voltage since ∆I_Lincreases

with input voltage. With ∆I_L= 0.4I_OUT(MAX)the output

ripplewillbelessthan100mVatmaximumV_IN,assuming:

T_J= 70°C + (17mA)(30V)(110°C/W) = 126°C

To prevent maximum junction temperature from being

exceeded, the input supply current must be checked

operating in continuous mode at maximum V_IN.

C_OUTRequired ESR < 2R_SENSE

Manufacturers such as Nichicon, United Chemicon and

SANYO should be considered for high performance

through-hole capacitors. The OS-CON semiconductor

dielectric capacitor available from SANYO has the lowest

ESR(size)productofanyaluminumelectrolyticatasome-

what higher price. Once the ESR requirement for C_OUThas

been met, the RMS current rating generally far exceeds

the I_RIPPLE(P-P)requirement.

Step-Down Converter: Topside MOSFET Driver

Supply (C_B, D_B)

AnexternalbootstrapcapacitorC_BconnectedtotheBOOST

pinsuppliesthegatedrivevoltageforthetopsideMOSFET.

Capacitor C_Bin the functional diagram is charged through

internal diode D_Bfrom INTV_CCwhen the SW pin is low.

When the topside MOSFET is to be turned on, the driver

places the C_Bvoltage across the gate to source of the

MOSFET. This enhances the MOSFET and turns on the

topside switch. The switch node voltage SW rises to V_IN

and the BOOST pin rises to V_IN+ INTV_CC. The value of the

boost capacitor C_Bneeds to be 50 times greater than the

total input capacitance of the topside MOSFET. In most

applications 0.1µF is adequate.

In surface mount applications multiple capacitors may

have to be paralleled to meet the ESR or RMS current

handling requirements of the application. Aluminum elec-

trolytic and dry tantalum capacitors are both available in

surface mount configurations. In the case of tantalum it is

critical that the capacitors are surge tested for use in

switching power supplies. An excellent choice is the AVX

TPS series of surface mount tantalums, available in case

heightsrangingfrom2mmto4mm. Othercapacitortypes

include SANYO OS-CON, Nichicon WF series and Sprague

595Dseriesandthenewceramics.Ceramiccapacitorsare

nowavailableinextremelylowESRandhighripplecurrent

Significant efficiency gains can be realized by supplying

topsidedriveroperatingvoltage fromtheoutput,sincethe

V_INcurrent resulting from the driver and control currents

will be scaled by a factor of (Duty Cycle)/(Efficiency). For

5V regulators this simply means connecting the BOOST

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3.3V

OR 5V

pin through a small Schottky diode (like a Central

CMDSH-3) to V_OUTas shown in Figure 10. However, for

3.3V and other lower voltage regulators, additional cir-

cuitry is required to derive boost supply power from the

output.

I

/RUN

I

/RUN

TH

D1

C

R

C

R

C

For low input voltage operation (V_IN< 7V), a Schottky

diode can be connected from V_INto BOOST to increase the

external MOSFET gate drive voltage. Be careful not to

exceed the maximum voltage on BOOST to SW pins

of 7.8V.

(a)

(b)

I

/RUN

TH

R1

D1

C

Output Voltage Programming

C1

R

C

The output voltage is set by a resistive divider according

to the following formula:

(c)

Figure 3. I_TH/RUN Pin Interfacing

1624 F03

R2

R1

V

= 1.19V 1+

OUT

Soft start can be implemented by ramping the voltage on

I_TH/RUN during start-up as shown in Figure 3(c). As the

voltage on I_TH/RUNramps from 1.19V to 2.4V the internal

peak current limit is also ramped at a proportional linear

rate. The peak current limit begins at approximately

10mV/R_SENSE(at V_ITH/RUN= 1.4V) and ends at:

The external resistive divider is connected to the output as

showninFigure2, allowingremotevoltagesensing. When

using remote sensing, a local 100Ω resistor should be

connected from L1 to R2 to prevent V_OUTfrom running

away if the sense lead is disconnected.

V

OUT

L1

160mV/R_SENSE(V_ITH/RUN= 2.4V)

R2

R1

V

FB

The output current thus ramps up slowly, charging the

outputcapacitor.Thepeakinductorcurrentandmaximum

output current are as follows:

LTC1624

GND

100pF

1624 F02

I_L(PEAK)= (V_ITH/RUN– 1.3V)/(6.8R_SENSE

)

Figure 2. Setting the LTC1624 Output Voltage

I_OUT(MAX)= I_LPEAK– ∆I_L/2

I_TH/RUN Function

with ∆I_L= ripple current in the inductor.

The I_TH/RUN pin is a dual purpose pin that provides the

loopcompensationandameanstoshutdowntheLTC1624.

Soft start can also be implemented with this pin. Soft start

reduces surge currents from V_INby gradually increasing

the internal current limit. Power supply sequencing can

also be accomplished using this pin.

During normal operation the voltage on the I_TH/RUN pin

willvaryfrom1.19Vto2.4Vdependingontheloadcurrent.

Pulling the I_TH/RUN pin below 0.8V puts the LTC1624 into

alowquiescentcurrentshutdown(I_Q<30µA).Thispincan

be driven directly from logic as shown in Figures 3(a)

and 3(b).

An internal 2.5µA current source charges up the external

capacitor C_C.When the voltage on I_TH/RUN reaches 0.8V

the LTC1624 begins operating. At this point the error

amplifier pulls up the I_TH/RUN pin to its maximum of 2.4V

(assuming V_OUTis starting low).

Efficiency Considerations

The percent efficiency of a switching regulator is equal to

the output power divided by the input power times 100%.

It is often useful to analyze individual losses to determine

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what is limiting the efficiency and which change would

producethemostimprovement. Percentefficiencycanbe

expressed as:

loss is thus reduced by the duty cycle.) For example, at

50% DC, if R_DS(ON)= 0.05Ω, R_L= 0.15Ω and R_SENSE

=

0.05Ω, then the effective total resistance is 0.2Ω. This

results in losses ranging from 2% to 8% for V_OUT= 5V

as the output current increases from 0.5A to 2A. I²R

losses cause the efficiency to drop at high output

currents.

%Efficiency = 100% – (L1 + L2 + L3 + ...)

whereL1, L2, etc. aretheindividuallossesasapercentage

of input power.

Although all dissipative elements in the circuit produce

losses, four main sources usually account for most of the

losses in LTC1624 circuits:

3. Transition losses apply only to the topside MOSFET(s),

andonlywhenoperatingathighinputvoltages(typically

20V or greater). Transition losses can be estimated

from:

1. LTC1624 V_INcurrent

2. I²R losses

Transition Loss = 2.5(V_IN)^1.85(I_MAX)(C_RSS)(f)

3. Topside MOSFET transition losses

4. Voltage drop of the Schottky diode

4. The Schottky diode is a major source of power loss at

high currents and gets worse at high input voltages.

The diode loss is calculated by multiplying the forward

voltage drop times the diode duty cycle multiplied by

the load current. For example, assuming a duty cycle of

50% with a Schottky diode forward voltage drop of

0.5V, the loss is a relatively constant 5%.

1. The V_INcurrent is the sum of the DC supply current I_Q,

given in the Electrical Characteristics table, and the

MOSFET driver and control currents. The MOSFET

driver current results from switching the gate

capacitanceofthepowerMOSFET. EachtimeaMOSFET

gate is switched from low to high to low again, a packet

of charge dQ moves from INTV_CCto ground. The

resulting dQ/dt is a current out of V_INwhich is typically

much larger than the control circuit current. In

continuous mode, I_GATECHG= f (Q_T+ Q_B), where Q_Tand

Q_Bare the gate charges of the topside and internal

bottom side MOSFETs.

As expected, the I²R losses and Schottky diode loss

dominate at high load currents. Other losses including

C_INand C_OUTESR dissipative losses and inductor core

lossesgenerallyaccountforlessthan2%totaladditional

loss.

Checking Transient Response

By powering BOOST from an output-derived source

(Figure 10 application), the additional V_INcurrent

resulting from the topside driver will be scaled by a

factor of (Duty Cycle)/(Efficiency). For example, in a

20V to 5V application, 5mA of INTV_CCcurrent results in

approximately 1.5mA of V_INcurrent. This reduces the

midcurrent loss from 5% or more (if the driver was

powered directly from V_IN) to only a few percent.

The regulator loop response can be checked by looking at

the load transient response. Switching regulators take

several cycles to respond to a step in DC (resistive) load

current. Whenaloadstepoccurs, V_OUTimmediatelyshifts

by an amount equal to (∆I_LOAD• ESR), where ESR is the

effective series resistance of C_OUT. ∆I_LOADalso begins to

charge or discharge C_OUTwhich generates a feedback

error signal. The regulator loop then acts to return V_OUTto

its steady-state value. During this recovery time V_OUTcan

be monitored for overshoot or ringing that would indicate

astabilityproblem. TheI_THexternalcomponentsshownin

theFigure1circuitwillprovideadequatecompensationfor

most applications.

2. I²R losses are predicted from the DC resistances of the

MOSFET, inductor and current shunt. In continuous

mode the average output current flows through L but is

“chopped” between the topside main MOSFET/current

shunt and the Schottky diode. The resistances of the

topside MOSFET and R_SENSEmultiplied by the duty

cycle can simply be summed with the resistance of L to

obtain I²R losses. (Power is dissipated in the sense

resistor only when the topside MOSFET is on. The I²R

Asecond, moreseveretransient, iscausedbyswitchingin

loads with large (>1µF) supply bypass capacitors. The

dischargedbypasscapacitorsareeffectivelyputinparallel

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with C_OUT, causing a rapid drop in V_OUT. No regulator can

deliver enough current to prevent this problem if the load

switch resistance is low and it is driven quickly. The only

solution is to limit the rise time of the switch drive so that

the load rise time is limited to approximately (25 • C_LOAD).

Thus a 10µF capacitor would require a 250µs rise time,

limiting the charging current to about 200mA.

36V, most applications will be limited to 30V by the

MOSFET BV_DSS

.

Modifying Burst Mode Operation

The LTC1624 automatically enters Burst Mode operation

at low output currents to boost efficiency. The point when

continuous mode operation changes to Burst Mode op-

eration scales as a function of maximum output current.

The output current when Burst Mode operation com-

mences is approximately 8mV/R_SENSE(8% of maximum

output current).

Automotive Considerations: Plugging into the

Cigarette Lighter

As battery-powered devices go mobile there is a natural

interest in plugging into the cigarette lighter in order to

conserveorevenrechargebatterypacksduringoperation.

But before you connect, be advised: you are plugging into

the supply from hell. The main battery line in an automo-

bileisthesourceofanumberofnastypotentialtransients,

including load dump, reverse battery and double battery.

WiththeadditionalcircuitryshowninFigure5theLTC1624

can be forced to stay in continuous mode longer at low

output currents. Since the LTC1624 is not a fully synchro-

nous architecture, it will eventually start to skip cycles as

the load current drops low enough. The point when the

minimum on-time (450ns) is reached determines the load

current when cycle skipping begins at approximately 1%

of maximum output current. Using the circuit in Figure 5

the LTC1624 will begin to skip cycles but stays in regula-

Load dump is the result of a loose battery cable. When the

cablebreaksconnection,thefieldcollapseinthealternator

can cause a positive spike as high as 60V which takes

several hundred milliseconds to decay. Reverse battery is

just what it says, while double battery is a consequence of

tow-truck operators finding that a 24V jump start cranks

cold engines faster than 12V.

tion when I_OUTis less than I_OUT(MIN)

:

2

)

t

f

ON MIN

(

V + V

IN

D

I

=

V − V

IN OUT

(

)

OUT MIN

(

)

2L

V

+ V

OUT D

ThenetworkshowninFigure4isthemoststraightforward

approach to protect a DC/DC converter from the ravages

of an automotive battery line. The series diode prevents

current from flowing during reverse battery, while the

transient suppressor clamps the input voltage during load

dump. Note that the transient suppressor should not

conduct during double battery operation, but must still

clamptheinputvoltagebelowbreakdownoftheconverter.

Although the LTC1624 has a maximum input voltage of

where t_ON(MIN)= 450ns, f = 200kHz.

The transistor Q1 in the circuit of Figure 5 operates as a

current source developing an 18mV offset across the

V

IN

+

C

R

SENSE

IN

1000pF

100Ω

18mV

V

IN

–

SENSE

–

+

12V

50A I

PK

Q1

2N2222

RATING

LTC1624

V

IN

TG

LTC1624

R*

TRANSIENT VOLTAGE

SUPPRESSOR

L1

V

SW

– 0.7V)

OUT

GENERAL INSTRUMENT

1.5KA24A

+

D1

MBRS340T3

C

OUT

(V

OUT

180µA

*R =

1624 F04

1624 F05

Figure 4. Plugging into the Cigarette Lighter

Figure 5. Modifying Burst Mode Operation

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With the 0.05Ω sense resistor I_SC(AVG)= 2A will result,

increasing the 0.5V Schottky diode dissipation to 0.98W.

100Ω resistor in series with the SENSE^–pin. This offset

cancels the internal offset in current comparator I₂(refer

to Functional Diagram). This comparator in conjunction

with the voltage on the I_TH/RUN pin determines when to

enter into Burst Mode operation (refer to Low Current

Operation in Operation section). With the additional exter-

nal offset present, the drive to the topside MOSFET is

alwaysenabledeverycycleandconstantfrequencyopera-

C_INis chosen for an RMS current rating of at least 1.0A at

temperature. C_OUTis chosen with an ESR of 0.03Ω for low

outputripple. Theoutputrippleincontinuousmodewillbe

highest at the maximum input voltage. The output voltage

ripple due to ESR is approximately:

V

_ORIPPLE= R_ESR(∆I_L) = 0.03Ω (1.58A_P-P) = 47mV_P-P

tion occurs for I_OUT> I_OUT(MIN)

.

Step-Down Converter: Duty Cycle Limitations

Step-Down Converter: Design Example

At high input to output differential voltages the on-time

gets very small. Due to internal gate delays and response

times of the internal circuitry the minimum recommended

on-time is 450ns. Since the LTC1624’s frequency is inter-

nally set to 200kHz a potential duty cycle limitation exists.

When the duty cycle is less than 9%, cycle skipping may

occurwhichincreasestheinductorripplecurrentbutdoes

not cause V_OUTto lose regulation. Avoiding cycle skipping

imposes a limit on the input voltage for a given output

voltage only when V_OUT< 2.2V using 30V MOSFETs.

(Remember not to exceed the absolute maximum voltage

of 36V.)

As a design example, assume V_IN= 12V(nominal),

V_IN= 22V(max), V_OUT= 3.3V and I_MAX= 2A. R_SENSEcan

immediately be calculated:

R_SENSE= 100mV/2A = 0.05Ω

Assume a 10µH inductor. To check the actual value of the

ripple current the following equation is used:

V − V

V

+ V

IN

OUT OUT D

∆I =

L

V + V

f L

( )( )

IN

D

The highest value of the ripple current occurs at the

maximum input voltage:

V_IN(MAX)= 11.1V_OUT+ 5V

For DC > 9%

22V − 3.3V 3.3V + 0.5V

Boost Converter Applications

∆I =

= 1.58A

P-P

L

22V + 0.5V

200kHz 10µH

(

)

The LTC1624 is also well-suited to boost converter appli-

cations. A boost converter steps up the input voltage to a

higher voltage as shown in Figure 6.

The power dissipation on the topside MOSFET can be

easily estimated. Choosing a Siliconix Si4412DY results

in: R_DS(ON)= 0.042Ω, C_RSS= 100pF. At maximum input

voltage with T(estimated) = 50°C:

V

IN

+

R

SENSE

C

IN

P

=

MAIN

V

IN

–

2

SENSE

3.3V + 0.5V

22V + 0.5V

2A 1+ 0.005 50°C − 25°C 0.042Ω

(

)

(

)(

) (

]

)

[

L1

BOOST

D1

1.85

V

OUT

LTC1624

GND

+ 2.5 22V

2A 100pF 200kHz = 62mW

(

)

(

)(

)

M1

TG

R2

C

B

+

The most stringent requirement for the Schottky diode

occurswhenV_OUT=0V(i.e.shortcircuit)atmaximumV_IN.

In this case the worst-case dissipation rises to:

V

SW

C

FB

OUT

R1

1624 F06

V

IN

P = I

V

( )

D

SC AVG

D

(

)

Figure 6. Boost Converter

V + V

IN

D

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Boost Converters: Power MOSFET Selection

Boost Converter: Inductor Selection

One external N-channel power MOSFET must be selected

for use with the LTC1624 for the switch. In boost applica-

tions the source of the power MOSFET is grounded along

with the SW pin. The peak-to-peak gate drive levels are set

by the INTV_CCvoltage. The gate drive voltage is equal to

approximately 5V for V_IN> 5.6V and a logic level MOSFET

can be used. At V_INvoltages below 5V the gate drive

voltage is equal to V_IN– 0.6V and a sublogic level MOSFET

should be used.

For most applications the inductor will fall in the range of

10µH to 100µH. Higher values reduce the input ripple

voltage and reduce core loss. Lower inductor values are

chosen to reduce physical size.

Theinputcurrentoftheboostconverteriscalculatedatfull

load current. Peak inductor current can be significantly

higher than output current, especially with smaller induc-

tors and lighter loads. The following formula assumes

continuousmodeoperationandcalculatesmaximumpeak

inductor current at minimum V_IN:

Selection criteria for the power MOSFET include the “ON”

resistance R_DS(ON), reverse transfer capacitance C_RSS

,

∆I

input voltage and maximum output current. When the

LTC1624 is operating in continuous mode the duty cycle

for the MOSFET is given by:

L MAX

(

)

V

OUT

I

=I

+

L PEAK

OUT MAX

(

)

(

)

2

V

IN MIN

(

)

V

IN

+ V

D

The ripple current in the inductor (∆I_L) is typically 20% to

30% of the peak inductor current occuring at V_IN(MIN)and

Main Switch Duty Cycle = 1−

V

OUT

I_OUT(MAX)

.

The MOSFET power dissipation at maximum output cur-

rent is given by:

V V

+ V − V

(

)

IN OUT

D

IN

∆I

=

L P-P

(

)

V

2

200kHz L V

+ V

IN MIN

(

)( )(

)

OUT

D

(

)

P

= I

1−

1+ δ R

+

(

)

MAIN

IN MAX

(

DS ON

)

(

)

V

+ V

OUT

D

with ∆I_L(MAX)= ∆I_L(P-P)at V_IN= V_IN(MIN)

.

1.85

)

Remember boost converters are not short-circuit pro-

tected, and that under output short conditions, inductor

current is limited only by the available current of the input

supply, I_{OUT(OVERLOAD)}. Specify the maximum inductor

current to safely handle the greater of I_L(PEAK)or

k V

I

C

200kHz

(

)(

)

OUT

IN MAX

RSS

(

)

V

+ V

D

OUT

where I

=I

IN MAX

(

OUT MAX

)

(

)

I

_{OUT(OVERLOAD)}. Make sure the inductor’s saturation cur-

V

IN MIN

(

)

rent rating (current when inductance begins to fall)

exceeds the maximum current rating set by R_SENSE

.

δ is the temperature dependency of R_DS(ON)and k is a

constant inversely related to the gate drive current.

Boost Converter: R_SENSESelection for Maximum

Output Current

MOSFETs have I²R losses, plus the P_MAINequation

includes an additional term for transition losses that are

highest at high output voltages. For V_OUT< 20V the high

currentefficiencygenerallyimproveswithlargerMOSFETs,

while for V_OUT> 20V the transition losses rapidly increase

to the point that the use of a higher R_DS(ON)device with

lower C_RSSactual provides higher efficiency. For addi-

tional information refer to Step-Down Converter: Power

MOSFET Selection in the Applications Information

section.

R_SENSEis chosen based on the required output current.

Remember the LTC1624 current comparator has a maxi-

mum threshold of 160mV/R_SENSE. The current compara-

torthresholdsetsthepeakoftheinductorcurrent,yielding

a maximum average output current I_OUT(MAX)equal to

I_L(PEAK)less half the peak-to-peak ripple current (∆I_L),

divided by the output-input voltage ratio (see equation for

I_{L(PEAK) .}

)

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Allowing a margin for variations in the LTC1624 (without

considering variation in R_SENSE), assuming 30% ripple

current in the inductor, yields:

0.3 V

V

− V

(

)(

)

IN OUT

IN

C I

≈

IN RIPPLE

200kHz L V

(

)( )(

)

OUT

V

Theinputcapacitorcanseeaveryhighsurgecurrentwhen

abatteryissuddenlyconnectedandsolidtantalumcapaci-

tors can fail under this condition. Be sure to specify surge

tested capacitors.

IN MIN

(

)

100mV

R

=

SENSE

V

+ V

I

OUT

D

OUT MAX

(

)

Boost Converter: Output Diode

Boost Converter: Duty Cycle Limitations

The output diode conducts current only during the switch

off-time. Peak reverse voltage for boost converters is

equal to the regulator output voltage. Average forward

current in normal operation is equal to output current.

Remember boost converters are not short-circuit pro-

tected. Checktobesurethediode’scurrentratingexceeds

themaximumcurrentsetbyR_SENSE.Schottkydiodessuch

as Motorola MBR130LT3 are recommended.

The minimum on-time of 450ns sets a limit on how close

V_INcan approach V_OUTwithout the output voltage over-

shooting and tripping the overvoltage comparator. Unless

very low values of inductances are used, this should never

be a problem. The maximum input voltage in continuous

mode is:

V_IN(MAX)= 0.91V_OUT+ 0.5V

For DC = 9%

Boost Converter: Output Capacitors

SEPIC Converter Applications

The output capacitor is normally chosen by its effective

series resistance (ESR), because this is what determines

output ripple voltage.

The LTC1624 is also well-suited to SEPIC (Single Ended

Primary Inductance Converter) converter applications.

The SEPIC converter shown in Figure 7 uses two induc-

tors. The advantage of the SEPIC converter is the input

voltage may be higher or lower than the output voltage.

Since the output capacitor’s ESR affects efficiency, use

low ESR capacitors for best performance. Boost regula-

tors have large RMS ripple current in the output capacitor

that must be rated to handle the current. The output

capacitor ripple current (RMS) is:

The first inductor L1 together with the main N-channel

MOSFET switch resemble a boost converter. The second

inductor L2 and output diode D1 resemble a flyback or

buck-boostconverter. ThetwoinductorsL1andL2canbe

independentbutalsocanbewoundonthesamecoresince

V

− V

IN

OUT

C

I

≈I

OUT RIPPLE RMS

OUT

(

)

V

IN

V

IN

Output ripple is then simply: V_OUT= R_ESR(∆I_L(RMS)).

+

R

SENSE

C

IN

Boost Converter: Input Capacitors

V

IN

SENSE

–

The input capacitor of a boost converter is less critical due

to the fact that the input current waveform is triangular,

and does not contain large square wave currents as found

in the output capacitor. The input voltage source imped-

ance determines the size of the capacitor that is typically

10µF to 100µF. A low ESR is recommended although not

as critical as the output capacitor and can be on the order

of 0.3Ω. Input capacitor ripple current for the LTC1624

used as a boost converter is:

L1

C1

D1

BOOST

V

OUT

+

LTC1624

GND

L2

V

M1

TG

R2

R1

C

+

B

C

SW

OUT

FB

1624 F07

Figure 7. SEPIC Converter

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identical voltages are applied to L1 and L2 throughout the

switching cycle. By making L1 = L2 and wound on the

same core the input ripple is reduced along with cost and

size. All SEPIC applications information that follows

assumes L1 = L2 = L.

highest at high total input plus output voltages. For

(V_IN+ V_OUT) < 20V the high current efficiency generally

improves with larger MOSFETs, while for (V_IN+ V_OUT) >

20V the transition losses rapidly increase to the point that

the use of a higher R_DS(ON)device with lower C_RSSactual

provideshigherefficiency.Foradditionalinformationrefer

to the Step-Down Converter: Power MOSFET Selection in

the Applications Information section.

SEPIC Converter: Power MOSFET Selection

One external N-channel power MOSFET must be selected

for use with the LTC1624 for the switch. As in boost

applications the source of the power MOSFET is grounded

along with the SW pin. The peak-to-peak gate drive levels

are set by the INTV_CCvoltage. This voltage is equal to

approximately 5V for V_IN> 5.6V and a logic level MOSFET

can be used. At V_INvoltages below 5V the INTV_CCvoltage

is equal to V_IN– 0.6V and a sublogic level MOSFET should

be used.

SEPIC Converter: Inductor Selection

For most applications the equal inductor values will fall in

the range of 10µH to 100µH. Higher values reduce the

input ripple voltage and reduce core loss. Lower inductor

values are chosen to reduce physical size and improve

transient response.

Like the boost converter the input current of the SEPIC

converter is calculated at full load current. Peak inductor

current can be significantly higher than output current,

especially with smaller inductors and lighter loads. The

following formula assumes continuous mode operation

and calculates maximum peak inductor current at mini-

mum V_IN:

Selection criteria for the power MOSFET include the “ON”

resistance R_DS(ON), reverse transfer capacitance C_RSS

,

input voltage and maximum output current. When the

LTC1624 is operating in continuous mode the duty cycle

for the MOSFET is given by:

V

+ V

D

OUT

Main Switch Duty Cycle =

V + V

+ V

D

IN

OUT

∆I

2

V

L1

OUT

I

=I

+

The MOSFET power dissipation and maximum switch

current at maximum output current are given by:

L1 PEAK

OUT MAX

(

)

(

)

V

IN MIN

(

IN MIN

)

V

+ V

D

P

=

∆I

(

)

MAIN

L2

I

=I

+

L2 PEAK

OUT MAX

(

)

(

)

2

V

IN MIN

V

+ V

D

(

)

OUT

I

1+ δ R

+

(

)

SW MAX

DS ON

(

)

(

)

V

+V

+ V

IN MIN

OUT D

(

)

The ripple current in the inductor (∆I_L) is typically 20% to

1.85

30%ofthepeakcurrentoccuringatV_IN(MIN)andI_OUT(MAX)

,

k V

+V

I

C

( )(

RSS

200kHz

)

IN MIN

OUT

SW MAX

(

)

(

)

and ∆I_L1= ∆I_L2. Maximum ∆I_Loccurs at maximum V_IN.

V

+ V

(

)(

)

IN OUT

D

∆I

=

L P-P

(

)

V

+ V

D

OUT

200kHz L V + V

+ V

D

where I

=I

+1

(

)( )(

)

IN

OUT

SW MAX

OUT MAX

(

)

(

)

V

IN MIN

(

)

By making L1 = L2 and wound on the same core the value

of inductance in all the above equations are replaced by

2L due to their mutual inductance. Doing this maintains

thesameripplecurrentandinductiveenergystorageinthe

inductors. For example a Coiltronix CTX10-4 is a 10µH

inductor with two windings. With the windings in parallel

δ is the temperature dependency of R_DS(ON)and k is a

constant inversely related to the gate drive current. The

peak switch current is I_SW(MAX)+ ∆I_L.

MOSFETs have I²R losses plus the P_MAINequation

includes an additional term for transition losses that are

16

LTC1624

U

W U U

APPLICATIONS INFORMATION

10µH inductance is obtained with a current rating of 4A.

Splitting the two windings creates two 10µH inductors

with a current rating of 2A each. Therefore substitute

(2)(10µH) = 20µH for L in the equations.

outputripplevoltage. Theinputcapacitorneedstobesized

to handle the ripple current safely.

Since the output capacitor’s ESR affects efficiency, use

low ESR capacitors for best performance. SEPIC regula-

tors, like step-down regulators, have a triangular current

waveform but have maximum ripple at V_IN(MAX). The input

capacitor ripple current is:

Specify the maximum inductor current to safely handle

I_L(PEAK). Make sure the inductor’s saturation current rat-

ing (current when inductance begins to fall) exceeds the

maximum current rating set by R_SENSE

.

∆I

L

I

=

RIPPLE RMS

(

)

SEPIC Converter: R_SENSESelection for Maximum

Output Current

12

The output capacitor ripple current is:

R_SENSEis chosen based on the required output current.

Remember the LTC1624 current comparator has a maxi-

mum threshold of 160mV/R_SENSE. The current compara-

torthresholdsetsthepeakoftheinductorcurrent,yielding

a maximum average output current I_OUT(MAX)equal to

I_L1(PEAK)less half the peak-to-peak ripple current, ∆I_L,

divided by the output-input voltage ratio (see equation for

V

OUT

I

=I

OUT

RIPPLE RMS

(

)

IN

The output capacitor ripple voltage (RMS) is:

V_OUT(RIPPLE)= 2(∆I_L)(ESR)

I_{L1(PEAK) .}

)

Theinputcapacitorcanseeaveryhighsurgecurrentwhen

a battery is suddenly connected, and solid tantalum

capacitors can fail under this condition. Be sure to specify

surge tested capacitors.

Allowing a margin for variations in the LTC1624 (without

considering variation in R_SENSE), assuming 30% ripple

current in the inductor, yields:

SEPIC Converter: Coupling Capacitor (C1)

V

IN MIN

(

)

D

100mV

R

=

SENSE

The coupling capacitor C1 in Figure 7 sees a nearly

rectangular current waveform. During the off-time the

current through C1 is I_OUT(V_OUT/V_IN) while approximately

–I_OUTflows though C1 during the on-time. This current

waveform creates a triangular ripple voltage on C1:

I

V

+ V

OUT MAX

OUT

(

)

SEPIC Converter: Output Diode

The output diode conducts current only during the switch

off-time. Peak reverse voltage for SEPIC converters is

equal to V_OUT+ V_IN. Average forward current in normal

operation is equal to output current. Peak current is:

V

I

OUT

∆V =

C1

V + V

+ V

D

200kHz C1

IN

OUT

(

)( )

V

+ V

D

The maximum voltage on C1 is then:

OUT

I

=I

+1 + ∆I

L

D1 PEAK

OUT MAX

(

)

(

)

V

IN MIN

(

)

V_C1(MAX)= V_IN+ ∆V_C1/2 (typically close to V_IN(MAX)).

The ripple current though C1 is:

Schottky diodes such as MBR130LT3 are recommended.

V

OUT

SEPIC Converter: Input and Output Capacitors

I

=I

OUT

RIPPLE C1

( )

IN

The output capacitor is normally chosen by its effective

series resistance (ESR), because this is what determines

The maximum ripple current occurs at I_OUT(MAX)and

V_IN(MIN). The capacitance of C1 should be large enough so

17

LTC1624

U

W U U

APPLICATIONS INFORMATION

Positive-to-Negative Converter: Output Voltage

Programming

that the voltage across C1 is constant such that V_C1= V_IN

at full load over the entire V_INrange. Assuming the enegry

storage in the coupling capacitor C1 must be equal to the

enegry stored in L1, the minimum capacitance of C1 is:

Setting the output voltage for a positive-to-negative con-

verter is different from other architectures since the feed-

back voltage is referenced to the LTC1624 ground pin and

the ground pin is referenced to –V_OUT. The output voltage

is set by a resistive divider according to the following

formula:

2

) (

2

)

L1I

V

OUT

(

OUT

C1

=

MIN

(

)

4

)

V

IN MIN

(

R1

R2

DC

1−DC

SEPIC Converter: Duty Cycle Limitations

V

= 1.19V 1+

≈ −V

IN

OUT

The minimum on-time of 450ns sets a limit on how high

an input-to-output ratio can be tolerated while not skip-

ping cycles. This only impacts designs when very low

output voltages (V_OUT< 2.5V) are needed. Note that a

SEPIC converter would not be appropriate at these low

output voltages. The maximum input voltage is (remem-

ber not to exceed the absolute maximum limit of 36V):

The external resistive divider is connected to the output as

shown in Figure 8.

Positive-to-Negative Converter: Power

MOSFET Selection

One external N-channel power MOSFET must be selected

for use with the LTC1624 for the switch. As in step-down

applications the source of the power MOSFET is con-

nected to the Schottky diode and inductor. The peak-to-

peak gate drive levels are set by the INTV_CCvoltage. The

gate drive voltage is equal to approximately 5V for V_IN>

5.6VandalogiclevelMOSFETcanbeused. AtV_INvoltages

below 5V the INTV_CCvoltage is equal to V_IN– 0.6V and a

sublogic level MOSFET should be used.

V_IN(MAX)= 10.1V_OUT+ 5V

For DC > 9%

Positive-to-Negative Converter Applications

The LTC1624 can also be used as a positive-to-negative

converter with a grounded inductor shown in Figure 8.

Since the LTC1624 requires a positive feedback signal

relative to device ground, Pin 4 must be tied to the

regulated negative output. A resistive divider from the

negative output to ground sets the output voltage.

Selection criteria for the power MOSFET include the “ON”

resistance R_DS(ON), reverse transfer capacitance C_RSS

input voltage and maximum output current. When the

LTC1624 is operating in continuous mode the duty cycle

for the MOSFET is given by:

Remember not to exceed maximum V_INratings V_IN

V_OUT≤ 36V.

+

,

1000pF

8

V

IN

V

+ V

D

1

2

3

4

OUT

–

V

IN

SENSE

Main Switch Duty Cycle =

+

C

R

C

IN

SENSE

C

V + V

+ V

D

7

IN

OUT

BOOST

LTC1624

I

/RUN

TH

6

5

M1

TG

with V_OUTbeing the absolute value of V_OUT.

V

FB

C

B

100pF

L1

The MOSFET power dissipation and maximum switch

current are given by:

SW

GND

D1

R1

P

MAIN

= I

×

C

OUT

SW MAX

(

)

R2

I

I + δ R

+

(

)

OUT MAX

DS ON

(

)

(

)

{

–V

OUT

1.85

1624 F08

k V

(

+ V

C

(

200kHz

)

)(

RSS

)

IN MAX

OUT

Figure 8. Positive-to-Negative Converter

(

)

{

18

LTC1624

U

W U U

APPLICATIONS INFORMATION

V

+ V

(

)

V + V

+ V

D

IN

(

OUT D

)

IN

OUT

∆I

=

where: I

=I

L P-P

(

SW MAX

OUT MAX

)

(

)

(

)

V

200kHz L V + V

+ V

D

(

)( )

IN

(

IN

OUT

)

Specify the maximum inductor current to safely handle

I_L(PEAK). Make sure the inductor’s saturation current rat-

ing (current when inductance begins to fall) exceeds the

δ is the temperature dependency of R_DS(ON)and k is a

constant inversely related to the gate drive current. The

maximum switch current occurs at V_IN(MIN)and the peak

switch current is I_SW(MAX)+ ∆I_L/2. The maximum voltage

maximum current rating set by R_SENSE

.

across the switch is V_IN(MAX)+ V_OUT

.

Positive-to-Negative Converter: R_SENSESelection for

Maximum Output Current

MOSFETs have I²R losses plus the P_MAINequation

includes an additional term for transition losses that are

highest at high total input plus output voltages. For

( V_OUT+ V_IN) < 20V the high current efficiency generally

improves with larger MOSFETs, while for ( V_OUT+ V_IN)

> 20V the transition losses rapidly increase to the point

that the use of a higher R_DS(ON)device with lower C_RSS

actual provides higher efficiency. For additional informa-

tion refer to the Step-Down Converter: Power MOSFET

Selection in the Applications Information section.

R_SENSEis chosen based on the required output current.

Remember the LTC1624 current comparator has a maxi-

mum threshold of 160mV/R_SENSE. The current compara-

torthresholdsetsthepeakoftheinductorcurrent,yielding

a maximum average output current I_OUT(MAX)equal to

I_L(PEAK)less half the peak-to-peak ripple current with the

remainder divided by the duty cycle.

Allowing a margin for variations in the LTC1624 (without

consideringvariationinR_SENSE)andassuming30%ripple

current in the inductor, yields:

Positive-to-Negative Converter: Inductor Selection

For most applications the inductor will fall in the range of

10µH to 100µH. Higher values reduce the input and output

ripple voltage (although not as much as step-down con-

verters) and also reduce core loss. Lower inductor values

are chosen to reduce physical size and improve transient

response but do increase output ripple.

V

IN MIN

(

)

100mV

R

=

SENSE

I

V

+ V

D

OUT MAX

IN MIN

OUT

(

)

(

)

Positive-to-Negative Converter: Output Diode

The output diode conducts current only during the switch

off-time. Peak reverse voltage for positive-to-negative

converters is equal to V_OUT+ V_IN. Average forward

current in normal operation is equal to I_D(PEAK)– ∆I_L/2.

Peak diode current (occurring at V_IN(MIN)) is:

Like the boost converter, the input current of the positive-

to-negative converter is calculated at full load current.

Peak inductor current can be significantly higher than

output current, especially with smaller inductors (with

high ∆I_Lvalues). The following formula assumes continu-

ous mode operation and calculates maximum peak induc-

tor current at minimum V_IN:

V

+V

D

(

OUT

)

∆I

2

L

I

=I

+1 +

D PEAK

OUT MAX

(

)

(

)

V

IN

V + V

+ V

D

∆I

2

IN

OUT

L

I

=I

+

L PEAK

OUT MAX

(

)

(

)

V

IN

Positive-to-Negative Converter: Input and

Output Capacitors

The ripple current in the inductor (∆I_L) is typically 20% to

50% of the peak inductor current occuring at V_IN(MIN)and

I_OUT(MAX)to minimize output ripple. Maximum ∆I_Loccurs

at minimum V_IN.

The output capacitor is normally chosen by its effective

series resistance (ESR), because this is what determines

output ripple voltage. Both input and output capacitors

need to be sized to handle the ripple current safely.

19

LTC1624

U

W U U

APPLICATIONS INFORMATION

I_TH/RUN pin below 0.8V relative to the LTC1624 ground

pin. With the LTC1624 ground pin referenced to –V_OUT

Positive-to-negativeconvertershavehighripplecurrentin

both the input and output capacitors. For long capacitor

lifetime, the RMS value of this current must be less than

the high frequency ripple rating of the capacitor.

,

the nonimal range on the I_TH/RUN pin is –V_OUT(in

shutdown) to (–V_OUT+ 2.4V)(at Max I_OUT). Referring to

Figure 15, M2, M3 and R3 provide a level shift from typical

TTL levels to the LTC1624 operating as positive-to-nega-

tive converter. MOSFET M3 supplies gate drive to M2

duringshutdown, whileM2pullstheI_TH/RUNpinvoltageto

–V_OUT, shutting down the LTC1624.

ThefollowingformulagivesanapproximatevalueforRMS

ripple current. This formula assumes continuous mode

andlowcurrentripple.Smallinductorswillgivesomewhat

higher ripple current, especially in discontinuous mode.

For the exact formulas refer to Application Note 44, pages

28 to 30. The input and output capacitor ripple current

(occurring at V_IN(MIN)) is:

Step-Down Converters: PC Board Layout Checklist

When laying out the printed circuit board, the following

checklist should be used to ensure proper operation of the

LTC1624. These items are also illustrated graphically in

the layout diagram of Figure 9. Check the following in your

layout:

V

OUT

Capacitor I

= ff I

( )(

)

RMS

OUT

V

IN

ff = Fudge factor (1.2 to 2.0)

The output peak-to-peak ripple voltage is:

V_OUT(P-P)= R_ESR(I_D(MAX)

1. Are the signal and power grounds segregated? The

LTC1624 ground (Pin 4) must return to the (–) plate

of C_OUT.

)

2. Does the V_FB(Pin 3) connect directly to the feedback

resistors? The resistive divider R1, R2 must be con-

nectedbetweenthe(+)plateofC_OUTandsignalground.

The 100pF capacitor should be as close as possible to

the LTC1624.

The input capacitor can also see a very high surge current

when a battery is suddenly connected, and solid tantalum

capacitors can fail under this condition. Be sure to specify

surge tested capacitors.

Positive-to-Negative Converter: Duty Cycle

Limitations

3. Does the V_INlead connect to the input voltage at the

samepointasR_SENSEandaretheSENSE^–andV_INleads

routed together with minimum PC trace spacing? The

filter capacitor between V_INand SENSE^–should be as

close as possible to the LTC1624.

The minimum on-time of 450ns sets a limit on how high

ofinput-to-outputratiocanbetoleratedwhilenotskipping

cycles. This only impacts designs when very low output

voltages ( V_OUT< 2.5V) are needed. The maximum input

voltage is:

4. Does the (+) plate of C_INconnect to R_SENSEas closely

as possible? This capacitor provides the AC current to

the MOSFET(s). Also, does C_INconnect as close as

possible to the V_INand ground pin of the LTC1624?

This capacitor also supplies the energy required to

recharge the bootstrap capacitor. Adequate input

decoupling is critical for proper operation.

V_IN(MAX)< 10.1V_OUT+ 5V

For DC > 9%

V_IN(MAX)<36V– V_OUTForabsolutemaximumratings

Positive-to-Negative Converter: Shutdown

Considerations

5. Keep the switch node SW away from sensitive small-

signal nodes. Ideally, M1, L1 and D1 should be con-

nected as closely as possible at the switch node.

Since the ground pin on the LTC1624 is referenced to

–V_OUT, additional circuitry is needed to put the LTC1624

into shutdown. Shutdown is enabled by pulling the

20

LTC1624

U

TYPICAL APPLICATIONS

1000pF

+

1

8

7

6

5

–

V

SENSE

+

IN

V

IN

C

IN

R

C

SENSE

2

BOOST

/RUN

LTC1624

I

TH

–

3

4

M1

TG

V

FB

L1

C

B

100pF

+

0.1µF

SW

GND

R2

R1

D1

+

C

V

OUT

BOLD LINES INDICATE

HIGH CURRENT PATHS

–

1624 F09

Figure 9. LTC1624 Layout Diagram (See Board Layout Checklist)

V

IN

5.3V TO 28V

C

IN

+

1000pF

0.1µF

22µF

35V

× 2

R

1

2

3

8

7

6

SENSE

–

V

SENSE

IN

0.033Ω

BOOST

/RUN

LTC1624

I

TH

C

560pF

D2

M1

TG

V

FB

R

CMDSH-3

Si4412DY

C

B

4.7k

100pF

V

5V

3A

0.1µF

OUT

4

5

SW

GND

L1*

10µH

D1

MBRS340T3

R2

35.7k

1%

C

OUT

+

100µF

10V

R1

11k

1%

*COILTRONICS CTX10-4

× 2

1624 F10

Figure 10. 5V/3A Converter with Output Derived Boost Voltage

21

LTC1624

U

TYPICAL APPLICATIONS

V

IN

4.8V TO 22V

1000pF

1

2

8

7

6

–

V

SENSE

IN

C

IN

+

R

22µF

35V

× 2

SENSE

0.1µF

0.068Ω

BOOST

/RUN

LTC1624

I

TH

C

470pF

3

4

M1

Si6436DY

TG

V

FB

R

C

L1*

10µH

C

B

6.8k

100pF

V

1.8V

1.5A

0.1µF

OUT

5

SW

GND

D1

R2

35.7k

1%

MBRS340T3

C

OUT

+

100µF

10V

R1

69.8k

1%

× 2

*SUMIDA CDR105B-100

1624 F11

Figure 11. Wide Input Range 1.8V/1.5A Converter

V

IN

12.3V TO 28V

1000pF

0.1µF

1

2

3

8

7

6

–

V

SENSE

IN

C

IN

+

R

SENSE

22µF

0.068Ω

35V

× 2

BOOST

/RUN

LTC1624

I

TH

C

470pF

M1

Si4412DY

TG

V

FB

R

C

L1*

47µH

C

B

6.8k

100pF

V

12V

1A

0.1µF

OUT

4

5

SW

GND

D1

R2

35.7k

1%

MBRS140T3

C

OUT

+

100µF

16V

R1

3.92k

1%

× 2

*SUMIDA CDRH125-470

Figure 12. 12V/1A Low Dropout Converter

1624 F12

V

IN

5.2V TO 11V

1000pF

0.1µF

1

2

3

8

7

6

–

V

SENSE

IN

C

IN

+

22µF

35V

× 2

R

SENSE

0.04Ω

D1

MBRS130LT3

L1*

22µH

BOOST

/RUN

LTC1624

I

TH

C

330pF

V

TG

OUT

V

FB

R

C

M1

Si4412DY

12V

C

B

3.3k

100pF

0.75A

0.1µF

4

5

R2

35.7k

1%

SW

GND

C

OUT

+

100µF

16V

R1

3.92k

1%

*SUMIDA CDRH125-220

× 2

Figure 13. 12V/0.75A Boost Converter

1624 F13

22

LTC1624

U

TYPICAL APPLICATIONS

V

IN

5V TO 15V

1000pF

0.1µF

1

8

7

6

+

C

–

IN

V

SENSE

IN

22µF

35V

R

D1

SENSE

0.068Ω

35V

2

MBRS130LT3

L1a*

BOOST

/RUN

LTC1624

I

TH

C

+

330pF

3

4

V

M1

Si4412DY

OUT

TG

V

FB

12V

R

4.7k

L1b*

C

B

0.5A

100pF

0.1µF

5

R2

35.7k

1%

SW

GND

C

OUT

+

100µF

16V

R1

3.92k

1%

*COILTRONICS CTX20-4

× 2

Figure 14. 12V/0.4A SEPIC Converter

1624 F14

V

IN

5V TO 22V

1000pF

1

2

3

8

7

6

–

V

C

SENSE

C

IN

C

IN

+

R

C

1000pF

22µF

35V

× 2

R

SENSE

3.3k

0.1µF

0.025Ω

V

CC

V

CC

BOOST

/RUN

LTC1624

I

TH

SHUTDOWN

M1

Si4410DY

M3

TP0610L

TG

V

D2

CMDSH-3

FB

L1*

33µH

C

B

100pF

0.1µF

4

5

M2

VN2222

SW

GND

R2

78.7k

1%

D1

MBRS340T3

C

OUT

+

100µF

10V

R3

R1

24.9k

1%

100k

× 2

V

–5V

2A

OUT

1624 F15

*COILCRAFT DO5022P-333

Figure 15. Inverting –5V/2A Converter

V

IN

3.5V TO 18V

1000pF

0.1µF

1

8

7

6

–

V

SENSE

IN

C

IN

+

R

SENSE

22µF

35V

× 2

2

0.068Ω

BOOST

/RUN

LTC1624

I

TH

C

470pF

3

4

M1

TG

V

FB

Si6426DQ

R

C

L1*

20µH

C

B

100pF

6.8k

V

0.1µF

OUT

5

SW

3.3V

GND

1.5A

D1

R2

35.7k

1%

MBRS340T3

C

OUT

+

100µF

10V

R1

20k

1%

× 2

*COILTRONICS CTX20-4

1624 F16

Figure 16. Low Dropout 3.3V/1.5A Converter

23

LTC1624

U

TYPICAL APPLICATIONS

V

IN

3.6V TO 18V

C

IN

+

1000pF

22µF

35V

× 2

R

SENSE

0.05Ω

1

8

7

6

–

V

SENSE

IN

22µF

35V

+

D1

0.1µF

D2

CMDSH-3

MBRS130LT3

2

3

L1a*

BOOST

/RUN

LTC1624

I

TH

C

V

OUT

330pF

M1

Si6426DQ

V

5V

1A

OUT

TG

V

FB

R

L1b*

C

B

100pF

6.8k

0.1µF

4

5

R2

35.7k

1%

SW

GND

C

OUT

+

100µF

16V

R1

11k

1%

× 2

* COILTRONICS CTX20-4

1624 F17

Figure 17. 5V/1A SEPIC Converter with Output Derived Boost Voltage

V

IN

13V TO

28V

+

C

, C

IN1 IN2

1000µF

35V

R

, 0.015Ω

SENSE1

× 2

SENSE2

C4, 0.1µF

C5

3.3µF

50V

C7

3.3µF

50V

LTC1624

1

2

3

8

7

6

–

V

C

SENSE

IN

C

100pF

BOOST

TG

I

/RUN

TH

C

B

0.1µF

M1*

V

FB

V

12V

10A

L1

R1

11k

1%

OUT

4

5

R

C

SW

GND

20k

D2

R5

MBR0540

C

220Ω

+

OUT

R2

100k

1%

C10

220pF

D1*

2700µF

16V

Z1

IN 755

C9

0.1µF

1624 F18

C

, C = SANYO 35MV1000GX

M1 = INTERNATIONAL RECTIFIER IRL3803

R , R = IRC LR2010-01-R015-F

SENSE1 SENSE2

* BOTH D1 AND M1 MOUNTED TO SAME

THERMALLOY #6399B HEAT SINK

IN1 IN2

C5, C7 = WIMA MKS2

C

= SANYO 16MV2700GX

OUT

D1 = MOTOROLA MBR2535CT

L1 = PULSE ENGINEERING PO472

Figure 18. 24V to 12V/10A Buck Converter with Output-Derived Boost Voltage

24

LTC1624

U

TYPICAL APPLICATIONS

V

IN

20V TO

32V

R

+

C

SENSE

IN

0.025Ω

22µF

L1

35V

47µH

C5

0.1µF

D1

V

OUT

90V

LTC1624

0.5A

1

2

3

8

7

6

–

V

C

SENSE

IN

C

820pF

BOOST

TG

I

/RUN

C

OUT

+

TH

C

B

100µF

0.1µF

100V

M1

V

FB

4

5

C3

100pF

R1

13.3k

R

SW

C

GND

6.8k

R2, 1M, 1%

1624 F19

C

= KEMET T495X226M035AS

L1 = COILCRAFT D05022P-473

M1 = INTERNATIONAL RECTIFIER IRL 540NS

R = IRC LR2010-01-R025-F

SENSE

IN

= SANYO 100MV100GX

OUT

D1 = MOTOROLA MBRS1100

Figure 19. 24V to 90V at 0.5A Boost Converter

V

IN

9V TO

15V

R

+

C

SENSE

IN

0.005Ω, 5%

100µF

L1

16V

10µH

C5

0.1µF

D1*

V

24V

5A

OUT

LTC1624

R5

1

2

3

8

7

6

–

750Ω

0.5W

V

C

SENSE

IN

C

4700pF

BOOST

TG

I

/RUN

TH

C

B

0.1µF

M1*

V

FB

C

OUT2

+

OUT1

Z1

4

5

C3

100pF

C4

1500pF

R1

52.3k

R

1000µF

SW

C

GND

IN755

27k

35V

7.5V

R2, 1M, 1%

1624 F20

C

= KEMET T495X107M016AS

L1 = MAGNETICS CORE #55930AZ WINDING = 8T#14BIF

M1 = INTERNATIONAL RECTIFIER IRL 3803

SENSE

*BOTH D1 AND Q1 MOUNTED ON

THERMALLOY MODEL 6399 HEAT SINK

IN

OUT1 OUT2

, C

= SANYO 35MV 1000GX

D1 = MOTOROLA MBR2535CT

R

= IRC OAR-3, 0.005Ω, 5%

Figure 20. 12V to 24V/5A Boost Converter

25

LTC1624

TYPICAL APPLICATIONS

U

V

IN

13V TO

28V

+V

IN

+

C

, C

IN1 IN2

22µF

35V

× 2

R

SENSE

C5, 0.1µF

0.033Ω

LTC1624

1

2

3

8

–

V

C

SENSE

IN

C

330pF

7

BOOST

I

/RUN

TH

C

B

0.1µF

6

M1

TG

V

FB

L1

27µH

R4

0.025Ω

V

12V

3A

OUT

4

5

C4

100pF

R1

3.92k

R

SW

C

GND

C

OUT

10k

+

D1

MBRS340

R2

35.7k

C9

100pF

100µF

16V

× 2

C10

0.1µF

Q2

C11

0.1µF

R6

1

2

3

4

8

1

8

7

6

5

10k

+V

IN

SENSE

AVE

OUT

IN

NC

C12

1µF

7

2

3

4

I

PROG

NC/ADJ

OUT

LTC1620

LT1121-5

6

5

R7

56k

C13

0.1µF

GND

–IN

V

GND

NC

CC

+IN

SHDN

R8

1M

CURRENT

ADJ

C14, 0.01µF

1624 F21

C

, C = KEMET T495X226M035AS

IN1 IN2

L1 = SUMIDA CDRH127-270

= IRC LR2010-01-R033-F

R

SENSE

R4 = IRC LR2010-01-R025-F

M1 = SILICONIX Si4412DY

Q2 = MOTOROLA MMBT A14

Figure 21. 12V/3A Adjustable Current Power Supply for Battery Charger or Current Source Applications

26

LTC1624

U

TYPICAL APPLICATIONS

V

IN

4.8V TO 28V

1000pF

0.1µF

1

2

3

8

7

6

–

V

SENSE

IN

C

IN

+

R

SENSE

22µF

35V

× 3

0.015Ω

BOOST

/RUN

LTC1624

I

TH

C

680pF

M1**

TG

V

FB

R

C

L1*

8µH

C

B

100pF

3.3k

V

3.3V

6.5A

0.1µF

OUT

4

5

SW

GND

D1

MBRD835L

R2

35.7k

1%

C

OUT

+

100µF

10V

R1

20k

1%

× 3

* PANASONIC 12TS-7ROLB

** SILICONIX SUD50N03-10

1624 F22

Figure 22. High Current 3.3V/6.5A Converter

U

PACKAGE DESCRIPTION ^{Dimensions in inches (millimeters) unless otherwise noted.}

S8 Package

8-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.189 – 0.197*

(4.801 – 5.004)

7

5

8

6

0.150 – 0.157**

(3.810 – 3.988)

0.228 – 0.244

(5.791 – 6.197)

1

0.053 – 0.069

3

4

2

0.010 – 0.020

(0.254 – 0.508)

× 45°

(1.346 – 1.752)

0.004 – 0.010

(0.101 – 0.254)

0.008 – 0.010

(0.203 – 0.254)

0°– 8° TYP

0.016 – 0.050

0.406 – 1.270

0.050

(1.270)

TYP

0.014 – 0.019

(0.355 – 0.483)

*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD

FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

SO8 0996

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.

27

LTC1624

TYPICAL APPLICATION

U

R3, 10Ω

+V

IN

4.5V TO

5.5V

R

C

IN

SENSE

+

0.0082Ω

100µF

10V

C4, 0.1µF

C3, 0.033µF

× 4

LTC1624

1

2

3

8

–

V

C

SENSE

IN

C

820pF

D2

7

BOOST

I

/RUN

TH

C

B

0.1µF

6

M1

D1

TG

V

FB

L1

1.68µH

+V

OUT

3.3V

10A

4

5

R

R1

20k

C

C2

100pF

SW

GND

6.8k

C

OUT

R2

35.7k

C8

100pF

+

470µF

6.3V

× 2

V

OUT

RTN

Q2

10k

1624 F23

L1 = PULSE ENGINEERING PE53691

C

(× 4) = KEMET T495D107M010AS

OUT

IN

M1 = INTERNATIONAL RECTIFIER IRL3803S

Q2 = MOTOROLA MMBTA14LT1

(× 2) = AVX TPSV477M006R0055

D1 = MOTOROLA MBRB2515L

D2 = MOTOROLA MBR0520

R

SENSE

= IRC OAR3-R0082

Figure 23. 5V to 3.3V/10A Converter (Surface Mount)

RELATED PARTS

PART NUMBER

DESCRIPTION

COMMENTS

LTC1147

High Efficiency Step-Down Controller

100% DC, Burst Mode Operation, 8-Pin SO

and PDIP

LTC1148HV/LTC1148

LTC1149

High Efficiency Synchronous Step-Down Controllers

High Efficiency Synchronous Step-Down Controller

Monolithic 0.6A Step-Down Switching Regulator

100% DC, Burst Mode Operation, V < 20V

IN

100% DC,Std Threshold MOSFETs, V < 48V

IN

LTC1159

100% DC, Logic Level MOSFETs, V < 40V

IN

LTC1174

100% DC, Burst Mode Operation, 8-Pin SO

100% DC, Burst Mode Operation, 14-Pin SO

LTC1265

1.2A Monolithic High Efficiency Step-Down Switching Regulator

High Efficiency Synchronous Step-Down Controller, N-Channel Drive

1.5A, 500kHz Step-Down Switching Regulators

LTC1266

100% DC, Burst Mode Operation, V < 20V

IN

LT^®1375/LT1376

LTC1433/LTC1434

LTC1435

High Frequency

Monolithic 0.45A Low Noise Current Mode Step-Down Switching Regulators 16- and 20-Pin Narrow SSOP

High Efficiency Low Noise Synchronous Step-Down Controller,

N-Channel Drive

Burst Mode Operation, 16-Pin Narrow SO

Adaptive Power^TMMode, 20- and 24-Pin SSOP

100% DC, 8-Pin MSOP, V < 20V

LTC1436/LTC1436-PLL High Efficiency Low Noise Synchronous Step-Down Controllers,

N-Channel Drive

LTC1474/LTC1475

Ultralow Quiesent Current Step-Down Monolithic Switching Regulators

IN

Adaptive Power is a trademark of Linear Technology Corporation.

1624f LT/TP 0198 4K • PRINTED IN USA

28 ^{Linear Technology Corporation}

●

1630McCarthyBlvd., Milpitas, CA95035-7417 (408)432-1900

●

FAX: (408) 434-0507 TELEX: 499-3977 www.linear-tech.com

LINEAR TECHNOLOGY CORPORATION 1997


型号：	LTC1624CS8
厂家：	Linear
描述：	High Efficiency SO-8 N-Channel Switching Regulator Controller 高英法fi效率的SO-8 N沟道开关稳压器控制器稳压器开关控制器
文件：	总28页 (文件大小：493K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

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