LTC1627CS8#TRPBF [Linear]
LTC1627 - Monolithic Synchronous Step-Down Switching Regulator; Package: SO; Pins: 8; Temperature Range: 0°C to 70°C;
| 型号： | LTC1627CS8#TRPBF |
| 厂家： | 
Linear |
| 描述： | LTC1627 - Monolithic Synchronous Step-Down Switching Regulator; Package: SO; Pins: 8; Temperature Range: 0°C to 70°C 开关 光电二极管 |
| 文件： | 总16页 (文件大小：231K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC1627
Monolithic Synchronous
Step-Down Switching Regulator
U
FEATURES
DESCRIPTIO
The LTC®1627 is a monolithic current mode synchronous
buck regulator using a fixed frequency architecture. The
operating supply range is from 8.5V down to 2.65V, making
it suitable for one or two lithium-ion battery-powered appli-
cations. Burst Mode operation provides high efficiency at
low load currents. 100% duty cycle provides low dropout
operation, which extends operating time in battery-operated
systems.
■
High Efficiency: Up to 96%
■
Constant Frequency 350kHz Operation
■
2.65V to 8.5V VIN Range
■
VOUT from 0.8V to VIN, IOUT to 500mA
■
No Schottky Diode Required
■
Synchronizable Up to 525kHz
Selectable Burst ModeTM Operation
■
■
Low Dropout Operation: 100% Duty Cycle
■
Precision 2.5V Undervoltage Lockout
The operating frequency is internally set at 350kHz, allowing
theuseofsmallsurfacemountinductors.Forswitchingnoise
sensitive applications it can be externally synchronized up to
525kHz. The SYNC/FCB control pin guarantees regulation of
secondarywindingsregardlessofloadonthemainoutputby
forcingcontinuousoperation. BurstModeoperationisinhib-
ited during synchronization or when the SYNC/FCB pin is
pulled low to reduce noise and RF interference. Soft-start is
provided by an external capacitor.
■
Secondary Winding Regulation
■
Current Mode Operation for Excellent Line and
Load Transient Response
■
Low Quiescent Current: 200µA
■
Shutdown Mode Draws Only 15µA Supply Current
±1.5% Reference Accuracy
Available in 8-Lead SO Package
U
■
■
APPLICATIO S
■
Cellular Telephones
Optionalbootstrappingenhancestheinternalswitchdrivefor
singlelithium-ioncellapplications.Theinternalsynchronous
switch increases efficiency and eliminates the need for an
external Schottky diode, saving components and board
space. The LTC1627 comes in an 8-lead SO package.
■
Portable Instruments
■
Wireless Modems
■
RF Communications
■
Distributed Power Systems
■
Scanners
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
■
Single and Dual Cell Lithium
U
100
TYPICAL APPLICATIO
V
= 3.3V
OUT
V
= 3.6V
IN
95
90
85
80
75
70
1
2
3
4
8
7
6
5
V
= 6V
IN
I
SYNC/FCB
TH
47pF
RUN/SS
LTC1627
V
DR
C
SS
0.1µF
V
IN
V
= 8.4V
IN
2.8V*
V
V
IN
FB
TO 8.5V
L1 15µH
+
C
IN
V
3.3V
OUT
GND
SW
22µF
+
C
OUT
16V
100µF
249k
6.3V
1
10
100
1000
*V
OUT
CONNECTED TO V FOR 2.8V < V < 3.3V
IN IN
80.6k
OUTPUT CURRENT (mA)
1627 F01a
1627 F01b
Figure 1a. High Efficiency Step-Down Converter
Figure 1b. Efficiency vs Output Load Current
1
LTC1627
W W
U W
U W
U
ABSOLUTE AXI U RATI GS
PACKAGE/ORDER I FOR ATIO
(Note 1)
Input Supply Voltage ................................ –0.3V to 10V
Driver Supply Voltage (VIN – VDR) ........... –0.3V to 10V
ITH Voltage .................................................. –0.3V to 5V
Run/SS, VFB Voltages ................................ –0.3V to VIN
Sync/FCB Voltage ...................................... –0.3V to VIN
VDR Voltage (VIN ≤ 5V) ............................... –5V to 0.3V
P-Channel Switch Source Current (DC) .............. 800mA
N-Channel Switch Sink Current (DC) .................. 800mA
Peak SW Sink and Source Current ......................... 1.5A
Operating Ambient Temperature Range
Commercial ............................................ 0°C to 70°C
Industrial ........................................... –40°C to 85°C
Junction Temperature (Note 2)............................. 125°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
I
1
2
3
4
8
7
6
5
SYNC/FCB
TH
RUN/SS
V
DR
LTC1627CS8
LTC1627IS8
V
V
FB
IN
GND
SW
S8 PART MARKING
S8 PACKAGE
8-LEAD PLASTIC SO
1627
1627I
TJMAX = 125°C, θJA = 110°C/ W
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature
range, otherwise specifications are at TA = 25°C. VIN = 5V unless otherwise specified.
SYMBOL
PARAMETER
CONDITIONS
(Note 3)
MIN
TYP
20
MAX
60
UNITS
nA
I
Feedback Current
VFB
V
Regulated Feedback Voltage
∆Output Overvoltage Lockout
Reference Voltage Line Regulation
Output Voltage Load Regulation
(Note 3)
●
0.788
20
0.80
60
0.812
110
V
FB
∆V
∆V
∆V
= V
– V
FB
mV
%/V
OVL
FB
OVL
OVL
V
= 2.8V to 8.5V (Note 3)
0.002
0.01
IN
V
I
I
Sinking 2µA (Note 3)
Sourcing 2µA (Note 3)
0.5
–0.5
0.8
–0.8
%
%
LOADREG
TH
TH
I
Input DC Bias Current
Synchronized
(Note 4)
S
V
V
V
V
= 8.5V, V
= 3.3V, Frequency = 525kHz
OUT
450
200
15
µA
µA
µA
µA
IN
ITH
RUN/SS
RUN/SS
Burst Mode Operation
Shutdown
Shutdown
= 0V, V = 8.5V, V = Open
320
35
IN
SYNC/FCB
IN
= 0V, 2.65V < V < 8.5V
= 0V, V < 2.65V
6
IN
V
Run/SS Threshold
0.4
1.2
0.7
2.25
0.8
1.0
3.3
V
µA
V
RUN/SS
I
Soft-Start Current Source
Auxiliary Feedback Threshold
Auxiliary Feedback Current
Oscillator Frequency
V
V
V
= 0V
RUN/SS
RUN/SS
V
Ramping Negative
= 0V
0.730
0.5
0.860
2.5
SYNC/FCB
SYNC/FCB
OSC
SYNC/FCB
SYNC/FCB
I
f
1.5
µA
V
V
= 0.8V
= 0V
315
350
35
385
kHz
kHz
FB
FB
V
Undervoltage Lockout
V
V
Ramping Down from 3V (–40°C to 85°C)
Ramping Up from 0V (–40°C to 85°C)
2.3
2.4
2.50
2.65
2.65
2.85
V
V
UVLO
IN
IN
V
V
Ramping Down from 3V (0°C to 70°C)
Ramping Up from 0V (0°C to 70°C)
2.50
2.65
2.65
2.80
V
V
IN
IN
R
R
R
R
of P-Channel FET
of N-Channel FET
(V – V ) = 5V, I = 100mA
0.5
0.6
±10
0.7
0.8
Ω
Ω
PFET
NFET
LSW
DS(ON)
DS(ON)
IN
DR
SW
I
= –100mA
SW
I
SW Leakage
V
= 0V
±1000
nA
RUN/SS
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 3: The LTC1627 is tested in a feedback loop that servos V to the
FB
balance point for the error amplifier (V = 0.8V).
ITH
Note 2: T is calculated from the ambient temperature T and power
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
J
A
dissipation P according to the following formula:
D
T = T + (P • 110°C/W)
J
A
D
2
LTC1627
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Input Voltage
Efficiency vs Load Current
Efficiency vs Load Current
100
95
90
85
80
75
70
100
95
90
85
80
75
70
100
95
90
85
80
75
V
V
= 3.6V
IN
OUT
Burst Mode
OPERATION
= 2.5V
L = 15µH
= 0V
V
V
DR
= 0V
DR
I
= 100mA
LOAD
I
= 300mA
LOAD
V
DR
= –V
IN
SYNCHRONIZED
AT 525kHz
I
= 10mA
LOAD
FORCED
CONTINUOUS
V
V
= 3.6V
V
= 2.5V
IN
OUT
OUT
= 2.5V
L = 15µH
= 0V
L = 15µH
V
DR
Burst Mode OPERATION
Burst Mode OPERATION
1
10
100
1000
1
10
100
1000
0
2
4
6
8
10
OUTPUT CURRENT (mA)
OUTPUT CURRENT (mA)
INPUT VOLTAGE (V)
1627 G02
1627 G03
1627 G01
Undervoltage Lockout Threshold
vs Temperature
DC Supply Current*
vs Input Voltage
Efficiency vs Load Current
100
2.75
2.70
2.65
2.60
2.55
2.50
2.45
2.40
2.35
550
500
450
400
350
300
250
200
150
T = 25°C
J
OUT
V
= 2.8V
= 3.6V
IN
V
= 1.8V
95
90
85
80
75
70
SYNCHRONIZED AT 525kHz
V
IN
V
IN
RAMPING UP
V
V
= 7.2V
IN
V
IN
= 2.5V
OUT
RAMPING DOWN
L = 15µH
= 0V
Burst Mode OPERATION
V
DR
Burst Mode OPERATION
2.30
100
1
10
100
1000
–50 –25
0
25
50
125
2.5
3.5
4.5
5.5
8.5
75 100
6.5
7.5
OUTPUT CURRENT (mA)
TEMPERATURE (°C)
INPUT VOLTAGE (V)
1627 G04
1627 G05
1627 G06
*DOES NOT INCLUDE GATE CHARGE CURRENT
Reference Voltage
vs Temperature
Forced Continuous Threshold
Voltage vs Temperature
Supply Current in Shutdown
vs Input Voltage
22
20
18
16
14
12
10
8
0.808
0.806
0.804
0.802
0.800
0.798
0.796
0.794
0.792
0.808
0.806
0.804
0.802
0.800
0.798
0.796
0.794
0.792
V
= 0V
V
IN
= 5V
V
IN
= 5V
RUN/SS
T = 85°C
J
T = 25°C
J
T = –40°C
J
6
4
0.790
0.790
2.5
3.5
4.5
5.5
8.5
–50 –25
0
25
50
125
–50 –25
0
25
50
125
6.5
7.5
75 100
75 100
INPUT VOLTAGE (V)
TEMPERATURE (°C)
TEMPERATURE (°C)
1627 G07
1627 G08
1627 G09
3
LTC1627
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Oscillator Frequency
vs Temperature
Oscillator Frequency
vs Input Voltage
Maximum Output Load Current
vs Input Voltage
390
380
370
360
350
340
330
320
310
390
380
370
360
350
340
330
320
310
1100
1000
900
800
700
600
500
400
300
V
V
= 5V
IN
SYNC/FCB
V
= 0V
SYNC/FCB
V
= –V
IN
DR
= 0V
V
= 0V
DR
V
= 2.5V
OUT
L = 15µH
300
300
200
–50 –25
0
25
50
125
2.5
3.5
4.5
5.5
8.5
2.5
3.5
4.5
5.5
8.5
75 100
6.5
7.5
6.5
7.5
TEMPERATURE (°C)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
1627 G10
1627 G11
1627 G12
Switch Leakage Current
vs Temperature
Switch Resistance
vs Temperature
Switch Resistance
vs Input Voltage
1800
1600
1400
1200
1000
800
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
V
= 0V
V
V
= 8.4V
= 0V
V
V
= 5V
= 0V
DR
IN
DR
IN
DR
SYNCHRONOUS
SWITCH
SYNCHRONOUS SWITCH
MAIN SWITCH
MAIN
SWITCH
SYNCHRONOUS
SWITCH
600
400
MAIN
SWITCH
200
0
0
0
–50 –25
0
25
50
125
–50 –25
0
25
50
125
2.5
3.5
4.5
5.5
6.5
7.5
8.5
75 100
75 100
TEMPERATURE (°C)
TEMPERATURE (°C)
INPUT VOLTAGE (V)
1627 G13
1627 G14
1627 G15
Burst Mode Operation
Load Step Transient Response
Load Step Transient Response
ITH
0.5V/DIV
ITH
0.5V/DIV
SW
5V/DIV
VOUT
50mV/DIV
AC COUPLED
VOUT
50mV/DIV
AC COUPLED
VOUT
20mV/DIV
AC COUPLED
ILOAD
500mA/DIV
ILOAD
500mA/DIV
ILOAD
200mA/DIV
1627 G16
1627 G17
1627 G18
25µs/DIV
25µs/DIV
10µs/DIV
V
IN = 5V
VIN = 5V
VOUT = 3.3V
L = 15µH
CIN = 22µF
COUT = 100µF
ILOAD = 0mA TO 500mA
FORCED CONTINUOUS MODE
V
IN = 5V
VOUT = 3.3V
L = 15µH
VOUT = 3.3V
L = 15µH
CIN = 22µF
COUT = 100µF
ILOAD = 0mA TO 500mA
Burst Mode OPERATION
CIN = 22µF
COUT = 100µF
ILOAD = 50mA
4
LTC1627
U
U
U
PI FU CTIO S
ITH (Pin 1): Error Amplifier Compensation Point. The
current comparator threshold increases with this control
voltage. Nominal voltage range for this pin is 0V to 1.2V.
VIN (Pin 6): Main Supply Pin. Must be closely decoupled
to GND, Pin 4.
VDR (Pin 7): Top Driver Return Pin. This pin can be
bootstrapped to go below ground to improve efficiency at
low VIN (see Applications Information).
RUN/SS (Pin 2): Combination of Soft-Start and Run
Control Inputs. A capacitor to ground at this pin sets the
ramptimetofullcurrentoutput. Thetimeisapproximately
0.5s/µF. Forcing this pin below 0.4V shuts down all the
circuitry.
SYNC/FCB (Pin 8): Multifunction Pin. This pin performs
three functions: 1) secondary winding feedback input, 2)
external clock synchronization and 3) Burst Mode opera-
tion or forced continuous mode select. For secondary
winding applications connect a resistive divider from the
secondary output. To synchronize with an external clock
apply a TTL/CMOS compatible clock with a frequency
between385kHzand525kHz. ToselectBurstModeopera-
tion, float the pin or tie it to VIN. Grounding Pin 8 forces
continuous operation (see Applications Information).
VFB (Pin 3): Feedback Pin. Receives the feedback voltage
from an external resistive divider across the output.
GND (Pin 4): Ground Pin.
SW (Pin 5): Switch Node Connection to Inductor. This pin
connects to the drains of the internal main and synchro-
nous power MOSFET switches.
U
U
W
FU CTIO AL DIAGRA
BURST
DEFEAT
X
Y = “0” ONLY WHEN X IS A CONSTANT “1”
V
IN
V
IN
Y
V
IN
1.5µA
SLOPE
COMP
SYNC/FCB
8
OSC
0.4V
–
+
0.6V
V
FB
6
V
IN
3
–
+
FREQ
SHIFT
EN
–
+
V
IN
SLEEP
6Ω
–
+
I
COMP
+
0.8V
0.12V
EA
–
BURST
0.8V
REF
2.25µA
I
TH
1
V
IN
S
R
Q
Q
SWITCHING
LOGIC
RUN/SOFT
START
RUN/SS
2
7
V
DR
UVLO
TRIP = 2.5V
AND
ANTI-
SHOOT-THRU
BLANKING
CIRCUIT
+
OVDET
–
0.86V
+
SW
5
4
SHUTDOWN
I
RCMP
–
GND
–
0.8V
FCB
1627 BD
+
5
LTC1627
U
OPERATIO
(Refer to Functional Diagram)
Main Control Loop
output load exceeds 100mA. The threshold voltage be-
tween Burst Mode operation and forced continuous mode
is 0.8V. This can be used to assist in secondary winding
regulationasdescribedinAuxiliaryWindingControlUsing
SYNC/FCB Pin in the Applications Information section.
The LTC1627 uses a constant frequency, current mode
step-down architecture. Both the main and synchronous
switches, consisting of top P-channel and bottom
N-channel power MOSFETs, are internal. During normal
operation, the internal top power MOSFET is turned on
each cycle when the oscillator sets the RS latch, and
turned off when the current comparator, ICOMP, resets the
RS latch. The peak inductor current at which ICOMP resets
the RS latch is controlled by the voltage on the ITH pin,
which is the output of error amplifier EA. The VFB pin,
described in the Pin Functions section, allows EA to
receive an output feedback voltage from an external resis-
tive divider. When the load current increases, it causes a
slight decrease in the feedback voltage relative to the 0.8V
reference, which, in turn, causes the ITH voltage to in-
crease until the average inductor current matches the new
load current. While the top MOSFET is off, the bottom
MOSFET is turned on until either the inductor current
starts to reverse as indicated by the current reversal
comparator IRCMP, or the beginning of the next cycle.
When the converter is in Burst Mode operation, the peak
current of the inductor is set to approximately 200mA,
even though the voltage at the ITH pin indicates a lower
value. The voltage at the ITH pin drops when the inductor’s
average current is greater than the load requirement. As
theITH voltagedropsbelow0.12V, theBURSTcomparator
trips, causing the internal sleep line to go high and turn off
both power MOSFETs.
In sleep mode, both power MOSFETs are held off and the
internal circuitry is partially turned off, reducing the quies-
cent current to 200µA. The load current is now being
supplied from the output capacitor. When the output
voltage drops, causing ITH to rise above 0.22V, the top
MOSFET is again turned on and this process repeats.
Short-Circuit Protection
The main control loop is shut down by pulling the RUN/SS
pin low. Releasing RUN/SS allows an internal 2.25µA
current source to charge soft-start capacitor CSS. When
CSS reaches0.7V,themaincontrolloopisenabledwiththe
Whentheoutputisshortedtoground, thefrequencyofthe
oscillator is reduced to about 35kHz, 1/10 the nominal
frequency. This frequency foldback ensures that the
inductor current has more time to decay, thereby prevent-
ing runaway. The oscillator’s frequency will progressively
increase to 350kHz (or the synchronized frequency) when
I
TH voltage clamped at approximately 5% of its maximum
value. As CSS continues to charge, ITH is gradually
released, allowing normal operation to resume.
V
FB rises above 0.3V.
Comparator OVDET guards against transient overshoots
>7.5% by turning the main switch off and turning the
synchronous switch on. With the synchronous switch
turned on, the output is crowbarred. This may cause a
large amount of current to flow from VIN if the main switch
is damaged, blowing the system fuse.
Frequency Synchronization
The LTC1627 can be synchronized with an external
TTL/CMOS compatible clock signal. The frequency range
of this signal must be from 385kHz to 525kHz. Do not
attempt to synchronize the LTC1627 below 385kHz as this
may cause abnormal operation and an undesired fre-
quency spectrum. The top MOSFET turn-on follows the
rising edge of the external source.
Burst Mode Operation
The LTC1627 is capable of Burst Mode operation in which
the internal power MOSFETs operate intermittently based
on load demand. To enable Burst Mode operation, simply
allow the SYNC/FCB pin to float or connect it to a logic
high. To disable Burst Mode operation and enable forced
continuous mode, connect the SYNC/FCB pin to GND. In
this mode, the efficiency is lowest at light loads, but
becomes comparable to Burst Mode operation when the
When the LTC1627 is clocked by an external source, Burst
Mode operation is disabled; the LTC1627 then operates in
PWM pulse skipping mode. In this mode, when the output
loadisverylow,currentcomparatorICOMP remainstripped
for more than one cycle and forces the main switch to stay
off for the same number of cycles. Increasing the output
6
LTC1627
U
OPERATIO
load slightly allows constant frequency PWM operation
to resume.
V
DR
C1
0.1µF
V
V
V
< 4.5V
IN
LTC1627
IN
D1
D2
Frequency synchronization is inhibited when the feedback
voltage VFB is below 0.6V. This prevents the external clock
from interfering with the frequency foldback for short-
circuit protection.
L1
SW
OUT
C2
0.1µF
+
C
OUT
100µF
1627 F02
Dropout Operation
Figure 2. Using a Charge Pump to Bias VDR
When the input supply voltage decreases toward the out-
put voltage, the duty cycle increases toward the maximum
on-time. Further reduction of the supply voltage forces the
main switch to remain on for more than one cycle until it
reaches 100% duty cycle. The output voltage will then be
determined by the input voltage minus the voltage drop
across the P-channel MOSFET and the inductor.
the charge pump at VIN ≥ 4.5V is not recommended to
ensure that (VIN – VDR) does not exceed its absolute
maximum voltage.
When VIN decreases to a voltage close to VOUT, the loop
may enter dropout and attempt to turn on the P-channel
MOSFET continuously. When the VDR charge pump is
enabled, a dropout detector counts the number of oscilla-
tor cycles that the P-channel MOSFET remains on, and
periodically forces a brief off period to allow C1 to
recharge.100%dutycycleisallowedwhenVDR isgrounded.
InBurstModeoperationorpulseskippingmodeoperation
(externally synchronized) with the output lightly loaded,
the LTC1627 transitions through continuous mode as it
enters dropout.
Slope Compensation and Inductor Peak Current
Undervoltage Lockout
Slope compensation provides stability by preventing
subharmonic oscillations. It works by internally adding a
ramptotheinductorcurrentsignalatdutycyclesinexcess
of 40%. As a result, the maximum inductor peak current
is lower for VOUT/VIN > 0.4 than when VOUT/VIN < 0.4. See
the inductor peak current as a function of duty cycle graph
in Figure 3. The worst-case peak current reduction occurs
withtheoscillatorsynchronizedatitsminimumfrequency,
i.e., to a clock just above the oscillator free-running
AprecisionundervoltagelockoutshutsdowntheLTC1627
when VIN drops below 2.5V, making it ideal for single
lithium-ion battery applications. In lockout, the LTC1627
draws only several microamperes, which is low enough to
preventdeepdischargeandpossibledamagetothelithium-
ion battery nearing its end of charge. A 150mV hysteresis
ensures reliable operation with noisy supplies.
Low Supply Operation
TheLTC1627isdesignedtooperatedownto2.65Vsupply
voltage. At this voltage the converter is most likely to be
running at high duty cycles or in dropout where the main
switch is on continuously. Hence, the I2R loss is due
mainly to the RDS(ON) of the P-channel MOSFET. See
Efficiency Considerations in the Applications Information
section.
950
900
850
800
750
700
650
600
550
500
WITHOUT
EXTERNAL
CLOCK SYNC
WORST CASE
EXTERNAL
CLOCK SYNC
When VIN is low (<4.5V) the RDS(ON) of the P-channel
MOSFET can be lowered by driving its gate below ground.
The top P-channel MOSFET driver makes use of a floating
return pin, VDR, to allow biasing below GND. A simple
charge pump bootstrapped to the SW pin realizes a
negativevoltageattheVDR pinasshowninFigure2. Using
V
IN
= 5V
0
10 20 30 40
70 80 90 100
50 60
DUTY CYCLE (%)
1627 F03
Figure 3. Maximum Inductor Peak Current vs Duty Cycle
7
LTC1627
W U U
U
APPLICATIO S I FOR ATIO
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
more turns of wire and therefore copper losses will
increase.
frequency. The actual reduction in average current is less
than for peak current.
The basic LTC1627 application circuit is shown in Figure
1. External component selection is driven by the load
requirementandbeginswiththeselectionofLfollowedby
CIN and COUT.
Ferritedesignshaveverylowcorelossesandarepreferred
at high switching frequencies, so design goals can con-
centrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that induc-
tance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Inductor Value Calculation
The inductor selection will depend on the operating fre-
quency of the LTC1627. The internal preset frequency is
350kHz, but can be externally synchronized up to 525kHz.
The operating frequency and inductor selection are inter-
related in that higher operating frequencies allow the use
of smaller inductor and capacitor values. However, oper-
ating at a higher frequency generally results in lower
efficiency because of internal gate charge losses.
Kool Mµ (from Magnetics, Inc.) is a very good, low loss
corematerialfortoroidswitha“soft”saturationcharacter-
istic. Molypermalloy is slightly more efficient at high
(>200kHz) switching frequencies but quite a bit more
expensive. Toroids are very space efficient, especially
when you can use several layers of wire, while inductors
wound on bobbins are generally easier to surface mount.
New designs for surface mount are available from
Coiltronics, Coilcraft and Sumida.
Theinductorvaluehasadirecteffectonripplecurrent.The
ripple current ∆IL decreases with higher inductance or
frequency and increases with higher VIN or VOUT
.
1
V
OUT
∆I =
V
1−
L
OUT
(1)
V
f L
( )( )
IN
CIN and COUT Selection
Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL = 0.4(IMAX).
Incontinuousmode,thesourcecurrentofthetopMOSFET
is a square wave of duty cycle VOUT/VIN. To prevent large
voltage transients, a low ESR input capacitor sized for the
maximum RMS current must be used. The maximum
RMS capacitor current is given by:
The inductor value also has an effect on Burst Mode
operation. The transition to low current operation begins
when the inductor current peaks fall to approximately
200mA. Lower inductor values (higher ∆IL) will cause this
to occur at lower load currents, which can cause a dip in
efficiency in the upper range of low current operation. In
Burst Mode operation, lower inductance values will cause
the burst frequency to increase.
1/2
]
V
V − V
OUT
(
)
OUT IN
[
C required I
I
IN
RMS MAX
V
IN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is com-
monlyusedfordesignbecauseevensignificantdeviations
donotoffermuchrelief.Notethatcapacitormanufacturer’s
ripplecurrentratingsareoftenbasedon2000hoursoflife.
This makes it advisable to further derate the capacitor, or
choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to meet
size or height requirements in the design. Always consult the
manufacturer if there is any question.
Inductor Core Selection
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot
affordthecorelossfoundinlowcostpowderedironcores,
forcing the use of more expensive ferrite, molypermalloy,
orKoolMµ® cores. Actualcorelossisindependentofcore
Kool Mµ is a registered trademark of Magnetics, Inc.
8
LTC1627
W U U
APPLICATIO S I FOR ATIO
TheselectionofCOUT isdrivenbytherequiredeffectiveseries
resistance (ESR). Typically, once the ESR requirement is
satisfied, the capacitance is adequate for filtering. The output
ripple ∆VOUT is determined by:
U
0.8V ≤ V
≤ 8.5V
OUT
R2
V
FB
LTC1627
GND
R1
1
∆VOUT ∆IL ESR +
8fCOUT
1627 F04
where f = operating frequency, COUT = output capacitance
and ∆IL = ripple current in the inductor. The output ripple
is highest at maximum input voltage since ∆IL increases
with input voltage. For the LTC1627, the general rule for
proper operation is:
Figure 4. Setting the LTC1627 Output Voltage
Run/Soft-Start Function
The RUN/SS pin is a dual purpose pin that provides the
soft-startfunctionandameanstoshutdowntheLTC1627.
Soft-start reduces surge currents from VIN by gradually
increasing the internal current limit. Power supply
sequencing can also be accomplished using this pin.
C
OUT required ESR < 0.25Ω
Manufacturers such as Nichicon, United Chemicon and
Sanyoshouldbeconsideredforhighperformancethrough-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR/size
ratio of any aluminum electrolytic at a somewhat higher
price. Once the ESR requirement for COUT has been met,
the RMS current rating generally far exceeds the
IRIPPLE(P-P) requirement.
An internal 2.25µA current source charges up an external
capacitor CSS. When the voltage on RUN/SS reaches 0.7V
the LTC1627 begins operating. As the voltage on RUN/SS
continues to ramp from 0.7V to 1.8V, the internal current
limit is also ramped at a proportional linear rate. The
current limit begins at 25mA (at VRUN/SS ≤ 0.7V) and ends
at the Figure 3 value (VRUN/SS ≈ 1.8V). The output current
thus ramps up slowly, charging the output capacitor. If
RUN/SS has been pulled all the way to ground, there will
be a delay before the current starts increasing and is given
by:
In surface mount applications multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
surfacemountconfigurations. Inthecaseoftantalum, itis
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS series of surface mount tantalum, available in case
heights ranging from 2mm to 4mm. Other capacitor types
include Sanyo POSCAP, KEMET T510 and T495 series,
Nichicon PL series and Sprague 593D and 595D series.
Consult the manufacturer for other specific recommenda-
tions.
0.7C
2.25µA
SS
t
=
DELAY
Pulling the RUN/SS pin below 0.4V puts the LTC1627 into
alowquiescentcurrentshutdown(IQ <15µA).Thispincan
be driven directly from logic as shown in Figure 5. Diode
D1 in Figure 5 reduces the start delay but allows CSS to
ramp up slowly providing the soft-start function. This
diode can be deleted if soft-start is not needed.
Output Voltage Programming
The output voltage is set by a resistive divider according
to the following formula:
3.3V OR 5V
RUN/SS
RUN/SS
D1
C
R2
R1
C
SS
SS
V
= 0.8V 1+
OUT
(2)
1627 F05
The external resistive divider is connected to the output,
allowing remote voltage sensing as shown in Figure 4.
Figure 5. RUN/SS Pin Interfacing
9
LTC1627
W U U
U
APPLICATIO S I FOR ATIO
Efficiency = 100% – (L1 + L2 + L3 + ...)
Auxiliary Winding Control Using SYNC/FCB Pin
whereL1, L2, etc. aretheindividuallossesasapercentage
of input power.
The SYNC/FCB pin can be used as a secondary feedback
input to provide a means of regulating a flyback winding
output. When this pin drops below its ground referenced
0.8V threshold, continuous mode operation is forced. In
continuous mode, the main and synchronous MOSFETs
are switched continuously regardless of the load on the
main output.
Although all dissipative elements in the circuit produce
losses, two main sources usually account for most of the
losses in LTC1627 circuits: VIN quiescent current and I2R
losses.
1. The VIN quiescent current is due to two components:
the DC bias current as given in the electrical character-
istics and the internal main switch and synchronous
switch gate charge currents. The gate charge current
results from switching the gate capacitance of the
internal power MOSFET switches. Each time the gate is
switched from high to low to high again, a packet of
charge dQ moves from VIN to ground. The resulting
dQ/dt is the current out of VIN that is typically larger
thantheDCbiascurrent. Incontinuousmode, IGATECHG
= f(QT + QB) where QT and QB are the gate charges of
the internal top and bottom switches. Both the DC bias
and gate charge losses are proportional to VIN and thus
their effects will be more pronounced at higher supply
voltages.
2. I2R losses are calculated from the resistances of the
internal switches RSW and external inductor RL. In
continuous mode the average output current flowing
through inductor L is “chopped” between the main
switch and the synchronous switch. Thus, the series
resistance looking into SW pin from L is a function of
both top and bottom MOSFET RDS(ON) and the duty
cycle (DC) as follows:
Synchronous switching removes the normal limitation
that power must be drawn from the inductor primary
winding in order to extract power from auxiliary windings.
With continuous synchronous operation power can be
drawn from the auxiliary windings without regard to the
primary output load.
Thesecondaryoutputvoltageissetbytheturnsratioofthe
transformerinconjunctionwithapairofexternalresistors
returned to the SYNC/FCB pin as shown in Figure 6. The
secondary regulated voltage VSEC in Figure 6 is given by:
R4
R3
V
N +1 V
− V
> 0.8V 1+
(
)(
)
SEC
OUT
DIODE
where N is the turns ratio of the transformer, VOUT is the
main output voltage sensed by VFB and VDIODE is the
voltage drop across the Schottky diode.
R4
V
SEC
SYNC/FCB
+
R3
L1
1:N
1µF
LTC1627
V
OUT
SW
+
RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC)
C
OUT
1627 F06
The RDS(ON) for both the top and bottom MOSFETs can
be obtained from the Typical Performance Characteris-
tics curves. Thus, to obtain I2R losses, simply add RSW
to RL and multiply by the square of the average output
current.
Figure 6. Secondary Output Loop Connection
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
oftenusefultoanalyzeindividuallossestodeterminewhat
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Other losses including CIN and COUT ESR dissipative losses,
MOSFETswitchinglossesandinductorcorelossesgenerally
account for less than 2% total additional loss.
10
LTC1627
W U U
U
APPLICATIO S I FOR ATIO
Checking Transient Response
the load rise time is limited to approximately (25 • CLOAD).
Thus, a 10µF capacitor would require a 250µs rise time,
limiting the charging current to about 130mA.
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to (∆ILOAD • ESR), where ESR is the effective series
resistance of COUT. ∆ILOAD also begins to charge or dis-
chargeCOUT, whichgeneratesafeedbackerrorsignal. The
regulator loop then acts to return VOUT to its steady-state
value.DuringthisrecoverytimeVOUT canbemonitoredfor
overshoot or ringing that would indicate a stability prob-
lem. The internal compensation provides adequate com-
pensationformostapplications. Butifadditionalcompen-
sation is required, the ITH pin can be used for external
compensation as shown in Figure 7.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1627. These items are also illustrated graphically in
the layout diagram of Figure 7. Check the following in your
layout:
1. Are the signal and power grounds segregated? The
LTC1627 signal ground consists of the resistive
divider, the optional compensation network (RC and
CC1), CSS and CC2. The power ground consists of the
(–) plate of CIN, the (–) plate of COUT and Pin 4 of the
LTC1627. The power ground traces should be kept
short, direct and wide. The signal ground and power
ground should converge to a common node in a star-
ground configuration.
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
dischargedbypasscapacitorsareeffectivelyputinparallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
2. Does the VFB pin connect directly to the feedback
resistors? The resistive divider R1/R2 must be con-
nectedbetweenthe(+)plateofCOUT andsignalground.
C
C2
R
C
C
1
2
3
4
C1
8
7
6
5
OPTIONAL
I
SYNC/FCB
TH
OPTIONAL
C
SS
RUN/SS
LTC1627
V
DR
C
V
V
V
FB
IN
+
L1
GND
SW
+
+
C
IN
D1
D2
R2
V
+
OUT
V
IN
C
OUT
R1
C
B
–
–
BOLD LINES INDICATE
HIGH CURRENT PATHS
1627 F07
Figure 7. LTC1627 Layout Diagram
11
LTC1627
W U U
U
APPLICATIO S I FOR ATIO
3. Does the (+) plate of CIN connect to VIN as closely as
possible? This capacitor provides the AC current to the
internal power MOSFETs.
2.5V
2.5V
4.2V
L =
1 −
= 14.5µH
350kHz 200mA
(
)(
)
A 15µH inductor works well for this application. For good
efficiency choose a 1A inductor with less than 0.25Ω
series resistance.
4. KeeptheswitchingnodeSWawayfromsensitivesmall-
signal nodes.
Design Example
CIN will require an RMS current rating of at least 0.25A at
temperature and COUT will require an ESR of less than
0.25Ω. In most applications, the requirements for these
capacitors are fairly similar.
As a design example, assume the LTC1627 is used in a
singlelithium-ionbattery-poweredcellularphoneapplica-
tion. The VIN will be operating from a maximum of 4.2V
down to about 2.7V. The load current requirement is a
maximumof0.5Abutmostofthetimeitwillbeonstandby
mode, requiring only 2mA. Efficiency at both low and high
load currents is important. Output voltage is 2.5V. With
this information we can calculate L using equation (1),
Forthefeedbackresistors,chooseR1=80.6k.R2canthen
be calculated from equation (2) to be:
V
0.8
OUT
R2 =
−1 • R1= 171k; use 169k
Figure 8 shows the complete circuit along with its effi-
ciency curve.
1
V
OUT
L =
V
1 −
OUT
(3)
V
f ∆I
( )(
IN
)
L
Substituting VOUT = 2.5V, VIN = 4.2V, ∆IL = 200mA and
f = 350kHz in equation (3) gives:
C
ITH
47pF
1
2
3
4
8
7
6
5
I
SYNC/FCB
TH
100
RUN/SS
LTC1627
V
V
95
90
85
80
75
70
65
60
55
50
45
V
V
= 3.6V
= 4.2V
DR
IN
IN
C1
C
V
IN
SS
0.1µF
0.1µF
V
2.8V TO
4.5V
FB
IN
15µH*
R2
V
2.5V
0.5A
OUT
GND
SW
††
IN
+
C
BAT54S**
D1
22µF
169k
1%
16V
†
+
C
OUT
100µF
6.3V
D2
R1
80.6k
1%
C2
0.1µF
V
= 2.5V
OUT
1
10
100
1000
* SUMIDA CD54-150
OUTPUT CURRENT (mA)
1627 F08a
** ZETEX BAT54S
†
1627 F08b
AVX TPSC107M006R0150
AVX TPSC226M016R0375
††
Figure 8. Single Lithium-Ion to 2.5V/0.5A Regulator
12
LTC1627
U
TYPICAL APPLICATIO S
5V Input to 3.3V/0.5A Regulator
C
ITH
47pF
* SUMIDA CD54-150
1
2
3
4
8
7
6
5
**
I
AVX TPSC107M006R0150
AVX TPSC226M016R0375
SYNC/FCB
TH
***
RUN/SS
LTC1627
V
V
DR
C
SS
V
IN
V
IN
= 5V
0.1µF
FB
15µH*
V
3.3V
0.5A
OUT
GND
SW
R2
C
***
22µF
+
IN
249k
1%
+
C
**
100µF
OUT
16V
R1
80.6k
1%
6.3V
1627 TA03
Double Lithium-Ion to 5V/0.5A Low Dropout Regulator
C
ITH
47pF
* SUMIDA CD54-330
1
2
3
4
8
7
6
5
**
I
AVX TPSD107M010R0100
AVX TPSC226M016R0375
SYNC/FCB
TH
***
RUN/SS
LTC1627
V
V
DR
C
SS
0.1µF
V
IN
V
IN
≤ 8.4V
FB
33µH*
V
OUT
GND
SW
5V
R2
C
***
22µF
+
IN
0.5A
422k
1%
+
C
**
100µF
OUT
16V
R1
80.6k
1%
10V
1627 TA04
13
LTC1627
U
TYPICAL APPLICATIO S
3.3V Input to 2.5V/0.5A Regulator
C
ITH
47pF
1
2
3
4
8
I
SYNC/FCB
TH
7
6
5
RUN/SS
LTC1627
V
V
DR
C1
C
SS
0.1µF
V
0.1µF
V
= 3.3V
FB
IN
IN
10µH*
R2
V
2.5V
0.5A
OUT
GND
SW
††
+
C
IN
BAT54S**
D1
22µF
169k
1%
16V
†
+
C
OUT
100µF
6.3V
D2
R1
80.6k
1%
C2
0.1µF
* SUMIDA CD54-100
1627 TA05
** ZETEX BAT54S
†
AVX TPSC107M006R0150
AVX TPSC226M016R0375
††
Single Lithium-Ion to 1.8V/0.3A Regulator
C
ITH
47pF
* SUMIDA CD54-150
1
2
3
4
8
7
6
5
**
I
AVX TPSC107M006R0150
AVX TPSC226M016R0375
SYNC/FCB
TH
***
RUN/SS
LTC1627
V
V
DR
C
SS
V
IN
V
IN
≤ 4.2V
0.1µF
FB
15µH*
V
1.8V
0.3A
OUT
GND
SW
R2
C
***
22µF
+
IN
100k
1%
+
C
**
100µF
OUT
16V
R1
80.6k
1%
6.3V
1627 TA01
14
LTC1627
U
TYPICAL APPLICATIO S
Double Lithium-Ion to 2.5V/0.5A Regulator
C
ITH
47pF
* SUMIDA CD54-250
1
2
3
4
8
7
6
5
**
I
AVX TPSC107M006R0150
AVX TPSC226M016R0375
SYNC/FCB
TH
***
RUN/SS
LTC1627
V
V
DR
C
SS
V
IN
V
IN
≤ 8.4V
0.1µF
FB
25µH*
V
2.5V
0.5A
OUT
GND
SW
R2
C
***
22µF
+
IN
169k
1%
+
C
**
100µF
OUT
16V
R1
80.6k
1%
6.3V
1627 TA01
U
PACKAGE DESCRIPTIO
Dimensions in inches (millimeters) unless otherwise noted.
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
7
5
8
6
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
3
4
2
0.010 – 0.020
(0.254 – 0.508)
× 45°
0.053 – 0.069
(1.346 – 1.752)
0.004 – 0.010
(0.101 – 0.254)
0.008 – 0.010
(0.203 – 0.254)
0°– 8° TYP
0.016 – 0.050
(0.406 – 1.270)
0.050
(1.270)
BSC
0.014 – 0.019
(0.355 – 0.483)
TYP
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
SO8 1298
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
15
LTC1627
U
TYPICAL APPLICATIO S
Dual Output 1.8V/300mA and 3.3V/100mA Application
†††
V
SEC
C
R3
249k
1%
ITH
3.3V
47pF
100mA
1
2
3
4
8
7
6
5
+
I
SYNC/FCB
TH
***22µF
R4
80.6k
1%
††
D2
6.3V
ZENER
1.8V
RUN/SS
LTC1627
V
V
DR
D1
†
25µH MBR0520LT1
C
SS
0.1µF
V
1.8V
0.3A
V
C
≤ 8.5V
V
IN
OUT
1:1
FB
IN
GND
SW
+
*
+
IN
C
**
OUT
R2
100k
1%
22µF
100µF
16V
6.3V
†
††
†††
R1
80.6k
1%
* AVX TPSC226M016R0375
** AVX TPSC107M006R0150
*** AVX TAJA226M006R
COILTRONICS CTX25-1
MMSZ4678T1
A 10mA MIN LOAD CURRENT
IS RECOMMENDED
1627 TA02
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1174/LTC1174-3.3
LTC1174-5
High Efficiency Step-Down and Inverting DC/DC Converters
Monolithic Switching Regulators, I
Burst Mode Operation
to 450mA,
OUT
LTC1265
LT®1375/LT1376
1.2A, High Efficiency Step-Down DC/DC Converter
1.5A, 500kHz Step-Down Switching Regulators
Constant Off-Time, Monolithic, Burst Mode Operation
High Frequency, Small Inductor, High Efficiency
LTC1436A/LTC1436A-PLL High Efficiency, Low Noise, Synchronous Step-Down Converters 24-Pin Narrow SSOP, Auxillary Output
LTC1474/LTC1475
LTC1504A
Low Quiescent Current Step-Down DC/DC Converters
Monolithic Synchronous Step-Down Switching Regulator
Monolithic, I
to 250mA, I = 10µA, 8-Pin MSOP
OUT Q
Low Cost, Voltage Mode I
to 500mA,
OUT
V
from 4V to 10V
IN
LTC1622
LTC1626
Low Input Voltage Current Mode Step-Down DC/DC Controller
Low Voltage, High Efficiency Step-Down DC/DC Converter
High Frequency, High Efficiency, 8-Pin MSOP
Monolithic, Constant Off-Time, I to 600mA,
OUT
Low Supply Voltage Range: 2.5V to 6V
LTC1701
LTC1707
Monolithic Current Mode Step-Down Switching Regulator
Monolithic Synchronous Step-Down Switching Regulator
Constant Off-Time, I to 500mA, 1MHz Operation,
OUT
V
from 2.5V to 5.5V
IN
1.19V V
Pin, Constant Frequency, I
to 600mA,
REF
OUT
V
from 2.65V to 8.5V
IN
LTC1735
LTC1772
High Efficiency Step-Down Converter
16-Pin SO and SSOP, V Up to 36V, Fault Protection
IN
Low Input Voltage Current Mode Step-Down DC/DC Controller
550kHz, 6-Pin SOT-23, I
Up to 5A,
OUT
V
from 2.2V to 10V
IN
LTC1877
LTC1878
High Efficiency Monolithic Step-Down Regulator
High Efficiency Monolithic Step-Down Regulator
550kHz, MS8, V Up to 12V, I = 10µA, I
to 600mA
IN
Q
OUT
550kHz, MS8, V Up to 7V, I = 10µA, I to 600mA
OUT
IN
Q
1627fa LT/TP 0600 2K REV A • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1998
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
16
●
●
(408)432-1900 FAX:(408)434-0507 www.linear-tech.com
相关型号：
LTC1627IS8#TR
LTC1627 - Monolithic Synchronous Step-Down Switching Regulator; Package: SO; Pins: 8; Temperature Range: -40°C to 85°C
Linear
LTC1627IS8#TRPBF
LTC1627 - Monolithic Synchronous Step-Down Switching Regulator; Package: SO; Pins: 8; Temperature Range: -40°C to 85°C
Linear
©2020 ICPDF网 联系我们和版权申明