LTC1709EG#TR [Linear]
LTC1709 - 2-Phase, 5-Bit Adjustable,High Efficiency, Synchronous Step-Down Switching Regulator; Package: SSOP; Pins: 36; Temperature Range: -40°C to 85°C;型号: | LTC1709EG#TR |
厂家: | Linear |
描述: | LTC1709 - 2-Phase, 5-Bit Adjustable,High Efficiency, Synchronous Step-Down Switching Regulator; Package: SSOP; Pins: 36; Temperature Range: -40°C to 85°C 开关 光电二极管 |
文件: | 总28页 (文件大小:291K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
LTC1709
2-Phase, 5-Bit Adjustable,
High Efficiency, Synchronous Step-Down
Switching Regulator
U
DESCRIPTIO
FEATURES
The LTC®1709 is a 2-phase, VID programmable, synchro-
nous step-down switching regulator controller that drives
all N-channel external power MOSFET stages in a fixed
frequency architecture. The 2-phase controller drives its
two output stages out of phase at frequencies up to
300kHz to minimize the RMS ripple currents in both input
and output capacitors. The 2-phase technique effectively
multiplies the fundamental frequency by two, improving
transient response while operating each channel at a
optimum frequency for efficiency. Thermal design is also
simplified.
■
Two Ouput Stages Operate Antiphase Reducing
Input Capacitance and Power Supply Noise
5-Bit VID Control (VRM 8.4 Compliant)
VOUT: 1.3V to 3.5V in 50mV/100mV Steps
Current Mode Control Ensures Current Sharing
True Remote Sensing Differential Amplifier
OPTI-LOOPTM Compensation Minimizes COUT
Programmable Fixed Frequency: 150kHz to
300kHz—Effective 300kHz to 600kHz Switching
Frequency
■
■
■
■
■
■
■
■
■
■
■
■
■
±1% Output Voltage Accuracy
Wide VIN Range: 4V to 36V Operation
Adjustable Soft-Start Current Ramping
Internal Current Foldback
An internal differential amplifier provides true remote
sensing of the regulated supply’s positive and negative
output terminals as required in high current applications.
Short-Circuit Shutdown Timer with Defeat Option
Overvoltage Soft-Latch Eliminates Nuisance Trips
Low Shutdown Current: 20µA
The RUN/SS pin provides soft-start and optional timed,
short-circuit shutdown. Current foldback limits MOSFET
dissipatonduringshort-circuitconditionswhenovercurrent
latchoff is disabled. OPTI-LOOP compensation allows the
transient response to be optimized for a wide range of
output capacitors and ESR values.
Small 36-Lead Narrow (0.209") SSOP Package
U
APPLICATIO S
■
Desktop Computers
■
Internet/Network Servers
■
Large Memory Arrays
■
DC Power Distribution Systems
, LTC and LT are registered trademarks of Linear Technology Corporation.
OPTI-LOOP is a trademark of Linear Technology Corporation.
■
Battery Chargers
U
TYPICAL APPLICATIO
V
IN
5V TO 28V
Efficiency Curve
10µF ×4
35V
0.1µF
100
90
Q1
Q2
V
TG1
BOOST 1
SW1
V
= 5V
OUT
= 200kHz
IN
RUN/SS
S
IN
0.002Ω
V
= 1.6V
0.47µF
f
S
1µH
S
LTC1709
1.2nF
BG1
15k
I
PGND
TH
SGND
80
+
S
SENSE1
SENSE1
–
5 VID BITS VID0–VID4
Q3
Q4
TG2
BOOST2
SW2
70
60
50
S
0.002Ω
EAIN
V
OUT
0.47µF
1.3V TO 3.5V
40A
1µH
FBOUT
SENSEIN
S
S
S
BG2
V
V
V
INTV
DIFFOUT
CC
+
C
10µF
OUT
+
–
SENSE 2
SENSE 2
1000µF
4V
OS
0
5
10 15 20 25 30 35 40 45
LOAD CURRENTS (A)
+
–
OS
×2
1709 TA01a
Q1–Q4 2× FAIRCHILD FDS7760A OR SILICONIX Si4874
1709 TA01
Figure 1. High Current 2-Phase Step-Down Converter
1
LTC1709
W W
U W
U
W
U
ABSOLUTE AXI U RATI GS
(Note 1)
PACKAGE/ORDER I FOR ATIO
TOP VIEW
ORDER PART
NUMBER
Input Supply Voltage (VIN).........................36V to –0.3V
Topside Driver Voltages (BOOST 1, 2).......42V to –0.3V
Switch Voltage (SW1, 2) .............................36V to –5 V
SENSE 1+, SENSE 2+, SENSE 1–,
1
2
NC
36
35
34
33
32
31
30
29
28
27
26
25
24
23
22
21
20
19
RUNN/SS
+
TG1
SENSE 1
–
LTC1709EG
3
SW1
BOOST 1
SENSE 1
4
EAIN
PLLFLTR
PLLIN
SENSE 2– Voltages....................... (1.1)INTVCC to –0.3V
EAIN, VOS+, VOS–, EXTVCC, INTVCC, RUN/SS,
AMPMD, VBIAS, ATTENIN, ATTENOUT,
5
V
IN
6
BG1
7
EXTV
CC
NC
VID0–VID4, Voltages ...................................7V to –0.3V
Boosted Driver Voltage (BOOST-SW) ..........7V to –0.3V
PLLFLTR, PLLIN, VDIFFOUT Voltages .... INTVCC to –0.3V
8
INTV
CC
I
TH
9
PGND
BG2
SGND
10
11
12
13
14
15
16
17
18
V
DIFFOUT
BOOST 2
SW2
V
V
–
ITH Voltage................................................2.7V to –0.3V
OS
+
–
Peak Output Current <1µs(TGL1, 2; BG1, 2).............. 3A
INTVCC RMS Output Current................................ 50mA
Operating Ambient Temperature Range
OS
TG2
SENSE 2
SENSE 2
+
AMPMD
V
BIAS
ATTENOUT
ATTENIN
VID0
(Note 2) .................................................. –40°C to 85°C
Junction Temperature (Note 3)............................. 125°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
VID4
VID3
VID2
VID1
G PACKAGE
36-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 85°C/W
Consult factory for Industrial and Military grade parts.
ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Main Control Loop
V
V
Regulated Feedback Voltage
Maximum Current Sense Threshold
Feedback Current
(Note 4); I Voltage = 1.2V
●
●
0.792
62
0.800
75
0.808
88
V
mV
nA
EAIN
TH
–
V
= 5V
SENSEMAX
INEAIN
SENSE
I
(Note 4)
(Note 4)
–5
–50
V
Output Voltage Load Regulation
LOADREG
Measured in Servo Loop; ∆I Voltage: 1.2V to 0.7V
●
●
0.1
–0.1
0.5
–0.5
%
%
TH
Measured in Servo Loop; ∆I Voltage: 1.2V to 2V
TH
V
V
Reference Voltage Line Regulation
Output Overvoltage Threshold
Undervoltage Lockout
V
= 3.6V to 30V (Note 4)
IN
0.002
0.86
3.5
0.02
0.88
4
%/V
V
REFLNREG
OVL
Measured at V
●
0.84
3
EAIN
UVLO
V
Ramping Down
V
IN
TH
TH
g
g
Transconductance Amplifier g
I
I
= 1.2V; Sink/Source 5µA; (Note 4)
3
mmho
V/mV
m
m
Transconductance Amplifier Gain
= 1.2V; (g xZ ; No Ext Load); (Note 4)
1.5
mOL
m
L
I
Input DC Supply Current
Normal Mode
Shutdown
(Note 5)
Q
EXTV Tied to V ; V
= 5V
470
20
µA
µA
CC
OUT OUT
V
V
V
= 0V
40
RUN/SS
I
Soft-Start Charge Current
RUN/SS Pin ON Arming
= 1.9V
Rising
–0.5
1.0
–1.2
1.5
µA
RUN/SS
RUN/SS
RUN/SS
V
1.9
V
RUN/SS
2
LTC1709
The ● denotes the specifications which apply over the full operating
ELECTRICAL CHARACTERISTICS
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
Rising from 3V
MIN
TYP
4.1
2
MAX
4.5
4
UNITS
V
V
RUN/SS Pin Latchoff Arming
RUN/SS Discharge Current
V
RUN/SS
RUN/SSLO
SCL
I
Soft Short Condition V
= 0.5V;
0.5
µA
EAIN
V
= 4.5V
RUN/SS
I
I
Shutdown Latch Disable Current
Total Sense Pins Source Current
Maximum Duty Factor
V
= 0.5V
EAIN
1.6
–60
99.5
5
µA
µA
%
SDLHO
SENSE
Each Channel: V
In Dropout
(Note 6)
C
C
–
– = V + + = 0V
SENSE1 , 2
–85
98
SENSE1 , 2
DF
MAX
Top Gate Transition Time:
Rise Time
Fall Time
TG1, 2 t
TG1, 2 t
= 3300pF
= 3300pF
30
40
90
90
ns
ns
r
f
LOAD
LOAD
Bottom Gate Transition Time:
Rise Time
Fall Time
(Note 6)
BG1, 2 t
BG1, 2 t
C
C
= 3300pF
= 3300pF
30
20
90
90
ns
ns
r
f
LOAD
LOAD
TG/BG t
Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
(Note 6)
= 3300pF Each Driver
1D
C
90
ns
LOAD
BG/TG t
Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
(Note 6)
= 3300pF Each Driver
2D
C
90
ns
ns
LOAD
t
Minimum On-Time
Tested with a Square Wave (Note 7)
180
200
ON(MIN)
Internal V Regulator
CC
V
V
V
V
V
Internal V Voltage
6V < V < 30V; V = 4V
EXTVCC
4.8
4.5
5.0
0.2
120
4.7
0.2
5.2
1.0
240
V
%
INTVCC
CC
IN
INT
INTV Load Regulation
I
I
I
I
= 0 to 20mA; V
= 4V
EXTVCC
LDO
LDO
CC
CC
CC
CC
CC
EXT
EXTV Voltage Drop
= 20mA; V
= 5V
mV
V
CC
EXTVCC
EXTV Switchover Voltage
= 20mA, EXTV Ramping Positive
●
EXTVCC
LDOHYS
CC
CC
EXTV Switchover Hysteresis
= 20mA, EXTV Ramping Negative
V
CC
CC
VID Parameters
R
ATTEN
Resistance Between ATTENIN and
ATTENOUT Pins
20
kΩ
ATTEN
Resistive Divider Worst-Case Error
Programmed from 1.3V to 2.05V (VID4 = 0)
Programmed from 2.1V to 3.5V (VID4 = 1)
●
●
–0.25
–0.35
+0.25
+0.25
%
%
ERR
R
VID0–VID4 Pull-Up Resistance
VID0–VID4 Logic Threshold Low
VID0–VID4 Logic Threshold High
VID0–VID4 Leakage
(Note 8)
40
kΩ
V
PULLUP
VID
VID
VID
0.4
1
THLOW
THHIGH
LEAK
1.6
V
V
< VID0–VID4 < 7V
µA
BIAS
Oscillator and Phase-Locked Loop
f
f
f
Nominal Frequency
Lowest Frequency
Highest Frequency
PLLIN Input Resistance
V
V
V
= 1.2V
= 0V
190
120
280
220
140
310
50
250
160
360
kHz
kHz
kHz
kΩ
NOM
LOW
HIGH
PLLFLTR
PLLFLTR
PLLFLTR
≥ 2.4V
R
PLLIN
I
Phase Detector Output Current
Sinking Capability
Sourcing Capability
PLLFLTR
f
f
< f
> f
–15
15
µA
µA
PLLIN
PLLIN
OSC
OSC
R
Controller 2-Controller 1 Phase
180
Deg
RELPHS
Differential Amplifier/Op Amp Gain Block (Note 9)
A
Gain
Differential Amp Mode
Differential Amp Mode; 0V < V < 5V
0.995
46
1
1.005
V/V
dB
DA
CMRR
Common Mode Rejection Ratio
55
DA
CM
3
LTC1709
The ● denotes the specifications which apply over the full operating
ELECTRICAL CHARACTERISTICS
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
Differential Amp Mode; Measured at V + Input
MIN
TYP
MAX
UNITS
kΩ
R
IN
Input Resistance
Input Offset Voltage
80
OS
V
Op Amp Mode; V = 2.5V; V = 5V;
DIFFOUT
6
mV
OS
CM
I
= 1mA
DIFFOUT
I
Input Bias Current
Op Amp Mode
30
200
nA
V/mV
V
B
A
V
Open Loop DC Gain
Op Amp Mode; 0.7V ≤ V
Op Amp Mode
< 10V
DIFFOUT
5000
OL
Common Mode Input Voltage Range
Common Mode Rejection Ratio
Power Supply Rejection Ratio
Maximum Output Current
Maximum Output Voltage
Gain-Bandwidth Product
Slew Rate
0
3
CM
CMRR
Op Amp Mode; 0V < V < 3V
70
70
10
10
90
90
35
11
2
dB
OA
CM
PSRR
Op Amp Mode; 6V < V < 30V
dB
OA
IN
I
Op Amp Mode; V
= 0V
mA
V
CL
DIFFOUT
DIFFOUT
DIFFOUT
V
Op Amp Mode; I
Op Amp Mode; I
= 1mA
= 1mA
O(MAX)
GBW
SR
MHz
V/µs
Op Amp Mode; R = 2k
5
L
Note 1: Absolute Maximum Ratings are those values beyond which the
life of a device may be impaired.
Note 6: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 2: The LTC1709EG is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 7: The minimum on-time condition corresponds to the on inductor
peak-to-peak ripple current ≥40% I
(see Minimum On-Time
MAX
Considerations in the Applications Information section).
Note 8: Each built-in pull-up resistor attached to the VID inputs also has a
Note 3: T is calculated from the ambient temperature T and power
series diode to allow input voltages higher than the VIDV supply without
damage or clamping (see the Applications Information section).
J
A
CC
dissipation P according to the following formulas:
D
LTC1709EG: T = T + (P • 85°C/W)
Note 4: The LTC1709 is tested in a feedback loop that servos V to a
specified voltage and measures the resultant V
Note 5: Dynamic supply current is higher due to the gate charge being
Note 9: When the AMPMD pin is high, the IC pins are connected directly to
the internal op amp inputs. When the AMPMD pin is low, internal MOSFET
switches connect four 40k resistors around the op amp to create a
standard unity-gain differential amp.
J
A
D
ITH
.
EAIN
delivered at the switching frequency. See Applications Information.
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Output Current
(Figure 12)
Efficiency vs Output Current
(Figure 12)
Efficiency vs Input Voltage
(Figure 12)
100
80
60
40
20
0
100
90
100
80
60
40
20
0
V
V
OUT
= 3.3V
= 5V
= 20A
V
V
= 2V
OUT
EXTVCC
OUT
IN
= 12V
I
f = 200kHz
V
V
V
V
= 5V
IN
IN
IN
IN
= 8V
= 12V
= 20V
V
V
= 5V
= 0V
EXTVCC
EXTVCC
80
V
V
= 2V
OUT
EXTVCC
INTERNAL LDO VS EXTERNALLY
APPLIED 5V OVERALL EFFICIENCY
(FIGURE 12)
= 0V
f = 200kHz
70
0.1
1
10
100
0.1
1
10
100
5
10
15
20
OUTPUT CURRENT (A)
V
IN
(V)
OUTPUT CURRENT (A)
1709 G01
1709 G02
1709 G03
4
LTC1709
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Supply Current vs Input Voltage
and Mode
INTVCC and EXTVCC Switch
Voltage vs Temperature
EXTVCC Voltage Drop
1000
800
600
400
200
0
250
200
150
100
50
5.05
5.00
4.95
4.90
4.85
4.80
4.75
4.70
INTV VOLTAGE
CC
ON
EXTV SWITCHOVER THRESHOLD
CC
SHUTDOWN
10 15
INPUT VOLTAGE (V)
0
0
5
20
25
30
35
0
10
20
30
40
50
50
TEMPERATURE (°C)
100 125
–50 –25
0
25
75
CURRENT (mA)
1709 G04
1709 G05
1709 G06
Maximum Current Sense Threshold
vs Percent of Nominal Output
Voltage (Foldback)
Maximum Current Sense Threshold
vs Duty Factor
Internal 5V LDO Line Reg
75
5.1
5.0
80
70
60
50
40
30
20
10
0
I
= 1mA
LOAD
4.9
4.8
4.7
4.6
4.5
50
25
0
4.4
0
20
40
60
80
100
20
10
INPUT VOLTAGE (V)
30
35
0
25
50
75
100
0
5
15
25
DUTY FACTOR (%)
PERCENT ON NOMINAL OUTPUT VOLTAGE (%)
1709 G08
1709 G07
1709 G09
Current Sense Threshold
vs ITH Voltage
Maximum Current Sense Threshold
vs VRUN/SS (Soft-Start)
Maximum Current Sense Threshold
vs Sense Common Mode Voltage
80
90
80
80
60
40
20
V
= 1.6V
SENSE(CM)
70
76
72
68
64
60
60
50
40
30
20
10
0
–10
–20
–30
0
0
1
2
3
4
5
0
1
2
3
4
5
6
0
0.5
1
1.5
(V)
2
2.5
V
(V)
COMMON MODE VOLTAGE (V)
V
RUN/SS
ITH
1709 G11
1709 G10
1709 G12
5
LTC1709
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Load Regulation
VITH vs VRUN/SS
SENSE Pins Total Source Current
0.0
–0.1
–0.2
–0.3
–0.4
2.5
2.0
1.5
1.0
100
50
FCB = 0V
= 15V
V
= 0.7V
OSENSE
V
IN
FIGURE 1
0
–50
–100
0.5
0
0
1
2
3
4
5
0
2
3
4
5
6
0
2
4
6
1
V
(V)
LOAD CURRENT (A)
V
COMMON MODE VOLTAGE (V)
RUN/SS
SENSE
1709 G13
1709 G14
1709 G15
Maximum Current Sense
Threshold vs Temperature
Soft-Start Up (Figure 12)
RUN/SS Current vs Temperature
80
78
76
74
72
70
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
VITH
1V/DIV
VOUT
2V/DIV
VRUNSS
2V/DIV
100ms/DIV
1629 G19
0
–50 –25
0
25
50
75 100 125
–50 –25
0
25
125
50
75 100
TEMPERATURE (°C)
TEMPERATURE (°C)
1709 G17
1709 G18
Load Step Response Using Active
Voltage Positioning (Figure 12)
Current Sense Pin Input Current
vs Temperature
EXTVCC Switch Resistance
vs Temperature
35
33
31
29
27
25
10
8
EXTV = 5V
CC
VOUT
50mV/DIV
6
20A
IOUT
10A/DIV
4
0A
2
20µs/DIV
1709 G20
0
–50 –25
0
25
50
75 100 125
–50 –25
0
25
50
75 100 125
TEMPERATURE (°C)
TEMPERATURE (°C)
1709 G21
1709 G22
6
LTC1709
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Oscillator Frequency
vs Temperature
Undervoltage Lockout
vs Temperature
VRUN/SS Shutdown Latch
Thresholds vs Temperature
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
350
300
3.50
3.45
3.40
3.35
V
= 5V
FREQSET
LATCH ARMING
250
200
150
100
50
LATCHOFF
THRESHOLD
V
= OPEN
= 0V
FREQSET
V
FREQSET
3.30
3.25
3.20
0
0
50
100 125
50
TEMPERATURE (°C)
100 125
–50 –25
0
25
TEMPERATURE (°C)
125
–50 –25
0
25
75
–50 –25
0
25
75
50
75 100
TEMPERATURE (°C)
1709 G23
1709 G24
1709 G25
U
U
U
PI FU CTIO S
RUN/SS (Pin 1): Combination of Soft-Start, Run Control
Input and Short-Circuit Detection Timer. A capacitor to
groundatthispinsetstheramptimetofullcurrentoutput.
Forcing this pin below 0.8V causes the IC to shut down all
internal circuitry. All functions are disabled in shutdown.
SENSE 1+, SENSE 2+ (Pins 2,14): The (+) Input to Each
Differential Current Comparator. The ITH pin voltage and
built-in offsets between SENSE– and SENSE+ pins in
conjunction with RSENSE set the current trip threshold.
NC (Pins 7, 36): Do not connect.
TH (Pin 8): Error Amplifier Output and Switching Regula-
torCompensationPoint.Bothcurrentcomparator’sthresh-
oldsincreasewiththiscontrolvoltage. Thenormalvoltage
range of this pin is from 0V to 2.4V
I
SGND (Pin 9): Signal Ground, common to both control-
lers. Route separately to the PGND pin.
VDIFFOUT (Pin 10): Output of a Differential Amplifier that
provides true remote output voltage sensing. This pin
normally drives an external resistive divider that sets the
output voltage.
SENSE 1–, SENSE 2– (Pins 3, 13): The (–) Input to the
Differential Current Comparators.
EAIN (Pin 4): Input to the Error Amplifier that compares
thefeedbackvoltagetotheinternal0.8Vreferencevoltage.
This pin is normally connected to a resistive divider from
the output of the differential amplifier (DIFFOUT).
VOS–, VOS+ (Pins 11, 12): Inputs to an Operational Ampli-
fier. Internal precision resistors capable of being elec-
tronically switched in or out can configure it as a differen-
tial amplifier or an uncommitted Op Amp.
PLLFLTR (Pin 5): The Phase-Locked Loop’s Low Pass
Filter is tied to this pin. Alternatively, this pin can be driven
with an AC or DC voltage source to vary the frequency of
the internal oscillator.
ATTENOUT (Pin 15): Voltage Feedback Signal Resistively
Divided According to the VID Programming Code.
ATTENIN (Pin 16): The Input to the VID Controlled Resis-
tive Divider.
PLLIN (Pin 6): External Synchronization Input to Phase
Detector. This pin is internally terminated to SGND with
50kΩ. The phase-locked loop will force the rising top gate
signal of controller 1 to be synchronized with the rising
edge of the PLLIN signal.
VID0–VID4 (Pins 17,18, 19, 20, 21): VID Control Logic
Input Pins.
V
BIAS (Pin 22): Supply Pin for the VID Control Circuit.
7
LTC1709
U
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PI FU CTIO S
AMPMD (Pin 23): This Logic Input pin controls the
connections of internal precision resistors that configure
the operational amplifier as a unity-gain differential
amplifier.
PGND (Pin 28): Driver Power Ground, connect to sources
of bottom N-channel MOSFETS and the (–) terminals of
CIN.
INTVCC (Pin 29): Output of the Internal 5V Linear Low
Dropout Regulator and the EXTVCC Switch. The driver and
control circuits are powered from this voltage source.
Decouple to power ground with a 1µF ceramic capacitor
placed directly adjacent to the IC and minimum of 4.7µF
additional tantalum or other low ESR capacitor.
TG2, TG1 (Pins 24, 35): High Current Gate Drives for Top
N-Channel MOSFETS. These are the outputs of floating
drivers with a voltage swing equal to INTVCC superim-
posed on the switch node voltage SW.
SW2, SW1 (Pins 25, 34): Switch Node Connections to
Inductors. Voltage swing at these pins is from a Schottky
diode (external) voltage drop below ground to VIN.
EXTVCC (Pin 30): External Power Input to an Internal
Switch . This switch closes and supplies INTVCC, bypass-
ing the internallow dropout regulator whenever EXTVCC is
higher than 4.7V. See EXTVCC Connection in the Applica-
tions Information section. Do not exceed 7V on this pin
and ensure VEXTVCC ≤ VIN.
BOOST 2, BOOST 1 (Pins 26, 33): Bootstrapped Supplies
to the Topside Floating Drivers. External capacitors are
connectedbetweentheBoostandSwitchpins,andSchottky
diodes are connected between the Boost and INTVCC pins.
VIN(Pin32):MainSupplyPin.Shouldbecloselydecoupled
to the IC’s signal ground pin.
BG2, BG1 (Pins 27, 31): High Current Gate Drives for
Bottom N-Channel MOSFETS. Voltage swing at these pins
is from ground to INTVCC.
8
LTC1709
U
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FU CTIO AL DIAGRA
PLLIN
INTV
CC
V
IN
PHASE DET
F
IN
50k
PLLFLTR
D
C
DUPLICATE FOR
SECOND CHANNEL
B
BOOST
TG
R
C
LP
B
DROP
OUT
DET
+
CLK1
CLK2
TOP
BOT
C
LP
IN
OSCILLATOR
BOT
FORCE BOT
SW
S
R
Q
Q
TO
SWITCH
LOGIC
INTV
CC
SECOND
BG
CHANNEL
PGND*
–
+
V
V
OS
OS
SHDN
A1
–
+
INTV
CC
I1
–
L
–
+
+
+
SENSE
SENSE
30k
30k
4(V
)
FB
–
C
OUT
R
SENSE
AMPMD
DIFFOUT
SLOPE
COMP
0V POSITION
45k
45k
2.4V
V
OUT
EAIN
V
V
0.8V
REF
FB
V
IN
–
EA
V
IN
+
0.80V
0.86V
+
–
4.7V
OV
5V
LDO
REG
+
–
EXTV
CC
V
IN
C
C
I
TH
1.2µA
INTV
CC
5V
+
SHDN
RST
RUN
R
C
SOFT-
4(V
FB
)
START
INTERNAL
SUPPLY
SGND
6V
RUN/SS
R2
20k
C
SS
ATTENIN
5-BIT VID DECODER
ATTENOUT
TYPICAL ALL
VID PINS
40k
R1
R1 VARIABLE
VID0
VID1
VID2
VID3
VID4
V
BIAS
1709 FBD
9
LTC1709
U
(Refer to Functional Diagram)
OPERATIO
Main Control Loop
Low Current Operation
The LTC1709 uses a constant frequency, current mode
step-down architecture with inherent current sharing.
During normal operation, the top MOSFET is turned on
each cycle when the oscillator sets the RS latch, and
turned off when the main current comparator, I1, resets
the RS latch. The peak inductor current at which I1 resets
the RS latch is controlled by the voltage on the ITH pin,
which is the output of the error amplifier EA. The differen-
tialamplifier,A1,producesasignalequaltothedifferential
voltage sensed across the output capacitor but re-refer-
ences it to the internal signal ground (SGND) reference.
The EAIN pin receives a portion of this voltage feedback
signal at the DIFFOUT as determined by VID logic input
pins (VID0 to VID4) and is compared to the internal
reference voltage by the EA. When the load current in-
creases, it causes a slight decrease in the EAIN pin voltage
relative to the 0.8V reference, which in turn causes the ITH
voltage to increase until the average inductor current
matches the new load current. After the top MOSFET has
turned off, the bottom MOSFET is turned on for the rest of
the period.
The LTC1709 operates in a continuous, PWM control
mode. The resulting operation at low output currents
optimizes transient response at the expense of substantial
negative inductor current during the latter part of the
period. The level of ripple current is determined by the
inductor value, input voltage, output voltage, and fre-
quency of operation.
Frequency Synchronization
The phase-locked loop allows the internal oscillator to be
synchronized to an external source via the PLLIN pin. The
output of the phase detector at the PLLFLTR pin is also the
DC frequency control input of the oscillator that operates
over a 140kHz to 310kHz range corresponding to a DC
voltageinputfrom0Vto2.4V.Whenlocked,thePLLaligns
the turn on of the top MOSFET to the rising edge of the
synchronizingsignal.WhenPLLINisleftopen,thePLLFLTR
pingoeslow,forcingtheoscillatortominimumfrequency.
InputcapacitanceESRrequirementsandefficiencylosses
are substantially reduced because the peak current drawn
from the input capacitor is effectively divided by two and
power loss is proportional to the RMS current squared. A
two stage, single output voltage implementation can re-
duce input path power loss by 75% and radically reduce
the required RMS current rating of the input capacitor(s).
The top MOSFET drivers are biased from floating boot-
strap capacitor CB, which normally is recharged during
each off cycle through an external Schottky diode. When
VIN decreasestoavoltageclosetoVOUT,however,theloop
may enter dropout and attempt to turn on the top MOSFET
continuously. A dropout detector detects this condition
and forces the top MOSFET to turn off for about 400ns
every 10th cycle to recharge the bootstrap capacitor, CB.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
of the IC circuitry is derived from INTVCC. When the
EXTVCC pin is left open, an internal 5V low dropout
regulator supplies INTVCC power. If the EXTVCC pin is
taken above 4.7V, the 5V regulator is turned off and an
internalswitchisturnedonconnectingEXTVCC toINTVCC.
This allows the INTVCC power to be derived from a high
efficiency external source such as the output of the regu-
lator itself or a secondary winding, as described in the
Applications Information section. An external Schottky
diode can be used to minimize the voltage drop from
EXTVCC to INTVCC in applications requiring greater than
the specified INTVCC current. Voltages up to 7V can be
applied to EXTVCC for additional gate drive capability.
The main control loop is shut down by pulling Pin 1 (RUN/
SS) low. Releasing RUN/SS allows an internal 1.2µA
current source to charge soft-start capacitor CSS. When
CSS reaches1.5V,themaincontrolloopisenabledwiththe
ITH voltageclampedatapproximately30%ofitsmaximum
value. As CSS continues to charge, ITH is gradually re-
leased allowing normal operation to resume. When the
RUN/SS pin is low, all LTC1709 functions are shut down.
IfVOUT hasnotreached70%ofitsnominalvaluewhenCSS
has charged to 4.1V, an overcurrent latchoff can be
invoked as described in the Applications Information
section.
10
LTC1709
U
(Refer to Functional Diagram)
OPERATIO
Differential Amplifier
Short-Circuit Detection
This amplifier provides true differential output voltage
sensing. Sensing both VOUT+ and VOUT– benefits regula-
tion in high current applications and/or applications hav-
ing electrical interconnection losses. The AMPMD pin
allows selection of internal, precision feedback resistors
for high common mode rejection differencing applica-
tions, or direct access to the actual amplifier inputs
withouttheseinternalfeedbackresistorsforotherapplica-
tions. The AMPMD pin is grounded to connect the internal
precisionresistorsinaunity-gaindifferencingapplication,
or tied to the INTVCC pin to bypass the internal resistors
and make the amplifier inputs directly available. The
amplifier is a unity-gain stable, 2MHz gain-bandwidth,
>120dB open-loop gain design. The amplifier has an
output slew rate of 5V/µs and is capable of driving capaci-
tive loads with an output RMS current typically up to
35mA. The amplifier is not capable of sinking current and
therefore must be resistively loaded to do so.
The RUN/SS capacitor is used initially to limit the inrush
current from the input power source. Once the controllers
have been given time, as determined by the capacitor on
the RUN/SS pin, to charge up the output capacitors and
provide full-load current, the RUN/SS capacitor is then
usedasashort-circuittimeoutcircuit.Iftheoutputvoltage
falls to less than 70% of its nominal output voltage the
RUN/SS capacitor begins discharging assuming that the
output is in a severe overcurrent and/or short-circuit
condition. If the condition lasts for a long enough period
as determined by the size of the RUN/SS capacitor, the
controller will be shut down until the RUN/SS pin voltage
is recycled. This built-in latchoff can be overidden by
providing a current >5µA at a compliance of 5V to the
RUN/SS pin. This current shortens the soft-start period
but also prevents net discharge of the RUN/SS capacitor
during a severe overcurrent and/or short-circuit condi-
tion.Foldbackcurrentlimitingisactivatedwhentheoutput
voltage falls below 70% of its nominal level whether or not
the short-circuit latchoff circuit is enabled.
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APPLICATIO S I FOR ATIO
ThebasicLTC1709applicationcircuitisshowninFigure 1
on the first page. External component selection begins
with the selection of the inductor(s) based on ripple
current requirements and continues with the RSENSE1, 2
resistor selection using the calculated peak inductor cur-
rent and/or maximum current limit. Next, the power
MOSFETs and D1 and D2 are selected. The operating
frequency and the inductor are chosen based mainly on
the amount of ripple current. Finally, CIN is selected for its
ability to handle the input ripple current (that PolyPhaseTM
operationminimizes)andCOUT ischosenwithlowenough
ESR to meet the output ripple voltage and load step
specifications (also minimized with PolyPhase). Current
mode architecture provides inherent current sharing be-
tween output stages. The circuit shown in Figure 1 can be
configured for operation up to an input voltage of 28V
(limited by the external MOSFETs).
current. The LTC1709 current comparator has a maxi-
mum threshold of 75mV/RSENSE and an input common
mode range of SGND to 1.1( INTVCC). The current com-
parator threshold sets the peak inductor current, yielding
a maximum average output current IMAX equal to the peak
value less half the peak-to-peak ripple current, ∆IL.
Allowing a margin for variations in the LTC1709 and
external component values yields:
R
SENSE = 2(50mV/IMAX
)
Operating Frequency
The LTC1709 uses a constant frequency, phase-lockable
architecture with the frequency determined by an internal
capacitor. This capacitor is charged by a fixed current plus
an additional current which is proportional to the voltage
applied to the PLLFLTR pin. Refer to Phase-Locked Loop
and Frequency Synchronization in the Applications Infor-
mation section for additional information.
RSENSE Selection For Output Current
RSENSE1, 2 are chosen based on the required peak output
PolyPhase is a registered trademark of Linear Technology Corporation.
11
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A graph for the voltage applied to the PLLFLTR pin vs
frequency is given in Figure 2. As the operating frequency
isincreasedthegatechargelosseswillbehigher,reducing
efficiency (see Efficiency Considerations). The maximum
switching frequency is approximately 310kHz.
In a 2-phase converter, the net ripple current seen by the
output capacitor is much smaller than the individual
inductor ripple currents due to ripple cancellation. The
details on how to calculate the net output ripple current
can be found in Application Note 77.
Figure 3 shows the net ripple current seen by the output
capacitors for the 1- and 2- phase configurations. The
outputripplecurrentisplottedforafixedoutputvoltageas
the duty factor is varied between 10% and 90% on the
x-axis. The output ripple current is normalized against the
inductor ripple current at zero duty factor. The graph can
be used in place of tedious calculations, simplifying the
design process.
2.5
2.0
1.5
1.0
0.5
0
Accepting larger values of ∆IL allows the use of low
inductances, butcanresultinhigheroutputvoltageripple.
A reasonable starting point for setting ripple current is ∆IL
=0.4(IOUT)/2,whereIOUT isthetotalloadcurrent.Remem-
ber, the maximum ∆IL occurs at the maximum input
voltage. The individual inductor ripple currents are deter-
mined by the inductor, input and output voltages.
1.0
120
170
220
270
320
OPERATING FREQUENCY (kHz)
1709 F02
Figure 2. Operating Frequency vs VPLLFLTR
Inductor Value Calculation and Output Ripple Current
1-PHASE
2-PHASE
0.9
0.8
The operating frequency and inductor selection are inter-
related in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because
MOSFET gate charge and transition losses increase di-
rectly with frequency. In addition to this basic tradeoff, the
effect of inductor value on ripple current and low current
operation must also be considered. The PolyPhase ap-
proach reduces both input and output ripple currents
while optimizing individual output stages to run at a lower
fundamental frequency, enhancing efficiency.
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
DUTY FACTOR (V /V
)
OUT IN
1709 F03
Figure 3. Normalized Output Ripple Current vs
Duty Factor [IRMS ≈ 0.3 (∆IO(P–P))]
Theinductorvaluehasadirecteffectonripplecurrent.The
inductor ripple current ∆IL per individual section, N,
decreases with higher inductance or frequency and in-
Inductor Core Selection
Once the values for L1 and L2 are known, the type of
inductor must be selected. High efficiency converters
generally cannot afford the core loss found in low cost
powdered iron cores, forcing the use of more expensive
ferrite, molypermalloy, or Kool Mµ® cores. Actual core
loss is independent of core size for a fixed inductor value,
creases with higher VIN or VOUT
:
VOUT
fL
VOUT
V
IN
∆IL =
1−
where f is the individual output stage operating frequency.
Kool Mµ is a registered trademark of Magnetics, Inc.
12
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but it is very dependent on inductance selected. As induc-
tance increases, core losses go down. Unfortunately,
increased inductance requires more turns of wire and
therefore copper losses will increase.
VOUT
V
IN
Main SwitchDuty Cycle =
V – VOUT
IN
Synchronous SwitchDuty Cycle =
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can con-
centrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that induc-
tance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
V
IN
The MOSFET power dissipations at maximum output
current are given by:
2
VOUT IMAX
PMAIN
=
1+ δ RDS(ON)
+
(
)
V
IN
2
Molypermalloy (from Magnetics, Inc.) is a very good, low
losscorematerialfortoroids,butitismoreexpensivethan
ferrite. A reasonable compromise from the same manu-
facturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire. Be-
cause they lack a bobbin, mounting is more difficult.
However, designs for surface mount are available which
do not increase the height significantly.
2
)
IMAX
2
k V
CRSS
f
(
IN
(
)( )
2
V – VOUT IMAX
IN
PSYNC
=
1+ δ RDS(ON)
(
)
V
IN
2
where δ is the temperature dependency of RDS(ON) and k
is a constant inversely related to the gate drive current.
Power MOSFET, D1 and D2 Selection
Both MOSFETs have I2R losses but the topside N-channel
equation includes an additional term for transition losses,
which peak at the highest input voltage. For VIN < 20V the
high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V the transition losses rapidly
increasetothepointthattheuseofahigherRDS(ON)device
with lower CRSS actual provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during a
short-circuit when the synchronous switch is on close to
100% of the period.
Two external power MOSFETs must be selected for each
output stage for the LTC1709: One N-channel MOSFET for
the top (main) switch, and one N-channel MOSFET for the
bottom (synchronous) switch.
The peak-to-peak drive levels are set by the INTVCC volt-
age. This voltage is typically 5V during start-up (see
EXTVCC PinConnection).Consequently,logic-levelthresh-
old MOSFETs must be used in most applications. The only
exception is if low input voltage is expected (VIN < 5V);
then, sublogic-level threshold MOSFETs (VGS(TH) < 1V)
should be used. Pay close attention to the BVDSS specifi-
cation for the MOSFETs as well; most of the logic-level
MOSFETs are limited to 30V or less.
The term (1 + δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs. Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs. CRSS is usually specified in the MOS-
FET characteristics. The constant k = 1.7 can be used to
estimate the contributions of the two terms in the main
switch dissipation equation.
SelectioncriteriaforthepowerMOSFETsincludethe“ON”
resistance RDS(ON), reverse transfer capacitance CRSS
,
input voltage, and maximum output current. When the
LTC1709isoperatingincontinuousmodethedutyfactors
for the top and bottom MOSFETs of each output stage are
given by:
TheSchottkydiodes,D1andD2showninFigure1conduct
during the dead-time between the conduction of the two
large power MOSFETs. This helps prevent the body diode
13
LTC1709
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APPLICATIO S I FOR ATIO
of the bottom MOSFET from turning on, storing charge
during the dead-time, and requiring a reverse recovery
period which would reduce efficiency. A 1A to 3A Schottky
(depending on output current) diode is generally a good
compromise for both regions of operation due to the
relatively small average current. Larger diodes result in
additional transition losses due to their larger junction
capacitance.
These worst-case conditions are commonly used for
design because even significant deviations do not offer
much relief. Note that capacitor manufacturer’s ripple
current ratings are often based on only 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to
meet size or height requirements in the design. Always
consult the capacitor manufacturer if there is any
question.
CIN and COUT Selection
In continuous mode, the source current of each top
It is important to note that the efficiency loss is propor-
tional to the input RMS current squared and therefore a
2-phase implementation results in 75% less power loss
when compared to a single phase design. Battery/input
protection fuse resistance (if used), PC board trace and
connector resistance losses are also reduced by the re-
ductionoftheinputripplecurrentina2-phasesystem.The
requiredamountofinputcapacitanceisfurtherreducedby
the factor, 2, due to the effective increase in the frequency
of the current pulses.
N-channel MOSFET is a square wave of duty cycle VOUT
/
VIN. A low ESR input capacitor sized for the maximum
RMS current must be used. The details of a closed form
equation can be found in Application Note 77. Figure 4
shows the input capacitor ripple current for a 2-phase
configuration with the output voltage fixed and input
voltage varied. The input ripple current is normalized
against the DC output current. The graph can be used in
place of tedious calculations. The minimum input ripple
currentcanbeachievedwhentheinputvoltageistwicethe
output voltage
The selection of COUT is driven by the required effective
series resistance (ESR). Typically once the ESR require-
ment has been met, the RMS current rating generally far
exceeds the IRIPPLE(P-P) requirements. The steady state
output ripple (∆VOUT) is determined by:
In the graph of Figure 4, the 2-phase local maximum input
RMS capacitor currents are reached when:
VOUT 2k − 1
=
1
V
IN
4
∆VOUT ≈ ∆IRIPPLE ESR +
16fCOUT
where k = 1, 2.
Where f = operating frequency of each stage, COUT
=
0.6
output capacitance and ∆IRIPPLE = combined inductor
ripple currents.
0.5
0.4
0.3
0.2
0.1
0
The output ripple varies with input voltage since ∆IL is a
functionofinputvoltage.Theoutputripplewillbelessthan
50mV at max VIN with ∆IL = 0.4IOUT(MAX)/2 assuming:
1-PHASE
2-PHASE
COUT required ESR < 4(RSENSE) and
COUT > 1/(16f)(RSENSE
)
The emergence of very low ESR capacitors in small,
surface mount packages makes very physically small
implementations possible. The ability to externally com-
pensatetheswitchingregulatorloopusingtheITHpin(OPTI-
LOOP compensation) allows a much wider selection of
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
DUTY FACTOR (V /V
)
OUT IN
1709 F04
Figure 4. Normalized RMS Input Ripple Current vs
Duty Factor for 1 and 2 Output Stages
14
LTC1709
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output capacitor types. OPTI-LOOP compensation effec-
tively removes constraints on output capacitor ESR. The
impedance characteristics of each capacitor type are sig-
nificantly different than an ideal capacitor and therefore
require accurate modeling or bench evaluation during
design.
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maxi-
mum junction temperature rating for the LTC1709 to be
exceeded. The supply current is dominated by the gate
charge supply current, in addition to the current drawn
from the differential amplifier output. The gate charge is
dependent on operating frequency as discussed in the
Efficiency Considerations section. The supply current can
either be supplied by the internal 5V regulator or via the
EXTVCC pin. When the voltage applied to the EXTVCC pin
is less than 4.7V, all of the INTVCC load current is supplied
by the internal 5V linear regulator. Power dissipation for
the IC is higher in this case by (IIN)(VIN – INTVCC) and
efficiency is lowered. The junction temperature can be
estimated by using the equations given in Note 1 of the
Electrical Characteristics. For example, the LTC1709 VIN
current is limited to less than 24mA from a 24V supply:
Manufacturers such as Nichicon, United Chemicon and
Sanyoshouldbeconsideredforhighperformancethrough-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo and the Panasonic SP
surface mount types have the lowest (ESR)(size) product
of any aluminum electrolytic at a somewhat higher price.
An additional ceramic capacitor in parallel with OS-CON
type capacitors is recommended to reduce the inductance
effects.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
surface mount configurations. New special polymer sur-
face mount capacitors offer very low ESR also but have
muchlowercapacitivedensityperunitvolume. Inthecase
oftantalum,itiscriticalthatthecapacitorsaresurgetested
for use in switching power supplies. Several excellent
choices are the AVX TPS, AVX TPSV or the KEMET T510
seriesofsurfacemounttantalums,availableincaseheights
ranging from 2mm to 4mm. Other capacitor types include
Sanyo OS-CON, Nichicon PL series and Sprague 595D
series. Consultthemanufacturerforotherspecificrecom-
mendations. A combination of capacitors will often result
in maximizing performance and minimizing overall cost
and size.
TJ = 70°C + (24mA)(24V)(85°C/W) = 119°C
Use of the EXTVCC pin reduces the junction temperature
to:
TJ = 70°C + (24mA)(5V)(85°C/W) = 80.2°C
The input supply current should be measured while the
controller is operating in continuous mode at maximum
VIN and the power dissipation calculated in order to pre-
vent the maximum junction temperature from being ex-
ceeded.
EXTVCC Connection
The LTC1709 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
When the voltage applied to EXTVCC rises above 4.7V, the
internal regulator is turned off and an internal switch
closes, connecting the EXTVCC pin to the INTVCC pin
therebysupplyinginternalandMOSFETgatedrivingpower
to the IC. The switch remains closed as long as the voltage
applied to EXTVCC remains above 4.5V. This allows the
MOSFET driver and control power to be derived from the
output during normal operation (4.7V < VEXTVCC < 7V) and
from the internal regulator when the output is out of
regulation (start-up, short-circuit). Do not apply greater
than 7V to the EXTVCC pin and ensure that EXTVCC < VIN +
0.3V when using the application circuits shown. If an
INTVCC Regulator
An internal P-channel low dropout regulator produces 5V
at the INTVCC pin from the VIN supply pin. The INTVCC
regulator powers the drivers and internal circuitry of the
LTC1709.TheINTVCC pinregulatorcansupplyupto50mA
peak and must be bypassed to power ground with a
minimum of 4.7µF tantalum or electrolytic capacitor. An
additional 1µF ceramic capacitor placed very close to the
IC is recommended due to the extremely high instanta-
neous currents required by the MOSFET gate drivers.
15
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external voltage source is applied to the EXTVCC pin when
the VIN supply is not present, a diode can be placed in
series with the LTC1709’s VIN pin and a Schottky diode
between the EXTVCC and the VIN pin, to prevent current
from backfeeding VIN.
4.7V but less than 7V. This can be done with either the
inductive boost winding as shown in Figure 5a or the
capacitive charge pump shown in Figure 5b. The charge
pump has the advantage of simple magnetics.
Topside MOSFET Driver Supply (CB,DB) (Refer to
Functional Diagram)
Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current resulting
from the driver and control currents will be scaled by the
ratio: (Duty Factor)/(Efficiency). For 5V regulators this
means connecting the EXTVCC pin directly to VOUT. How-
ever, for 3.3V and other lower voltage regulators, addi-
tionalcircuitryisrequiredtoderiveINTVCC powerfromthe
output.
External bootstrap capacitors CB1 and CB2 connected to
the BOOST 1 and BOOST 2 pins supply the gate drive
voltages for the topside MOSFETs. Capacitor CB in the
Functional Diagram is charged though diode DB from
INTVCC whentheSWpinislow.WhenthetopsideMOSFET
turns on, the driver places the CB voltage across the gate-
sourceofthedesiredMOSFET.ThisenhancestheMOSFET
and turns on the topside switch. The switch node voltage,
The following list summarizes the four possible connec-
tions for EXTVCC:
SW, rises to VIN and the BOOST pin rises to VIN + VINTVCC
.
1. EXTVCC left open (or grounded). This will cause INTVCC
to be powered from the internal 5V regulator resulting in
a significant efficiency penalty at high input voltages.
The value of the boost capacitor CB needs to be 30 to 100
times that of the total input capacitance of the topside
MOSFET(s). ThereversebreakdownofDB mustbegreater
than VIN(MAX).
2. EXTVCC connected directly to VOUT. This is the normal
connection for a 5V regulator and provides the highest
efficiency.
The final arbiter when defining the best gate drive ampli-
tude level will be the input supply current. If a change is
made that decreases input current, the efficiency has
improved. If the input current does not change then the
efficiency has not changed either.
3. EXTVCC connected to an external supply. If an external
supply is available in the 5V to 7V range, it may be used to
powerEXTVCC providingitiscompatiblewiththeMOSFET
gate drive requirements.
Output Voltage
4. EXTVCC connected to an output-derived boost network.
For 3.3V and other low voltage regulators, efficiency gains
can still be realized by connecting EXTVCC to an output-
derived voltage which has been boosted to greater than
The LTC1709 has a true remote voltage sense capablity.
Thesensingconnectionsshouldbereturnedfromtheload
back to the differential amplifier’s inputs through a com-
+
OPTIONAL EXTV CONNECTION
CC
V
IN
+
5V < V
< 7V
V
SEC
+
C
IN
IN
C
IN
V
IN
BAT85
0.22µF
BAT85
BAT85
V
LTC1709
IN
1N4148
V
TG1
SEC
TG1
+
LTC1709
N-CH
VN2222LL
R
1µF
EXTV
CC
N-CH
EXTV
CC
R
SENSE
SENSE
V
OUT
V
SW1
BG1
SW1
BG1
OUT
L1
T1
+
+
C
C
OUT
OUT
N-CH
N-CH
PGND
PGND
1709 F05b
1709 F05a
Figure 5a. Secondary Output Loop with EXTVCC Connection
Figure 5b. Capacitive Charge Pump for EXTVCC
16
LTC1709
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mon, tightly coupled pair of PC traces. The differential
amplifier corrects for DC drops in both the power and
ground paths. The differential amplifier output signal is
divided down and compared with the internal precision
0.8V voltage reference by the error amplifier.
Table 1. VID Output Voltage Programming
VID4
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
VID3
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
VID2
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
VID1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
VID0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
V
(V)
OUT
3.50V
3.40V
3.30V
3.20V
3.10V
3.00V
2.90V
2.80V
2.70V
2.60V
2.50V
2.40V
2.30V
2.20V
2.10V
*
The differential amplifier can be used in either of two
configurations according to the voltage applied to the
AMPMD pin. The first configuration with the connections
illustrated in the Functional Diagram, utilizes a set of
internal, precision resistors to enable precision instru-
mentation-type measurement of the output voltage. This
configuration is activated when the AMPMD pin is tied to
ground. When the AMPMD pin is tied to INTVCC, the
resistors are disconnected and the amplifier inputs are
made directly available. It can be used for general uses if
the amplifier is not required for true remote sensing. The
amplifier has a 0V to 3V common mode input range
limitation due to the internal switching of its inputs. The
output uses an NPN emitter follower without any internal
pull-down current. A DC resistive load to ground is re-
quired in order to sink current. The output will swing from
0V to 10V (VIN ≥ VDIFFOUT + 2V).
2.05V
2.00V
1.95V
1.90V
1.85V
1.80V
1.75V
1.70V
1.65V
1.60V
1.55V
1.50V
1.45V
1.40V
1.35V
1.30V
Output Voltage Programming
The output voltage is digitally set to levels between 1.3V
and3.5Vusingthevoltageidentification(VID)logicinputs
VID0 to VID4. The internal 5-bit DAC configured as a
precision resistive voltage divider sets the output voltage
in 100mV or 50mV increments according to Table 1.
The VID codes are engineered to be compatible with Intel
Pentium® II and Pentium III processor specifications for
output voltages from 1.3V to 3.5V.
The LSB (VID0) represents 50mV or 100mV increments
depending on the MSB. The MSB is VID4.
Between the ATTENOUT pin and ground is a variable
resistor,R1,whosevalueiscontrolledbythefiveVIDinput
pins (VID0 to VID4). Another resistor, R2, between the
ATTENIN and the ATTENOUT pins completes the resistive
divider. The output voltage is thus set by the ratio of
(R1 + R2) to R1.
* Represents codes without a defined output voltage as specified in Intel
specifications. The LTC1709 interprets these codes as a valid input and
produces an output voltage as follows: (11111) = 2V
Each VID digital input is pulled up by a 40k resistor in
series with a diode from VBIAS. Therefore, it must be
grounded to get a digital low input, and can be either
Pentium is a registered trademark of Intel Corporation.
17
LTC1709
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floatedorconnectedtoVBIAS togetadigitalhighinput.The
series diode is used to prevent the digital inputs from
being damaged or clamped if they are driven higher than
Diode D1 in Figure 6 reduces the start delay but allows CSS
to ramp up slowly providing the soft-start function. The
RUN/SS pin has an internal 6V zener clamp (see Func-
tional Diagram).
V
BIAS. The digital inputs accept CMOS voltage levels.
VBIAS is the supply voltage for the VID section. It is
normally connected to INTVCC but can be driven from
other sources. If it is driven from another source, that
source MUST be in the range of 2.7V to 5.5V and MUST be
alive prior to enabling the LTC1709.
V
INTV
IN
CC
R
3.3V OR 5V
RUN/SS
*
R
*
SS
SS
D1
RUN/SS
D1*
C
SS
C
SS
Soft-Start/Run Function
*OPTIONAL TO DEFEAT OVERCURRENT LATCHOFF
1709 F06
The RUN/SS pin provides three functions: 1) Run/Shut-
down,2)soft-startand3)adefeatableshort-circuitlatchoff
timer. Soft-start reduces the input power sources’ surge
currents by gradually increasing the controller’s current
limit ITH(MAX). The latchoff timer prevents very short,
extreme load transients from tripping the overcurrent
latch. A small pull-up current (>5µA) supplied to the RUN/
SS pin will prevent the overcurrent latch from operating.
The following explanation describes how the functions
operate.
Figure 6. RUN/SS Pin Interfacing
Fault Conditions: Overcurrent Latchoff
The RUN/SS pin also provides the ability to latch off the
controllerswhenanovercurrentconditionisdetected.The
RUN/SS capacitor, CSS, is used initially to limit the inrush
current of both controllers. After the controllers have been
started and been given adequate time to charge up the
output capacitors and provide full load current, the RUN/
SS capacitor is used for a short-circuit timer. If the output
voltagefallstolessthan70%ofitsnominalvalueafterCSS
reaches 4.1V, CSS begins discharging on the assumption
that the output is in an overcurrent condition. If the
condition lasts for a long enough period as determined by
the size of CSS, the controller will be shut down until the
RUN/SS pin voltage is recycled. If the overload occurs
during start-up, the time can be approximated by:
An internal 1.2µA current source charges up the soft-start
capacitor, CSS. When the voltage on RUN/SS reaches
1.5V, the controller is permitted to start operating. As the
voltage on RUN/SS increases from 1.5V to 3.0V, the
internal current limit is increased from 25mV/RSENSE to
75mV/RSENSE. The output current limit ramps up slowly,
taking an additional 1.25s/µF to reach full current. The
outputcurrentthusrampsupslowly,reducingthestarting
surge current required from the input power supply. If
RUN/SS has been pulled all the way to ground there is a
delay before starting of approximately:
t
LO1 ≈ (CSS • 0.6V)/(1.2µA) = 5 • 105 (CSS)
Iftheoverloadoccursafterstart-up,thevoltageonCSS will
continue charging and will provide additional time before
latching off:
1.5V
1.2µA
tDELAY
=
CSS = 1.25s / µF C
SS
(
)
t
LO2 ≈ (CSS • 3V)/(1.2µA) = 2.5 • 106 (CSS)
The time for the output current to ramp up is then:
This built-in overcurrent latchoff can be overridden by
providing a pull-up resistor, RSS, to the RUN/SS pin as
shown in Figure 6. This resistance shortens the soft-start
period and prevents the discharge of the RUN/SS capaci-
tor during a severe overcurrent and/or short-circuit con-
dition. When deriving the 5µA current from VIN as in the
figure, current latchoff is always defeated. Diode connect-
ing this pull-up resistor to INTVCC, as in
3V − 1.5V
1.2µA
tRAMP
=
CSS = 1.25s / µF C
SS
(
)
By pulling the RUN/SS pin below 0.8V the LTC1709 is put
into low current shutdown (IQ < 40µA). The RUN/SS pins
can be driven directly from logic as shown in Figure 6.
18
LTC1709
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Figure 6, eliminates any extra supply current during shut-
down while eliminating the INTVCC loading from prevent-
ing controller start-up.
The output of the phase detector is a complementary pair
of current sources charging or discharging the external
filter network on the PLLFLTR pin. A simplified block
diagram is shown in Figure 7.
Why should you defeat current latchoff? During the
prototypingstageofadesign,theremaybeaproblemwith
noise pickup or poor layout causing the protection circuit
to latch off the controller. Defeating this feature allows
troubleshooting of the circuit and PC layout. The internal
short-circuit and foldback current limiting still remains
active, thereby protecting the power supply system from
failure. A decision can be made after the design is com-
plete whether to rely solely on foldback current limiting or
to enable the latchoff feature by removing the pull-up
resistor.
If the external frequency (fPLLIN) is greater than the oscil-
lator frequency f0SC, current is sourced continuously,
pulling up the PLLFLTR pin. When the external frequency
is less than f0SC, current is sunk continuously, pulling
down the PLLFLTR pin. If the external and internal fre-
quencies are the same but exhibit a phase difference, the
currentsourcesturnonforanamountoftimecorrespond-
ing to the phase difference. Thus the voltage on the
PLLFLTR pin is adjusted until the phase and frequency of
the external and internal oscillators are identical. At this
stable operating point the phase comparator output is
open and the filter capacitor CLP holds the voltage. The
LTC1709 PLLIN pin must be driven from a low impedance
source such as a logic gate located close to the pin.
The value of the soft-start capacitor CSS may need to be
scaled with output voltage, output capacitance and load
current characteristics. The minimum soft-start capaci-
tance is given by:
The loop filter components (CLP, RLP) smooth out the
current pulses from the phase detector and provide a
stable input to the voltage controlled oscillator. The filter
components CLP and RLP determine how fast the loop
acquires lock. Typically RLP =10kΩ and CLP is 0.01µF to
0.1µF.
CSS > (COUT )(VOUT)(10-4)(RSENSE
)
The minimum recommended soft-start capacitor of CSS
0.1µF will be sufficient for most applications.
=
Phase-Locked Loop and Frequency Synchronization
The LTC1709 has a phase-locked loop comprised of an
internal voltage controlled oscillator and phase detector.
This allows the top MOSFET turn-on to be locked to the
rising edge of an external source. The frequency range of
the voltage controlled oscillator is ±50% around the
center frequency fO. A voltage applied to the PLLFLTR pin
of 1.2V corresponds to a frequency of approximately
220kHz. The nominal operating frequency range of the
LTC1709 is 140kHz to 310kHz.
2.4V
R
LP
10k
PHASE
DETECTOR
C
LP
EXTERNAL
OSC
PLLFLTR
PLLIN
DIGITAL
PHASE/
FREQUENCY
DETECTOR
OSC
50k
The phase detector used is an edge sensitive digital type
which provides zero degrees phase shift between the
external and internal oscillators. This type of phase detec-
tor will not lock up on input frequencies close to the
harmonics of the VCO center frequency. The PLL hold-in
range, ∆fH, is equal to the capture range, ∆fC:
1709 F07
Figure 7. Phase-Locked Loop Block Diagram
Minimum On-Time Considerations
Minimum on-time tON(MIN) is the smallest time duration
thattheLTC1709iscapableofturningonthetopMOSFET.
It is determined by internal timing delays and the gate
chargerequiredtoturnonthetopMOSFET.Lowdutycycle
∆fH = ∆fC = ±0.5 fO
(150kHz-300kHz)
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INTV
CC
applications may approach this minimum on-time limit
and care should be taken to ensure that:
R
R
T2
T1
I
TH
LTC1709
VOUT
R
C
tON MIN
<
(
)
C
C
V f
IN( )
1709 F08
Ifthedutycyclefallsbelowwhatcanbeaccommodatedby
the minimum on-time, the LTC1709 will begin to skip
cycles resulting in variable frequency operation. The out-
put voltage will continue to be regulated, but the ripple
current and ripple voltage will increase.
Figure 8. Active Voltage Positioning Applied to the LTC1709
tion is included in Design Solutions 10 or the LTC1736
data sheet. (See www.linear-tech.com)
Efficiency Considerations
The minimum on-time for the LTC1709 is generally less
than 200ns. However, as the peak sense voltage de-
creases,theminimumon-timegraduallyincreases.Thisis
of particular concern in forced continuous applications
withlowripplecurrentatlightloads.Ifthedutycycledrops
below the minimum on-time limit in this situation, a
significant amount of cycle skipping can occur with corre-
spondingly larger ripple current and voltage ripple.
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can be
expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
If an application can operate close to the minimum on-
time limit, an inductor must be chosen that has a low
enough inductance to provide sufficient ripple amplitude
to meet the minimum on-time requirement. As a general
rule, keep the inductor ripple current of each phase equal
whereL1, L2, etc. aretheindividuallossesasapercentage
of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1709 circuits: 1) I2R losses, 2) Topside
MOSFET transition losses, 3) INTVCC regulator current
and 4) LTC1709 VIN current (including loading on the
differential amplifier output).
to or greater than 15% of IOUT(MAX) at VIN(MAX)
.
Voltage Positioning
Voltage positioning can be used to minimize peak-to-peak
outputvoltageexcursionunderworst-casetransientload-
ing conditions. The open-loop DC gain of the control loop
is reduced depending upon the maximum load step speci-
fication. Voltage positioning can easily be added to the
LTC1709 by loading the ITH pin with a resistive divider
having a Thevenin equivalent voltage source equal to the
midpoint operating voltage of the error amplifier, or 1.2V
(see Figure 8).
1) I2R losses are predicted from the DC resistances of the
fuse (if used), MOSFET, inductor, current sense resistor,
and input and output capacitor ESR. In continuous mode
the average output current flows through L and RSENSE
,
but is “chopped” between the topside MOSFET and the
synchronous MOSFET. If the two MOSFETs have approxi-
mately the same RDS(ON), then the resistance of one
MOSFET can simply be summed with the resistances of L,
RSENSE and ESR to obtain I2R losses. For example, if each
RDS(ON)=10mΩ, RL=10mΩ, and RSENSE=5mΩ, then the
total resistance is 25mΩ. This results in losses ranging
from 2% to 8% as the output current increases from 3A to
15A per output stage for a 5V output, or a 3% to 12% loss
per output stage for a 3.3V output. Efficiency varies as the
inverse square of VOUT for the same external components
The resistive load reduces the DC loop gain while main-
taining the linear control range of the error amplifier. The
worst-case peak-to-peak output voltage deviation due to
transient loading can theoretically be reduced to half or
alternatively the amount of output capacitance can be
reduced for a particular application. A complete explana-
20
LTC1709
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and output power level. The combined effects of increas-
ingly lower output voltages and higher currents required
by high performance digital systems is not doubling but
quadrupling the importance of loss terms in the switching
regulator system!
minimum of 200µF to 300µF of output capacitance having
a maximum of 10mΩ to 20mΩ of ESR. The LTC1709
2-phase architecture typically halves the input and output
capacitance requirement over competing solutions. Other
lossesincludingSchottkyconductionlossesduringdead-
time and inductor core losses generally account for less
than 2% total additional loss.
2) Transition losses apply only to the topside MOSFET(s),
and are significant only when operating at high input
voltages (typically 12V or greater). Transition losses can
be estimated from:
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current. When a load step occurs, VOUT shifts by an
amount equal to ∆ILOAD(ESR), where ESR is the effective
series resistance of COUT • (∆ILOAD) also begins to charge
or discharge COUT generating the feedback error signal
thatforcestheregulatortoadapttothecurrentchangeand
return VOUT to its steady-state value. During this recovery
time VOUT can be monitored for excessive overshoot or
ringing, which would indicate a stability problem. The
availability of the ITH pin not only allows optimization of
control loop behavior but also provides a DC coupled and
AC filtered closed loop response test point. The DC step,
rise time, and settling at this test point truly reflects the
closed loop response. Assuming a predominantly second
order system, phase margin and/or damping factor can be
estimated using the percentage of overshoot seen at this
pin. The bandwidth can also be estimated by examining
the rise time at the pin. The ITH external components
shown in the Figure 1 circuit will provide an adequate
starting point for most applications.
2
Transition Loss = (1.7) VIN IO(MAX) CRSS
f
3) INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results from
switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched from low to high to
low again, a packet of charge dQ moves from INTVCC to
ground. The resulting dQ/dt is a current out of INTVCC that
is typically much larger than the control circuit current. In
continuous mode, IGATECHG = (QT + QB), where QT and QB
are the gate charges of the topside and bottom side
MOSFETs.
SupplyingINTVCC powerthroughtheEXTVCC switchinput
from an output-derived source will scale the VIN current
required for the driver and control circuits by the ratio
(Duty Factor)/(Efficiency). For example, in a 20V to 5V
application, 10mA of INTVCC current results in approxi-
mately 3mA of VIN current. This reduces the mid-current
loss from 10% or more (if the driver was powered directly
from VIN) to only a few percent.
4) The VIN current has two components: the first is the
DC supply current given in the Electrical Characteristics
table, which excludes MOSFET driver and control cur-
rents; the second is the current drawn from the differential
amplifier output. VIN current typically results in a small
(<0.1%) loss.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.2 to 5 times their suggested values) to optimize
transient response once the final PC layout is done and the
particular output capacitor type and value have been
determined. The output capacitors need to be decided
upon because the various types and values determine the
loop gain and phase. An output current pulse of 20% to
80% of full-load current having a rise time of <2µs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking
the feedback loop. The initial output voltage step resulting
from the step change in output current may not be within
Other “hidden” losses such as copper trace and internal
battery resistances can account for an additional 5% to
10% efficiency degradation in portable systems. It is very
important to include these “system” level losses in the
design of a system. The internal battery and input fuse
resistance losses can be minimized by making sure that
CIN has adequate charge storage and a very low ESR at the
switching frequency. A 50W supply will typically require a
21
LTC1709
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50A I RATING
PK
the bandwidth of the feedback loop, so this signal cannot
be used to determine phase margin. This is why it is better
to look at the Ith pin signal which is in the feedback loop
and is the filtered and compensated control loop re-
sponse. The gain of the loop will be increased by increas-
ing RC and the bandwidth of the loop will be increased by
decreasing CC. If RC is increased by the same factor that
CC is decreased, the zero frequency will be kept the same,
thereby keeping the phase the same in the most critical
frequency range of the feedback loop. The output voltage
settling behavior is related to the stability of the closed-
loopsystemandwilldemonstratetheactualoverallsupply
performance.
V
IN
12V
LTC1709
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
1709 F09
Figure 9. Automotive Application Protection
Design Example
Asadesignexample,assumeVIN=5V(nominal),VIN = 5.5V
(max), VOUT =1.8V, IMAX =20A, TA =70°Candf = 300kHz.
Theinductancevalueischosenfirstbasedona30%ripple
current assumption. The highest value of ripple current
occursatthemaximuminputvoltage. TiethePLLFLTRpin
to the INTVCC pin for 300kHz operation. The minimum
inductance for 30% ripple current is:
Automotive Considerations: Plugging into the
Cigarette Lighter
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserveorevenrechargebatterypacksduringoperation.
But before you connect, be advised: you are plugging into
the supply from hell. The main battery line in an automo-
bileisthesourceofanumberofnastypotentialtransients,
including load-dump, reverse-battery, and double-bat-
tery.
VOUT
f ∆I
( )
VOUT
V
IN
L ≥
1−
1.8V
1.8V
5.5V
≥
1−
300kHz 30% 10A
(
)( )(
)
≥ 1.35µH
Load-dump is the result of a loose battery cable. When the
cablebreaksconnection,thefieldcollapseinthealternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse-battery is
just what it says, while double-battery is a consequence of
tow truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
A 1.5µH inductor will produce 27% ripple current. The
peak inductor current will be the maximum DC value plus
one half the ripple current, or 11.4A. The minimum on-
time occurs at maximum VIN:
VOUT
V f
IN
1.8V
)(
tON MIN
=
=
= 1.1µs
(
)
ThenetworkshowninFigure9isthemoststraightforward
approach to protect a DC/DC converter from the ravages
of an automotive power line. The series diode prevents
current from flowing during reverse-battery, while the
transient suppressor clamps the input voltage during
load-dump. Note that the transient suppressor should not
conduct during double-battery operation, but must still
clamptheinputvoltagebelowbreakdownoftheconverter.
AlthoughtheLT1709hasamaximuminputvoltageof36V,
most applications will be limited to 30V by the MOSFET
5.5V 300kHz
(
)
The RSENSE resistors value can be calculated by using the
maximum current sense voltage specification with some
accomodation for tolerances:
50mV
11.4A
RSENSE
=
≈ 0.004Ω
BVDSS
.
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The power dissipation on the topside MOSFET can be
easily estimated. Using a Siliconix Si4420DY for example;
RDS(ON) = 0.013Ω, CRSS = 300pF. At maximum input
voltage with Tj (estimated) = 110°C at an elevated ambient
temperature:
The duty factor for this application is:
VO 1.8V
D.F. =
=
= 0.36
V
IN
5V
Using Figure 4, the RMS ripple current will be:
IINRMS = (20A)(0.23) = 4.6ARMS
1.8V
5.5V
PMAIN
=
10 2 1+ 0.005 110°C − 25°C
( ) )(
(
)
]
[
An input capacitor(s) with a 4.6ARMS ripple current rating
is required.
2
0.013Ω + 1.7 5.5V 10A 300pF
(
) (
)(
)
300kHz = 0.65W
(
)
The output capacitor ripple current is calculated by using
the inductor ripple already calculated for each inductor
andmultiplyingbythefactorobtainedfromFigure 3along
with the calculated duty factor. The output ripple in con-
tinuous mode will be highest at the maximum input
voltage since the duty factor is <50%. The maximum
output current ripple is:
The worst-case power disipated by the synchronous
MOSFET under normal operating conditions at elevated
ambient temperature and estimated 50°C junction tem-
perature rise is:
2
) (
5.5V − 1.8V
5.5V
= 1.29W
PSYNC
=
10A 1.48 0.013Ω
(
)(
)
VOUT
∆ICOUT
=
0.3 at 33%D. F.
(
)
fL
1.8V
Ashort-circuittogroundwillresultinafoldedbackcurrent
of about:
∆ICOUTMAX
=
0.3
300kHz 1.5µH
)(
= 1.2ARMS
(
)
200ns 5.5V
25mV
1
2
(
)
VOUTRIPPLE = 20mΩ 1.2A
= 24mV
RMS
(
)
ISC
=
+
= 7A
RMS
0.004Ω
1.5µH
The worst-case power disipated by the synchronous
MOSFET under short-circuit conditions at elevated ambi-
ent temperature and estimated 50°C junction temperature
rise is:
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1709. These items are also illustrated graphically in
the layout diagram of Figure 11. Check the following in
your layout:
5.5V − 1.8V
2
P
SYNC
=
7A 1.48 0.013Ω
(
) (
)(
)
5.5V
1) Are the signal and power grounds segregated? The
LTC1709 signal ground pin should return to the (–) plate
of COUT separately. The power ground returns to the
sources of the bottom N-channel MOSFETs, anodes of the
Schottky diodes, and (–) plates of CIN, which should have
as short lead lengths as possible.
= 630mW
which is less than half of the normal, full-load conditions.
Incidentally, since the load no longer dissipates power in
the shorted condition, total system power dissipation is
decreased by over 99%.
+
2) Does the LTC1709 VOS pin connect to the point of
–
load? Does the LTC1709 VOS pin connect to the load
return?
23
LTC1709
U
W
U U
APPLICATIO S I FOR ATIO
3)AretheSENSE– andSENSE+ leadsroutedtogetherwith
minimum PC trace spacing? The filter capacitors between
SENSE+ and SENSE– pin pairs should be as close as
possible to the LTC1709. Ensure accurate current sensing
with Kelvin connections at the current sense resistor.
finite impedances during the total period of the switching
regulator.ExternalOPTI-LOOPcompensationallowsover-
compensation for PC layouts which are not optimized but
this is not the recommended design procedure.
Simplified Visual Explanation of How a 2-Phase
Controller Reduces Both Input and Output RMS Ripple
Current
4) Does the (+) plate of CIN connect to the drains of the
topside MOSFETs and the (–) plate of CIN to the sources of
the bottom MOSFETS as closely as possible? This capaci-
tor provides the AC current to the MOSFETs. Keep the
input current path formed by the input capacitor, top and
bottom MOSFETs, and the Schottky diode on the same
side of the PC board in a tight loop to minimize conducted
and radiated EMI.
A multiphase power supply significantly reduces the
amount of ripple current in both the input and output
capacitors. Theeffectiveinputandoutputripplefrequency
is multiplied up by the number of phases used. Figure 11
graphically illustrates the principle.
The worst-case RMS ripple current for a single stage
design peaks at an input voltage of twice the output
voltage.Theworst-caseRMSripplecurrentforatwostage
design results in peak outputs of 1/4 and 3/4 of input
voltage. When the RMS current is calculated, higher
effective duty factor results and the peak current levels are
divided as long as the currents in each stage are balanced.
Refer to Application Note 77 for a detailed description of
howtocalculateRMScurrentforthemultiphaseswitching
regulator. Figures 3 and 4 help to illustrate how the input
and output currents are reduced by using an additional
phase. The input current peaks drop in half and the
frequency is doubled for this 2-phase converter. The input
capacity requirement is thus reduced theoretically by a
factor of four! Ceramic input capacitors with their
unbeatably low ESR characteristics can be used.
5) Is the INTVCC 1µF ceramic decoupling capacitor con-
nected closely between INTVCC and the PGND pin? This
capacitor carries the MOSFET driver peak currents. A
small value is recommended to allow placement immedi-
ately adjacent to the IC.
6) Keep the switching nodes, SW1 (SW2), away from
sensitive small-signal nodes. Ideally the switch nodes
should be placed at the furthest point from the LTC1709.
7)Usealowimpedancesourcesuchasalogicgatetodrive
the PLLIN pin and keep the lead as short as possible.
The diagram in Figure 10 illustrates all branch currents in
a 2-phase switching regulator. It becomes very clear after
studying the current waveforms why it is critical to keep
the high-switching-current paths to a small physical size.
High electric and magnetic fields will radiate from these
“loops” just as radio stations transmit signals. The output
capacitor ground should return to the negative terminal of
the input capacitor and not share a common ground path
with any switched current paths. The left half of the circuit
gives rise to the “noise” generated by a switching regula-
tor. The ground terminations of the sychronous MOSFETs
and Schottky diodes should return to the negative plate(s)
of the input capacitor(s) with a short isolated PC trace
since very high switched currents are present. A separate
isolated path from the negative plate(s) of the input
capacitor(s) should be used to tie in the IC power ground
pin (PGND) and the signal ground pin (SGND). This
technique keeps inherent signals generated by high cur-
rent pulses from taking alternate current paths that have
Figure 4 illustrates the RMS input current drawn from the
input capacitance vs the duty cycle as determined by the
ratio of input and output voltage. The peak input RMS
currentlevelofthesinglephasesystemisreducedby50%
in a 2-phase solution due to the current splitting between
the two stages.
An interesting result of the 2-phase solution is that the VIN
which produces worst-case ripple current for the input
capacitor, VOUT = VIN/2, in the single phase design pro-
duces zero input current ripple in the 2-phase design.
24
LTC1709
U
W U U
APPLICATIO S I FOR ATIO
The output ripple current is reduced significantly when
compared to the single phase solution using the same
inductance value because the VOUT/L discharge current
term from the stage that has its bottom MOSFET on
subtracts current from the (VIN - VOUT)/L charging current
resultingfromthestagewhichhasitstopMOSFETon. The
output ripple current is:
The input and output ripple frequency is increased by the
number of stages used, reducing the output capacity
requirements.WhenVIN isapproximatelyequalto2(VOUT
)
as illustrated in Figures 3 and 4, very low input and output
ripple currents result.
1− 2D 1−D
(
)
2VOUT
fL
∆IRIPPLE
=
1− 2D +1
where D is duty factor.
SW1
L1
R
SENSE1
D1
V
V
OUT
IN
R
IN
C
OUT
+
+
C
R
L
IN
SW2
L2
R
SENSE2
D2
BOLD LINES INDICATE
HIGH, SWITCHING
CURRENT LINES.
KEEP LINES TO A
MINIMUM LENGTH.
1709 F10
Figure 10. Instantaneous Current Path Flow in a Multiple Phase Switching Regulator
25
LTC1709
U
W
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APPLICATIO S I FOR ATIO
SINGLE PHASE
DUAL PHASE
SW V
SW1 V
SW2 V
I
CIN
I
L1
L2
I
COUT
I
I
CIN
I
COUT
RIPPLE
1709 F11
Figure 11. Single and 2-Phase Current Waveforms
26
LTC1709
U
PACKAGE DESCRIPTIO
Dimensions in inches (millimeters) unless otherwise noted.
G Package
36-Lead Plastic SSOP (0.209)
(LTC DWG # 05-08-1640)
12.67 – 12.93*
(0.499 – 0.509)
36 35 34 33 32 31 30 29 28 27 26 25 24 23 22 21 20 19
7.65 – 7.90
(0.301 – 0.311)
5
7
8
1
2
3
4
6
9 10 11 12 13 14 15 16 17 18
5.20 – 5.38**
(0.205 – 0.212)
1.73 – 1.99
(0.068 – 0.078)
0° – 8°
0.65
(0.0256)
BSC
0.13 – 0.22
0.55 – 0.95
(0.005 – 0.009)
(0.022 – 0.037)
0.05 – 0.21
(0.002 – 0.008)
0.25 – 0.38
(0.010 – 0.015)
NOTE: DIMENSIONS ARE IN MILLIMETERS
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.152mm (0.006") PER SIDE
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.254mm (0.010") PER SIDE
G36 SSOP 1098
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
27
LTC1709
U
TYPICAL APPLICATIO
L1
LTC1709
+
1
2
36
35
34
33
32
31
30
29
28
27
26
25
24
23
1.5µH
1000pF
0.004Ω
RUN/SS
NC
TG1
0.1µF
SENSE 1
SENSE 1
EAIN
0.22µF
D3
M1
M2
3
–
D1
SW1
10k
INTV
CC
4
MBRM
140T3
BOOST 1
10Ω
5
2.7k
51k
PLLFLTR
PLLIN
NC
V
IN
BG1
100pF
6
C
OUT
0.1µF
7
GND
4×180µF
4V
3.3nF
EXTV
CC
47k
15k
47µF×2
35V
8
1µF,25V
I
INTV
TH
SGND
CC
4.7µF
6.3V
9
V
IN
5V TO 28V
PGND
BG2
10
11
12
13
14
V
V
V
DIFFOUT
–
BOOST 2
SW2
OS
D2
MBRM
140T3
D4
+
10Ω
OS
–
+
SENSE 2
SENSE 2
TG2
0.22µF
M3
M4
AMPMD
0.004Ω
1000pF
V
OUT
15
16
17
18
22
21
20
19
1.3V TO 3.5V
L2
1.5µH
ATTENOUT
ATTENIN
VID0
V
BIAS
VID4
VID3
VID2
0.1µF
470pF
VID1
VID INPUTS
SWITCHING FREQUENCY = 310kHz
MI – M4: FAIRCHILD FDS7760A
L1 – L2: SUMIDA CEP125-1R5M
C
OUTPUT CAPACITORS: PANASONIC EEFUE0G181R
OUT
D3, D4: CENTRAL CMDSH-3TR
1709 TA02
Figure 12. 5V Input, 1.8V/20A Power Supply with Active Voltage Positioning
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DESCRIPTION
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COMMENTS
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LTC1438-ADJ
LTC1538-AUX
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LTC1702/LTC1703
Dual High Efficiency, 2-Phase Synchronous Step-Down Switching Regulator
LTC1709-7/
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High Efficiency, 2-Phase Synchronous Step-Down Switching Regulator with
5-Bit VID and Power Good Indication
1.3V ≤ V
1.1V ≤ V
≤ 3.5V (LTC1709-8),
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Current Mode Ensures Accurate Current Sharing,
3.5V ≤ V ≤ 36V
IN
LTC1708-PG
LTC1735
Dual High Efficiency, 2-Phase Synchronous Step-Down Switching Regulator
with 5-Bit VID and Power Good Indication
1.3V ≤ V
≤ 3.5V, Current Mode Ensures
OUT
Accurate Current Sharing, 3.5V ≤ V ≤ 36V
Burst ModeTM Operation, 16-Pin Narrow SSOP,
Fault Protection, 3.5V ≤ V ≤ 36V
IN
High Efficiency Synchronous Step-Down Controller
IN
LTC1736
High Efficiency Synchronous Step-Down Controller with 5-Bit VID
Output Fault Protection, Power Good, GN-24,
3.5V ≤ V ≤ 36V, 0.925V ≤ V
≤ 2V
IN
OUT
Adaptive Power and Burst Mode are trademarks of Linear Technology Corporation.
1709f LT/TP 0500 4K • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
28
LINEAR TECHNOLOGY CORPORATION 1999
●
●
(408)432-1900 FAX:(408)434-0507 www.linear-tech.com
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