LTC3112 [Linear]
15V, 200mA Synchronous Buck-Boost DC/DC Converter with 1.3μA Quiescent Current; 15V型,200mA同步降压 - 升压型DC / DC转换器1.3μA静态电流型号: | LTC3112 |
厂家: | Linear |
描述: | 15V, 200mA Synchronous Buck-Boost DC/DC Converter with 1.3μA Quiescent Current |
文件: | 总28页 (文件大小:382K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
LTC3129
15V, 200mA Synchronous
Buck-Boost DC/DC Converter
with 1.3µA Quiescent Current
FeaTures
DescripTion
n
Regulates V
Above, Below or Equal to V
The LTC®3129 is a high efficiency, 200mA buck-boost
OUT
IN
n
Wide V Range: 2.42V to 15V, 1.92V to 15V After
DC/DCconverterwithawideV andV range.Itincludes
IN
IN OUT
Start-Up (Bootstrapped)
anaccurateRUNpinthresholdtoallowpredictableregula-
tor turn-on and a maximum power point control (MPPC)
capability that ensures maximum power extraction from
non-ideal power sources such as photovoltaic panels.
n
n
n
n
n
n
n
n
n
n
n
n
n
Wide V
Range: 1.4V to 15.75V
OUT
200mA Output Current in Buck Mode
Single Inductor
1.3µA Quiescent Current
The LTC3129 employs an ultralow noise, 1.2MHz PWM
switchingarchitecturethatminimizessolutionfootprintby
allowing the use of tiny, low profile inductors and ceramic
capacitors. Built-in loop compensation and soft-start
simplify the design. For high efficiency operation at light
loads, automatic Burst Mode operation can be selected,
reducing the quiescent current to just 1.3µA.
Programmable Maximum Power Point Control
1.2MHz Ultralow Noise PWM
Current Mode Control
Pin Selectable Burst Mode® Operation
Up to 95% Efficiency
Accurate RUN Pin Threshold
Power Good Indicator
10nA Shutdown Current
Thermally Enhanced 3mm × 3mm QFN and
16-Lead MSOP Packages
Additionalfeaturesincludeapowergoodoutput, lessthan
10nA of shutdown current and thermal shutdown.
The LTC3129 is available in thermally enhanced 3mm ×
3mm QFN and 16-lead MSOP packages. For fixed output
voltageoptions,seethefunctionallyequivalentLTC3129-1,
which eliminates the need for an external feedback divider.
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and PowerPath is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.
applicaTions
n
Industrial Wireless Sensor Nodes
n
Post-Regulator for Harvested Energy
n
Solar Panel Post-Regulator/Charger
n
Intrinsically Safe Power Supplies
n
Wireless Microphones
n
Avionics-Grade Wireless Headsets
Typical applicaTion
22nF
22nF
Efficiency and Power Loss vs Load
10µH
100
90
80
70
60
50
40
30
20
10
0
1000
100
10
5V AT 200mA, V > 5V
IN
EFFICIENCY
5V AT 100mA, V < 5V
IN
BST1 SW1
SW2 BST2
2.42V TO 15V
V
V
V
OUT
V
IN
IN
OUT
10µF
10pF
LTC3129
3.32M
RUN
10µF
POWER LOSS
MPPC
PWM
FB
V
1
CC
PGOOD
1.02M
V
V
V
V
= 2.5V
= 3.6V
= 5V
IN
IN
IN
IN
0.1
V
CC
= 15V
V
= 5V
OUT
0.01
2.2µF
GND
PGND
0.01
0.1
1
10
100
1000
OUTPUT CURRENT (mA)
3129 TA01b
3129 TA01a
3129f
1
For more information www.linear.com/3129
LTC3129
absoluTe MaxiMuM raTings
(Notes 1, 8)
V , V
Voltages .................................... –0.3V to 18V
V , FB, PWM, MPPC Voltages.................... –0.3V to 6V
IN OUT
CC
SW1 DC Voltage............................ –0.3V to (V + 0.3V)
PGOOD Sink Current .............................................15mA
IN
SW2 DC Voltage..........................–0.3V to (V
+ 0.3V)
Operating Junction Temperature Range
OUT
SW1, SW2 Pulsed (<100ns) Voltage ..............–1V to 19V
BST1 Voltage ....................(SW1 – 0.3V) to (SW1 + 6V)
BST2 Voltage ....................(SW2 – 0.3V) to (SW2 + 6V)
RUN, PGOOD Voltages............................... –0.3V to 18V
(Notes 2, 5)............................................ –40°C to 125°C
Storage Temperature Range .................. –65°C to 150°C
MSE Lead Temperature (Soldering, 10 sec) ..........300°C
pin conFiguraTion
TOP VIEW
TOP VIEW
16 15 14 13
1
2
3
4
5
6
7
8
V
16 V
IN
CC
RUN
MPPC
GND
FB
15 BST1
14 SW1
13 PGND
12 SW2
11 BST2
BST1
1
2
3
4
12
V
OUT
V
11 PGOOD
IN
17
PGND
17
PGND
V
PWM
NC
10
9
CC
NC
RUN
NC
10
9
V
OUT
PGOOD
PWM
5
6
7
8
MSE PACKAGE
16-LEAD PLASTIC MSOP
T
= 125°C, θ = 10°C/W, θ = 40°C/W (NOTE 6)
JC JA
JMAX
UD PACKAGE
16-LEAD (3mm × 3mm) PLASTIC QFN
EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB
T
= 125°C, θ = 7.5°C/W, θ = 68°C/W (NOTE 6)
JC JA
JMAX
EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB
orDer inForMaTion
LEAD FREE FINISH
LTC3129EUD#PBF
LTC3129IUD#PBF
LTC3129EMSE#PBF
LTC3129IMSE#PBF
TAPE AND REEL
PART MARKING*
LGDR
PACKAGE DESCRIPTION
16-Lead (3mm × 3mm) Plastic QFN
TEMPERATURE RANGE
–40°C to 125°C
LTC3129EUD#TRPBF
LTC3129IUD#TRPBF
LTC3129EMSE#TRPBF
LTC3129IMSE#TRPBF
LGDR
16-Lead (3mm × 3mm) Plastic QFN
16-Lead Plastic MSOP
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
3129
3129
16-Lead Plastic MSOP
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3129f
2
For more information www.linear.com/3129
LTC3129
elecTrical characTerisTics The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). Unless otherwise noted, VIN = 12V, VOUT = 5V.
PARAMETER
Start-Up Voltage
CONDITIONS
MIN
TYP
MAX
2.42
15
UNITS
V
l
l
l
l
l
l
V
2.25
IN
Input Voltage Range
V
V
> 2.42V (Back-Driven)
> 2.42V (Back-Driven)
1.92
1.8
V
CC
CC
V
V
UVLO Threshold (Rising)
UVLO Hysteresis
1.9
2.0
V
IN
IN
80
100
130
15.75
1.199
10
mV
V
Output Voltage Adjust Range
Feedback Voltage
1.4
1.151
1.175
0.1
V
Feedback Input Current
FB = 1.25V
nA
nA
µA
Quiescent Current (V ) – Shutdown
RUN = 0V, Including Switch Leakage
10
100
3
IN
Quiescent Current (V ) UVLO
Either V or V Below Their UVLO Threshold, or
1.9
IN
IN
CC
RUN Below the Threshold to Enable Switching
Quiescent Current – Burst Mode Operation
Measured on V , FB > 1.25V
1.3
10
2.0
50
µA
nA
IN
PWM = 0V, RUN = V
IN
N-Channel Switch Leakage on V and V
SW1 = 0V, V = 15V
IN
IN
OUT
SW2 = 0V, V
RUN = 0V
= 15V
OUT
N-Channel Switch On-Resistance
Inductor Average Current Limit
V
= 4V
0.75
Ω
CC
l
l
V
V
> UV Threshold (Note 4)
< UV Threshold (Note 4)
220
80
275
130
350
200
mA
mA
OUT
OUT
l
l
Inductor Peak Current Limit
Maximum Boost Duty Cycle
(Note 4)
400
85
500
89
680
95
mA
%
FB = 1.10V. Percentage of Period SW2 is Low in
Boost Mode (Note 7)
l
l
Minimum Duty Cycle
FB = 1.25V. Percentage of Period SW1 is High in
Buck Mode (Note 7)
0
%
Switching Frequency
SW1 and SW2 Minimum Low Time
MPPC Voltage
PWM = V
(Note 3)
1.0
1.2
90
1.4
MHz
ns
V
CC
l
1.12
1.175
1
1.22
10
MPPC Input Current
MPPC = 5V
> 2.4V
nA
V
l
l
RUN Threshold to Enable V
0.5
1.16
50
0.9
1.22
80
1.15
1.28
120
10
CC
RUN Threshold to Enable Switching (Rising)
RUN (Switching) Threshold Hysteresis
RUN Input Current
V
V
CC
mV
nA
V
RUN = 15V
PWM = 5V
1
l
l
PWM Input High
1.6
PWM Input Low
0.5
1
V
PWM Input Current
0.1
3
µA
ms
V
Soft-Start Time
l
l
V
V
Voltage
V
IN
> 4.85V
3.4
4.1
4.7
CC
CC
Dropout Voltage (V – V
)
V
IN
V
IN
= 3.0V, Switching
35
0
60
2
mV
mV
IN
CC
= 2.0V (V in UVLO)
CC
V
V
V
V
V
V
V
UVLO Threshold (Rising)
UVLO Hysteresis
Current Limit
2.1
4
2.25
60
2.42
V
mV
mA
V
CC
CC
CC
CC
CC
CC
OUT
l
l
V
CC
= 0V
20
40
5.5
4
Back-Drive Voltage (Maximum)
Input Current (Back-Driven)
V
V
= 5.5V (Switching)
2
mA
µA
V
CC
Leakage to V if V > V
= 5.5V, V = 1.8V, Measured on V
IN
–7
IN
CC
IN
CC
IN
l
UV Threshold (Rising)
0.95
1.15
1.35
3129f
3
For more information www.linear.com/3129
LTC3129
elecTrical characTerisTics The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). Unless otherwise noted, VIN = 12V, VOUT = 5V.
PARAMETER
CONDITIONS
MIN
TYP
150
10
MAX
UNITS
mV
nA
V
OUT
V
OUT
V
OUT
V
OUT
UV Hysteresis
Current – Shutdown
Current – Sleep
Current – Active
RUN = 0V, V
= 15V Including Switch Leakage
100
OUT
PWM = 0V, FB = 1.25V
PWM = V , V = 15V (Note 4), FB = 1.25V
V
/27
OUT
µA
5
9
µA
CC OUT
PGOOD Threshold, Falling
PGOOD Hysteresis
PGOOD Voltage Low
PGOOD Leakage
Referenced to Programmed V
Referenced to Programmed V
Voltage
Voltage
–5.5
–7.5
2.5
250
1
–10
%
OUT
OUT
%
I
= 1mA
300
50
mV
nA
SINK
PGOOD = 15V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 4: Current measurements are made when the output is not switching.
Note 5: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may result in device degradation or failure.
Note 2: The LTC3129 is tested under pulsed load conditions such that
T ≈ T . The LTC3129E is guaranteed to meet specifications from
J
A
0°C to 85°C junction temperature. Specifications over the –40°C to
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC3129I is guaranteed over the full –40°C to 125°C operating junction
Note 6: Failure to solder the exposed backside of the package to the PC
board ground plane will result in a much higher thermal resistance.
Note 7: Switch timing measurements are made in an open-loop test
configuration. Timing in the application may vary somewhat from these
values due to differences in the switch pin voltage during non-overlap
durations when switch pin voltage is influenced by the magnitude and
duration of the inductor current.
Note 8: Voltage transients on the switch pin(s) beyond the DC limits
specified in the Absolute Maximum Ratings are non-disruptive to normal
operation when using good layout practices as described elsewhere in the
data sheet and application notes and as seen on the product demo board.
temperature range. The junction temperature (T ) is calculated from the
J
ambient temperature (T ) and power dissipation (P ) according to the
A
D
formula: T = T + (P • θ °C/W), where θ is the package thermal
J
A
D
JA
JA
impedance. Note that the maximum ambient temperature consistent
with these specifications, is determined by specific operating conditions
in conjunction with board layout, the rated thermal package thermal
resistance and other environmental factors.
Note 3: Specification is guaranteed by design and not 100% tested in
production.
TA = 25°C, unless otherwise noted.
Typical perForMance characTerisTics
Efficiency, VOUT = 2.5V
Power Loss, VOUT = 2.5V
Efficiency, VOUT = 3.3V
100
90
80
70
60
50
40
30
20
10
0
1000
100
10
100
90
80
70
60
50
40
30
20
10
0
BURST
BURST
PWM
PWM
PWM
1
V
V
V
V
V
= 2.5V
= 3.6V
= 5V
= 10V
= 15V
V
V
V
V
V
= 2.5V
= 3.6V
= 5V
= 10V
= 15V
V
IN
V
IN
V
IN
V
IN
V
IN
= 2.5V
= 3.6V
= 5V
= 10V
= 15V
IN
IN
IN
IN
IN
IN
IN
IN
IN
IN
BURST
0.1
0.01
0.01
0.1
1
10
100
1000
0.01
0.1
1
10
100
1000
0.01
0.1
1
10
100
1000
OUTPUT CURRENT (mA)
OUTPUT CURRENT (mA)
OUTPUT CURRENT (mA)
3129 G01
3129 G02
3129 G03
3129f
4
For more information www.linear.com/3129
LTC3129
TA = 25°C, unless otherwise noted.
Typical perForMance characTerisTics
Power Loss, VOUT = 3.3V
Efficiency, VOUT = 5V
Power Loss, VOUT = 5V
100
90
80
70
60
50
40
30
20
10
0
1000
100
10
1000
100
10
BURST
PWM
PWM
PWM
1
1
V
= 2.5V
= 3.6V
= 5V
= 10V
= 15V
V
V
V
V
V
= 2.5V
= 3.6V
= 5V
= 10V
= 15V
V
V
V
V
V
= 2.5V
= 3.6V
= 5V
= 10V
= 15V
IN
IN
IN
IN
IN
IN
IN
IN
IN
IN
IN
IN
IN
IN
IN
BURST
BURST
V
V
V
V
0.1
0.1
0.01
0.01
0.01
0.1
1
10
100
1000
0.01
0.1
1
10
100
1000
0.01
0.1
1
10
100
1000
OUTPUT CURRENT (mA)
OUTPUT CURRENT (mA)
OUTPUT CURRENT (mA)
3129 G05
3129 G06
3129 G04
Efficiency, VOUT = 12V
Power Loss, VOUT = 12V
Efficiency, VOUT = 15V
100
90
80
70
60
50
40
30
20
10
0
1000
100
10
100
90
80
70
60
50
40
30
20
10
0
BURST
BURST
PWM
PWM
BURST
PWM
1
V
= 2.5V
= 3.6V
= 5V
= 10V
= 15V
V
= 2.5V
= 3.6V
= 5V
= 10V
= 15V
V
= 2.5V
= 3.6V
= 5V
= 10V
= 15V
IN
IN
IN
IN
IN
IN
IN
IN
IN
IN
IN
IN
V
V
V
0.1
V
V
V
V
V
V
V
V
V
IN
IN
IN
0.01
0.01
0.1
1
10
100
1000
0.01
0.1
1
10
100
1000
0.01
0.1
1
10
100
1000
OUTPUT CURRENT (mA)
OUTPUT CURRENT (mA)
OUTPUT CURRENT (mA)
3129 G07
3129 G08
3129 G09
Maximum Output Current
vs VIN and VOUT
No Load Input Current
vs VIN and VOUT (PWM = 0V)
Power Loss, VOUT = 15V
250
200
150
100
50
30
25
1000
100
10
V
V
V
V
= 2.5V
OUT
OUT
OUT
OUT
= 5V
= 10V
= 15V
PWM
FB DIVIDER CURRENT = 2µA
20
15
10
5
V
V
V
V
V
V
V
V
= 2.5V
OUT
OUT
OUT
OUT
OUT
OUT
OUT
OUT
BURST
= 3.3V
= 4.1V
= 5V
1
V
= 2.5V
= 3.6V
= 5V
= 10V
= 15V
IN
IN
IN
IN
= 6.9V
= 8.2V
= 12V
= 15V
V
0.1
V
V
V
IN
0
0
0.01
2
3
4
5
6
7
8
9
10 11 12 13 14 15
2.5
4.5
6.5
8.5 10.5 12.5 14.5
(V)
0.01
0.1
1
10
100
1000
V
(V)
V
OUTPUT CURRENT (mA)
IN
IN
3129 G11
3129 G12
3129 G10
3129f
5
For more information www.linear.com/3129
LTC3129
TA = 25°C, unless otherwise noted.
Typical perForMance characTerisTics
FB Voltage vs Temperature
(Normalized to 25°C)
Burst Mode Threshold
vs VIN and VOUT
Switch RDS(ON) vs Temperature
1.00
0.75
0.50
0.25
0
80
70
60
50
40
30
1.3
1.2
1.1
1.0
0.9
0.8
0.7
0.6
0.5
0.4
V
CC
V
CC
V
CC
V
CC
= 2.5V
= 3V
= 4V
= 5V
V
V
V
V
V
V
V
V
= 2.5V
= 3.3V
= 4.1V
= 5V
OUT
OUT
OUT
OUT
OUT
OUT
OUT
OUT
–0.25
–0.50
–0.75
–1.00
20
10
= 6.9V
= 8.2V
= 12V
= 15V
0
–45 –20
5
30
55
80 105 130
–45 –20
5
30
55
80 105 130
2
4
6
8
10
(V)
12
14
16
TEMPERATURE (°C)
TEMPERATURE (°C)
V
IN
3129 G15
3129 G14
3129 G13
Maximum Output Current
vs Temperature (Normalized to 25°C)
Average Input Current Limit
vs MPPC Voltage
Accurate RUN Threshold
vs Temperature (Normalized to 25°C)
2
1
100
90
80
70
60
50
40
30
20
10
0
15
10
5
0
0
–5
–10
–15
–1
–2
–45 –20
5
30
55
80 105 130
1.13 1.135 1.14 1.145 1.15 1.155 1.16 1.165 1.17
–45 –20
5
30
55
80 105 130
TEMPERATURE (°C)
MPPC PIN VOLTAGE (V)
TEMPERATURE (°C)
3129 G16
3129 G17
3129 G18
VCC Dropout Voltage vs Temperature
(PWM Mode, Switching)
VCC Dropout Voltage vs VIN
(PWM Mode, Switching)
Fixed Frequency PWM
Waveforms
60
50
40
30
20
10
0
60
SW2
5V/DIV
50
40
30
20
10
0
SW1
5V/DIV
I
L
200mA/DIV
3129 G21
500ns/DIV
L = 10µH
V
V
= 7V
IN
= 5V
OUT
OUT
–45 –20
5
30
55
80 105 130
2
2.25 2.5 2.75
3
3.25 3.5 3.75
4
I
= 200mA
TEMPERATURE (°C)
V
(V)
IN
3129 G19
3129 G20
3129f
6
For more information www.linear.com/3129
LTC3129
TA = 25°C, unless otherwise noted.
Typical perForMance characTerisTics
Burst Mode Ripple on VOUT
Fixed Frequency Ripple on VOUT
Burst Mode Waveforms
SW1
5V/DIV
V
OUT
V
OUT
100mV/DIV
20mV/DIV
SW2
5V/DIV
I
L
200mA/DIV
I
I
L
L
200mA/DIV
100mA/DIV
3129 G23
3129 G22
3129 G24
50µs/DIV
L = 10µH
200ns/DIV
100µs/DIV
L = 10µH
L = 10µH
V
V
I
= 7V
V
V
I
= 7V
V
V
I
= 7V
IN
OUT
IN
OUT
IN
OUT
OUT
= 5V
= 5V
= 5mA
= 22µF
= 5V
= 5mA
= 200mA
= 10µF
OUT
OUT
OUT
C
C
C
= 22µF (WITH THE RECOMMENDED
FEEDFORWARD CAPACITOR)
OUT
OUT
Step Load Transient Response in
Burst Mode Operation
Step Load Transient Response in
Fixed Frequency
Start-Up Waveforms
V
OUT
5V/DIV
V
V
OUT
100mV/DIV
OUT
100mV/DIV
V
CC
5V/DIV
RUN
5V/DIV
I
VOUT
I
I
VOUT
VIN
100mA/DIV
100mA/DIV
200mA/DIV
3129 G25
3129 G26
3129 G27
1ms/DIV
500µs/DIV
500µs/DIV
V
V
I
= 7V
L = 10µH
IN
OUT
= 5V
V
V
C
I
= 7V
= 7V
IN
OUT
OUT
= 50mA
= 22µF
= 5V
= 10µF
OUT
OUT
C
= 22µF (WITH THE RECOMMENDED
= 50mA to 150mA STEP
FEEDFORWARD CAPACITOR)
OUT
= 5mA to 125mA STEP
PGOOD Response to a Drop
On VOUT
MPPC Response to a Step Load
V
OUT
2V/DIV
PGOOD
2V/DIV
V
IN
2V/DIV
V
OUT
2V/DIV
I
VOUT
100mA/DIV
3129 G28
3129 G29
1ms/DIV
2ms/DIV
SET TO 3.5V
V
= 5V
V
V
C
V
= 5V
OC
MPPC
= 22µF, R = 10Ω,
OUT
OUT
OUT
IN
IN
IN
= 5V, C
= 22µF
OUT
I
= 25mA to 125mA STEP
3129f
7
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LTC3129
pin FuncTions (QFN/MSOP)
BST1 (Pin 1/Pin 15): Bootstrapped Floating Supply for
High Side NMOS Gate Drive. Connect to SW1 through a
22nF capacitor, as close to the part as possible. The value
is not critical. Any value from 4.7nF to 47nF may be used.
V
= 1.175V • [1+(R1/R2)]. Note this pin is very noise
OUT
sensitive, therefore minimize trace length and stray ca-
pacitance.
NC (Pins 8, 9/Pins 6, 7): Unused. These pins should be
grounded.
V (Pin2/Pin16):InputVoltagefortheConverter.Connect
IN
a minimum of 4.7µF ceramic decoupling capacitor from
PWM (Pin 10/Pin 8): Mode Select Pin.
thispintothegroundplane, asclosetothepinaspossible.
PWM = Low (ground): Enables automatic Burst Mode
operation.
V
CC
(Pin 3/Pin 1): Output voltage of the internal voltage
regulator. This is the supply pin for the internal circuitry.
Bypass this output with a minimum of 2.2µF ceramic
capacitor close to the pin. This pin may be back-driven
by an external supply, up to a maximum of 5.5V.
PWM = High (tie to V ): Fixed frequency PMW opera-
CC
tion.
This pin should not be allowed to float. It has an internal
5M pull-down resistor.
RUN (Pin 4/Pin 2): Input to the Run Comparator. Pull this
pinabove1.1VtoenabletheV regulatorandabove1.28V
CC
PGOOD (Pin 11/Pin 9): Open drain output that pulls to
ground when FB drops too far below its regulated volt-
age. Connect a pull-up resistor from this pin to a positive
supply. This pin can sink up to the absolute maximum
rating of 15mA when low. Note that this pin is forced low
to enable the converter. Connecting this pin to a resistor
divider from V to ground allows programming a V
IN
IN
start threshold higher than the 1.8V (typical) V UVLO
IN
threshold. In this case, the typical V turn-on threshold is
IN
determined by V = 1.22V • [1+(R3/R4)] (see Figure 2).
IN
in shutdown or V UVLO.
CC
MPPC (Pin 5/Pin 3): Maximum Power Point Control Pro-
V
(Pin 12/Pin 10): Output voltage of the converter.
OUT
gramming Pin. Connect this pin to a resistor divider from
Connectaminimumvalueof4.7µFceramiccapacitorfrom
V togroundtoenabletheMPPCfunctionality. IftheV
IN
OUT
thispintothegroundplane, asclosetothepinaspossible.
load is greater than what the power source can provide,
BST2 (Pin 13/Pin 11): Bootstrapped floating supply for
high side NMOS gate drive. Connect to SW2 through a
22nF capacitor, as close to the part as possible. The value
is not critical. Any value from 4.7nF to 47nF may be used.
the MPPC will reduce the inductor current to regulate V
IN
to a voltage determined by: V = 1.175V • [1+(R5/R6)]
IN
(see Figure 3). By setting the V regulation voltage appro-
IN
priately, maximum power transfer from the limited source
is assured. Note this pin is very noise sensitive, therefore
minimize trace length and stray capacitance. Please refer
to the Applications Information section for more detail
on programming the MPPC for different sources. If this
SW2 (Pin 14/Pin 12): Switch Pin. Connect to one side of
the inductor. Keep PCB trace lengths as short and wide
as possible to reduce EMI.
PGND (Pin 15, Exposed Pad Pin 17/Pin 13, Exposed
Pad Pin 17): Power Ground. Provide a short direct PCB
path between PGND and the ground plane. The exposed
pad must also be soldered to the PCB ground plane. It
serves as a power ground connection, and as a means of
conducting heat away from the die.
function is not needed, tie the pin to V .
CC
GND (Pin 6/Pin 4): Signal Ground. Provide a short direct
PCB path between GND and the ground plane where the
exposed pad is soldered.
FB (Pin 7/Pin 5): Feedback Input to the Error Amplifier.
Connect to a resistor divider from V
to ground. The
SW1 (Pin 16/Pin 14): Switch Pin. Connect to one side of
the inductor. Keep PCB trace lengths as short and wide
as possible to reduce EMI.
OUT
output voltage can be adjusted from 1.4V to 15.75V by:
3129f
8
For more information www.linear.com/3129
LTC3129
block DiagraM
BST1
SW1
SW2
BST2
V
IN
V
IN
V
CC
V
REF
LDO
V
CC_GD
V
OUT
START
V
OUT
A
B
DRIVER
DRIVER
V
CC
V
4.1V
CC
V
CC
I
I
SENSE
SENSE
D
DRIVER
1.175V
V
START
RUN
V
REF
REF
NC
NC
V
REF_GD
C
DRIVER
DRV_C
+
–
START
DRV_B
0.9V
DRV_A
DRV_D
+
–
SD
I
SENSE
+
–
–
+
UV
FB
1.22V
500mA
I
V
IN
1.1V
LIM
1.175V
LOGIC
ENABLE
I
SENSE
–
+
UVLO
–
+
V
C
I
–
+
I
ZERO
SENSE
–
+
+
–
PWM
1.175V
20mA
THERMAL
SHUTDOWN
RESET
SOFT-START
OSC
MPPC
PWM
+
–
1.175V
PGOOD
–
+
–
+
600mV
CLAMP
5M
–
+
SLEEP
–7.5%
100mV
GND
PGND
3129 BD
3129f
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For more information www.linear.com/3129
LTC3129
operaTion
INTRODUCTION
PWM MODE OPERATION
IfthePWMpinishighoriftheloadcurrentontheconverter
is high enough to command PWM mode operation with
PWM low, the LTC3129 operates in a fixed 1.2MHz PWM
mode using an internally compensated average current
mode control loop. PWM mode minimizes output voltage
ripple and yields a low noise switching frequency spec-
trum. A proprietary switching algorithm provides seam-
less transitions between operating modes and eliminates
discontinuities in the average inductor current, inductor
ripple current and loop transfer function throughout all
modes of operation. These advantages result in increased
efficiency,improvedloopstabilityandloweroutputvoltage
rippleincomparisontothetraditionalbuck-boostconverter.
TheLTC3129isa1.3µAquiescentcurrent,monolithic,cur-
rent mode, buck-boost DC/DC converter that can operate
overawideinputvoltagerangeof1.92Vto15Vandprovide
up to 200mA to the load. Internal, low R
N-channel
DS(ON)
power switches reduce solution complexity and maximize
efficiency.Aproprietaryswitchcontrolalgorithmallowsthe
buck-boostconvertertomaintainoutputvoltageregulation
with input voltages that are above, below or equal to the
output voltage. Transitions between the step-up or step-
downoperatingmodesareseamlessandfreeoftransients
and sub-harmonic switching, making this product ideal
for noise sensitive applications. The LTC3129 operates
at a fixed nominal switching frequency of 1.2MHz, which
providesanidealtrade-offbetweensmallsolutionsizeand
high efficiency. Current mode control provides inherent
input line voltage rejection, simplified compensation and
rapid response to load transients.
Figure 1 shows the topology of the LTC3129 power stage
which is comprised of four N-channel DMOS switches
and their associated gate drivers. In PWM mode operation
both switch pins transition on every cycle independent of
the input and output voltages. In response to the internal
control loop command, an internal pulse width modulator
generates the appropriate switch duty cycle to maintain
regulation of the output voltage.
Burst Mode capability is also included in the LTC3129 and
is user-selected via the PWM input pin. In Burst Mode
operation, the LTC3129 provides exceptional efficiency at
light output loading conditions by operating the converter
only when necessary to maintain voltage regulation. The
BurstModequiescentcurrentisamiserly1.3µA. Athigher
loads, the LTC3129 automatically switches to fixed fre-
quencyPWMmodewhenBurstModeoperationisselected.
(Please refer to the Typical Performance Characteristics
curves for the mode transition point at different input and
output voltages.) If the application requires extremely low
noise, continuous PWM operation can also be selected
via the PWM pin.
C
BST1
C
BST2
L
BST1
V
A
SW1
SW2
D
V
OUT
BST2
IN
V
CC
V
CC
V
CC
V
CC
B
C
PGND
PGND
A MPPC (maximum power point control) function is also
provided that allows the input voltage to the converter to
be servo'd to a programmable point for maximum power
when operating from various non-ideal power sources
such as photovoltaic cells. The LTC3129 also features
an accurate RUN comparator threshold with hysteresis,
allowing the buck-boost DC/DC converter to turn on and
LTC3129
3129 F01
Figure 1. Power Stage Schematic
off at user-selected V voltage thresholds. With a wide
IN
voltage range, 1.3µA Burst Mode current and program-
mable RUN and MPPC pins, the LTC3129 is well suited
for many diverse applications.
3129f
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For more information www.linear.com/3129
LTC3129
operaTion
When stepping down from a high input voltage to a lower
output voltage, the converter operates in buck mode and
switch D remains on for the entire switching cycle except
fortheminimumswitchlowduration(typically90ns).Dur-
ing the switch low duration, switch C is turned on which
outputs are used to control the duty cycle of the switch
pins on a cycle-by-cycle basis.
The voltage error amplifier monitors the output voltage,
V
OUT
through a voltage divider and makes adjustments to
thecurrentcommandasnecessarytomaintainregulation.
The voltage error amplifier therefore controls the outer
voltage regulation loop. The average current amplifier
makes adjustments to the inductor current as directed by
forces SW2 low and charges the flying capacitor, C
.
BST2
This ensures that the switch D gate driver power supply
rail on BST2 is maintained. The duty cycle of switches A
and B are adjusted to maintain output voltage regulation
in buck mode.
the voltage error amplifier output via V and is commonly
C
referred to as the inner current loop amplifier.
If the input voltage is lower than the output voltage, the
converter operates in boost mode. Switch A remains on
for the entire switching cycle except for the minimum
switch low duration (typically 90ns). During the switch
low duration, switch B is turned on which forces SW1
The average current mode control technique is similar to
peak current mode control except that the average current
amplifier, by virtue of its configuration as an integrator,
controls average current instead of the peak current. This
difference eliminates the peak to average current error
inherent to peak current mode control, while maintaining
most of the advantages inherent to peak current mode
control.
low and charges the flying capacitor, C
. This ensures
BST1
that the switch A gate driver power supply rail on BST1 is
maintained.ThedutycycleofswitchesCandDareadjusted
to maintain output voltage regulation in boost mode.
Average current mode control requires appropriate com-
pensation for the inner current loop, unlike peak current
mode control. The compensation network must have high
DC gain to minimize errors between the actual and com-
manded average current level, high bandwidth to quickly
change the commanded current level following transient
load steps and a controlled mid-band gain to provide a
form of slope compensation unique to average current
mode control. The compensation components required
to ensure proper operation have been carefully selected
and are integrated within the LTC3129.
Oscillator
The LTC3129 operates from an internal oscillator with a
nominalfixedfrequencyof1.2MHz. ThisallowstheDC/DC
converterefficiencytobemaximizedwhilestillusingsmall
external components.
Current Mode Control
The LTC3129 utilizes average current mode control for the
pulsewidthmodulator.Currentmodecontrol,bothaverage
and the better known peak method, enjoy some benefits
compared to other control methods including: simplified
loop compensation, rapid response to load transients and
inherent line voltage rejection.
Inductor Current Sense and Maximum Output Current
As part of the current control loop required for current
mode control, the LTC3129 includes a pair of current
sensing circuits that measure the buck-boost converter
inductor current.
Referring to the Block Diagram, a high gain, internally
compensated transconductance amplifier monitors Vout
through a voltage divider connected to the FB pin. The
error amplifier output is used by the current mode control
loop to command the appropriate inductor current level.
The inverting input of the internally compensated average
current amplifier is connected to the inductor current
sense circuit. The average current amplifier's output is
compared to the oscillator ramps, and the comparator
Thevoltageerroramplifieroutput,V ,isinternallyclamped
C
to a nominal level of 0.6V. Since the average inductor
current is proportional to V , the 0.6V clamp level sets
C
the maximum average inductor current that can be pro-
grammed by the inner current loop. Taking into account
the current sense amplifier's gain, the maximum average
3129f
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LTC3129
operaTion
inductor current is approximately 275mA (typical). In
Overload Current Limit and I
Comparator
ZERO
Buck mode, the output current is approximately equal to
The internal current sense waveform is also used by the
peakoverloadcurrent(I )andzerocurrent(I )com-
the inductor current I .
L
PEAK
ZERO
I
≈ I • 0.89
parators. The I
current comparator monitors Isense
OUT(BUCK)
L
PEAK
and turns off switch A if the inductor current level exceeds
its maximum internal threshold, which is approximately
500mA. An inductor current level of this magnitude will
occur during a fault, such as an output short-circuit, or
during large load or input voltage transients.
The 90ns SW1/SW2 forced low time on each switching
cycle briefly disconnects the inductor from V and V
resulting in about 11% less output current in either buck
orboostmodeforagiveninductorcurrent.Inboostmode,
the output current is related to average inductor current
and duty cycle by:
OUT
IN
TheLTC3129featuresneardiscontinuousinductorcurrent
operation at light output loads by virtue of the I
com-
ZERO
I
≈ I • (1 – D) • Efficiency,
L
OUT(BOOST)
parator circuit. By limiting the reverse current magnitude
in PWM mode, a balance between low noise operation and
where D is the converter duty cycle.
improved efficiency at light loads is achieved. The I
ZERO
Since the output current in boost mode is reduced by the
duty cycle (D), the output current rating in buck mode is
always greater than in boost mode. Also, because boost
mode operation requires a higher inductor current for a
givenoutputcurrentcomparedtobuckmode,theefficiency
comparator threshold is set near the zero current level in
PWMmode,andasaresult,thereversecurrentmagnitude
will be a function of inductance value and output voltage
due to the comparator's propagation delay. In general,
higher output voltages and lower inductor values will
result in increased reverse current magnitude.
in boost mode will be lower due to higher I ² • R
L
DS(ON)
losses in the power switches. This will further reduce the
outputcurrentcapabilityinboostmode.Ineitheroperating
mode, however, the inductor peak-to-peak ripple current
does not play a major role in determining the output cur-
rent capability, unlike peak current mode control.
In automatic Burst Mode operation (PWM pin low), the
I
comparator threshold is increased so that reverse
ZERO
inductor current does not normally occur. This maximizes
efficiency at very light loads.
With peak current mode control, the maximum output
current capability is reduced by the magnitude of inductor
ripplecurrentbecausethepeakinductorcurrentlevelisthe
control variable, but the average inductor current is what
determines the output current. The LTC3129 measures
and controls average inductor current, and therefore, the
inductor ripple current magnitude has little effect on the
maximum current capability in contrast to an equivalent
peak current mode converter. Under most conditions in
buck mode, the LTC3129 is capable of providing a mini-
mum of 200mA to the load. In boost mode, as described
previously, the output current capability is related to the
Burst Mode OPERATION
When the PWM pin is held low, the LTC3129 is configured
for automatic Burst Mode operation. As a result, the buck-
boost DC/DC converter will operate with normal continu-
ous PWM switching above a predetermined minimum
output load and will automatically transition to power
saving Burst Mode operation below this output load level.
Note that if the PWM pin is low, reverse inductor current is
not allowed at any load. Refer to the Typical Performance
Characteristics section to determine the Burst Mode
transition threshold for various combinations of V and
IN
boost ratio or duty cycle (D). For example, for a 3.6V V
IN
V
. If PWM is low, at light output loads, the LTC3129
OUT
to 5V output application, the LTC3129 can provide up
to 150mA to the load. Refer to the Typical Performance
characteristics section for more detail on output current
capability.
will go into a standby or sleep state when the output volt-
age achieves its nominal regulation level. The sleep state
halts PWM switching and powers down all non-essential
3129f
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LTC3129
operaTion
functions of the IC, significantly reducing the quiescent
current of the LTC3129 to just 1.3µA typical. This greatly
improves overall power conversion efficiency when the
output load is light. Since the converter is not operating
in sleep, the output voltage will slowly decay at a rate
determined by the output load resistance and the output
capacitor value. When the output voltage has decayed by
a small amount, typically 1%, the LTC3129 will wake and
resume normal PWM switching operation until the volt-
logic threshold. The V regulator includes current-limit
CC
protectiontosafeguardagainstaccidentalshort-circuiting
of the V rail.
CC
Undervoltage Lockout (UVLO)
Therearetwoundervoltagelockout(UVLO)circuitswithin
the LTC3129 that inhibit switching; one that monitors V
IN
and another that monitors V . Either UVLO will disable
CC
operation of the internal power switches and keep other
age on V
is restored to the previous level. If the load
OUT
IC functions in a reset state if either V or V are below
IN
CC
is very light, the LTC3129 may only need to switch for a
few cycles to restore V and may sleep for extended
their respective UVLO thresholds.
OUT
The V UVLO comparator has a falling voltage threshold
IN
periods of time, significantly improving efficiency. If the
load is suddenly increased above the burst transition
threshold, the part will automatically resume continuous
PWM operation until the load is once again reduced.
of 1.8V (typical). If V falls below this level, IC operation
IN
is disabled until V rises above 1.9V (typical), as long as
IN
the V voltage is above its UVLO threshold.
CC
The V UVLO has a falling voltage threshold of 2.19V
CC
A feedforward capacitor on the feedback divider can be
(typical). If the V voltage falls below this threshold, IC
CC
used to reduce Burst Mode V
ripple. This is discussed
OUT
operation is disabled until V rises above 2.25V (typical)
CC
in more detail in the Applications Information section of
this data sheet.
as long as V is above its nominal UVLO threshold level.
IN
Depending on the particular application, either of these
UVLO thresholds could be the limiting factor affecting the
minimuminputvoltagerequiredforoperation.Becausethe
Note that Burst Mode operation is inhibited until soft-start
is done, the MPPC pin is greater than 1.175V and V
has reached regulation.
OUT
V
regulator uses V for its power input, the minimum
CC
IN
input voltage required for operation is determined by the
Soft-Start
V
minimum voltage, as input voltage (V ) will always
CC
IN
The LTC3129 soft-start circuit minimizes input current
transientsandoutputvoltageovershootoninitialpowerup.
The required timing components for soft-start are internal
to the LTC3129 and produce a nominal soft-start dura-
tion of approximately 3ms. The internal soft-start circuit
be higher than V in the normal (non-bootstrapped)
CC
configuration. Therefore, the minimum V for the part
to startup is 2.25V (typical).
IN
In applications where V is bootstrapped (powered
CC
through a Schottky diode by either V
or an auxiliary
slowly ramps the error amplifier output, V . In doing so,
OUT
C
power rail), the minimum input voltage for operation will
the current command of the IC is also slowly increased,
starting from zero. It is unaffected by output loading or
output capacitor value. Soft-start is reset by the UVLO on
both V and V , the RUN pin and thermal shutdown.
be limited only by the V UVLO threshold (1.8V typical).
IN
Please note that if the bootstrap voltage is derived from
theLTC3129V
andnotanindependentpowerrail, then
OUT
IN
CC
the minimum input voltage required for initial startup is
still 2.25V (typical).
V
Regulator
CC
Aninternallowdropoutregulator(LDO)generatesanomi-
nal 4.1V V rail from V . The V rail powers the internal
Note that if either V or V are below their UVLO thresh-
IN CC
olds, or if RUN is below its accurate threshold of 1.22V
(typical), then the LTC3129 will remain in a soft shutdown
CC
IN
CC
control circuitry and the gate drivers of the LTC3129. The
regulator is disabled in shutdown to reduce quiescent
V
state, where the V quiescent current will be only 1.9µA
CC
IN
current and is enabled by raising the RUN pin above its
typical.
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LTC3129
operaTion
V
OUT
Undervoltage
With the addition of an optional resistor divider as shown
in Figure 2, the RUN pin can be used to establish a user-
programmableturn-onandturn-offthreshold.Thisfeature
can be utilized to minimize battery drain below a certain
input voltage, or to operate the converter in a hiccup mode
from very low current sources.
There is also an undervoltage comparator that monitors
the output voltage. Until V reaches 1.15V (typical), the
OUT
average current limit is reduced by a factor of two. This
reduces power dissipation in the device in the event of a
shorted output. In addition, N-channel switch D, which
feeds V , will be disabled until V
exceeds 1.15V.
OUT
OUT
LTC3129
V
ACCURATE THRESHOLD
IN
RUN Pin Comparator
1.22V
–
+
ENABLE SWITCHING
R3
RUN
In addition to serving as a logic level input to enable cer-
tain functions of the IC, the RUN pin includes an accurate
internal comparator that allows it to be used to set custom
rising and falling ON/OFF thresholds with the addition of
an optional external resistor divider. When RUN is driven
+
–
R4
ENABLE LDO AND
CONTROL CIRCUITS
0.9V
LOGIC THRESHOLD
above its logic threshold (0.9V typical), the V regulator
CC
3129 F02
is enabled, which provides power to the internal control
circuitryoftheIC.IfthevoltageonRUNisincreasedfurther
so that it exceeds the RUN comparator's accurate analog
threshold (1.22V typical), all functions of the buck-boost
converterwillbeenabledandastart-upsequencewillensue,
Figure 2. Accurate RUN Pin Comparator
Note that once RUN is above 0.9V typical, the quiescent
input current on V (or V if back-driven) will increase to
IN
CC
about 1.9µA typical until the V and V UVLO thresholds
IN
CC
assuming the V and V UVLO thresholds are satisfied.
IN
CC
are satisfied.
IfRUNisbroughtbelowtheaccuratecomparatorthreshold,
thebuck-boostconverterwillinhibitswitching,buttheV
TheconverterisenabledwhenthevoltageonRUNexceeds
1.22V (nominal). Therefore, the turn-on voltage threshold
on V is given by:
CC
regulator and control circuitry will remain powered unless
RUN is brought below its logic threshold. Therefore, in
order to completely shut down the IC and reduce the Vin
current to 10nA (typical), it is necessary to ensure that
RUNisbroughtbelowitsworstcaselowlogicthresholdof
0.5V. RUN is a high voltage input and can be tied directly
IN
V
= 1.22V • (1 + R3/R4)
IN(TURN-ON)
The RUN comparator includes a built-in hysteresis of
approximately 80mV, so that the turn off threshold will
be 1.14V.
to V to continuously enable the IC when the input supply
IN
There may be cases due to PCB layout, very large value
resistorsforR3andR4, orproximitytonoisycomponents
wherenoisepickupmaycausetheturn-onorturn-offofthe
IC to be intermittent. In these cases, a small filter capaci-
tor can be added across R4 to ensure proper operation.
is present. Also note that RUN can be driven above V
IN
or V
as long as it stays within the operating range of
OUT
the IC (up to 15V).
3129f
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LTC3129
operaTion
PGOOD Comparator
a minimum input voltage when using high resistance
sources, such as photovoltaic panels, so as to maximize
The LTC3129 provides an open-drain PGOOD output that
input power transfer and prevent V from dropping too
IN
pulls low if V
falls more than 7.5% (typical) below its
OUT
low under load. Referring to Figure 3, the MPPC pin is
programmedvalue.WhenV risestowithin5%(typical)
OUT
internally connected to the non-inverting input of a g
m
of its programmed value, the internal PGOOD pull-down
will turn off and PGOOD will go high if an external pull-
up resistor has been provided. An internal filter prevents
amplifier,whoseinvertinginputisconnectedtothe1.175V
reference. If the voltage at MPPC, using the external volt-
age divider, falls below the reference voltage, the output of
nuisance trips of PGOOD due to short transients on V
.
OUT
the amplifier pulls the internal V node low. This reduces
C
Note that PGOOD can be pulled up to any voltage, as long
as the absolute maximum rating of 18V is not exceeded,
and as long as the maximum sink current rating is not
exceeded when PGOOD is low. Note that PGOOD will
the commanded average inductor current so as to reduce
the input current and regulate V to the programmed
IN
minimum voltage, as given by:
also be driven low if V is below its UVLO threshold or
V
= 1.175V • (1 + R5/R6)
CC
IN(MPPC)
if the part is in shutdown (RUN below its logic threshold)
Note that external compensation should not be required
while V is being held up (or back-driven). PGOOD is
CC
for MPPC loop stability if input filter capacitor, C , is at
IN
not affected by V UVLO or the accurate RUN threshold.
IN
least 22µF.
Maximum Power-Point Control (MPPC)
The divider resistor values can be in the MΩ range to
minimize the input current in very low power applications.
However, straycapacitanceandnoisepickupontheMPPC
pin must also be minimized.
The MPPC input of the LTC3129 can be used with an
optional external voltage divider to dynamically adjust
the commanded inductor current in order to maintain
*C
IN
V
IN
R
R5
R6
LTC3129
S
MPPC
+
–
+
V
SOURCE
–
1.175V
+
–
V
* C SHOULD BE AT
IN
C
FB
CURRENT
LEAST 22µF FOR
COMMAND
MPPC APPLICATIONS
VOLTAGE
ERROR AMP
3129 F03
Figure 3. MPPC Amplifier with External Resistor Divider
3129f
15
For more information www.linear.com/3129
LTC3129
operaTion
The MPPC pin controls the converter in a linear fashion
when using sources that can provide a minimum of 5mA
to 10mA of continuous input current. For operation from
weaker input sources, refer to the Applications Informa-
tion section to see how the programmable RUN pin can
be used to control the converter in a hysteretic manner to
provide an effective MPPC function for sources that can
provide as little as 5µA or less. If the MPPC function is not
and significantly improve efficiency. As a result, careful
consideration must be given to the thermal environment
of the IC in order to provide a means to remove heat from
the IC and ensure that the LTC3129 is able to provide its
full rated output current. Specifically, the exposed die
attach pad of both the QFN and MSE packages must be
soldered to a copper layer on the PCB to maximize the
conduction of heat out of the IC package. This can be ac-
complished by utilizing multiple vias from the die attach
pad connection underneath the IC package to other PCB
layer(s) containing a large copper plane. A typical board
layout incorporating these concepts is shown in Figure 4.
required, the MPPC pin should be tied to V .
CC
Thermal Considerations
The power switches of the LTC3129 are designed to oper-
ate continuously with currents up to the internal current
limit thresholds. However, when operating at high current
levels, there may be significant heat generated within the
IftheICdietemperatureexceedsapproximately180°C,over
temperature shutdown will be invoked and all switching
will be inhibited. The part will remain disabled until the die
temperature cools by approximately 10°C. The soft-start
circuit is re-initialized in overtemperature shutdown to
provide a smooth recovery when the IC die temperature
cools enough to resume operation.
IC. In addition, the V regulator can also generate wasted
CC
heat when V is very high, adding to the total power
IN
dissipation of the IC. As described elsewhere in this data
sheet, bootstrapping of the V for 5V output applications
CC
CC
can essentially eliminate the V power dissipation term
GND
V
IN
C
IN
V
CC
L
C
OUT
3129 F04
GND
V
OUT
Figure 4. Typical 2-Layer PC Board Layout (MSE Package)
3129f
16
For more information www.linear.com/3129
LTC3129
applicaTions inForMaTion
A standard application circuit for the LTC3129 is shown on
the front page of this data sheet. The appropriate selection
of external components is dependent upon the required
performance of the IC in each particular application given
considerations and trade-offs such as PCB area, input
and output voltage range, output voltage ripple, transient
response, required efficiency, thermal considerations and
cost. This section of the data sheet provides some basic
guidelines and considerations to aid in the selection of
external components and the design of the applications
circuit, as well as more application circuit examples.
sible. V is the regulator output and is also the internal
CC
supply pin for the LTC3129 control circuitry as well as the
gate drivers and boost rail charging diodes. The V pin is
CC
not intended to supply current to other external circuitry.
Inductor Selection
ThechoiceofinductorusedinLTC3129applicationcircuits
influences the maximum deliverable output current, the
converterbandwidth,themagnitudeoftheinductorcurrent
ripple and the overall converter efficiency. The inductor
must have a low DC series resistance, when compared to
the internal switch resistance, or output current capabil-
ity and efficiency will be compromised. Larger inductor
valuesreduceinductorcurrentripplebutmaynotincrease
output current capability as is the case with peak current
mode control as described in the Maximum Output Cur-
rent section. Larger value inductors also tend to have a
higher DC series resistance for a given case size, which
will have a negative impact on efficiency. Larger values
of inductance will also lower the right half plane (RHP)
zero frequency when operating in boost mode, which can
compromise loop stability. Nearly all LTC3129 application
circuitsdeliverthebestperformancewithaninductorvalue
between 3.3µH and 10µH. Buck mode-only applications
can use the larger inductor values as they are unaffected
bytheRHPzero,whilemostlyboostapplicationsgenerally
require inductance on the low end of this range depending
on how large the step-up ratio is.
Programming V
OUT
The output voltage of the LTC3129 is set by connecting
the FB pin to an external resistor divider from V to
ground, as shown in Figure 5, according to the equation:
OUT
V
= 1.175V • (1+ R1/R2)
OUT
V
OUT
V
OUT
LTC3129
C
OUT
R1
R2
C
FF
FB
3129 F05
Figure 5. VOUT Feedback Divider
Asmallfeedforwardcapacitorcanbeaddedinparallelwith
R1 (in Figure 5) to reduce Burst Mode ripple and improve
transient response. Details on selecting a feedforward
capacitor are provided later in this data sheet.
Regardless of inductor value, the saturation current rating
shouldbeselectedsuchthatitisgreaterthantheworst-case
averageinductorcurrentplushalfoftheripplecurrent.The
peak-to-peak inductor current ripple for each operational
modecanbecalculatedfromthefollowingformula, where
f is the switching frequency (1.2MHz), L is the inductance
V
Capacitor Selection
CC
in µH and t
is the switch pin minimum low time in
LOW
The V output of the LTC3129 is generated from V by a
CC
IN
µs. The switch pin minimum low time is typically 0.09µs.
low dropout linear regulator. The V regulator has been
CC
designed for stable operation with a wide range of output
capacitors. For most applications, a low ESR capacitor of
at least 2.2µF should be used. The capacitor should be
VOUT V – V
1
f
IN
OUT
∆IL(P−P)(BUCK)
=
– t
A
LOW
L
V
IN
located as close to the V pin as possible and connected
V
L
VOUT – V
1
CC
IN
IN
∆IL(P−P)(BOOST)
=
– t
A
f
LOW
totheV pinandgroundthroughtheshortesttracespos-
CC
VOUT
3129f
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LTC3129
applicaTions inForMaTion
It should be noted that the worst-case peak-to-peak in-
Differentinductorcorematerialsandstyleshaveanimpact
on the size and price of an inductor at any given current
rating. Shielded construction is generally preferred as it
minimizes the chances of interference with other circuitry.
Thechoiceofinductorstyledependsupontheprice,sizing,
and EMI requirements of a particular application. Table 1
provides a wide sampling of inductors that are well suited
to many LTC3129 applications.
ductor ripple current occurs when the duty cycle in buck
mode is minimum (highest V ) and in boost mode when
IN
the duty cycle is 50% (V
IN
= 2 • V ). As an example, if
OUT
IN
V
(minimum) = 2.5V and V (maximum) = 15V, V
IN OUT
= 5V and L = 10µH, the peak-to-peak inductor ripples at
the voltage extremes (15V V for buck and 2.5V V for
IN
IN
boost) are:
Buck = 248mA peak-to-peak
Boost = 93mA peak-to-peak
Table 1. Recommended Inductors
VENDOR
PART
Coilcraft
www.coilcraft.com
EPL2014, EPL3012, EPL3015, LPS3015,
LPS3314, XFL3012
One half of this inductor ripple current must be added to
the highest expected average inductor current in order to
select the proper saturation current rating for the inductor.
Coiltronics
www.cooperindustries.com
SDH3812, SD3814, SD3114, SD3118
Murata
www.murata.com
LQH3NP, LQH32P, LQH44P
To avoid the possibility of inductor saturation during load
transients, an inductor with a saturation current rating of
at least 600mA is recommended for all applications.
Sumida
www.sumida.com
CDRH2D16, CDRH2D18, CDRH3D14,
CDRH3D16
Taiyo-Yuden
www.t-yuden.com
NR3012T, NR3015T, NRS4012T,
BRC2518
Inadditiontoitsinfluenceonpowerconversionefficiency,
the inductor DC resistance can also impact the maximum
output current capability of the buck-boost converter
particularly at low input voltages. In buck mode, the
output current of the buck-boost converter is primarily
limited by the inductor current reaching the average cur-
rent limit threshold. However, in boost mode, especially
at large step-up ratios, the output current capability can
also be limited by the total resistive losses in the power
stage. These losses include, switch resistances, inductor
DC resistance and PCB trace resistance. Avoid inductors
with a high DC resistance (DCR) as they can degrade the
maximum output current capability from what is shown
in the Typical Performance Characteristics section and
from the Typical Application circuits.
TDK
www.tdk.com
VLS3012, VLS3015, VLF302510MT,
VLF302512MT
Toko
www.tokoam.com
DB3015C, DB3018C, DB3020C, DP418C,
DP420C, DEM2815C, DFE322512C,
DFE252012C
Würth
www.we-online.com
WE-TPC 2813, WE-TPC 3816,
WE-TPC 2828
Recommended inductor values for different operating
voltage ranges are given in Table 2. These values were
chosen to minimize inductor size while maintaining an
acceptable amount of inductor ripple current for a given
V and V
range.
IN
OUT
Table 2. Recommended Inductor Values
V
V
V
V
V
AND V
RANGE
RECOMMENDED INDUCTOR VALUES
3.3µH to 4.7µH
IN
IN
IN
IN
IN
OUT
and V
and V
and V
and V
Both < 4.5V
Both < 8V
OUT
OUT
OUT
OUT
As a guideline, the inductor DCR should be significantly
less than the typical power switch resistance of 750mΩ
each. The only exceptions are applications that have a
maximumoutputcurrentrequirementmuchlessthanwhat
the LTC3129 is capable of delivering. Generally speaking,
inductors with a DCR in the range of 0.15Ω to 0.3Ω are
recommended. Lower values of DCR will improve the ef-
ficiency at the expense of size, while higher DCR values
will reduce efficiency (typically by a few percent) while
allowing the use of a physically smaller inductor.
4.7µH to 6.8µH
Both < 11V
Up to 15.75V
6.8µH to 8.2µH
8.2µH to 10µH
Feedforward Capacitor
The use of a voltage feedforward capacitor, as shown in
Figure 5, offers a number of performance advantages. A
feedforward capacitor will reduce output voltage ripple in
3129f
18
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LTC3129
applicaTions inForMaTion
Examining the previous equations reveals that the output
voltage ripple increases with load current and is gener-
ally higher in boost mode than in buck mode. Note that
these equations only take into account the voltage ripple
that occurs from the inductor current to the output being
discontinuous. They provide a good approximation to the
rippleatanysignificantloadcurrentbutunderestimatethe
output voltage ripple at very light loads where the output
voltage ripple is dominated by the inductor current ripple.
Burst Mode operation and improve transient response. In
addition, due to the wide V and V
operating range
IN
OUT
of the LTC3129 and its fixed internal loop compensation,
some applications may require the use of a feedforward
capacitor to assure light-load stability (less than ~15mA)
when operating in PWM mode (PWM pin pulled high).
Therefore, to reduce Burst Mode ripple and improve
phase margin at light load when PWM mode operation is
selected, a feedforward capacitor is recommended for all
applications. The recommended feedforward capacitor
value can be calculated by:
In addition to the output voltage ripple generated across
the output capacitance, there is also output voltage ripple
produced across the internal resistance of the output
capacitor. The ESR-generated output voltage ripple is
proportionaltotheseriesresistanceoftheoutputcapacitor
C = 66/R1
FF
Where R1 is the top feedback divider resistor value in MΩ
and C is the recommended feedforward capacitor value
and is given by the following expressions where R
is
FF
ESR
in picofarads (use the nearest standard value). Refer to
the series resistance of the output capacitor and all other
terms as previously defined.
the application circuits for examples.
ILOAD ESR
R
Output Capacitor Selection
∆V
=
≅ ILOAD ESR
R
V
P−P(BUCK)
1– tLOW
f
A low effective series resistance (ESR) output capacitor
of 4.7µF minimum should be connected at the output of
the buck-boost converter in order to minimize output volt-
age ripple. Multilayer ceramic capacitors are an excellent
option as they have low ESR and are available in small
footprints. The capacitor value should be chosen large
enough to reduce the output voltage ripple to acceptable
levels. Neglecting the capacitor's ESR and ESL (effec-
tive series inductance), the peak-to-peak output voltage
ripple in PWM mode can be calculated by the following
ILOAD ESR OUT
R
V
∆V
=
P−P(BOOST)
V 1– t
f
(
)
IN
LOW
VOUT
V
≅ ILOAD ESR
R
V
IN
InmostLTC3129applications,anoutputcapacitorbetween
10µF and 22µF will work well. To minimize output ripple
in Burst Mode operation, or transients incurred by large
step loads, values of 22µF or larger are recommended.
formula, where f is the frequency in MHz (1.2MHz), C
OUT
is the capacitance in µF, t
is the switch pin minimum
LOW
low time in µs (0.09µs typical) and I
current in amperes.
is the output
LOAD
Input Capacitor Selection
The V pin carries the full inductor current and provides
IN
ILOAD LOW
t
power to internal control circuits in the IC. To minimize
input voltage ripple and ensure proper operation of the IC,
a low ESR bypass capacitor with a value of at least 4.7µF
∆V
=
V
P−P(BUCK)
COUT
ILOAD
V
– VIN + tLOWfV
IN
OUT
should be located as close to the V pin as possible. The
IN
∆V
=
V
P−P(BOOST)
fC
OUT
VOUT
traces connecting this capacitor to V and the ground
IN
plane should be made as short as possible.
3129f
19
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LTC3129
applicaTions inForMaTion
Whenpoweredthroughlongleadsorfromapowersource
with significant resistance, a larger value bulk input ca-
pacitor may be required and is generally recommended.
Insuchapplications, a47µFto100µFlow-ESRelectrolytic
capacitorinparallelwitha1µFceramiccapacitorgenerally
yields a high performance, low cost solution.
of its rated capacitance when operated at even half of its
maximum rated voltage. This effect is generally reduced
as the case size is increased for the same nominal value
capacitor. As a result, it is often necessary to use a larger
value capacitance or a higher voltage rated capacitor than
wouldordinarilyberequiredtoactuallyrealizetheintended
capacitanceattheoperatingvoltageoftheapplication.X5R
and X7R dielectric types are recommended as they exhibit
the best performance over the wide operating range and
temperature of the LTC3129. To verify that the intended
capacitance is achieved in the application circuit, be sure
to consult the capacitor vendor's curve of capacitance
versus DC bias voltage.
Note that applications using the MPPC feature should
use a minimum C of 22µF. Larger values can be used
IN
without limitation.
Recommended Input and Output Capacitor Types
The capacitors used to filter the input and output of the
LTC3129 must have low ESR and must be rated to handle
the AC currents generated by the switching converter.
This is important to maintain proper functioning of the
IC and to reduce output voltage ripple. There are many
capacitor types that are well suited to these applications
including multilayer ceramic, low ESR tantalum, OS-CON
and POSCAP technologies. In addition, there are certain
types of electrolytic capacitors such as solid aluminum
organic polymer capacitors that are designed for low
ESR and high AC currents and these are also well suited
to some LTC3129 applications. The choice of capacitor
technology is primarily dictated by a trade-off between
size, leakage current and cost. In backup power applica-
tions, the input or output capacitor might be a super or
ultra capacitor with a capacitance value measuring in the
Farad range. The selection criteria in these applications
are generally similar except that voltage ripple is generally
not a concern. Some capacitors exhibit a high DC leak-
age current which may preclude their consideration for
applications that require a very low quiescent current in
BurstModeoperation. Notethatultracapacitorsmayhave
a rather high ESR, therefore a 4.7µF (minimum) ceramic
capacitor is recommended in parallel, close to the IC pins.
Using the Programmable RUN Function to Operate
from Extremely Weak Input Sources
Another application of the programmable RUN pin is that
it can be used to operate the converter in a hiccup mode
fromextremelylowcurrentsources.Thisallowsoperation
from sources that can only generate microamps of output
current,andwouldbefartooweaktosustainnormalsteady-
stateoperation,evenwiththeuseoftheMPPCpin.Because
until it is
the LTC3129 draws only 1.9µA typical from V
IN
enabled, the RUN pin can be programmed to keep the IC
disabled until V reaches the programmed voltage level.
IN
Inthismanner,theinputsourcecantrickle-chargeaninput
storage capacitor, even if it can only supply microamps of
current, until V reaches the turn-on threshold set by the
IN
RUN pin divider. The converter will then be enabled using
the stored charge in the input capacitor, until Vin drops
below the turn-off threshold, at which point the converter
will turn off and the process will repeat.
This approach allows the converter to run from weak
sources such as thin-film solar cells using indoor light-
ing. Although the converter will be operating in bursts,
it is enough to charge an output capacitor to power low
duty cycle loads, such as wireless sensor applications,
or to trickle charge a battery. In addition, note that the
input voltage will be cycling (with a small ripple as set by
the RUN hysteresis) about a fixed voltage, as determined
by the divider. This allows the high impedance source to
operate at the programmed optimal voltage for maximum
power transfer.
Ceramic capacitors are often utilized in switching con-
verter applications due to their small size, low ESR and
low leakage currents. However, many ceramic capacitors
intended for power applications experience a significant
loss in capacitance from their rated value as the DC bias
voltage on the capacitor increases. It is not uncommon for
a small surface mount capacitor to lose more than 50%
3129f
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LTC3129
applicaTions inForMaTion
When using high value divider resistors (in the MΩ
MPPC Compensation and Gain
range) to minimize current draw on V , a small noise
IN
When using MPPC, there are a number of variables that
affect the gain and phase of the input voltage control
loop. Primarily these are the input capacitance, the MPPC
filter capacitor may be necessary across the lower divider
resistor to prevent noise from erroneously tripping the
RUNcomparator.Thecapacitorvalueshouldbeminimized
so as not to introduce a time delay long enough for the
divider ratio and the V source resistance (or current). To
IN
simplify the design of the application circuit, the MPPC
control loop in the LTC3129 is designed with a relatively
low gain, such that external MPPC loop compensation is
input voltage to drop significantly below the desired V
IN
threshold before the converter is turned off. Note that
larger V decoupling capacitor values will minimize this
IN
generally not required when using a V capacitor value
IN
effect by providing more holdup time on V .
IN
of at least 22µF. The gain from the MPPC pin to the in-
ternal VC control voltage is about 12, so a drop of 50mV
on the MPPC pin (below the 1.175V MPPC threshold),
corresponds to a 600mV drop on the internal VC voltage,
which reduces the average inductor current all the way
to zero. Therefore, the programmed input MPPC voltage
will be maintained within about 4% over the load range.
Programming the MPPC Voltage
As discussed in the previous section, the LTC3129 in-
cludes an MPPC function to optimize performance when
operatingfromvoltagesourceswithrelativelyhighsource
resistance. Using an external voltage divider from V , the
IN
MPPCfunctiontakescontroloftheaverageinductorcurrent
when necessary to maintain a minimum input voltage, as
programmed by the user. Referring to Figure 3:
Note that if large value V capacitors are used (which may
IN
have a relatively high ESR) a small ceramic capacitor of
at least 4.7µF should be placed in parallel across the V
IN
V
= 1.175V • (1 + R5/R6)
input, near the V pin of the IC.
IN(MPPC)
IN
This is useful for such applications as photovoltaic
powered converters, since the maximum power transfer
point occurs when the photovoltaic panel is operated at
about 75% of its open-circuit voltage. For example, when
operating from a photovoltaic panel with an open-circuit
voltage of 5V, the maximum power transfer point will be
when the panel is loaded such that its output voltage is
about 3.75V. Choosing values of 2MΩ for R5 and 909kΩ
for R6 will program the MPPC function to regulate the
Bootstrapping the V Regulator
CC
The high and low side gate drivers are powered through
theV rail,whichisgeneratedfromtheinputvoltage,V ,
CC
IN
through an internal linear regulator. In some applications,
especially at high input voltages, the power dissipation
in the linear regulator can become a major contributor to
thermalheatingoftheICandoverallefficiency. TheTypical
Performance Characteristics section provides data on the
maximuminputcurrentsoastomaintainV ataminimum
V
CC
current and resulting power loss versus V and V
.
IN
IN
OUT
of 3.74V (typical). Note that if the panel can provide more
power than the LTC3129 can draw, the input voltage will
rise above the programmed MPPC point. This is fine as
long as the input voltage doesn't exceed 15V.
Asignificantperformanceadvantagecanbeattainedinhigh
V applications where converter output voltage (V ) is
IN
OUT
programmed to 5V, if V
is used to power the V rail.
OUT
CC
Powering V in this manner is referred to as bootstrap-
CC
For weak input sources with very high resistance (hun-
dreds of Ohms or more), the LTC3129 may still draw more
ping. This can be done by connecting a Schottky diode
(such as a BAT54) from V
to V as shown in Figure 6.
OUT
CC
current than the source can provide, causing V to drop
Withthebootstrapdiodeinstalled, thegatedrivercurrents
aresuppliedbythebuck-boostconverterathighefficiency
rather than through the internal linear regulator. The in-
ternal linear regulator contains reverse blocking circuitry
IN
below the UVLO threshold. For these applications, it is
recommended that the programmable RUN feature be
used, as described in the previous section.
3129f
21
For more information www.linear.com/3129
LTC3129
applicaTions inForMaTion
that allows V to be driven above its nominal regulation
Sources of Small Photovoltaic Panels
CC
level with only a very slight amount of reverse current.
A list of companies that manufacture small solar panels
(sometimes referred to as modules or solar cell arrays)
suitable for use with the LTC3129 is provided in Table 3.
Please note that the bootstrapping supply (either V
or
OUT
a separate regulator) must be limited to less than 5.7V so
as not to exceed the maximum V voltage of 5.5V after
CC
Table 3. Small Photovoltaic Panel Manufacturers
the diode drop.
Sanyo
http://panasonic.net/energy/amorton/en/
http://www.powerfilmsolar.com/
By maintaining V above its UVLO threshold, bootstrap-
CC
PowerFilm
ping, even to a 3.3V output, also allows operation down
Ixys
Corporation
http://www.ixys.com/ProductPortfolio/GreenEnergy.aspx
to the V UVLO threshold of 1.8V (typical).
IN
G24
Innovations
http://www.g24i.com/
V
OUT
V
OUT
SolarPrint
http://www.solarprint.ie/
LTC3129
C
OUT
BAT54
V
CC
2.2µF
3129 F06
Figure 6. Example of VCC Bootstrap
3129f
22
For more information www.linear.com/3129
LTC3129
Typical applicaTions
Hiccup Converter Powers Wireless Sensor from Indoor Lighting
Transmit Rate vs Light Level
(Fluorescent)
22nF
22nF
4.7µH
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
PULSED I
OUT
25mA FOR 5ms
BST1 SW1
SW2 BST2
UVLO = 3.5V
V
IN
V
OUT
V
V
OUT
IN
3.6V
4.7µF
22µF
2M
LTC3129
4.42M
1M
RUN
+
470µF
6.3V
FB
976k
V
MPPC
PWM
CC
PGOOD
PGOOD
PV PANEL
SANYO AM-1815
V
CC
10pF
NC
NC
0.5
0
4.9cm × 5.8cm
2.37M
2.2µF
0
400
800
1200
1600
2000
GND
PGND
LIGHT LEVEL (Lx)
3129 TA02b
3129 TA02a
Low Noise 3.6V Converter Using Bootstrap Diode to Extend Lower VIN Range
22nF
22nF
6.8µH
V
V
< 3.6V, I
> 3.6V, I
= 100mA
= 200mA
IN
IN
OUT
OUT
BST1 SW1
SW2 BST2
V
V
OUT
IN
1.8V TO 15V
V
V
OUT
IN
3.6V
10µF
LTC3129
2M
33pF
BAT54
RUN
FB
V
CC
MPPC
976k
PGOOD
PWM
V
CC
10µF
NC
NC
2.2µF
GND
PGND
3129 TA03
3129f
23
For more information www.linear.com/3129
LTC3129
Typical applicaTions
Solar Powered Converter with MPPC Charges Storage Capacitor
Average Output Current
vs Light Level (Daylight)
22nF
22nF
4.7µH
100.0
10.0
1.0
BST1 SW1
SW2 BST2
UVLO = 4.3V
1M
V
IN
V
OUT
4.8V
V
IN
V
OUT
+
4.7µF
LTC3129
1F
3.09M
1M
47µF
CERAMIC
RUN
V
CC
COOPER BUSSMANN
PB-5R0V105-R
MPPC
FB
PowerFilm
SP4.2-37
SOLAR
PGOOD
PGOOD
PWM
NC
V
CC
MODULE
392k
NC
8.4cm × 3.7cm
0.1
2.2µF
1000
10000
100000
1000000
GND
PGND
LIGHT LEVEL (Lx)
3129 TA04b
3129 TA04a
Li-Ion Powered 3V Converter with 3.1V Input UVLO Reduces Low Battery IQ to 3µA
22nF
22nF
4.7µH
BST1 SW1
SW2 BST2
UVLO = 3.1V
2M
V
OUT
V
IN
V
OUT
3V
200mA
10µF
LTC3129
+
33pF
1.58M
1.02M
Li-Ion
RUN
FB
4.7µF
V
CC
MPPC
PGOOD
PWM
NC
V
CC
1.27M
NC
2.2µF
10pF
GND
PGND
3129 TA05
15V Converter Operates from Three to Eight AA or AAA Cells
22nF
22nF
10µH
V
IN
2.42V TO 15V
BST1 SW1
SW2 BST2
V
OUT
15V
V
IN
V
OUT
10µF
25V
25mA MINIMUM
LTC3129
3.01M
255k
22pF
RUN
10µF
V
FB
THREE TO EIGHT
AA OR AAA
MPPC
PWM
NC
CC
PGOOD
BATTERIES
V
CC
NC
2.2µF
GND
PGND
3129 TA06
3129f
24
For more information www.linear.com/3129
LTC3129
Typical applicaTions
Energy Harvesting Converter Operates from a Variety of Weak Sources
22nF
22nF
4.7µH
BAS 70-05
BST1 SW1
SW2 BST2
UVLO = 3.3V
V
OUT
V
V
OUT
IN
5V
10µF
INPUT SOURCES:
• RF
LTC3129
4.99M
V
3.32M
1.02M
22pF
• AC
RUN
• PIEZO
• COIL-MAGNET
FB
100µF
CERAMIC
MPPC
PWM
NC
CC
PGOOD
BAS 70-06
3.01M
V
CC
NC
10pF
2.2µF
GND
PGND
3129 TA07
Solar Powered Converter Extends Battery Life in Low Power 3V Primary Battery Applications
TOKO DEM2812C
22nF
22nF
3.3µH
FDC6312P
DUAL PMOS
V
OUT
3V TO 3.2V
S1
S2
BST1 SW1
SW2 BST2
V
UVLO = 3.7V
IN
3.20V
4.22M
D1
D2
2.2µF
V
V
OUT
IN
22µF
15pF
LTC3129
4.7µF 4.99M
G1
G2
2.43M
RUN
FB
CR2032
3V COIN CELL
V
OUT
R4
2.43M
PV PANEL
SANYO AM-1815
OR
+
470µF
6.3V
V
CC
MPPC
PowerFilm SP4.2-37
PWM
NC
PGOOD
2.43M
BAT54
NC
V
CC
74LVC2G04
10pF
GND
PGND
2.2µF
3129 TA09
Percentage of Added Battery Life vs Light Level and Load
(PowerFilm SP4.2-37, 30sq cm Panel)
1000
100
10
AVERAGE LOAD = 165µW
AVERAGE LOAD = 330µW
AVERAGE LOAD = 660µW
AVERAGE LOAD = 1650µW
AVERAGE LOAD = 3300µW
1
100
1,000
10,000
LIGHT LEVEL (Lx)
3129 TA09b
3129f
25
For more information www.linear.com/3129
LTC3129
package DescripTion
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
UD Package
16-Lead Plastic QFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1700 Rev A)
Exposed Pad Variation AA
0.70 ±0.05
3.50 ±0.05
2.10 ±0.05
1.65 ±0.05
(4 SIDES)
PACKAGE OUTLINE
0.25 ±0.05
0.50 BSC
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
BOTTOM VIEW—EXPOSED PAD
PIN 1 NOTCH R = 0.20 TYP
OR 0.25 × 45° CHAMFER
R = 0.115
TYP
0.75 ±0.05
3.00 ±0.10
(4 SIDES)
15 16
PIN 1
TOP MARK
(NOTE 6)
0.40 ±0.10
1
2
1.65 ±0.10
(4-SIDES)
(UD16 VAR A) QFN 1207 REV A
0.200 REF
0.25 ±0.05
0.50 BSC
0.00 – 0.05
NOTE:
1. DRAWING CONFORMS TO JEDEC PACKAGE OUTLINE MO-220 VARIATION (WEED-4)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
3129f
26
For more information www.linear.com/3129
LTC3129
package DescripTion
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev E)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ±0.102
(.112 ±.004)
2.845 ±0.102
(.112 ±.004)
0.889 ±0.127
(.035 ±.005)
1
8
0.35
REF
5.23
(.206)
MIN
1.651 ±0.102
(.065 ±.004)
1.651 ±0.102
(.065 ±.004)
3.20 – 3.45
(.126 – .136)
0.12 REF
DETAIL “B”
CORNER TAIL IS PART OF
THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
DETAIL “B”
16
9
0.305 ±0.038
0.50
(.0197)
BSC
NO MEASUREMENT PURPOSE
4.039 ±0.102
(.159 ±.004)
(NOTE 3)
(.0120 ±.0015)
TYP
0.280 ±0.076
(.011 ±.003)
RECOMMENDED SOLDER PAD LAYOUT
16151413121110
9
REF
DETAIL “A”
0.254
(.010)
3.00 ±0.102
(.118 ±.004)
(NOTE 4)
0° – 6° TYP
4.90 ±0.152
(.193 ±.006)
GAUGE PLANE
0.53 ±0.152
(.021 ±.006)
1 2 3 4 5 6 7 8
DETAIL “A”
0.86
(.034)
REF
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
0.1016 ±0.0508
(.004 ±.002)
MSOP (MSE16) 0911 REV E
0.50
(.0197)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
3129f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
27
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
LTC3129
Typical applicaTion
Dual VIN Application, Using the LTC4412 PowerPath™ Controller
22nF
22nF
MBR0520
10µH
12V WALL ADAPTER INPUT
V
V
= 12V, I
= 200mA
= 50mA
IN
IN
OUT
OUT
FDN338
BSS314
= 3.6V, I
BST1 SW1
SW2 BST2
V
OUT
V
V
OUT
IN
12V
10µF
25V
LTC3129
10pF
3.01M
LTC4412
GATE
10µF
RUN
V
FB
IN
V
CC
MPPC
+
SENSE
STAT
324k
Li-Ion
PGOOD
PWM
NC
V
CC
CTL
NC
GND
2.2µF
GND
PGND
3129 TA08
relaTeD parTs
PART NUMBER DESCRIPTION
COMMENTS
= 2.2V, V
LTC3103
LTC3104
LTC3105
LTC3112
LTC3115-1
LTC3531
15V, 300mA Synchronous Step-Down DC/DC Converter with
Ultralow Quiescent Current
V
I
= 15V, V
= 0.8V, I = 1.8µA,
OUT(MIN) Q
IN(MIN)
IN(MAX)
<1µA, 3mm × 3mm DFN-10, MSOP-10 Packages
SD
15V, 300mA Synchronous Step-Down DC/DC Converter with
Ultralow Quiescent Current and 10mA LDO
V
SD
= 2.2V, V
= 15V, V
= 0.8V, I = 2.8µA,
OUT(MIN) Q
IN(MIN)
IN(MAX)
I
<1µA, 4mm × 3mm DFN-14, MSOP-16 Packages
400mA Step-up Converter with MPPC and 250mV Start-Up
V
SD
= 0.2V, V
= 5V, V
= 0 5.25V , I = 22µA,
MAX Q
IN(MIN)
IN(MAX)
OUT(MIN)
I
<1µA, 3mm × 3mm DFN-10/MSOP-12 Packages
15V, 2.5A, 750kHz Monolithic Synch Buck/Boost
V
SD
= 2.7V, V
= 15V, V
= 2.7V to 14V, I = 50µA,
OUT(MIN) Q
IN(MIN)
IN(MAX)
I
<1µA, 4mm × 5mm DFN-16 TSSOP-20E Packages
40V, 2A, 2MHz Monolithic Synch Buck/Boost
V
SD
= 2.7V, V
= 40V, V
= 2.7V to 40V, I = 50µA,
OUT(MIN) Q
IN(MIN)
IN(MAX)
I
<1µA, 4mm × 5mm DFN-16 and TSSOP-20E Packages
5.5V, 200mA, 600kHz Monolithic Synch Buck/Boost
20V, 50mA High Efficiency Nano Power Step-Down Regulator
Ultralow Voltage Step-Up Converter and Power Manager
V
SD
= 1.8V, V
= 5.5V, V
= 2V to 5V, I = 16µA,
OUT(MIN) Q
IN(MIN)
IN(MAX)
I
<1µA, 3mm × 3mm DFN-8 and ThinSOT Packages
LTC3388-1/
LTC3388-3
V
= 2.7V, V
=20V, V
= Fixed 1.1V to 5.5V,
OUT(MIN)
IN(MIN)
IN(MAX)
I = 720nA, I = 400nA, 3mm × 3mm DFN-10, MSOP-10 Packages
Q
SD
LTC3108/
LTC3108-1
V
= 0.02V, V
= 1V, V
= Fixed 2.35V to 5V,
OUT(MIN)
IN(MIN)
IN(MAX)
I = 6µA, I <1µA, 3mm × 4mm DFN-12, SSOP-16 Packages
Q
SD
LTC3109
LTC3588-1
LTC4070
Auto-Polarity, Ultralow Voltage Step-Up Converter and Power
Manager
V
Q
= 0.03V, V
SD
= 1V, V
= Fixed 2.35V to 5V,
OUT(MIN)
IN(MIN)
IN(MAX)
I = 7µA, I <1µA, 4mm × 4mm QFN-20, SSOP-20 Packages
Piezo Electric Energy Harvesting Power Supply
V
= 2.7V, V
= 20V, V
= Fixed 1.8V to 3.6V,
OUT(MIN)
IN(MIN)
IN(MAX)
I = 950nA, I 450nA, 3mm × 3mm DFN-10, MSOP-10E Packages
Q
SD
Li-Ion/Polymer Low Current Shunt Battery Charger System
V
= 450nA to 50mA, V
+ 4.0V, 4.1V, 4.2V, I = 300nA,
IN(MIN)
FLOAT Q
2mm × 3mm DFN-8, MSOP-8 Packages
3129f
LT 0213 • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
28
●
●
LINEAR TECHNOLOGY CORPORATION 2013
(408)432-1900 FAX: (408) 434-0507 www.linear.com/3129
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