LTC3422EDD#TRPBF [Linear]
LTC3422 - 1.5A, 3MHz Synchronous Step-Up DC/DC Converter with Output Disconnect; Package: DFN; Pins: 10; Temperature Range: -40°C to 85°C;
| 型号： | LTC3422EDD#TRPBF |
| 厂家： | 
Linear |
| 描述： | LTC3422 - 1.5A, 3MHz Synchronous Step-Up DC/DC Converter with Output Disconnect; Package: DFN; Pins: 10; Temperature Range: -40°C to 85°C 转换器 |
| 文件： | 总24页 (文件大小：522K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC3122
15V, 2.5A Synchronous
Step-Up DC/DC Converter
with Output Disconnect
DescripTion
FeaTures
The LTC®3122 is a synchronous step-up DC/DC converter
withtrueoutputdisconnectandinrushcurrentlimiting.The
2.5A current limit along with the ability to program output
voltages up to 15V makes the LTC3122 well suited for a
variety of demanding applications. Once started, opera-
tion will continue with inputs down to 500mV, extending
runtime in many applications.
n
V Range: 1.8V to 5.5V, 500mV After Start-Up
IN
n
Output Voltage Range: 2.2V to 15V
n
800mA Output Current for V = 5V and V
= 12V
IN
OUT
n
n
n
n
Output Disconnects from Input When Shut Down
Synchronous Rectification: Up to 95% Efficiency
Inrush Current Limit
Up to 3MHz Adjustable Switching Frequency
Synchronizable to External Clock
The LTC3122 features output disconnect in shutdown,
dramatically reducing input power drain and enabling
Selectable Burst Mode® Operation: 25µA I
n
n
n
n
n
Q
Output Overvoltage Protection
Soft-Start
V
OUT
to completely discharge. Adjustable PWM switching
from 100kHz to 3MHz optimizes applications for highest
efficiency or smallest solution footprint. The oscillator
can also be synchronized to an external clock for noise
sensitive applications. Selectable Burst Mode operation
reducesquiescentcurrentto25µA,ensuringhighefficiency
across the entire load range. An internal soft-start limits
inrush current during start-up.
<1µA I in Shutdown
Q
12-Lead, 3mm × 4mm × 0.75mm Thermally
Enhanced DFN and MSOP Packages
applicaTions
n
RF Power
n
Piezo Actuators
Other features include a <1µA shutdown current and ro-
bust protection under short-circuit, thermal overload, and
output overvoltage conditions. The LTC3122 is offered in
both a low profile 12-lead (3mm × 4mm × 0.75 mm) DFN
packageanda12-leadthermallyenhancedMSOPpackage.
n
Small DC Motors
12V Analog Rail From Battery, 5V, or Backup Capacitor
n
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.
Typical applicaTion
5V to 12V Synchronous Boost Converter with Output Disconnect
Efficiency Curve
100
90
80
70
60
50
40
30
20
10
0
10
1
3.3µH
V
IN
Burst Mode
OPERATION
5V
SW
V
OUT
V
V
OUT
12V
IN
800mA
LTC3122
PWM
SD
OFF ON
100nF
4.7µF
BURST PWM
PWM/SYNC
RT
CAP
FB
1.02M
113k
22µF
0.1
PWM POWER LOSS
V
V
C
CC
SGND
PGND
210k
390pF
57.6k
10pF
4.7µF
0
0.01
0.1
1
10
100
1000
LOAD CURRENT (mA)
3122 TA01b
3122 TA01a
3122f
1
LTC3122
absoluTe MaxiMuM raTings
(Note 1)
V Voltage .................................................. –0.3V to 6V
OUT
CAP Voltage
< 5.7V ............................–0.3V to (V
IN
V
Voltage ............................................ –0.3V to 18V
V
+ 0.3V)
+ 0.3V)
OUT
5.7V ≤ V
OUT
OUT
SW Voltage (Note 2) .................................. –0.3V to 18V
SW Voltage (Pulsed < 100ns) (Note 2)....... –0.3V to 19V
≤ 11.7V......(V
– 6V) to (V
OUT
OUT
V
OUT
> 11.7V.................................(V
All Other Pins............................................... –0.3V to 6V
– 6V) to 12V
OUT
V , RT Voltage .......................................... –0.3V to V
C
CC
Operating Junction Temperature Range
(Notes 3, 4)............................................ –40°C to 125°C
Storage Temperature Range .................. –65°C to 150°C
MSE Lead Temperature (Soldering, 10sec) ...........300°C
pin conFiguraTion
TOP VIEW
TOP VIEW
SW
1
2
3
4
5
6
12 CAP
11
10 SGND
1
2
3
4
5
6
SW
PGND
12 CAP
11
10 SGND
PGND
V
OUT
V
OUT
V
IN
V
13
13
PGND
IN
PGND
PWM/SYNC
9
8
7
SD
FB
PWM/SYNC
9
8
7
SD
V
CC
RT
V
CC
FB
V
C
RT
V
C
MSE PACKAGE
12-LEAD PLASTIC MSOP
DE PACKAGE
T
= 125°C, θ = 40°C/W (NOTE 5), θ = 10°C/W
JA JC
EXPOSED PAD (PIN 13) IS PGND,
MUST BE SOLDERED TO PCB FOR RATED THERMAL PERFORMANCE
JMAX
12-LEAD (4mm × 3mm) PLASTIC DFN
T
JMAX
= 125°C, θ = 43°C/W (NOTE 5), θ = 5°C/W
JA JC
EXPOSED PAD (PIN 13) IS PGND,
MUST BE SOLDERED TO PCB FOR RATED THERMAL PERFORMANCE
orDer inForMaTion
LEAD FREE FINISH
LTC3122EDE#PBF
LTC3122IDE#PBF
LTC3122EMSE#PBF
LTC3122IMSE#PBF
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3122EDE#TRPBF
LTC3122IDE#TRPBF
LTC3122EMSE#TRPBF
LTC3122IMSE#TRPBF
3122
3122
3122
3122
12-Lead (4mm × 3mm) Plastic DFN
12-Lead (4mm × 3mm) Plastic DFN
12-Lead Plastic MSOP
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
12-Lead Plastic MSOP
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3122f
2
LTC3122
elecTrical characTerisTics The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C (Note 3). VIN = 3.6V, VOUT = 12V, RT = 57.6k unless otherwise noted.
PARAMETER
CONDITIONS
= 0V
MIN.
TYP
MAX
1.8
5.5
15
UNITS
V
l
l
l
l
Minimum Start-Up Voltage
Input Voltage Range
V
OUT
1.7
After V
≥ 2.2V
0.5
2.2
V
OUT
Output Voltage Adjust Range
Feedback Voltage
V
1.178
1.202
1
1.225
50
V
Feedback Input Current
Quiescent Current, Shutdown
Quiescent Current, Active
Quiescent Current, Burst
V
V
= 1.4V
= 0V, V
nA
µA
µA
FB
= 0V, Not Including Switch Leakage
OUT
0.01
500
1
SD
V = 0V, Measured On V , Non-Switching
C
700
IN
Measured on V , V > 1.4V
Measured on V , V > 1.4V
25
10
40
20
µA
µA
IN FB
OUT FB
N-channel MOSFET Switch Leakage Current
P-channel MOSFET Switch Leakage Current
N-channel MOSFET Switch On-Resistance
P-channel MOSFET Switch On-Resistance
N-channel MOSFET Current Limit
Maximum Duty Cycle
V
V
= 15V, V
= 15V, V = 0V
0.1
0.1
20
20
µA
µA
Ω
SW
OUT
C
= 0V, V
= 15V, V = 0V
SD
SW
OUT
0.121
0.188
3.5
Ω
l
l
l
l
l
l
l
2.5
90
4.5
A
V
V
= 1.0V
= 1.4V
94
%
FB
Minimum Duty Cycle
0
1.15
3
%
FB
Switching Frequency
0.85
0.1
1
MHz
MHz
V
SYNC Frequency Range
PWM/SYNC Input High
0.9•V
CC
PWM/SYNC Input Low
0.1•V
V
CC
PWM/SYNC Input Current
V
V
V
= 5.5V
0.01
–5.6
4.25
95
1
µA
V
PWM/SYNC
CAP Clamp Voltage
> 6.1V, Referenced to V
–5.2
4
–6.0
4.5
OUT
OUT
V
CC
Regulation Voltage
< 2.8V, V > 5V
OUT
V
IN
l
Error Amplifier Transconductance
Error Amplifier Output Current
Soft-Start Time
70
120
µS
µA
ms
V
25
10
l
l
SD Input High
1.6
SD Input Low
0.25
2
V
SD Input Current
V
SD
= 5.5V
1
µA
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: Voltage transients on the SW pin beyond the DC limit specified in
the Absolute Maximum Ratings are non-disruptive to normal operations
when using good layout practices, as shown on the demo board or
described in the data sheet or application notes.
to meet specifications over the full –40°C to 125°C operating junction
temperature range. The junction temperature (T in °C) is calculated from
J
the ambient temperature (T in °C) and power dissipation (P in Watts)
A
D
according to the formula: T = T + (P • θ ) where θ is the thermal
J
A
D
JA
JA
impedance of the package.
Note 4: The LTC3122 includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature shutdown is active.
Continuous operation above the specified maximum operating junction
temperature may result in device degradation or failure.
Note 3: The LTC3122 is tested under pulsed load conditions such that
T ≈ T . The LTC3122E is guaranteed to meet performance specifications
A
J
from 0°C to 85°C. Specifications over the –40°C to 125°C operating
junction temperature range are assured by design, characterization and
correlation with statistical process controls. The LTC3122I is guaranteed
Note 5: Failure to solder the exposed backside of the package to the PC
board ground plane will result in a thermal impedance much higher than
the rated package specifications.
3122f
3
LTC3122
Typical perForMance characTerisTics
Configured as front page application unless otherwise specified.
Efficiency vs Load Current,
VOUT = 5V
Efficiency vs Load Current,
VOUT = 7.5V
Efficiency vs Load Current,
VOUT = 12V
100
90
80
70
60
50
40
30
20
10
0
100
90
80
70
60
50
40
30
20
10
0
100
90
80
70
60
50
40
30
20
10
0
BURST
BURST
BURST
PWM
V
PWM
PWM
= 5.4V
= 4.2V
= 2.6V
V
V
V
= 5.4V
= 3.8V
= 2.3V
V
V
V
= 4.2V
= 3.3V
= 0.6V
IN
IN
IN
IN
IN
IN
IN
IN
IN
V
V
0.01
0.1
1
10
100
1000
0.01
0.1
1
10
100 1000 10000
0.01
0.1
1
10
100 1000 10000
LOAD CURRENT (mA)
LOAD CURRENT (mA)
LOAD CURRENT (mA)
3122 G03
3122 G02
3122 G01
PWM Mode Operation
Load Transient Response
Inrush Current Control
V
V
OUT
SD
5V/DIV
OUT
20mV/DIV
500mV/DIV
AC-COUPLED
AC-COUPLED
V
OUT
INDUCTOR
CURRENT
1A/DIV
800mA
5V/DIV
I
OUT
500mA/DIV
INPUT
CURRENT
1A/DIV
80mA
80mA
I
= 200mA
LOAD
3122 G06
3122 G04
3122 G05
2ms/DIV
1µs/DIV
500µs/DIV
RDS(ON) vs Temperature,
Both NMOS and PMOS
Oscillator Frequency
vs Temperature
Feedback vs Temperature
0.2
0.1
80
60
1.0
0.5
0
40
0
–0.1
–0.2
–0.3
–0.4
–0.5
–0.6
20
–0.5
–1.0
–1.5
–2.0
0
–20
–40
–60
–10
40
90
140
–50
–10
30
70
110
150
–60
–10
40
90
140
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
3122 G07
3122 G08
3122 G09
3122f
4
LTC3122
Typical perForMance characTerisTics
PWM Mode Maximum Output
Current vs VIN
Peak Current Limit Change
vs Temperature
PWM Operation No Load Input
Current vs VIN
2
1
2.0
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
70
60
50
40
30
20
10
0
V
V
V
= 5V
= 7.5V
= 12V
V
OUT
V
OUT
V
OUT
= 5V
= 7.5V
= 12V
OUT
OUT
OUT
0
–1
–2
–3
–4
–50
–10
30
70
110
150
0.5
1.5
2.5
V
3.5
(V)
4.5
5.5
0
1
2
3
4
5
6
TEMPERATURE (°C)
V
(V)
IN
IN
3122 G11
3122 G10
3122 G12
Burst Mode Maximum Output
Current vs VIN
Burst Mode Quiescent Current
Change vs Temperature
Burst Mode IZERO Current vs VIN
75
60
45
30
15
0
350
300
250
200
150
100
50
V
V
V
V
= 2.2V
= 5V
= 7.5V
= 12V
V
V
V
V
= 2.2V
OUT
OUT
OUT
OUT
OUT
OUT
OUT
OUT
140
120
100
80
= 5V
= 7.5V
= 12V
60
40
20
–15
0
0
–50
–10
30
70
110
150
0.5
1.5
2.5
3.5
4.5
5.5
0.5
1.5
2.5
V , RISING(V)
IN
3.5
4.5
5.5
TEMPERATURE (°C)
V
, FALLING (V)
IN
3122 G14
3122 G13
3122 G14
SD Pin Threshold
Frequency vs RT
Frequency Accuracy
4
3
3.0
2.5
2.0
1.5
1.0
0.5
0
12
10
8
V
V
V
= 15V
= 3.6V
= 2.2V
FREQUENCY
PERIOD
OUT
OUT
OUT
V
OUT
2
5V/DIV
1
0
6
900mV
–1
–2
–3
–4
400mV
V
SD
4
500mV/DIV
3122 G16
2
1s/DIV
0
0
1
2
3
4
5
6
0
100
200
300
(kΩ)
400
500
600
V
FALLING (V)
R
T
IN
3122 G18
3122 G17
3122f
5
LTC3122
Typical perForMance characTerisTics
Efficiency vs Frequency
CAP Pin Voltage vs VOUT
VCC vs VIN
0
–1
–2
–3
–4
–5
–6
–7
4.5
4.0
3.5
3.0
2.5
100
90
80
70
60
50
40
30
20
10
0
f
f
f
= 200kHz
= 1MHz
= 3MHz
OSC
OSC
OSC
V
V
FALLING
RISING
IN
IN
0
2
4
6
8
10
12
14
0
1
2
3
4
5
6
10
100
1000
V
(V)
V
(V)
IN
LOAD CURRENT (mA)
OUT
3122 G20
3122 G21
3122 G19
Burst Mode Operation
to PWM Mode
PWM Mode to Burst Mode
Operation
Burst Mode Operation
V
OUT
V
V
OUT
100mV/DIV
AC-COUPLED
OUT
100mV/DIV
100mV/DIV
AC-COUPLED
AC-COUPLED
V
SW
V
10V/DIV
PWM/SYNC
2V/DIV
INDUCTOR
CURRENT
0.5A/DIV
V
PWM/SYNC
2V/DIV
I
= 70mA
I
= 70mA
I
= 50mA
LOAD
LOAD
LOAD
3122 G22
3122 G23
3122 G24
5µs/DIV
20µs/DIV
20µs/DIV
Burst Mode Transient
Synchronized Operation
Short-Circuit Response
SHORT-CIRCUIT APPLIED
V
OUT
V
OUT
5V/DIV
200mV/DIV
AC-COUPLED
V
SW
SHORT-CIRCUIT
REMOVED
5V/DIV
100mA
I
LOAD
SYNCHRONIZED TO 1.3MHz
INPUT
CURRENT
2A/DIV
100mA/DIV
V
PWM/SYNC
5V/DIV
10mA
10mA
3122 G25
3122 G26
3122 G27
200µs/DIV
1µs/DIV
100µs/DIV
3122f
6
LTC3122
pin FuncTions
SW (Pin 1): Switch Pin. Connect an inductor from this
V
(Pin 5): V Regulator Output. Connect a low-ESR
CC CC
filter capacitor of at least 4.7µF from this pin to GND to
providearegulatedrailapproximatelyequaltothelowerof
pin to V . Keep PCB trace lengths as short and wide as
IN
possible to reduce EMI and voltage overshoot. An internal
V and 4.25V. When V
below 3V, V will regulate to the lower of approximately
is higher than V , and V falls
anti-ringing resistor is connected between SW and V
IN
OUT
IN IN
IN
after the inductor current has dropped to near zero, to
minimize EMI. The anti-ringing resistor is also activated
in shutdown and during the sleep periods of Burst Mode
operation.
CC
V
and 4.25V. A UVLO event occurs if V drops below
OUT
CC
1.6V. Switching is inhibited, and a soft-start is initiated
when V returns above 1.7V.
CC
RT (Pin 6): Frequency Adjust Pin. Connect an external
PGND (Pins 2, 13): Power Ground. When laying out your
PCB, provide a short, direct path between PGND and the
output capacitor and tie directly to the ground plane. The
exposed pad is ground and must be soldered to the PCB
ground plane for rated thermal performance.
resistor (R ) from this pin to SGND to program the oscil-
T
lator frequency according to the formula:
R = 57.6/ƒ
T
OSC
where ƒ
is in MHz and R is in kΩ.
T
OSC
V (Pin 3): Input Supply Pin. The device is powered from
IN
VC (Pin 7): Error Amplifier Output. A frequency compen-
sation network is connected to this pin to compensate
the control loop. See Compensating the Feedback Loop
section for guidelines.
V unless V
exceeds V and V is less than 3V. Place
IN
OUT
IN IN
a low ESR ceramic bypass capacitor of at least 4.7µF from
V
to PGND. X5R and X7R dielectrics are preferred for
IN
their superior voltage and temperature characteristics.
FB (Pin 8): Feedback Input to the Error Amplifier. Connect
PWM/SYNC (Pin 4): Burst Mode Operation Select and
OscillatorSynchronization. Donotleavethispinfloating.
the resistor divider tap to this pin. Connect the top of the
divider to V
and the bottom of the divider to SGND.
OUT
• PWM/SYNC = High. Disable Burst Mode Operation and
The output voltage can be adjusted from 2.2V to 15V ac-
cording to this formula:
maintain low noise, constant frequency operation.
• PWM/SYNC = Low. Enable Burst Mode operation.
V
OUT
= 1.202V • (1 + R1/R2)
• PWM/SYNC = External CLK. The internal oscillator is
synchronized to the external CLK signal. Burst Mode
operation is disabled. A clock pulse width between
100ns and 2µs is required to synchronize the oscillator.
An external resistor must be connected between RT
and GND to program the oscillator slightly below the
desired synchronization frequency.
SD(Pin9):LogicControlledShutdownInput.Bringingthis
pin above 1.6V enables normal, free-running operation,
forcingthispinbelow0.25VshutstheLTC3122down, with
quiescentcurrentbelow1μA.Donotleavethispinfloating.
SGND (Pin 10): Signal Ground. When laying out a PC
board, provide a short, direct path between SGND and
the (–) side of the output capacitor.
In non-synchronized applications, repeated clocking of
the PWM/SYNC pin to affect an operating mode change
is supported with these restrictions:
V
(Pin 11): Output Voltage Sense and the Source of
OUT
the Internal Synchronous Rectifier MOSFET. Driver bias
is derived from V . Connect the output filter capacitor
OUT
from V
to PGND, as close to the IC as possible. A
• Boost Mode (V
> V ): I
<500µA: ƒ
≤
PWM/SYNC
OUT
OUT
≥ 500µA: ƒ
IN OUT
minimum value of 10µF ceramic is recommended. V
100Hz, I
≤ 5kHz
OUT
OUT
PWM/SYNC
is disconnected from V when SD is low.
IN
• Buck Mode (V
< V ): I
PWM/SYNC
<5mA: ƒ
≤ 5Hz,
OUT
≥ 5mA: ƒ
IN OUT
PWM/SYNC
CAP (Pin 12): Serves as the Low Reference for the Syn-
I
≤ 5kHz
OUT
chronous Rectifier Gate Drive. Connect a low ESR filter
capacitor(typically100nF)fromthispintoV
toprovide
OUT
an elevated ground rail, approximately 5.6V below V
used to drive the synchronous rectifier.
,
OUT
3122f
7
LTC3122
block DiagraM
BULK CONTROL
SIGNALS
SW
1
V
IN
ANTI-RING
V
OUT
L1
V
OUT
11
2.2V TO 15V
V
IN
V
IN
3
C
OUT
1.8V TO 5.5V
TSD
C
PWM
LOGIC
AND
IN
V
REF_UP
OSC
C1
100nF
DRIVERS
SD
16.2V
OVLO
+ –
CURRENT
SENSE
SD
SHUTDOWN
SD
9
4
+ –
I
ZERO
PGND
COMP
OVLO
R
CAP
FB
PL
12
8
PWM
BURST
SYNC
C
PL
PWM/SYNC
R1
R2
CONTROL
+
– –
V
–
+
C
1.202V
V
V
IN OUT
g
ERROR
AMPLIFIER
m
I
LIM
REF
V
BEST
VC
7
ADAPTIVE SLOPE COMPENSATION
V
CC
R
C
F
C
5
LDO
SD
TSD
OVLO
SOFT-START
VC CLAMP
C
VCC
C
C
4.7µF
V
REFERENCE
UVLO
REF_UP
1.202V
OSCILLATOR
OSC
RT
SGND
PGND
6
10
2
TSD
THERMAL SD
R
T
EXPOSED PAD 13
LTC3122
3122 BD
THE VALUES OF RC, CC, AND CF ARE BASED UPON OPERATING CONDITIONS.
PLEASE REFER TO COMPENSATING THE FEEDBACK LOOP SECTION FOR
GUIDELINES TO DETERMINE OPTIMAL VALUES OF THESE COMPONENTS.
3122f
8
LTC3122
operaTion
The LTC3122 is an adjustable frequency, 100kHz to 3MHz
synchronous boost converter housed in either a 12-lead
4mm × 3mm DFN or a thermally enhanced MSOP pack-
age. The LTC3122 offers the unique ability to start-up
and regulate the output from inputs as low as 1.8V and
continue to operate from inputs as low as 0.5V. Output
voltages can be programmed between 2.2V and 15V. The
device also features fixed frequency, current mode PWM
control for exceptional line and load regulation. The cur-
rent mode architecture with adaptive slope compensation
provides excellent transient load response and requires
minimal output filtering. An internal 10ms closed loop
soft-start simplifies the design process while minimizing
the number of external components.
zero to its final programmed value. This limits the inrush
current drawn from the input source. As a result, the du-
ration of the soft-start is largely unaffected by the size of
the output capacitor or the output regulation voltage. The
closed loop nature of the soft-start allows the converter
to respond to load transients that might occur during
the soft-start interval. The soft-start period is reset by a
shutdown command on SD, a UVLO event on V (V
<
CC CC
≥ 16.2V), or
1.6V), an overvoltage event on V
(V
OUT OUT
an overtemperature event (thermal shutdown is invoked
when the die temperature exceeds 170°C). Upon removal
of these fault conditions, the LTC3122 will soft-start the
output voltage.
Error Amplifier
WithitslowR
andlowgatechargeinternalN-channel
DS(ON)
The non-inverting input of the transconductance error
amplifier is internally connected to the 1.202V reference
and the inverting input is connected to FB. An external
MOSFET switch and P-channel MOSFET synchronous
rectifier, the LTC3122 achieves high efficiency over a wide
range of load current. High efficiency is achieved at light
loadswhenBurstModeoperationiscommanded.Operation
can be best understood by referring to the Block Diagram.
resistive voltage divider from V
to ground programs
OUT
the output voltage from 2.2V to 15V via FB as shown in
Figure 1.
R1
R2
LOW VOLTAGE OPERATION
VOUT = 1.202V 1+
The LTC3122 is designed to allow start-up from input
voltages as low as 1.8V. When V
exceeds 2.2V, the
OUT
Selecting an R2 value of 121kΩ to have approximately
10µA of bias current in the V
the formula:
LTC3122 continues to regulate its output, even when V
IN
resistor divider yields
OUT
falls to as low as 0.5V. The limiting factors for the applica-
tion become the availability of the input source to supply
sufficient power to the output at the low voltages, and
the maximum duty cycle. Note that at low input voltages,
small voltage drops due to series resistance become
critical and greatly limit the power delivery capability of
the converter. This feature extends operating times by
maximizing the amount of energy that can be extracted
from the input source.
R1 = 100.67•(V
– 1.202V)
OUT
where R1 is in kΩ.
Power converter control loop compensation is set by a
simple RC network between V and ground.
C
V
OUT
LTC3122
R1
–
+
FB
1.202V
LOW NOISE FIXED FREQUENCY OPERATION
Soft-Start
R2
The LTC3122 contains internal circuitry to provide closed-
loop soft-start operation. The soft-start utilizes a linearly
increasing ramp of the error amplifier reference voltage
from zero to its nominal value of 1.202V in approximately
3122 F01
Figure 1. Programming the Output Voltage
10ms, with the internal control loop driving V
from
OUT
3122f
9
LTC3122
operaTion
Internal Current Limit
shutdown, and draw no current from the input source. It
also allows for inrush current limiting at turn-on, minimiz-
ing surge currents seen by the input supply. Note that to
obtain the advantages of output disconnect, there must
not be an external Schottky diode connected between SW
The current limit comparator shuts off the N-channel
MOSFETswitchonceitsthresholdisreached. Peakswitch
current is limited to 3.5A, independent of input or output
voltage, except when V
currentlimitbeingapproximatelyhalfofthenominalpeak.
is below 1.5V, resulting in the
OUT
and V . The output disconnect feature also allows V
OUT
OUT
to be pulled high, without reverse current being backfed
Lossless current sensing converts the peak current sig-
nal of the N-channel MOSFET switch into a voltage that
is summed with the internal slope compensation. The
summed signal is compared to the error amplifier output
to provide a peak current control command for the PWM.
into the power source connected to V .
IN
Shutdown
The boost converter is disabled by pulling SD below 0.25V
and enabled by pulling SD above 1.6V. Note that SD can
be driven above V or V , as long as it is limited to less
IN
OUT
Zero Current Comparator
than the absolute maximum rating.
Thezerocurrentcomparatormonitorstheinductorcurrent
being delivered to the output and shuts off the synchro-
nous rectifier when this current reduces to approximately
50mA. This prevents the inductor current from reversing
in polarity, improving efficiency at light loads.
Thermal Shutdown
If the die temperature exceeds 170°C typical, the LTC3122
will go into thermal shutdown (TSD). All switches will be
turnedoffuntilthedietemperaturedropsbyapproximately
7°C, whenthedevicere-initiatesasoft-startandswitching
can resume.
Oscillator
Theinternaloscillatorisprogrammedtothedesiredswitch-
ing frequency with an external resistor from the RT pin to
GND according to the following formula:
Boost Anti-Ringing Control
The anti-ringing control connects a resistor across the
inductor to damp high frequency ringing on the SW pin
during discontinuous current mode operation when the
inductor current has dropped to near zero. Although
57.6
ƒOSC (MHz) =
R (kΩ)
T
The oscillator also can be synchronized to an external
frequency by applying a pulse train to the PWM/SYNC pin.
An external resistor must be connected between RT and
GNDtoprogramtheoscillatortoafrequencyapproximately
25% below that of the externally applied pulse train used
the ringing of the resonant circuit formed by L and C
SW
(capacitance on SW pin) is low energy, it can cause EMI
radiation.
V
CC
Regulator
for synchronization. R is selected in this case according
T
An internal low dropout regulator generates the 4.25V
(nominal) V rail from V or V , depending upon
to this formula:
CC
IN
OUT
operating conditions. V is supplied from V when V
CC
IN
IN
IN
OUT
73.2
(MHz)
RT(kΩ) =
is greater than 3.5V, otherwise the greater of V and V
ƒ
SYNC
is used. The V rail powers the internal control circuitry
CC
and power MOSFET gate drivers of the LTC3122. The V
CC
Output Disconnect
regulator is disabled in shutdown to reduce quiescent
current and is enabled by forcing the SD pin above its
threshold. A 4.7µF or larger capacitor must be connected
The LTC3122’s output disconnect feature eliminates body
diode conduction of the internal P-channel MOSFET
between V and SGND.
CC
rectifier. This allows for V
to discharge to 0V during
OUT
3122f
10
LTC3122
applicaTions inForMaTion
Overvoltage Lockout
InBurstModeoperation, theLTC3122switchesasynchro-
nously. The inductor current is first charged to 600mA
by turning on the N-channel MOSFET switch. Once this
current threshold is reached, the N-channel is turned off
and the P-channel synchronous switch is turned on, de-
livering current to the output. When the inductor current
discharges to approximately zero, the cycle repeats. In
Burst Mode operation, energy is delivered to the output
until the nominal regulation value is reached, at which
point the LTC3122 transitions to sleep mode. In sleep, the
outputswitchesareturnedoffandtheLTC3122consumes
only 25μA of quiescent current. When the output voltage
droopsapproximately1%, switchingresumes. Thismaxi-
mizesefficiencyatverylightloadsbyminimizingswitching
and quiescent losses. Output voltage ripple in Burst Mode
operation is typically 1% peak-to-peak. Additional output
capacitance (10μF or greater), or the addition of a small
feed-forwardcapacitor(10pFto50pF)connectedbetween
An overvoltage condition occurs when V
exceeds ap-
OUT
proximately 16.2V. Switching is disabled and the internal
soft-start ramp is reset. Once V drops below approxi-
OUT
mately 15.6V, a soft-start cycle is initiated and switching
is enabled. If the boost converter output is lightly loaded
so that the time constant product of the output capaci-
tance, C , and the output load resistance, R
is near
OUT
OUT
or greater than the soft-start time of approximately 10ms,
thesoft-startrampmayendbeforeorsoonafterswitching
resumes,defeatingtheinrushcurrentlimitingoftheclosed
loop soft-start following an overvoltage event.
Short-Circuit Protection
The LTC3122 output disconnect feature allows output
short-circuitprotection.Toreducepowerdissipationunder
overload and short-circuit conditions, the peak switch
current limit is reduced to 1.6A. Once V
> 1.5V, the
OUT
V
and FB can help further reduce the output ripple.
OUT
current limit is set to its nominal value of 3.5A.
The maximum output current (I ) capability in Burst
OUT
Mode operation varies with V and V , as shown in
V > V Operation
IN
OUT
IN
OUT
Figure 2.
TheLTC3122step-upconverterwillmaintainvoltageregu-
lation even when the input voltage is above the desired
outputvoltage.Notethatoperatinginthismodewillexhibit
lower efficiency and a reduced output current capability.
Refer to the Typical Performance Characteristics section
for details.
350
300
250
200
150
100
50
V
V
V
V
= 2.2V
= 5V
= 7.5V
= 12V
OUT
OUT
OUT
OUT
Burst Mode OPERATION
When the PWM/SYNC pin is held low, the boost converter
operates in Burst Mode operation to improve efficiency
at light loads and reduce standby current at no load. The
0
0.5
1.5
2.5
3.5
4.5
5.5
input thresholds for this pin are determined relative to V
CC
V
, FALLING (V)
IN
3122 F02
with a low being less than 10% of V and a high being
CC
greater than 90% of V . The LTC3122 will operate in
Figure 2. Burst Mode Maximum Output Current vs VIN
CC
fixed frequency PWM mode even if Burst Mode operation
is commanded during soft-start.
3122f
11
LTC3122
applicaTions inForMaTion
PCB LAYOUT GUIDELINES
rent capability by reducingthe inductorripplecurrent. The
minimum inductance value, L, is inversely proportional to
operatingfrequencyandisgivenbythefollowingequation:
The high switching frequency of the LTC3122 demands
careful attention to board layout. A careless layout will
result in reduced performance. Maximizing the copper
area for ground will help to minimize die temperature rise.
A multilayer board with a separate ground plane is ideal,
but not absolutely necessary. See Figure 3 for an example
of a two-layer board layout.
V • V
ƒ •Ripple • VOUT
− V
IN
(
)
3
ƒ
IN
OUT
L >
µH and L >
where:
Ripple = Allowable inductor current ripple (amps
peak-to-peak)
ƒ = Switching Frequency in MHz
PGND
The inductor current ripple is typically set for 20% to
40% of the maximum inductor current. High frequency
ferrite core inductor materials reduce frequency depen-
dent power losses compared to cheaper powdered iron
types, improving efficiency. The inductor should have
low ESR (series resistance of the windings) to reduce the
PGND
CAP
V
SW
1
2
3
4
5
6
12
OUT
V
IN
11
10 SGND
9
13
PGND
2
V
CC
8
7
FB
I R power losses, and must be able to support the peak
inductor current without saturating. Molded chokes and
some chip inductors usually do not have enough core
area to support the peak inductor currents of 3A to 4A
seen on the LTC3122. To minimize radiated noise, use a
shielded inductor.
V
C
RT
See Table 1 for suggested components and suppliers.
3122 F02
Table 1. Recommended Inductors
MAX DC
Figure 3. Traces Carrying High Current Are Direct (PGND, SW, VIN
and VOUT). Trace Area at FB and VC Are Kept Low. Trace Length to
Input Supply Should Be Kept Short. VIN and VOUT Ceramic Capacitors
Should Be Placed as Close to the LTC3122 Pins as Possible
VALUE DCR CURRENT
SIZE (mm)
W × L × H
PART NUMBER
(µH) (mΩ)
(A)
Coilcraft LPS4018
Coilcraft MSS7341
Coilcraft MSS1260T
1
3.3
33
42
20
54.9
3.8
3.72
4.34
4 × 4 × 1.8
7.3 × 7.3 × 4.1
12.3 × 12.3 × 6.2
SCHOTTKY DIODE
Coiltronics DRQ73
Coiltronics SD7030
Coiltronics DR125
0.992
3.3
33
24
24
59
3.99
3
3.84
7.6 × 7.6 × 3.55
7 × 7 × 3
12.5 × 12.5 × 6
Although it is not required, adding a Schottky diode from
SW to V
can improve the converter efficiency by about
OUT
4%.Notethatthisdefeatstheoutputdisconnectandshort-
circuit protection features of the LTC3122.
Murata LQH6PP
Murata LQH6PP
1
9
4.3
3.8
6 × 6 × 4.3
6 × 6 × 4.3
3.3
16
Sumida CDRH50D28RNP 1.2
Sumida CDRH8D28NP
Sumida CDRH129HF
13
18
53
4.8
4
4.25
5 × 5 × 2.8
8 × 8 × 3
12 × 12 × 10
3.3
33
COMPONENT SELECTION
Inductor Selection
Taiyo-Yuden NR6045
3
31
3.2
6 × 6 × 4.5
TDK LTF5022T
TDK SPM6530T
TDK VLF12060T
1.2
3.3
33
25
20
53
4.2
4.1
3.4
5 × 5.2 × 2.2
7 × 7 × 3.2
11.7 × 12 × 6
The LTC3122 can utilize small surface mount inductors
due to its high switching frequency (up to 3MHz). Larger
values of inductance will allow slightly greater output cur-
Würth WE-PD
3.3
32.5
3.1
7.3 × 7.3 × 2
3122f
12
LTC3122
applicaTions inForMaTion
Output and Input Capacitor Selection
of its rated capacitance when operated near its rated volt-
age. As a result it is sometimes necessary to use a larger
capacitor value or a capacitor with a larger value and case
size, such as 1812 rather than 1206, in order to actually
realize the intended capacitance at the full operating volt-
age. Be sure to consult the vendor’s curve of capacitance
vsDCbiasvoltage. Table2showsasamplingofcapacitors
suited for LTC3122 applications.
Low ESR (equivalent series resistance) capacitors should
be used to minimize the output voltage ripple. Multilayer
ceramic capacitors are an excellent choice as they have
extremely low ESR and are available in small footprints.
X5R and X7R dielectric materials are preferred for their
ability to maintain capacitance over wide voltage and tem-
perature ranges. Y5V types should not be used. Although
ceramic capacitors are recommended, low ESR tantalum
capacitors may be used as well.
Table 2. Representative Output Capacitors
MANUFACTURER,
PART NUMBER
VALUE VOLTAGE SIZE L × W × H (mm)
(µF)
(V)
TYPE, ESR (mΩ)
When selecting output capacitors, the magnitude of the
peak inductor current, together with the ripple voltage
specification, determine the choice of the capacitor. Both
theESR(equivalentseriesresistance)ofthecapacitorand
the charge stored in the capacitor each cycle contribute
to the output voltage ripple.
AVX,
22
25
3.2 × 2.5 × 2.79,
X5R Ceramic
12103D226MAT2A
Kemet,
C2220X226K3RACTU
22
22
25
16
25
25
25
25
16
25
25
25
4.5
2.5
2.5
5.7 × 5.0 × 2.4,
X7R Ceramic
Kemet,
A700D226M016ATE030
7.3 × 4.3 × 2.8,
Alum. Polymer, 30mΩ
Murata,
GRM32ER71E226KE15L
22
3.2 × 2.5 × 2.5,
X7R Ceramic
The ripple due to the charge is approximately:
Nichicon,
PLV1E121MDL1
82
8 × 8 × 12,
Alum. Polymer, 25mΩ
IP • VIN
VRIPPLE(CHARGE)
≈
COUT • VOUT • ƒ
where I is the peak inductor current.
Panasonic,
ECJ-4YB1E226M
22
3.2 × 2.5 × 2.5,
X5R Ceramic
P
Sanyo,
25TQC22MV
22
7.3 × 4.3 × 3.1,
POSCAP, 50mΩ
The ESR of C
is usually the most dominant factor for
OUT
Sanyo,
16TQC100M
100
7.3 × 4.3 × 1.9,
POSCAP, 45mΩ
ripple in most power converters. The ripple due to the
capacitor ESR is:
Sanyo,
25SVPF47M
47
6.6 × 6.6 × 5.9,
OS-CON, 30mΩ
VOUT
VRIPPLE(ESR) =ILOAD •RESR
where R
•
Taiyo Yuden,
TMK325BJ226MM-T
22
3.2 × 2.5 × 2.5,
X5R Ceramic
V
IN
TDK,
47
6.5 × 5.5 × 5.5,
X5R Ceramic
= capacitor equivalent series resistance.
ESR
CKG57NX5R1E476M
Cap-XX
GS230F
1.2Farads
1.5Farads
50Farads
39 × 17 × 3.8
28mΩ
Theinputfiltercapacitorreducespeakcurrentsdrawnfrom
the input source and reduces input switching noise. A low
ESR bypass capacitor with a value of at least 4.7µF should
Cooper
A1030-2R5155
Ø = 10, L = 30
60mΩ
be located as close to the V pin as possible.
IN
Maxwell
BCAP0050-P270
Ø = 18, L = 40
20mΩ
Low ESR and high capacitance are critical to maintain low
output voltage ripple. Capacitors can be used in parallel
for even larger capacitance values and lower effective
ESR. Ceramic capacitors are often utilized in switching
converterapplicationsduetotheirsmallsize, lowESRand
low leakage currents. However, many ceramic capacitors
experience significant loss in capacitance from their rated
value with increased DC bias voltage. It is not uncommon
forasmallsurfacemountcapacitortolosemorethan50%
For applications requiring a very low profile and very large
capacitance, the GS, GS2 and GW series from Cap-XX
and PowerStor Aerogel Capacitors from Cooper all offer
very high capacitance and low ESR in various low profile
packages.
Amethodforimprovingtheconverter’stransientresponse
usesasmallfeed-forwardseriesnetworkofacapacitorand
3122f
13
LTC3122
applicaTions inForMaTion
a resistor across the top resistor of the feedback divider
possible. If the junction temperature rises above ~170°C,
the part will go into thermal shutdown, and all switching
will stop until the temperature drops approximately 7°C.
(from V
to FB). This adds a phase-lead zero and pole
OUT
to the transfer function of the converter as calculated in
the Compensating the Feedback Loop section.
Compensating the Feedback Loop
OPERATING FREQUENCY SELECTION
The LTC3122 uses current mode control, with internal
adaptiveslopecompensation.Currentmodecontrolelimi-
natesthesecondorderfilterduetotheinductorandoutput
capacitorexhibitedinvoltagemodecontrol,andsimplifies
the power loop to a single pole filter response. Because
of this fast current control loop, the power stage of the IC
combined with the external inductor can be modeled by a
Thereareseveralconsiderationsinselectingtheoperating
frequencyoftheconverter.Typicallythefirstconsideration
is to stay clear of sensitive frequency bands, which cannot
tolerateanyspectralnoise.Forexample,inproductsincor-
porating RF communications, the 455kHz IF frequency is
sensitive to any noise, therefore switching above 600kHz
is desired. Some communications have sensitivity to
1.1MHz and in that case a 1.5MHz switching converter
frequencymaybeemployed.Asecondconsiderationisthe
physical size of the converter. As the operating frequency
is increased, the inductor and filter capacitors typically
can be reduced in value, leading to smaller sized external
components. The smaller solution size is typically traded
forefficiency,sincetheswitchinglossesduetogatecharge
increase with frequency.
transconductance amplifier g and a current controlled
mp
current source. Figure 4 shows the key equivalent small
signal elements of a boost converter.
The DC small-signal loop gain of the system shown in
Figure 4 is given by the following equation:
R2
R1+R2
GBOOST = GEA •GMP •GPOWER
•
where G is the DC gain of the error amplifier, G is
EA
MP
the modulator gain, and G
is the inductor current
POWER
Anotherconsiderationiswhethertheapplicationcanallow
pulse-skipping.Whentheboostconverterpulse-skips,the
minimum on-time of the converter is unable to support
the duty cycle. This results in a low frequency component
to the output ripple. In many applications where physical
size is the main criterion, running the converter in this
mode is acceptable. In applications where it is preferred
not to enter this mode, the maximum operating frequency
is given by:
to V
gain.
OUT
–
+
V
OUT
g
mp
I
L
η • V
IN
• I
R
R
L
L
ESR
2 • V
OUT
C
OUT
MODULATOR
R
PL
1.202V
REFERENCE
C
PL
+
–
V
C
R1
R2
g
ma
VOUT − VIN
FB
ƒMAX _NOSKIP
≤
Hz
R
R
O
C
ERROR
AMPLIFIER
C
F
VOUT • tON(MIN)
= minimum on-time = 100ns.
C
C
where t
3122 F04
ON(MIN)
C : COMPENSATION CAPACITOR
C
C
C
: OUTPUT CAPACITOR
OUT
: PHASE LEAD CAPACITOR
PL
Thermal Considerations
C : HIGH FREQUENCY FILTER CAPACITOR
F
g
g
: TRANSCONDUCTANCE AMPLIFIER INSIDE IC
: POWER STAGE TRANSCONDUCTANCE AMPLIFIER
ma
mp
For the LTC3122 to deliver its full power, it is imperative
that a good thermal path be provided to dissipate the heat
generated within the package. This can be accomplished
by taking advantage of the large thermal pad on the un-
derside of the IC. It is recommended that multiple vias in
the printed circuit board be used to conduct heat away
from the IC and into a copper plane with as much area as
R : COMPENSATION RESISTOR
C
R : OUTPUT RESISTANCE DEFINED AS V /I
L
OUT LOADMAX
R : OUTPUT RESISTANCE OF g
O
PL
ma
R
: PHASE LEAD RESISTOR
R1, R2: FEEDBACK RESISTOR DIVIDER NETWORK
R
: OUTPUT CAPACITOR ESR
ESR
η : CONVERTER EFFICIENCY (~90% AT HIGHER CURRENTS)
Figure 4. Boost Converter Equivalent Model
3122f
14
LTC3122
applicaTions inForMaTion
GEA = gma •RO ≈ 950V/V
The current mode zero (Z3) is a right half plane zero
which can be an issue in feedback control design, but is
manageable with proper external component selection.
As a general rule, the frequency at which the open-loop
gain of the converter is reduced to unity, known as the
(Not Adjustable; gma = 95µS, RO ≈ 10MΩ)
∆IL
GMP = gmp
=
≈ 3.4S (Not Adjustable)
∆VC
crossover frequency ƒ , should be set to less than one
C
η• VIN
third of the right half plane zero (Z3), and under one eighth
∆VOUT
∆IL
GPOWER
=
=
of the switching frequency ƒ . Once ƒ is selected, the
OSC
C
2 •IOUT
valuesforthecompensationcomponentscanbecalculated
using a bode plot of the power stage or two generally valid
assumptions: P1 dominates the gain of the power stage
Combining the two equations above yields:
1.7 • η• VIN
for frequencies lower than ƒ and ƒ is much higher than
C
C
GDC = GMP •GPOWER
≈
V/V
P2. First calculate the power stage gain at ƒ , G in V/V.
IOUT
C
ƒC
AssumingtheoutputpoleP1dominatesG forthisrange,
ƒC
it is expressed by:
Converter efficiency η will vary with I
and switching
OUT
frequency ƒ
as shown in the typical performance
OSC
GDC
GƒC
≈
V/V
characteristics curves.
2
ƒ
P1
C
1+
2
Output Pole: P1 =
Hz
1
2 • π •RL •COUT
Decide how much phase margin (Φ ) is desired. Greater
m
Error Amplifier Pole: P2 =
Error Amplifier Zero: Z1=
Hz
phasemargincanoffermorestabilitywhilelowerphasemar-
2 • π •RO •(CC +CF )
gincanyieldfastertransientresponse. Typically, Φ ≈60°
m
1
is optimal for minimizing transient response time while
Hz
allowing sufficient margin to account for component vari-
2 • π •RC •CC
ability. Φ is the phase boost of Z1, P2, and P5 while Φ is
1
2
1
ESR Zero: Z2 =
RHP Zero: Z3 =
Hz
the phase boost of Z5 and P4. Select Φ and Φ such that
1
2
2 • π •RESR •COUT
VOUT
1.2V
V
2 •RL
2 • π • VOUT2 •L
Φ1 ≤ 74°; Φ2 ≤ 2 • tan−1
− 90° and
IN
Hz
ƒ
Z3
C
Φ1 + Φ2 = Φm + tan−1
ƒOSC
3
High Frequency Pole: P3 >
ꢀ
1
where V
is in V and ƒ and Z3 are in kHz.
C
OUT
Phase Lead Zero: Z4 =
Hz
2 • π •(R1+RPL )•CPL
Setting Z1, P5, Z4, and P4 such that
ƒC ƒC
1
Phase Lead Pole: P4 =
Hz
Z1=
, P5 = ƒC a1, Z4 =
, P4 = ƒC a2
R1•R2
R1+R2
2 • π •
+R
•C
PL
a1
a2
PL
allows a and a to be determined using Φ and Φ
Error Amplifier Filter Pole:
1
2
1
2
1
Φ + 90°
Φ +90°
a1 = tan2
, a = tan2
1
2
P5 =
Hz
2
2
2
CC •CF
2 • π •RC •
C +C
F
C
3122f
15
LTC3122
applicaTions inForMaTion
The compensation will force the converter gain G
Once the compensation values have been calculated, ob-
taining a converter bode plot is strongly recommended to
verify calculations and adjust values as required.
BOOST
to unity at ƒ by using the following expression for C :
C
C
3
10 • g • R2 • G
a − 1
a
2
(
)
ma
1
ƒC
C =
pF
Using the circuit in Figure 5 as an example, Table 3 shows
the parameters used to generate the bode plot shown in
Figure 6.
C
2π • ƒ • R1+ R2
a
1
(
)
C
(g in µS, ƒ in kHz, G in V/V)
ma
C
ƒC
Table 3. Bode Plot Parameters for Type II Compensation
Once C is calculated, R and C are determined by:
C
C
F
PARAMETER
VALUE
5
UNITS
V
COMMENT
App Specific
App Specific
App Specific
App Specific
App Specific
App Specific
Adjustable
Adjustable
Adjustable
Fixed
106 • a1
2π • ƒC •CC
V
IN
RC =
kΩ (ƒC in kHz, CC in pF)
V
OUT
12
V
R
C
15
Ω
L
CC
a1 −1
22
µF
CF =
OUT
R
ESR
5
mΩ
µH
MHz
kΩ
kΩ
µS
MΩ
S
L
ƒ
3.3
1
The values of the phase lead components are given by
the expressions:
OSC
R1
R2
g
1020
113
95
R1•R2
R1+R2
a2 −1
R1− a •
2
ma
RPL
CPL
=
kΩ and
R
O
10
Fixed
g
mp
3.4
80
Fixed
106 a −1 R1+R2
(
)
)
pF
(
2
%
App Specific
Adjustable
Adjustable
Adjustable
Optional
η
=
2π • ƒC •R12 a2
R
210
390
10
kΩ
pF
C
C
F
C
C
where R1, R2, and R are in kΩ and ƒ is in kHz.
PL
C
pF
R
0
kΩ
pF
PL
PL
Note that selecting Φ = 0° forces a = 1, and so the
2
2
C
0
Optional
converter will have Type II compensation and therefore
no feedforward: R is open (infinite impedance) and C
PL
PL
From Figure 6, the phase is 60° when the gain reaches
0dB, so the phase margin of the converter is 60°. The
crossover frequency is 15kHz, which is more than three
times lower than the 108.4kHz frequency of the RHP zero
to achieve adequate phase margin.
= 0pF. If a = 0.833 • V
(its maximum), feedforward is
PL
2
OUT
maximized; R = 0 and C is maximized for this com-
PL
pensation method.
3122f
16
LTC3122
applicaTions inForMaTion
L1
3.3µH
V
IN
5V
SW
V
OUT
V
V
OUT
12V
IN
800mA
LTC3122
C1
100nF
C
IN
SD
OFF ON
4.7µF
R1
1.02M
BURST PWM
PWM/SYNC
RT
CAP
FB
C
OUT
22µF
R2
113k
V
V
C
CC
R
R
T
C
SGND
PGND
C
57.6k
210k
F
10pF
C
C
390pF
C
VCC
4.7µF
3122 F05a
Transient Response with 400mA
to 800mA Load Step
Switching Waveforms with 800mA Load
V
OUT
100mV/DIV
AC-COUPLED
V
OUT
500mV/DIV
AC-COUPLED
SW
10V/DIV
INDUCTOR
CURRENT
1A/DIV
I
LOAD
500mA/DIV
3122 F05b
3122 F05c
200ns/DIV
100µs/DIV
Figure 5. 1MHz, 5V to 12V, 800mA Boost Converter
170
150
130
110
90
180
140
100
60
PHASE
20
70
–20
–60
–100
–140
–180
–220
50
30
GAIN
10
10
–10
–30
0.01
0.1
1
100
1000
FREQUENCY (kHz)
3122 F06
Figure 6. Bode Plot for Example Converter
3122f
17
LTC3122
applicaTions inForMaTion
L1
3.3µH
V
IN
5V
V
OUT
SW
V
V
OUT
12V
IN
800mA
C1
100nF
LTC3122
C
IN
SD
OFF ON
R
PL
4.7µF
604k
BURST PWM
PWM/SYNC
CAP
FB
1.02M
C
PL
10pF
C
OUT
22µF
RT
V
V
C
CC
R2
113k
SGND
PGND
R
R
T
57.6k
C
C
F
127k
33pF
C
C
C
VCC
4.7µF
220pF
3122 F06
Figure 7. Boost Converter with Phase Lead
The circuit in Figure 7 shows the same application as
that in Figure 5 with Type III compensation. This is ac-
From Figure 8, the phase margin is still optimized at 60°
and the crossover frequency remains 15kHz. Adding C
PL
complished by adding C and R and adjusting C , C ,
and R provides some feedforward signal in Burst Mode
PL
PL
C
F
PL
and R accordingly. Table 4 shows the parameters used
operation, leading to lower output voltage ripple.
C
to generate the bode plot shown in Figure 8.
170
150
130
110
90
180
140
100
60
Table 4. Bode Plot Parameters for Type III Compensation
PARAMETER
VALUE
5
UNITS
V
COMMENT
App Specific
App Specific
App Specific
App Specific
App Specific
App Specific
Adjustable
Adjustable
Adjustable
Fixed
PHASE
V
V
IN
12
V
OUT
20
R
15
Ω
L
70
–20
–60
–100
–140
–180
–220
C
22
µF
OUT
50
GAIN
R
L
5
mΩ
µH
MHz
kΩ
kΩ
µS
MΩ
S
ESR
30
3.3
1
10
–10
ƒ
OSC
–30
R1
R2
113
1020
95
0.01
0.1
1
10
100
1000
FREQUENCY (kHz)
3122 F08
g
ma
Figure 8. Bode Plot Showing Phase Lead
R
10
Fixed
O
g
3.4
80
Fixed
mp
%
App Specific
Adjustable
Adjustable
Adjustable
Adjustable
Adjustable
η
R
127
220
33
kΩ
pF
C
C
F
C
C
pF
R
604
10
kΩ
pF
PL
C
PL
3122f
18
LTC3122
Typical applicaTions
Single Li-Cell to 6V, 5W Synchronous Boost Converter for RF Transmitter
L1
3.3µH
V
IN
2.5V TO 4.2V
V
= 3.6V
SW
IN
V
OUT
V
V
V
OUT
6V
IN
OUT
500mV/DIV
833mA
LTC3122
C1
100nF
C
IN
SD
OFF ON
AC-COUPLED
4.7µF
R1
487k
PWM/SYNC
RT
CAP
FB
C
OUT
22µF
833mA
R2
121k
V
V
C
CC
OUTPUT
CURRENT
500mA/DIV
80mA
80mA
R
R
T
C
SGND
PGND
C
57.6k
73.2k
F
47pF
C
C
560pF
C
3122 TA02b
VCC
4.7µF
100µs/DIV
C
, C : 4.7µF, 16V, X7R, 1206
3122 TA02a
IN VCC
C1: 100nF, 16V, X7R, 1206
C
: 22µF, 16V, X7R, 1812
OUT
L1: TDK SPM6530T-3R3M
2 AA Cell to 12V Synchronous Boost Converter, 180mA
L1
3.3µH
V
IN
1.8V TO 3V
2.3
100
90
80
70
60
50
40
30
20
10
0
SW
V
OUT
12V
2.1
1.9
1.7
1.5
1.3
1.1
0.9
0.7
0.5
V
V
OUT
IN
180mA
LTC3122
C1
SD
OFF ON
100nF
R1
C
IN
PWM/SYNC
RT
CAP
FB
1.02M
4.7µF
C
OUT
22µF
R2
113k
V
V
C
CC
R
R
T
C
SGND
PGND
57.6k
200k
C
F
10pF
C
VCC
4.7µF
C
C
560pF
EFFICIENCY
INPUT CURRENT
C
, C : 4.7µF, 16V, X7R, 1206
3122 TA03a
1.6 1.8
2
2.2 2.4 2.6 2.8
(V)
3
3.2
IN VCC
C1: 100nF, 16V, X7R, 1206
: 22µF, 25V, X7R, 1812
V
IN
C
3122 TA03b
OUT
L1: TDK SPM6530T-3R3M
3122f
19
LTC3122
Typical applicaTions
3.3V to 12V Synchronous Boost Converter with Output Disconnect, 500mA
L1
3.3µH
V
IN
3.3V
SW
V
OUT
V
V
OUT
12V
IN
500mA
LTC3122
C1
100nF
SD
OFF ON
R1
1.02M
C
IN
PWM/SYNC
RT
CAP
FB
SW
5V/DIV
4.7µF
C
OUT
22µF
R2
113k
V
V
C
INDUCTOR
CURRENT
1A/DIV
CC
R
R
T
C
SGND
PGND
57.6k
232k
C
F
10pF
C
VCC
4.7µF
C
C
3122 TA04b
500ns/DIV
470pF
C
, C : 4.7µF, 16V, X7R, 1206
3122 TA04a
IN VCC
C1: 100nF, 16V, X7R, 1206
C
: 22µF, 25V, X7R, 1812
OUT
L1: TDK SPM6530T-3R3M
USB/Battery Powered Synchronous Boost Converter, 4.3V to 5V, 1A
L1
3.3µH
V
IN
4.3V TO 5.5V
V
= 4.3V
SW
IN
V
5V
1A
OUT
V
V
OUT
IN
V
OUT
LTC3122
C1
100nF
500mV/DIV
SD
OFF ON
AC-COUPLED
R1
383k
C
IN
PWM/SYNC
RT
CAP
FB
4.7µF
C
OUT
100µF
1A
R2
121k
OUTPUT
CURRENT
500mA/DIV
V
V
C
CC
100mA
R
R
C
T
SGND
PGND
57.6k
43.2k
C
F
68pF
C
VCC
4.7µF
C
C
3122 TA05b
200µs/DIV
1000pF
C
, C : 4.7µF, 16V, X7R, 1206
3122 TA05a
IN VCC
C1: 100nF, 16V, X7R, 1206
: 100µF, 16V, X7R, 1812
C
OUT
L1: TDK SPM6530T-3R3M
3122f
20
LTC3122
Typical applicaTions
5V to Dual Output Synchronous Boost Converter, 15V
C2
470nF
L1
3.3µH
V
IN
–15.1
–15.0
–14.9
–14.8
–14.7
–14.6
–14.5
–14.4
–14.3
–14.2
–14.1
15.1
5V
15.0
SW
V
OUT1
14.9
V
OUT1
V
V
OUT
IN
15V
LTC3122
C1
100nF
OFF ON
U1
14.8
14.7
14.6
14.5
14.4
14.3
14.2
14.1
SD
R1
C
C
IN
PWM/SYNC
RT
CAP
FB
OUT1
22µF
1.3M
4.7µF
R2
113k
V
V
CC
C
V
OUT2
V
OUT2
R
R
C
T
SGND
PGND
–15V
57.6k
365k
C
F
C
Z1
OUT2
10pF
C
VCC
4.7µF
C
C
47µF
150pF
0
50
100
150
200
C
C
, C : 4.7µF, 16V, X7R, 1206
: 47µF, 25V, X7R, 1206
C1: 100nF, 16V, X7R, 1206
: 22µF, 25V, X7R, 1812
C2: 470nF, 25V, X7R, 1206
L1: TDK SPM6530T-3R3M
3122 TA06a
IN VCC
OUTPUT CURRENT (mA)
3122 TA06b
OUT2
C
OUT1
U1: CENTRAL SEMICONDUCTOR CBAT54S
Z1: DIODES, INC. DDZ16ASF-7
Single Li-Cell 3-LED Driver, 2.5V/4.2V to 350mA
L1
3.3µH
V
IN
2.5V TO
4.2V
SW
V
IN
= 3.6V
V
V
OUT
IN
LTC3122
D1
D2
D3
C1
SD
OFF ON
SD
100nF
5V/DIV
C
IN
C
OUT1
22µF
V
PWM/SYNC
RT
CAP
FB
CC
4.7µF
LT1006
+
–
V
V
C
CC
R
LED
CURRENT
100mA/DIV
S
0.1Ω
R
R
C
2k
T
SGND
PGND
57.6k
C
VCC
C
R1
1.02M
R2
30.9k
C
3122 TA07b
4.7µF
2ms/DIV
3.9nF
C
, C : 4.7µF, 6V, X7R, 1206
3122 TA07a
IN VCC
C1: 100nF, 6V, X7R, 1206
: 22µF, 16V, X7R, 1812
C
OUT
L1: TDK SPM6530T-3R3M
D1, D2, D3: CREE XPGWHT-L1-0000-00G51
3122f
21
LTC3122
package DescripTion
DE/UE Package
12-Lead Plastic DFN (4mm × 3mm)
(Reference LTC DWG # 05-08-1695 Rev D)
0.70 ±0.05
3.30 ±0.05
3.60 ±0.05
2.20 ±0.05
1.70 ± 0.05
PACKAGE OUTLINE
0.25 ± 0.05
0.50 BSC
2.50 REF
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
0.40 ± 0.10
4.00 ±0.10
(2 SIDES)
R = 0.115
TYP
7
12
R = 0.05
TYP
3.30 ±0.10
3.00 ±0.10
(2 SIDES)
1.70 ± 0.10
PIN 1
TOP MARK
(NOTE 6)
PIN 1 NOTCH
R = 0.20 OR
0.35 × 45°
CHAMFER
(UE12/DE12) DFN 0806 REV D
6
1
0.25 ± 0.05
0.75 ±0.05
0.200 REF
0.50 BSC
2.50 REF
BOTTOM VIEW—EXPOSED PAD
0.00 – 0.05
NOTE:
1. DRAWING PROPOSED TO BE A VARIATION OF VERSION
(WGED) IN JEDEC PACKAGE OUTLINE M0-229
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
3122f
22
LTC3122
package DescripTion
MSE Package
12-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1666 Rev F)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ±0.102
(.112 ±.004)
2.845 ±0.102
(.112 ±.004)
0.889 ±0.127
(.035 ±.005)
1
6
0.35
REF
1.651 ±0.102
(.065 ±.004)
5.23
(.206)
MIN
1.651 ±0.102
(.065 ±.004)
3.20 – 3.45
(.126 – .136)
0.12 REF
DETAIL “B”
CORNER TAIL IS PART OF
THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
DETAIL “B”
12
7
0.65
(.0256)
BSC
0.42 ±0.038
4.039 ±0.102
(.159 ±.004)
(NOTE 3)
(.0165 ±.0015)
TYP
0.406 ±0.076
RECOMMENDED SOLDER PAD LAYOUT
(.016 ±.003)
12 11 10 9 8 7
REF
DETAIL “A”
0.254
(.010)
3.00 ±0.102
(.118 ±.004)
(NOTE 4)
0° – 6° TYP
4.90 ±0.152
(.193 ±.006)
GAUGE PLANE
0.53 ±0.152
(.021 ±.006)
1
2 3 4 5 6
DETAIL “A”
0.86
(.034)
REF
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.22 – 0.38
(.009 – .015)
TYP
0.1016 ±0.0508
(.004 ±.002)
MSOP (MSE12) 0911 REV F
0.650
(.0256)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
3122f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LTC3122
Typical applicaTion
Dual Supercapacitor Backup Power Supply, 0.5V to 5V
L1
3.3µH
V
IN
0.5V TO 5V
V
OUT
SW
V
V
OUT
IN
V
OUT
5V
20mV/DIV
AC-COUPLED
C1
100nF
LTC3122
SC1
50F
OFF ON
SD
SW
R1
383k
SC2
50F
5V/DIV
PWM/SYNC
RT
CAP
FB
C
OUT
100µF
C
IN
V
V
C
R2
121k
4.7µF
CC
INDUCTOR
CURRENT
500mA/DIV
R
T
R
C
SGND
PGND
57.6k
43.2k
C
F
C
VCC
4.7µF
68pF
C
1nF
C
3122 TA08b
500ns/DIV
OUTPUT CURRENT = 50mA
V
IN
= 0.5V
C
, C : 4.7µF, 16V, X7R, 1206
IN VCC
3122 TA08a
C1: 100nF, 16V, X7R, 1206
: 100µF, 16V, X7R, 1812
C
OUT
L1: TDK SPM6530T-3R3M
SC1, SC2: MAXWELL BCAP0050-P270
relaTeD parTs
PART NUMBER
DESCRIPTION
COMMENTS
LTC3421
3A I , 3MHz, Synchronous Step-Up DC/DC Converter
95% Efficiency, V = 0.5V to 4.5V, V
SD
= 5.25V, I = 12μA,
Q
SW
IN
OUT(MAX)
OUT(MAX)
OUT(MAX)
with Output Disconnect
I
< 1μA, QFN24 Package
LTC3422
LTC3112
LTC3458
LTC3528
LTC3539
LTC3459
LTC3499
LTC3115-1
1.5A I , 3MHz Synchronous Step-Up DC/DC Converter
95% Efficiency, V = 0.5V to 4.5V, V
= 5.25V, I = 25μA,
Q
SW
IN
with Output Disconnect
I
SD
< 1μA, 3mm × 3mm DFN Package
2.5A I , 750kHz, Synchronous Buck-Boost DC/DC Converter 95% Efficiency, V = 2.7V to 15V, V
= 14V, I = 50μA,
Q
SW
IN
with Output Disconnect, Burst Mode Operation
I
SD
< 1μA, 4mm × 5mm DFN and TSSOP Packages
1.4A I , 1.5MHz, Synchronous Step-Up DC/DC Converter/
93% Efficiency, V = 1.5V to 6V, V
SD
= 7.5V, I = 15μA,
OUT(MAX) Q
SW
IN
Output Disconnect/Burst Mode Operation
I
< 1μA, DFN12 Package
1A I , 1MHz, Synchronous Step-Up DC/DC Converter
94% Efficiency, V = 700mV to 5.25V, V
= 5.25V, I = 12µA,
Q
SW
IN
OUT(MAX)
OUT(MAX)
with Output Disconnect/Burst Mode Operation
I
SD
< 1µA, 3mm × 2mm DFN Package
2A I , 1MHz/2MHz, Synchronous Step-Up DC/DC Converters 94% Efficiency, V = 700mV to 5.25V, V
= 5.25V, I = 10µA,
Q
SW
IN
with Output Disconnect/Burst Mode Operation
I
SD
< 1µA, 3mm × 2mm DFN Package
70mA I , 10V Micropower Synchronous Boost Converter/
V
= 1.5V to 5.5V, V
= 10V, I = 10μA, I < 1μA,
SW
IN
OUT(MAX)
Q
SD
Output Disconnect/Burst Mode Operation
ThinSOT™ Package
750mA Synchronous Step-Up DC/DC Converters with
Reverse-Battery Protection
94% Efficiency, V = 1.8V to 5.5V, V
= 6V, I = 20µA,
Q
IN
OUT(MAX)
I
SD
< 1µA, 3mm × 3mm DFN and MSOP Packages
40V, 2A Synchronous Buck-Boost DC/DC Converter
95% Efficiency, V = 2.7V to 40V, V
SD
= 40V, I = 50µA,
IN
OUT(MAX) Q
I
< 3µA, 4mm × 5mm DFN and TSSOP Packages
3122f
LT 0712 • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
24
●
●
ꢀLINEAR TECHNOLOGY CORPORATION 2012
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
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Linear
LTC3424EMS#PBF
LTC3424 - Low Output Voltage, 3MHz Micropower Synchronous Boost Converters; Package: MSOP; Pins: 10; Temperature Range: -40°C to 85°C
Linear
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