LTC3772BETS8#TRPBF
更新时间：2024-10-29 18:56:23
品牌：Linear
描述：LTC3772B - Micropower No RSENSE Constant Frequency Step-Down DC/DC Controller; Package: SOT; Pins: 8; Temperature Range: -40°C to 85°C
概述
规格参数
数据手册LTC3772BETS8#TRPBF 概述
LTC3772B - Micropower No RSENSE Constant Frequency Step-Down DC/DC Controller; Package: SOT; Pins: 8; Temperature Range: -40°C to 85°C 开关式稳压器或控制器
LTC3772BETS8#TRPBF 规格参数
| 是否Rohs认证： | 符合 | 生命周期： | Transferred |
| 零件包装代码： | SOT | 包装说明： | VSSOP, TSSOP8,.1 |
| 针数： | 8 | Reach Compliance Code： | compliant |
| ECCN代码： | EAR99 | HTS代码： | 8542.39.00.01 |
| 风险等级： | 5.16 | 模拟集成电路 - 其他类型： | SWITCHING CONTROLLER |
| 控制模式： | CURRENT-MODE | 最大输入电压： | 9.8 V |
| 最小输入电压： | 2.75 V | 标称输入电压： | 4.2 V |
| JESD-30 代码： | R-PDSO-G8 | JESD-609代码： | e3 |
| 长度： | 2.9 mm | 湿度敏感等级： | 1 |
| 功能数量： | 1 | 端子数量： | 8 |
| 最高工作温度： | 70 °C | 最低工作温度： | |
| 最大输出电流： | 1 A | 封装主体材料： | PLASTIC/EPOXY |
| 封装代码： | VSSOP | 封装等效代码： | TSSOP8,.1 |
| 封装形状： | RECTANGULAR | 封装形式： | SMALL OUTLINE, VERY THIN PROFILE, SHRINK PITCH |
| 峰值回流温度（摄氏度）： | 260 | 认证状态： | Not Qualified |
| 座面最大高度： | 1 mm | 子类别： | Switching Regulator or Controllers |
| 表面贴装： | YES | 切换器配置： | SINGLE |
| 最大切换频率： | 650 kHz | 技术： | CMOS |
| 温度等级： | COMMERCIAL | 端子面层： | Matte Tin (Sn) |
| 端子形式： | GULL WING | 端子节距： | 0.65 mm |
| 端子位置： | DUAL | 处于峰值回流温度下的最长时间： | 30 |
| 宽度： | 1.625 mm | Base Number Matches： | 1 |
LTC3772BETS8#TRPBF 数据手册
通过下载LTC3772BETS8#TRPBF数据手册来全面了解它。这个PDF文档包含了所有必要的细节，如产品概述、功能特性、引脚定义、引脚排列图等信息。
PDF下载LTC3772B
Micropower No RSENSE
Constant Frequency Step-Down
DC/DC Controller
U
FEATURES
DESCRIPTIO
The LTC®3772B is a constant frequency current mode
step-downDC/DCcontrollerinalowprofile8-leadSOT-23
■
No Current Sense Resistor Required
■
High Output Currents Easily Achieved
(ThinSOTTM
) and a 3mm × 2mm DFN package. The No
■
Internal Soft-Start Ramps VOUT
RSENSETM architecture eliminates the need for a current
sense resistor, improving efficiency and saving board
space.
■
Wide VIN Range: 2.75V to 9.8V
■
Low Dropout: 100% Duty Cycle
■
Constant Frequency 550kHz Operation
■
Low Ripple Pulse Skipping Operation at Light Load
The LTC3772B automatically switches into pulse skipping
operation at light loads. It consumes only 200µA of quies-
cent current under a no-load condition.
■
Output Voltage as Low as 0.8V
■
±1.5% Voltage Reference Accuracy
■
Current Mode Operation for Excellent Line and Load
The LTC3772B incorporates an undervoltage lockout fea-
ture that shuts down the device when the input voltage
falls below 2V. To maximize the runtime from a battery
source, the external P-channel MOSFET is turned on
continuously in dropout (100% duty cycle). High switch-
ing frequency of 550kHz allows the use of a small inductor
and capacitors. An internal soft-start smoothly ramps the
output voltage from zero to its regulation point.
Transient Response
■
■
Only 8µA Supply Current in Shutdown
Low Profile 8-Lead SOT-23 (1mm) and
(3mm × 2mm) DUFN (0.75mm) Packages
APPLICATIO S
■
1- or 2-Cell Li-Ion Battery-Powered Applications
■
Wireless Devices
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
■
Portable Computers
Distributed Power Systems
ThinSOT and No R
are trademarks of Linear Technology Corporation.
SENSE
All other trademarks are the property of their respective owners. Protected by
U.S. Patents including 5731694, 6127815.
■
U
TYPICAL APPLICATIO
Efficiency and Power Loss vs Load Current
(Figure 5 Circuit)
550kHz Micropower Step-Down DC/DC Converter
680pF
20k
100
90
80
70
60
50
40
30
20
10
0
10
V
IN
I
/RUN
V
TH
IN
PGATE
SW
2.75V TO 9.8V
LTC3772B
10µF
EFFICIENCY
1
GND
3.3µH
82.5k
V
2.5V
2A
OUT
V
FB
0.1
0.01
0.001
47µF
POWER LOSS
22pF 174k
3772B TA01
V
V
= 3.3V
= 5V
IN
IN
1
10
100
1000
10000
LOAD CURRENT (mA)
3772B TA01b
3772bfa
1
LTC3772B
W W
U W
ABSOLUTE MAXIMUM RATINGS (Note 1)
Operating Temperature Range (Note 2) .. – 40°C to 85°C
Junction Temperature (Note 3)............................ 125°C
Storage Temperature Range ................. –65°C to 125°C
Lead Temperature (Soldering, 10 sec)
Input Supply Voltage (VIN)........................ –0.3V to 10V
IPRG Voltage ............................... –0.3V to (VIN + 0.3V)
VFB, ITH/RUN Voltages ............................. –0.3V to 2.4V
SW Voltage ........... –2V to (VIN + 1V) or 10V Maximum
PGATE Peak Output Current (<10µs) ........................ 1A
TSOT-23 ........................................................... 300°C
U
W U
PACKAGE/ORDER INFORMATION
TOP VIEW
TOP VIEW
GND
1
2
3
4
8
7
6
5
PGATE
V
FB
V
I 1
PRG
/RUN 2
8 NC
IN
9
I
7 SW
6 V
IN
TH
I
TH
/RUN
SW
NC
V
FB
3
I
PRG
GND 4
5 PGATE
TS8 PACKAGE
DDB PACKAGE
8-LEAD PLASTIC TSOT-23
8-LEAD (3mm × 2mm) PLASTIC DFN
TJMAX = 125°C, θJA = 230°C/W
TJMAX = 125°C, θJA = 76°C/W
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
ORDER PART NUMBER
DDB PART MARKING
LBWP
ORDER PART NUMBER
LTC3772BETS8
TS8 PART MARKING
LTBWN
LTC3772BEDDB
Order Options Tape and Reel: Add #TR
Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF
Lead Free Part Marking: http://www.linear.com/leadfree/
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The
●
indicates specifications which apply over the full operating
temperature range, otherwise specifications are at T = 25°C. V = 4.2V unless otherwise noted. (Note 2)
A
IN
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input Voltage Range
●
2.75
9.8
V
Input DC Supply Current
No Load
Shutdown
(Note 4)
V
V
V
= 0.83V
200
8
1
325
20
5
µA
µA
µA
FB
/RUN = 0V
ITH
UVLO
< UVLO Threshold – 100mV
IN
Undervoltage Lockout (UVLO) Threshold
V
V
Rising
Falling
●
●
2.0
1.85
2.75
2.60
V
V
IN
IN
Start-Up Current Source
V
V
/RUN = 0V
0.7
0.3
1.2
0.6
1.7
µA
ITH
ITH
Shutdown Threshold (at I /RUN)
/RUN Rising
●
●
0.95
V
TH
Regulated Feedback Voltage
0°C ≤ T ≤ 85°C (Note 5)
–40°C ≤ T ≤ 85°C (Note 5)
0.788
0.780
0.800
0.800
0.812
0.812
V
V
A
A
Feedback Voltage Line Regulation
Feedback Voltage Load Regulation
2.75V ≤ V ≤ 9V (Note 5)
0.08
0.2
mV/V
IN
I
I
/RUN = 1.6V (Note 5)
TH
/RUN = 1V (Note 5)
TH
0.5
–0.5
0.2
–0.2
%
%
3772bfa
2
LTC3772B
ELECTRICAL CHARACTERISTICS
The
●
indicates specifications which apply over the full operating
temperature range, otherwise specifications are at T = 25°C. V = 4.2V unless otherwise noted. (Note 2)
A
IN
PARAMETER
Input Current
CONDITIONS
(Note 5)
MIN
–10
TYP
2
MAX
10
UNITS
nA
V
FB
Overvoltage Protect Threshold
Overvoltage Protect Hysteresis
Measured at V
0.850
0.880
40
0.910
V
FB
mV
Oscillator Frequency
Normal Operation
Output Short Circuit
V
V
= 0.8V
= 0V
500
550
200
650
kHz
kHz
FB
FB
Gate Drive Rise Time
Gate Drive Fall Time
C
C
= 3000pF
= 3000pF
40
40
ns
ns
LOAD
LOAD
Peak Current Sense Voltage
I
I
I
= GND (Note 6)
= Floating
●
●
●
55
120
190
70
138
208
85
155
225
mV
mV
mV
PRG
PRG
PRG
= V
IN
Default Soft-Start Time
Time for V to Ramp from 0.05V to 0.75V
0.8
ms
FB
dissipation P according to the following formula:
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
D
T = T + (P • θ °C/W)
J
A
D
JA
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 2: The LTC3772BETS8/LTC3772BEDDB are guaranteed to meet
specifications from 0°C to 70°C. Specifications over the –40°C to 85°C
operating temperature range are assured by design, characterization and
correlation with statistical process controls.
Note 5: The LTC3772B is tested in a feedback loop that servos V to the
FB
output of the error amplifier while maintaining I /RUN at the midpoint of
TH
the current limit range.
Note 6: Peak current sense voltage is reduced dependent on duty cycle as
given in Figure 1.
Note 3: T is calculated from the ambient temperature T and power
J
A
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Quiescent Current (No Load)
vs Input Voltage
Quiescent Current (No Load)
vs Temperature
Quiescent Current (Shutdown)
vs Input Voltage
225
220
215
210
205
200
195
250
230
210
190
170
150
25
V
= 5V
IN
20
15
10
5
0
7
8
2
3
4
5
6
9
10
20
TEMPERATURE (°C)
–60 –40 –20
0
40 60 80 100
6
2
3
4
5
7
8
9
10
V
(V)
IN
INPUT VOLTAGE (V)
3772B G01
3772B G02
3772B G03
3772bfa
3
LTC3772B
TYPICAL PERFOR A CE CHARACTERISTICS
U W
Quiescent Current (Shutdown)
vs Temperature
Shutdown Threshold
vs Temperature
Regulated Feedback Voltage
vs Temperature
14
12
800
700
600
500
812
808
804
800
V
IN
= 4.2V
V
= 4.2V
IN
10
8
6
4
2
796
792
788
0
400
–20
0
20 40
100
–60 –40
60 80
–50 –30 –10 10
30
50
70
90
30
TEMPERATURE (°C)
80
90
–50 –30 –10 10
50
TEMPERATURE (°C)
TEMPERATURE (°C)
3772B G04
3772B G05
3772B G06
Regulated Feedback Voltage
vs Input Voltage
Oscillator Frequency
vs Temperature
Oscillator Frequency
vs Input Voltage
600
590
580
570
560
550
540
530
520
510
500
560
555
550
545
0.812
0.808
0.804
0.800
0.796
0.792
0.788
T
A
= 25°C
V
= 4.2V
IN
540
7
8
2
3
4
5
6
9
10
2
3
4
5
6
7
8
9
10
–50
–10 10
30
50
70
90
–30
INPUT VOLTAGE (V)
V
(V)
TEMPERATURE (°C)
IN
3772B G07
3772B G09
3772B G08
I
/RUN Start-Up Current
I
/RUN Start-Up Current
Undervoltage Lockout Thresholds
vs Temperature
TH
TH
vs Temperature
vs Input Voltage
1.5
1.4
1.3
1.2
1.1
1.0
0.9
0.8
0.7
0.6
0.5
2.5
2.4
2.3
2.2
2.1
2.0
1.9
1.8
1.7
1.6
1.5
2.1
1.9
1.7
1.5
I
/RUN = 0V
I /RUN = 0V
TH
TH
RISING
1.3
1.1
FALLING
0.9
0.7
0.5
–60
60 80
6
80
–40 –20
20
TEMPERATURE (°C)
0
40
100
2
4
8
10
–60
20
TEMPERATURE (°C)
60
–40 –20
0
40
100
INPUT VOLTAGE (V)
3772B FG10
3772B G11
3772B G12
3772bfa
4
LTC3772B
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Maximum Current Sense
Threshold vs Temperature
Foldback Frequency
vs Temperature
Soft-Start Time vs Temperature
1100
1000
900
800
700
600
500
230
220
210
200
190
180
170
160
150
300
250
200
150
100
50
V
= 0V
FB
I
= V
IN
PRG
I
= FLOAT
PRG
I
= GND
PRG
0
40 60
TEMPERATURE (°C)
–60 –40 –20
0
20
80 100
20 40
0
TEMPERATURE (°C)
40 60
TEMPERATURE (°C)
–60 –40 –20
60 80 100
–60 –40 –20
0
20
80 100
3772B G14
3772B G15
3772B G13
Efficiency vs Load Current
Efficiency vs Load Current
100
100
V
= 3.3V
IN
V
= 3.3V
OUT
90
80
90
80
70
60
50
40
V
= 4.2V
IN
V
= 2.5V
OUT
V
= 7V
IN
V
= 1.8V
OUT
V
= 5V
IN
70
60
50
40
V
= 2.5V
V
= 5V
OUT
IN
FIGURE 5 CIRCUIT
FIGURE 5 CIRCUIT
1000 10000
LOAD CURRENT (mA)
1
10
100 1000 10000
10
100
LOAD CURRENT (mA)
3772B G16
3772B G17
Start-Up
Load Step
V
OUT
V
OUT
100mV/DIV
(AC)
1V/DIV
I
/RUN
I
TH
L
1V/DIV
2A/DIV
INDUCTOR
CURRENT
2A/DIV
I
LOAD
2A/DIV
3772B G19
3772B G18
V
V
I
= 5V
20µs/DIV
V
= 5V
500µs/DIV
IN
OUT
IN
= 2.5V
V
= 2.5V
OUT
= 100mA TO 1.5A
R
= 1.5Ω
LOAD
LOAD
FIGURE 5 CIRCUIT
FIGURE 5 CIRCUIT
3772bfa
5
LTC3772B
U
U
U
PI FU CTIO S
(DDB/TS8)
NC (Pin 5/Pin 8): No Connection Required.
GND (Pin 1/Pin 4): Ground Pin.
SW (Pin 6/Pin 7): Switch Node Connection to Inductor
and Current Sense Input Pin. Normally, the external
P-channel MOSFET’s drain is connected to this pin.
VFB (Pin 2/Pin 3): Receives the feedback voltage from an
external resistor divider across the output.
ITH/RUN (Pin 3/Pin 2): This pin performs two functions. It
servesastheerroramplifiercompensationpointaswellas
the run control input. Nominal voltage range for this pin is
0.7V to 1.9V. Forcing this pin below 0.6V causes the
device to be shut down. In shutdown, all functions are
disabled and the PGATE pin is held high.
VIN (Pin 7/Pin 6): Supply and Current Sense Input Pin.
This pin must be closely decoupled to GND (Pin 4).
Normally the external P-channel MOSFET’s source is
connected to this pin.
PGATE(Pin8/Pin5):GateDrivefortheExternalP-Channel
MOSFET. This pin swings from 0V to VIN.
IPRG (Pin 4/Pin 1): Current Sense Limit Pin. Three-state
pin selects maximum peak sense voltage threshold. The
pin selects the maximum voltage drop across the external
P-channel MOSFET. Tie to VIN, GND or float to select
208mV, 70mV or 138mV respectively.
Exposed Pad (Pin 9, DDB Only): The Exposed Pad is
ground and must be soldered to the PCB for electrical
connection and optimum thermal performance.
3772bfa
6
LTC3772B
U
U
W
FU CTIO AL DIAGRA
SW
V
IN
SLOPE
COMPENSATION
UV
UNDERVOLTAGE
LOCKOUT
VOLTAGE
REFERENCE
0.8V
–
+
I
PRG
1.2µA
SHUTDOWN
CURRENT
COMPARATOR
COMPARATOR
I
/RUN
TH
–
+
I
+
LIM
S
I
TH
BUFFER
SHDN
–
550kHz
OSCILLATOR
R
RS
LATCH
Q
V
IN
FREQUENCY
FOLDBACK
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
PGATE
0V
OVERVOLTAGE
COMPARATOR
SHORT-CIRCUIT
DETECT
+
+
–
–
ERROR
AMPLIFIER
V
0.88V
0.3V
–
FB
+
+
0.8V
SOFT-START
RAMP
1.2V
GND
3772B FD
3772bfa
7
LTC3772B
U
(Refer to the Functional Diagram)
OPERATIO
cause the external P-channel MOSFET to be turned on
100%; i.e., DC. The output voltage will then be determined
by the input voltage minus the voltage drop across the
sense resistor, the MOSFET and the inductor.
Main Control Loop (Normal Operation)
TheLTC3772Bisaconstantfrequencycurrentmodestep-
down switching regulator controller. During normal op-
eration, the external P-channel MOSFET is turned on each
cycle when the oscillator sets the RS latch and turned off
when the current comparator resets the latch. The peak
inductor current at which the current comparator trips is
controlled by the voltage on the ITH/RUN pin, which is the
outputoftheerroramplifier.Thenegativeinputtotheerror
amplifier is the output feedback voltage VFB, which is
generated by an external resistor divider connected be-
tween VOUT and ground. When the load current increases,
it causes a slight decrease in VFB relative to the 0.8V
reference, which in turn causes the ITH/RUN voltage to
increase until the average inductor current matches the
new load current.
Undervoltage Lockout Protection
To prevent operation of the external P-channel MOSFET
with insufficient gate drive, an undervoltage lockout cir-
cuit is incorporated into the LTC3772B. When the input
supply voltage drops below approximately 2V, the
P-channel MOSFET and all internal circuitry other than the
undervoltage block itself are turned off. Input supply
current in undervoltage is approximately 1µA.
Short-Circuit Protection
If the output is shorted to ground, the frequency of the
oscillator is folded back from 550kHz to approximately
200kHz while maintaining the same minimum on time.
This lower frequency allows the inductor current to safely
discharge, thereby preventing current runaway. After the
short is removed, the oscillator frequency will gradually
increase back to 550kHz as VFB rises through 0.3V on its
way back to 0.8V.
ThemaincontrolloopisshutdownbypullingtheITH/RUN
pin to ground. Releasing the ITH/RUN pin allows an
internal 1µA current source to charge up the external
compensation network. When the ITH/RUN pin voltage
reaches approximately 0.6V, the main control loop is
enabled and the ITH/RUN voltage is pulled up by a clamp
to its zero current level of approximately one diode
voltage drop (0.7V). As the external compensation net-
work continues to charge up, the corresponding peak
inductorcurrentlevelfollows, allowingnormaloperation.
The maximum peak inductor current attainable is set by a
clamp on the ITH/RUN pin at 1.2V above the zero current
level (approximately 1.9V).
Overvoltage Protection
If VFB exceeds its regulation point of 0.8V by more than
10% for any reason, such as an output short-circuit to a
higher voltage, the overvoltage comparator will hold the
external P-channel MOSFET off. This comparator has a
typical hysteresis of 40mV.
Dropout Operation
Peak Current Sense Voltage Selection and Slope
Compensation (IPRG Pins)
When the input supply voltage decreases towards the
output voltage, the rate of change of inductor current
during the on cycle decreases. This reduction means that
at some input-output differential, the external P-channel
MOSFET will remain on for more than one oscillator cycle
(start dropping off-cycles) since the inductor current has
not ramped up to the threshold set by the error amplifier.
Further reduction in input supply voltage will eventually
When a controller is operating below 20% duty cycle, the
maximum sense voltage allowed across the external
P-channel MOSFET is 138mV, 70mV or 208mV for the
three respective states of the IPRG pin.
3772bfa
8
LTC3772B
U
(Refer to the Functional Diagram)
OPERATIO
However, once the controller’s duty cycle exceeds 20%,
slope compensation begins and effectively reduces the
peak sense voltage by an amount given by the curve in
Figure 1.
voltage to rise smoothly from 0V to its final value, while
maintaining control of the inductor current. After the
soft-start is timed out, it is disabled until the part is put in
shutdown again or the input supply is cycled.
Thepeakinductorcurrentisdeterminedbythepeaksense
voltage and the on-resistance of the external P-channel
MOSFET:
Light Load Current Operation
Under very light load current conditions, the ITH/RUN pin
voltage will be very close to the zero current level of 0.85V.
As the load current decreases further, an internal offset at
the current comparator input will assure that the current
comparator remains tripped (even at zero load current)
and the regulator will start to skip cycles, as it must, in
order to maintain regulation. This behavior allows the
regulator to maintain constant frequency down to very
light loads, resulting in low output ripple as well as low
audio noise and reduced RF interference, while providing
high light load efficiency.
∆VSENSE(MAX)
IPEAK
=
RDS(ON)
Soft-Start
The start-up of VOUT is controlled by the LTC3772B inter-
nal soft-start. During soft-start, the error amplifier com-
pares the feedback signal VFB to the internal soft-start
ramp (instead of the 0.8V reference), which rises linearly
from 0V to 0.8V in about 0.6ms. This allows the output
100
90
80
70
60
50
40
30
20
10
0
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
3772B F01
Figure 1. Reduction in Sense Voltage Due to
Slope Compensation vs Duty Cycle
3772bfa
9
LTC3772B
W U U
U
APPLICATIO S I FOR ATIO
ThebasicLTC3772Bapplicationcircuitisshownonthefront
page of this data sheet. The load requirement drives the
selection of external components: the power MOSFET,
inductor and output diode, as well as the input bypass
However, for operation above 20% duty cycle, slope com-
pensation has to be taken into consideration to select the
appropriatevalueofRDS(ON)fortherequiredamountofload
current:
capacitor CIN and output bypass capacitor COUT
.
∆VSENSE(MAX) – SF
5
6
RDS(ON)(MAX)
=
•
Power MOSFET Selection
IOUT(MAX)
AnexternalP-channelpowerMOSFETmustbeselectedfor
use with the LTC3772B. The main selection criteria for the
powerMOSFETarethethresholdvoltageVGS(TH), the“on”
resistanceRDS(ON), reversetransfercapacitanceCRSS and
total gate charge.
whereSFisafactorwhosevalueisobtainedfromthecurve
in Figure 1.
These must be further derated to take into account the
significantvariationinon-resistancewithtemperature.The
following equation is a good guide for determining the
required RDS(ON)MAX at 25°C (manufacturer’s specifica-
tion),allowingsomemarginforvariationsintheLTC3772B
and external component values:
SincetheLTC3772Bisdesignedforoperationdowntolow
inputvoltages,asublogiclevelthresholdMOSFET(RDS(ON)
guaranteed at VGS = 2.5V) is required for applications that
workclosetothisvoltage.WhentheseMOSFETsareused,
makesurethattheinputsupplytotheLTC3772Bislessthan
the absolute maximum VGS rating.
∆VSENSE(MAX) – SF
5
6
RDS(ON)(MAX) = • 0.9 •
IOUT(MAX) • ρT
TheP-channelMOSFET’son-resistanceischosenbasedon
the required load current. The maximum average output
loadcurrentIOUT(MAX) isequaltothepeakinductorcurrent
minus half the peak-to-peak ripple current IRIPPLE. The
LTC3772B’s current comparator monitors the drain-to-
source voltage VDS of the P-channel MOSFET, which is
sensed between the VIN and SW pins. The peak inductor
current is limited by the current threshold, set by the volt-
age on the ITH pin of the current comparator. The voltage
on the ITH pin is internally clamped, which limits the maxi-
mum current sense threshold ∆VSENSE(MAX) to approxi-
mately 138mV when IPRG is floating (70mV when IPRG is
tied low; 208mV when IPRG is tied high).
The ρT is a normalizing term accounting for the tempera-
ture variation in on-resistance, which is typically about
0.4%/°C, as shown in Figure 2. Junction to case tempera-
tureTJC isabout10°Cinmostapplications.Foramaximum
ambienttemperatureof70°C,usingρ80°C≅1.3intheabove
equation is a reasonable choice.
The required minimum RDS(ON) of the MOSFET is also
governed by its allowable power dissipation. For applica-
tionsthatmayoperatetheLTC3772Bindropout–i.e.,100%
2.0
1.5
1.0
0.5
0
TheoutputcurrentthattheLTC3772Bcanprovideisgivenby:
∆VSENSE(MAX)
IRIPPLE
IOUT(MAX)
=
–
RDS(ON)
2
A reasonable starting point is setting ripple current IRIPPLE
to be 40% of IOUT(MAX). Rearranging the above equation
yields:
∆VSENSE(MAX)
IOUT(MAX)
5
RDS(ON)(MAX) = •
6
50
100
–50
150
0
JUNCTION TEMPERATURE (ϒC)
3772B F02
for Duty Cycle < 20%.
Figure 2. R
vs Temperature
DS(ON)
3772bfa
10
LTC3772B
W U U
APPLICATIO S I FOR ATIO
U
duty cycle–at its worst case the required RDS(ON) is given
by:
Inductor Core Selection
Oncetheinductancevalueisdetermined,thetypeofinduc-
tormustbeselected.Actualcorelossisindependentofcore
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
moreturnsofwireandthereforecopperlosseswillincrease.
P
P
RDS(ON)(DC=100%)
=
(IOUT(MAX))2(1+ δP)
where PP is the allowable power dissipation and δP is the
temperature dependency of RDS(ON). (1 + δP) is generally
given for a MOSFET in the form of a normalized RDS(ON) vs
temperature curve, but δP = 0.005/°C can be used as an
approximation for low voltage MOSFETs.
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can
concentrate on copper loss and preventing saturation.
Ferrite core material saturates “hard,” which means that
inductance collapses abruptly when the peak design cur-
rent is exceeded. This results in an abrupt increase in in-
ductorripplecurrentandconsequentoutputvoltageripple.
Do not allow the core to saturate!
In applications where the maximum duty cycle is less than
100%andtheLTC3772Bisincontinuousmode,theRDS(ON)
is governed by:
P
P
RDS(ON)
≅
(DC)IOUT2(1+ δP)
Different core materials and shapes will change the size/
currentandprice/currentrelationshipofaninductor.Toroid
or shielded pot cores in ferrite or permalloy materials are
small and don’t radiate much energy, but generally cost
more than powdered iron core inductors with similar
characteristics. The choice of which style inductor to use
mainlydependsonthepricevssizerequirementsandany
radiated field/EMI requirements. New designs for surface
mount inductors are available from Coiltronics, Coilcraft,
Toko and Sumida.
where DC is the maximum operating duty cycle of the
LTC3772B.
Inductor Value Calculation
The operating frequency and inductor selection are inter-
related in that higher operating frequencies permit the use
ofasmallerinductorforthesameamountofinductorripple
current. However, this is at the expense of efficiency due
to an increase in MOSFET gate charge losses.
Output Diode Selection
The inductance value also has a direct effect on ripple
current.Innormaloperation,theripplecurrent,IRIPPLE,de-
creaseswithhigherinductanceorfrequencyandincreases
with higher VIN or as VOUT approaches 1/2 VIN. The
inductor’s peak-to-peak ripple current is given by:
Thecatchdiodecarriesloadcurrentduringtheoff-time.The
average diode current is therefore dependent on the
P-channelswitchdutycycle.Athighinputvoltagesthediode
conducts most of the time. As VIN approaches VOUT the
diode conducts only a small fraction of the time. The most
stressfulconditionforthediodeiswhentheoutputisshort-
circuited.Underthisconditionthediodemustsafelyhandle
IPEAK at close to 100% duty cycle. Therefore, it is impor-
tant to adequately specify the diode peak current and av-
erage power dissipation so as not to exceed the diode
ratings.
V − VOUT ⎛ VOUT + VD ⎞
IN
IRIPPLE
=
⎜
⎟
f(L) ⎝ V + VD ⎠
IN
where f is the operating frequency. VD is the forward volt-
age drop of the catch diode, 0.5V typical. Accepting larger
values of IRIPPLE allows the use of low inductances, but re-
sultsinhigheroutputvoltagerippleandgreatercorelosses.
A reasonable starting point for setting ripple current is
IRIPPLE =0.4(IOUT(MAX)).Remember,themaximumIRIPPLE
occurs at the maximum input voltage.
3772bfa
11
LTC3772B
W U U
U
APPLICATIO S I FOR ATIO
Under normal load conditions, the average current con-
ducted by the diode is:
The output filtering capacitor C smooths out current flow
from the inductor to the load, help maintain a steady out-
putvoltageduringtransientloadchangesandreduceoutput
voltage ripple. The capacitors must be selected with suf-
ficiently low ESR to minimize voltage ripple and load step
transients and sufficiently bulk capacitance to ensure the
control loop stability.
⎛ V − VOUT
⎞
⎟
IN
ID
=
I
OUT
⎜
⎝ V + VD ⎠
IN
The allowable forward voltage drop in the diode is calcu-
lated from the maximum short-circuit current as:
The output ripple, ∆VOUT, is determined by:
PD
IPEAK
V ≅
⎛
⎝
1
⎞
F
∆VOUT ≤ ∆I ESR+
⎜
⎟
⎠
L
8fCOUT
where PD is the allowable power dissipation and will be
determined by efficiency and/or thermal requirements.
Theoutputrippleishighestatmaximuminputvoltagesince
∆ILincreaseswithinputvoltage.Multiplecapacitorsplaced
in parallel may be needed to meet the ESR and RMS cur-
renthandlingrequirements.Drytantalum,specialpolymer,
aluminumelectrolyticandceramiccapacitorsareallavail-
able in surface mount packages. Special polymer capaci-
torsofferverylowESRbuthavelowercapacitancedensity
than other types. Tantalum capacitors have the highest
capacitance density but it is important to only use types
thathavebeensurgetestedforuseinswitchingpowersup-
plies. Aluminum electrolytic capacitors have significantly
higher ESR but can be used in cost-sensitive applications
provided that consideration is given to ripple current rat-
ings and long term reliability. Ceramic capacitors have
excellent low ESR characteristics but can have a high
voltage coefficient and audible piezoelectric effects. The
highQofceramiccapacitorswithtraceinductancecanalso
lead to significant ringing.
A fast switching diode must also be used to optimize effi-
ciency. Schottky diodes are a good choice for low forward
drop and fast switching times. Remember to keep lead
length short and observe proper grounding to avoid ring-
ing and increased dissipation.
An additional consideration in applications where low no-
load quiescent current is critical is the reverse leakage
currentofthediodeattheregulatedoutputvoltage. Aleak-
age greater than several microamperes can represent a
significant percentage of the total input current.
CIN and COUT Selection
The input capacitance, CIN, is needed to filter the trapezoi-
dalcurrentatthesourceofthetopMOSFET.Topreventlarge
ripple voltage, a low ESR input capacitor sized for the
maximum RMS current should be used. RMS current is
given by:
Using Ceramic Input and Output Capacitors
Higher values, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them ideal
for switching regulator applications. However, care must
be taken when these capacitors are used at the input and
output. When a ceramic capacitor is used at the input and
thepowerissuppliedbyawalladapterthroughlongwires,
aloadstepattheoutputcaninduceringingattheinput,VIN.
At best, this ringing can couple to the output and be mis-
taken as loop instability. At worst, a sudden inrush of cur-
rent through the long wires can potentially cause a voltage
spike at VIN large enough to damage the part.
VOUT
V
IN
V
IN
VOUT
IRMS = IOUT(MAX)
–1
This formula has a maximum at VIN = 2VOUT, where IRMS
= IOUT/2. This simple worst-case condition is commonly
usedfordesignbecauseevensignificantdeviationsdonot
offer much relief. Note that ripple current ratings from
capacitormanufacturersareoftenbasedononly2000hours
oflifewhichmakesitadvisabletofurtherderatethecapaci-
tor,orchooseacapacitorratedatahighertemperaturethan
required.Severalcapacitorsmayalsobeparalleledtomeet
size or height requirements in the design.
3772bfa
12
LTC3772B
W U U
APPLICATIO S I FOR ATIO
U
1. The VIN current is the DC supply current, given in the
electrical characteristics, that excludes MOSFET driver
and control currents. VIN current results in a small loss
which increases with VIN.
For ceramic capacitors, use X7R or X5R types: do not use
Y5V. Manufacturers include AVX, Kemet, Murata, Taiyo
Yuden and TDK.
Setting Output Voltage
2. MOSFETgatechargecurrentresultsfromswitchingthe
gate capacitance of the power MOSFET. Each time a
MOSFET gate is switched from low to high to low again,
a packet of charge dQ moves from VIN to ground. The
resulting dQ/dt is a current out of VIN that is typically
much larger than the DC supply current. In continuous
mode, IGATECHG = (f)(dQ).
3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L but is
“chopped” between the P-channel MOSFET (in series
withRSENSE)andtheoutputdiode.TheMOSFETRDS(ON)
plusRSENSE multipliedbydutycyclecanbesummedwith
the resistances of L and RSENSE to obtain I2R losses.
The LTC3772B output voltages are each set by an external
feedback resistor divider carefully placed across the out-
put as shown in Figure 3. The regulated output voltage is
determined by:
⎛
⎝
RB ⎞
RA ⎠
VOUT = 0.8V • 1+
⎜
⎟
Toimprovethefrequencyresponse,afeed-forwardcapaci-
tor, CFF, may be used. Great care should be taken to route
the VFB line away from noise sources, such as the inductor
or the SW line.
V
OUT
LTC3772B
R
C
FF
B
A
4. Theoutputdiodeisamajorsourceofpowerlossathigh
currentsandgetsworseathighinputvoltages.Thediode
loss is calculated by multiplying the forward voltage
timesthediodedutycyclemultipliedbytheloadcurrent.
Forexample,assumingadutycycleof50%withaSchot-
tkydiodeforwardvoltagedropof0.4V,thelossincreases
from0.5%to8%astheloadcurrentincreasesfrom0.5A
to 2A.
V
FB
R
3772B F03
Figure 3. Setting Output Voltage
Efficiency Considerations
The efficiency of a switching regulator is equal to the out-
putpowerdividedbytheinputpowertimes100%.Itisoften
useful to analyze individual losses to determine what is
limitingtheefficiencyandwhichchangewouldproducethe
most improvement. Efficiency can be expressed as:
5. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input volt-
ages. Transition losses can be estimated from:
Transition Loss = 2(VIN)2IO(MAX) RSS
(f)
C
OtherlossesincludingCIN andCOUT ESRdissipativelosses
and inductor core losses, generally account for less than
2% total additional loss.
Efficiency = 100% – (η1 + η2 + η3 + ...)
where η1, η2, etc. are the individual losses as a percent-
age of input power.
Foldback Current Limiting
Although all dissipative elements in the circuit produce
losses, five main sources usually account for most of the
lossesinLTC3772Bcircuits:1)LTC3772BDCbiascurrent,
2) MOSFET gate charge current, 3) I2R losses, 4) voltage
drop of the output diode and 5) external MOSFET transi-
tion losses.
AsdescribedintheOutputDiodeSelection,theworst-case
dissipation occurs with a short-circuited output when the
diodeconductsthecurrentlimitvaluealmostcontinuously.
3772bfa
13
LTC3772B
W U U
U
APPLICATIO S I FOR ATIO
Topreventexcessiveheatinginthediode,foldbackcurrent
limiting can be added to reduce the current in proportion
to the severity of the fault.
and values determine the loop feedback factor gain and
phase. Anoutputcurrentpulseof20%to100%offullload
current having a rise time of 1µs to 10µs will produce
outputvoltageandITH pinwaveformsthatwillgiveasense
of the overall loop stability. The gain of the loop will be
increased by increasing RC and the bandwidth of the loop
will be increased by decreasing CC. The output voltage
settling behavior is related to the stability of the closed-
loopsystemandwilldemonstratetheactualoverallsupply
performance. For a detailed explanation of optimizing the
compensation components, including a review of control
loop theory, refer to Application Note 76.
Foldbackcurrentlimitingisimplementedbyaddingdiodes
DFB1 and DFB2 between the output and the ITH/RUN pin as
shown in Figure 4. In a hard short (VOUT = 0V), the current
will be reduced to approximately 50% of the maximum
output current.
V
LTC3772B
OUT
R
R
B
A
I
/RUN V
FB
TH
D
D
FB1
FB2
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
dischargedbypasscapacitorsareeffectivelyputinparallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
the load rise time is limited to approximately (25)(CLOAD).
Thus a 10µF capacitor would require a 250µs rise time,
limiting the charging current to about 200mA.
3772B F04
Figure 4. Foldback Current Limiting
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to (∆ILOAD)(ESR), where ESR is the effective series
resistance of COUT. ∆ILOAD also begins to charge or dis-
chargeCOUT, whichgeneratesafeedbackerrorsignal. The
regulator loop then returns VOUT to its steady-state value.
Duringthisrecoverytime,VOUT canbemonitoredforover-
shoot or ringing. OPTI-LOOP compensation allows the
transient response to be optimized over a wide range of
output capacitance and ESR values.
Minimum On-Time Considerations
Minimum on-time, tON(MIN), is the smallest amount of
time that the LTC3772B is capable of turning the top
MOSFET on and then off. It is determined by internal
timing delays and the gate charge required to turn on the
top MOSFET. The minimum on-time for the LTC3772B is
about 250ns. Low duty cycle and high frequency applica-
tions may approach this minimum on-time limit and care
should be taken to ensure that:
The ITH series RC-CC filter (see Functional Diagram) sets
the dominant pole-zero loop compensation. The ITH exter-
nal components shown in the Figure 5 circuit will provide
an adequate starting point for most applications. The
values can be modified slightly (from 0.2 to 5 times their
suggestedvalues)tooptimizetransientresponseoncethe
final PC layout is done and the particular output capacitor
type and value have been determined. The output capaci-
tors need to be decided upon because the various types
VOUT
tON(MIN)
<
f • V
IN
Ifthedutycyclefallsbelowwhatcanbeaccommodatedby
the minimum on-time, the LTC3772B will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple current and ripple voltage will increase.
3772bfa
14
LTC3772B
U
TYPICAL APPLICATIO S
550kHz Micropower, 1A, 2-Cell Li-Ion to 3.3V
OUT
Step-Down DC/DC Converter
100pF
15k
V
IN
I
/RUN
V
TH
IN
PGATE
SW
5V TO 8.4V
C
LTC3772B
IN
22µF
GND
Si2341DS
I
PRG
L1 4.7µH
56.2k
V
3.3V
1A
OUT
V
FB
C
OUT
UPS120
47µF
22pF 174k
L1: SUMIDA CR43-4R7
3772B TA02a
C
C
: MURATA GRM32ER61C226KA65B
IN
OUT
: MURATA GRM32ER60J476ME20B
Efficiency vs Load Current
100
V
IN
= 5.5V
90
80
V
= 7.2V
IN
V
= 8.4V
IN
70
60
50
40
1
10
100
1000
10000
LOAD CURRENT (mA)
3772B TA02b
Start-Up
Load Step
V
V
OUT
OUT
100mV/DIV
(AC)
2V/DIV
I
/RUN
I
TH
L
1V/DIV
500mA/DIV
I
L
I
LOAD
500mA/DIV
1A/DIV
3772B TA02c
3772B TA02d
V
V
= 5.5V
400µs/DIV
V
= 5.5V
20µs/DIV
IN
OUT
IN
= 3.3V
= 3Ω
V
= 3.3V
OUT
LOAD
R
I
= 40mA TO 500mA
LOAD
3772bfa
15
LTC3772B
U
TYPICAL APPLICATIO S
550kHz Micropower 3A Step-Down DC/DC Converter
220pF
34.8k
V
IN
I
/RUN
V
TH
IN
PGATE
SW
2.75V TO 9.8V
C
LTC3772B
IN
22µF
GND
NTMS5PO2R2
I
PRG
L1 2.2µH
82.5k
V
2.5V
3A
OUT
V
FB
C
OUT
B320A
100µF
×2
22pF 174k
L1: VISHAY IHLP-2525CZ-01
: GRM32ER61A220KA65B
: TAIYO YUDEN LDK375BJ107MM
3772B TA03a
C
C
IN
OUT
Efficiency vs Load Current
100
90
80
V
IN
= 3.3V
V
IN
= 5V
70
60
50
40
1
10
100
1000
10000
LOAD CURRENT (mA)
3772B TA03b
Start-Up
Load Step
V
OUT
100mV/DIV
(AC)
V
OUT
2V/DIV
I
L
I
/RUN
TH
2A/DIV
1V/DIV
I
I
LOAD
L
2A/DIV
2A/DIV
3772B TA03c
3772B TA03d
C
R
R
= 220pF
= 34.8k
= 1.5Ω
400µs/DIV
V
V
I
= 5V
20µs/DIV
ITH
ITH
LOAD
IN
OUT
= 2.5V
= 15OmA TO 2A
LOAD
3772bfa
16
LTC3772B
U
TYPICAL APPLICATIO S
550kHz Micropower 5V to 1.8V
at 8A DC/DC Converter
IN
OUT
470pF
15k
V
IN
5V
I
/RUN
V
TH
IN
PGATE
SW
C
LTC3772B
IN
22µF
Si9433DBY
GND
×2
V
I
PRG
IN
L1 1µH
140k
V
1.8V
8A
OUT
V
FB
C
OUT
CSHD10-45L
100µF
×2
22pF 174k
3772B TA04a
L1: TOKO FDV0630-1R0
C
C
: MURATA GRM32ER61C226K
OUT
IN
: MURATA GRM32ER60J107K
Efficiency vs Load Current
100
90
80
70
60
50
40
100
1000
10000
LOAD CURRENT (mA)
3772B TA04b
Start-Up
Load Step
V
OUT
V
OUT
200mV/DIV
1V/DIV
AC COUPLED
I
/RUN
TH
I
L
1V/DIV
10A/DIV
I
L
I
LOAD
10A/DIV
5A/DIV
3772B TA04d
3772B TA04c
V
V
I
= 5V
20µs/DIV
V
V
= 5V
500µs/DIV
IN
OUT
IN
OUT
= 1.8V
= 1.8V
= 800mA TO 8A
R
= 0.25Ω
LOAD
LOAD
3772bfa
17
LTC3772B
U
PACKAGE DESCRIPTIO
DDB Package
8-Lead Plastic DFN (3mm × 2mm)
(Reference LTC DWG # 05-08-1702)
0.61 ±0.05
(2 SIDES)
R = 0.115
0.40 ± 0.10
3.00 ±0.10
(2 SIDES)
TYP
5
R = 0.05
TYP
8
0.70 ±0.05
2.55 ±0.05
1.15 ±0.05
2.00 ±0.10
PIN 1 BAR
TOP MARK
PIN 1
(2 SIDES)
R = 0.20 OR
(SEE NOTE 6)
0.25 × 45°
PACKAGE
OUTLINE
0.56 ± 0.05
(2 SIDES)
CHAMFER
4
1
(DDB8) DFN 0905 REV B
0.25 ± 0.05
0.25 ± 0.05
0.75 ±0.05
0.200 REF
0.50 BSC
2.20 ±0.05
(2 SIDES)
0.50 BSC
2.15 ±0.05
(2 SIDES)
0 – 0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING CONFORMS TO VERSION (WECD-1) IN JEDEC PACKAGE OUTLINE M0-229
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
3772bfa
18
LTC3772B
U
PACKAGE DESCRIPTIO
TS8 Package
8-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1637)
2.90 BSC
(NOTE 4)
0.52
MAX
0.65
REF
1.22 REF
1.50 – 1.75
(NOTE 4)
2.80 BSC
1.4 MIN
3.85 MAX 2.62 REF
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.22 – 0.36
8 PLCS (NOTE 3)
0.65 BSC
0.80 – 0.90
0.20 BSC
DATUM ‘A’
0.01 – 0.10
1.00 MAX
0.30 – 0.50 REF
1.95 BSC
0.09 – 0.20
(NOTE 3)
TS8 TSOT-23 0802
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
3772bfa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
19
LTC3772B
U
TYPICAL APPLICATIO
220pF
15k
V
IN
I
/RUN
V
TH
IN
PGATE
SW
3V TO 8V
C
IN
LTC3772B
22µF
GND
FDC638P
L1 3.3µH
I
PRG
82.5k
V
2.5V
2A
OUT
V
FB
C
OUT
47µF
B220A
22pF
174k
3772B F05
L1: TOKO D53LC #A915AY-3R3M
C
C
: TAIYO YUDEN LMK316BJ226ML
OUT
IN
: TAIYO YUDEN JMK325BJ476MM
Figure 5. 550kHz Micropower Step-Down DC/DC Converter
RELATED PARTS
PART NUMBER
LTC1624
DESCRIPTION
High Efficiency SO-8 N-Channel Switching Regulator Controller
No R
TM Synchronous Step-Down Regulator
COMMENTS
N-Channel Drive, 3.5V ≤ V ≤ 36V
IN
LTC1625
97% Efficiency, No Sense Resistor
SENSE
LTC1772/LTC1772B 550kHz ThinSOT Step-Down DC/DC Controllers
LTC1778/LTC1778-1 No R Current Mode Synchronous Step-Down Controllers
2.5V ≤ V ≤ 9.8V, V
≥ 0.8V, I
≤ 6A
IN
OUT
OUT
4V ≤ V ≤ 36V, 0.8V ≤ V
≤ (0.9)(V ), I
Up to 20A
SENSE
IN
OUT
IN OUT
LTC1872/LTC1872B 550kHz ThinSOT Step-Up DC/DC Controllers
2.5V ≤ V ≤ 9.8V; 90% Efficiency
IN
LTC3411/LTC3412
1.25A/2.5A, 4MHz Monolithic Synchronous Step-Down Converter
95% Efficiency, 2.5V ≤ V ≤ 5.5V, V
≥ 0.8V,
≥ 0.8V,
≥ 0.8V,
IN
OUT
OUT
OUT
TSSOP16 Exposed Pad Package
LTC3414
4A, 4MHz Monolithic Synchronous Step-Down Converter
8A, 4MHz Monolithic Synchronous Step-Down Converter
95% Efficiency, 2.5V ≤ V ≤ 5.5V, V
IN
TSSOP20 Exposed Pad Package
LTC3418
95% Efficiency, 2.5V ≤ V ≤ 5.5V, V
IN
TSSOP20 Exposed Pad Package
LTC3440
600mA (I ), 2MHz Synchronous Buck-Boost DC/DC Converter
2.5V ≤ V ≤ 5.5V, Single Inductor
IN
OUT
LTC3736/LTC3736-2 Dual, 2-Phase, No R
Synchronous Controller
V : 2.75V to 9.8V, I
4mm × 4mm QFN Package
Up to 5A,
SENSE
IN
OUT
with Output Tracking
LTC3736-1
LTC3737
LTC3772
LTC3776
Dual, 2-Phase, No R
with Spread Spectrum
Synchronous Controller
V : 2.75V to 9.8V, Spread Spectrum Operation, Output
Voltage Tracking, 4mm × 4mm QFN Package
SENSE
IN
Dual, 2-Phase, No R
Controller with Output Tracking
V : 2.75V to 9.8V, I
Up to 5A,
SENSE
IN
OUT
4mm × 4mm QFN Package
Micropower No R
Constant Frequency Controller
V : 2.75V to 9.8V, I Up to 5A, ThinSOT,
SENSE
IN
OUT
3mm × 2mm DFN Package
Dual, 2-Phase, No R
DDR/QDR Memory Termination
Synchronous Controller for
Provides V and V with one IC, 2.75V ≤ V ≤ 9.8V,
SENSE
DDQ
TT
IN
Adjustable Constant Frequency with PLL Up to 850kHz,
Spread Spectrum Operation, 4mm × 4mm QFN and
16-Lead SSOP Packages
LTC3808
No R
Output Tracking
, Low EMI, Synchronous Step-Down Controller with
2.75V ≤ V ≤ 9.8V, Spread Spectrum Operation,
SENSE
IN
3mm × 4mm DFN and 16-Lead SSOP Packages
LTC3809/LTC3809-1 No R , Synchronous Step-Down Controller
2.75V ≤ V ≤ 9.8V, 3mm × 4mm DFN and 10-Lead MSOP
SENSE
IN
Packages
3772bfa
LT 0606 REV A • PRINTED IN THE USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
20
●
●
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2005
LTC3772BETS8#TRPBF 相关器件
| 型号 | 制造商 | 描述 | 价格 | 文档 |
| LTC3772EDDB | Linear | Micropower No RSENSE Constant Frequency Step-Down DC/DC Controller | 获取价格 |
|
| LTC3772EDDB#TRM | Linear | LTC3772 - Micropower No RSENSE Constant Frequency Step-Down DC/DC Controller; Package: DFN; Pins: 8; Temperature Range: -40&deg;C to 85&deg;C | 获取价格 |
|
| LTC3772EDDB#TRPBF | Linear | LTC3772 - Micropower No RSENSE Constant Frequency Step-Down DC/DC Controller; Package: DFN; Pins: 8; Temperature Range: -40&deg;C to 85&deg;C | 获取价格 |
|
| LTC3772ETS8 | Linear | Micropower No RSENSE Constant Frequency Step-Down DC/DC Controller | 获取价格 |
|
| LTC3772ETS8#PBF | Linear | 暂无描述 | 获取价格 |
|
| LTC3772ETS8#TR | Linear | LTC3772 - Micropower No RSENSE Constant Frequency Step-Down DC/DC Controller; Package: SOT; Pins: 8; Temperature Range: -40&deg;C to 85&deg;C | 获取价格 |
|
| LTC3772ETS8#TRM | Linear | LTC3772 - Micropower No RSENSE Constant Frequency Step-Down DC/DC Controller; Package: SOT; Pins: 8; Temperature Range: -40&deg;C to 85&deg;C | 获取价格 |
|
| LTC3772ETS8#TRMPBF | Linear | LTC3772 - Micropower No RSENSE Constant Frequency Step-Down DC/DC Controller; Package: SOT; Pins: 8; Temperature Range: -40&deg;C to 85&deg;C | 获取价格 |
|
| LTC3772ETS8#TRPBF | Linear | LTC3772 - Micropower No RSENSE Constant Frequency Step-Down DC/DC Controller; Package: SOT; Pins: 8; Temperature Range: -40&deg;C to 85&deg;C | 获取价格 |
|
| LTC3773 | Linear | Triple Output Synchronous 3-Phase DC/DC Controller with Up/Down Tracking | 获取价格 |
|
LTC3772BETS8#TRPBF 相关文章
HARTING(浩亭)圆形连接器产品选型手册
- 2024-10-31
- 6
HYCON(宏康科技)产品选型手册
- 2024-10-31
- 6
GREEGOO整流二极管和晶闸管产品选型手册
- 2024-10-31
- 7
西门子豪掷106亿美元，战略收购工程软件巨头Altair
- 2024-10-31
- 8
