LTC3801BES6_技术文档

LTC3801/LTC3801B

Micropower

Constant Frequency Step-Down

DC/DC Controllers in ThinSOT

U

FEATURES

DESCRIPTIO

The LTC^®3801/LTC3801B are constant frequency cur-

rent mode step-down DC/DC controllers in a low profile

■

High Efficiency: Up to 94%

■

Very Low No-Load Quiescent Current:

(1mm max) 6-lead SOT-23 (ThinSOT^TM

) package. The

Only 16µA (LTC3801)

■

High Output Currents Easily Achieved

parts provide excellent AC and DC load and line regula-

tion with ±1.5% output voltage accuracy. The LTC3801

consumes only 195µA of quiescent current in normal

operation, dropping to 16µA under no-load conditions.

■

Internal Soft-Start

■

Wide V_INRange: 2.4V to 9.8V

■

Low Dropout: 100% Duty Cycle

■

Constant Frequency 550kHz Operation

TheLTC3801/LTC3801Bincorporateanundervoltagelock-

out feature that shuts down the device when the input

voltage falls below 2.2V. The LTC3801 automatically

switches into Burst Mode operation at light loads which

enhancesefficiencyatlowoutputcurrent.IntheLTC3801B,

BurstModeoperationisdisabledforloweroutputrippleat

light loads.

Burst Mode^®Operation for High Efficiency

■

at Light Loads (LTC3801)

■

Burst Mode Operation Disabled for Lower Output

Ripple at Light Loads (LTC3801B)

■

Output Voltage as Low as 0.8V

■

±1.5% Voltage Reference Accuracy

■

Current Mode Operation for Excellent Line and Load

To further maximize the life of a battery source, the

external P-channel MOSFET is turned on continuously in

dropout (100% duty cycle). High switching frequency of

550kHz allows the use of a small inductor.

, LTC and LT are registered trademarks of Linear Technology Corporation.

Burst Mode is a registered trademark of Linear Technology Corporation.

ThinSOT is a trademark of Linear Technology Corporation.

Transient Response

■

Only 6µA Supply Current in Shutdown (LTC3801)

Low Profile (1mm) SOT-23 Package

U

APPLICATIO S

■

1- or 2-Cell Li-Ion Battery-Powered Applications

■

Wireless Devices

■

Portable Computers

Distributed Power Systems

■

U

LTC3801 Efficiency vs Load Current*

TYPICAL APPLICATIO

100

V

= 2.5V

OUT

V

= 3.3V

95

90

85

IN

V

= 4.2V

IN

550kHz Micropower Step-Down DC/DC Controller

220pF

V

= 6.6V

IN

V

IN

10k

80

75

2.7V TO 9.8V

I

/RUN

LTC3801/

V

TH

IN

–

10µF

0.025Ω

V

= 8.4V

V

= 9.8V

IN

LTC3801B

IN

70

65

60

55

50

GND

SENSE

V

PGATE

FB

402k

4.7µH

V

OUT

866k

2.5V

2A

+

0.1

1

10

100

1000 10000

47µF

LOAD CURRENT (mA)

3801 TA02

3801 TA01

*SEE NO-LOAD I_Qvs INPUT VOLTAGE ON THE LAST PAGE OF THIS DATA SHEET

3801f

1

LTC3801/LTC3801B

W W U W

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ABSOLUTE MAXIMUM RATINGS

PACKAGE/ORDER INFORMATION

(Note 1)

ORDER PART

NUMBER

Input Supply Voltage (V_IN)........................ –0.3V to 10V

SENSE^–, PGATE Voltages ............ –0.3V to (V_IN+ 0.3V)

V_FB, I_TH/RUN Voltages ............................. –0.3V to 2.4V

PGATE Peak Output Current (<10µs) ........................ 1A

Operating Temperature Range (Note 2) .. – 40°C to 85°C

Junction Temperature (Note 3)............................ 150°C

Storage Temperature Range ................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec).................. 300°C

TOP VIEW

I

/RUN 1

GND 2

6 PGATE

5 V

TH

LTC3801ES6

LTC3801BES6

IN

4 SENSE

–

V

3

FB

S6 PART MARKING

S6 PACKAGE

6-LEAD PLASTIC TSOT-23

T_JMAX= 150°C, θ_JA= 230°C/W

LTACR

LTAHN

Consult LTC Marketing for parts specified with wider operating temperature ranges.

ELECTRICAL CHARACTERISTICS

The ● indicates specifications which apply over the full operating

temperature range, otherwise specifications are at T_A= 25°C. V_IN= 4.2V unless otherwise noted. (Note 2)

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

Input Voltage Range

●

2.4

9.8

V

Input DC Supply Current

Normal Operation

SLEEP Mode

Typicals at V = 4.2V (Note 4)

IN

2.4V ≤ V ≤ 9.8V, V /RUN = 1.3V

195

16

6

300

30

15

17

2

µA

IN

ITH

2.4V ≤ V ≤ 9.8V (LTC3801 Only)

Shutdown

2.4V ≤ V ≤ 9.8V, V /RUN = 0V (LTC3801)

IN ITH

2.4V ≤ V ≤ 9.8V, V /RUN = 0V (LTC3801B)

8

IN

ITH

UVLO

V

< UVLO Threshold

1

IN

Undervoltage Lockout Threshold

V

Rising

Falling

●

1.8

1.7

2.3

2.2

V

IN

Start-Up Current Source

V

/RUN = 0V (LTC3801)

ITH

/RUN = 0V (LTC3801B)

ITH

0.5

1.0

1

2

1.5

3.0

µA

Shutdown Threshold (at I /RUN)

Regulated Feedback Voltage

V

/RUN Rising

●

0.3

0.788

0.780

0.6

0.800

0.95

0.812

V

TH

ITH

0°C ≤ T ≤ 85°C (Note 5)

A

–40°C ≤ T ≤ 85°C (Note 5)

A

Feedback Voltage Line Regulation

Feedback Voltage Load Regulation

2.4V ≤ V ≤ 9.8V (Note 5)

0.05

2

mV/V

mV/µA

IN

I

/RUN Sinking 5µA (Note 5)

TH

/RUN Sourcing 5µA (Note 5)

TH

V

Input Current

(Note 5)

Measured at V

2

10

0.910

nA

V

mV

FB

Overvoltage Protect Threshold

Overvoltage Protect Hysteresis

Oscillator Frequency

Normal Operation

Output Short Circuit

0.850

500

0.880

40

FB

V

= 0.8V

= 0V

550

210

650

kHz

FB

Gate Drive Rise Time

Gate Drive Fall Time

C

= 3000pF

40

ns

LOAD

Peak Current Sense Voltage

Duty Cycle < 40% (Note 6)

LTC3801

●

109

95

117

104

125

113

mV

LTC3801B

Peak Current Sense Voltage in Burst Mode Operation

Default Soft-Start Time

LTC3801 Only

26

0.6

mV

ms

T = T + (P • θ °C/W)

Note 4: Dynamic supply current is higher due to the gate charge being

delivered at the switching frequency.

Note 1: Absolute Maximum Ratings are those values beyond which the life

J

A

D

JA

of a device may be impaired.

Note 2: The LTC3801ES6/LTC3801BES6 are guaranteed to meet specifica-

tions from 0°C to 70°C. Specifications over the –40°C to 85°C operating

temperature range are assured by design, characterization and correlation

with statistical process controls.

Note 5: The LTC3801/LTC3801B are tested in a feedback loop that servos

V

to the output of the error amplifier while maintaining I /RUN at the

FB

TH

midpoint of the current limit range.

Note 3: T is calculated from the ambient temperature T and power

J

A

Note 6: Peak current sense voltage is reduced dependent on duty cycle as

dissipation P according to the following formula:

D

given in Figure 1.

3801f

2

LTC3801/LTC3801B

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TYPICAL PERFOR A CE CHARACTERISTICS

Input DC Supply Current (Normal)

vs Input Voltage

Input DC Supply Current (SLEEP)

vs Input Voltage (LTC3801 Only)

Input DC Supply Current

(Shutdown) vs Input Voltage

225

20

15

T

= 25°C

ITH

T

= 25°C

A

T

= 25°C

ITH

A

V

/RUN = 1.3V

V

/RUN = 0V

215

205

18

16

12

9

LTC3801B

LTC3801

6

195

185

175

14

12

10

3

0

6

2

3

4

5

7

8

9

10

2

3

4

5

7

8

9

10

6

2

3

4

5

7

8

9

10

V

(V)

V

IN

(V)

V

(V)

IN

3801 G01

3801 G02

3801 G03

Undervoltage Lockout Threshold

vs Temperature

Shutdown Threshold

vs Temperature

Regulated Feedback Voltage

vs Temperature

2.2

2.0

1.8

1.6

1.4

1.2

800

700

600

500

812

808

804

800

V

IN

= 4.2V

V

IN

= 4.2V

V

IN

RISING

V

IN

FALLING

796

792

788

400

–50 –30 –10 10

30

50

70

90

–50 –30 –10 10

30

50

70

90

30

TEMPERATURE (°C)

80

90

–50 –30 –10 10

50

TEMPERATURE (°C)

3801 G04

3801 G05

3801 G06

Regulated Feedback Voltage

vs Input Voltage

Oscillator Frequency

vs Temperature

Oscillator Frequency

vs Input Voltage

600

590

580

570

560

550

540

530

520

510

500

560

555

550

545

812

808

804

800

796

792

788

T

= 25°C

T

= 25°C

A

V

= 4.2V

A

IN

540

7

8

2

3

4

5

6

9

10

2

3

4

5

6

7

8

9

10

–50

–10 10

30

50

70

90

–30

V

IN

(V)

V

(V)

TEMPERATURE (°C)

IN

3801 G07

3801 G09

3801 G08

3801f

3

LTC3801/LTC3801B

U

PI FU CTIO S

SENSE^–(Pin 4): Current Sense Pin. An external sense

I_TH/RUN (Pin 1): This pin performs two functions. It

servesastheerroramplifiercompensationpointaswellas

the run control input. Nominal voltage range for this pin is

0.7V to 1.9V. Forcing this pin below 0.6V causes the

device to be shut down. In shutdown, all functions are

disabled and the PGATE pin is held high.

resistor is connected between this pin and V_IN(Pin 5).

V_IN(Pin 5): Supply Pin. This pin must be closely de-

coupled to GND (Pin 2).

PGATE (Pin 6): Gate Drive for the External P-Channel

MOSFET. This pin swings from 0V to V_IN.

GND (Pin 2): Ground Pin.

V

_FB(Pin 3): Receives the feedback voltage from an exter-

nal resistor divider across the output.

U

W

FU CTIO AL DIAGRA

4

–

SENSE

15mV (LTC3801B)

V

IN

5

BURST

SLOPE

COMPENSATION

UV

UNDERVOLTAGE

LOCKOUT

VOLTAGE

REFERENCE

DEFEAT

BURST

CLAMP

0.8V

–

(LTC3801B)

+

CURRENT

COMPARATOR

1µA (LTC3801)

2µA (LTC3801B)

SHUTDOWN

COMPARATOR

I

/RUN

–

+

0.3V

TH

1

I

+

LIM

I

TH

SHDN

BUFFER

–

550kHz

OSCILLATOR

R

S

RS

LATCH

Q

V

IN

FREQUENCY

FOLDBACK

SLEEP

SWITCHING

LOGIC AND

BLANKING

CIRCUIT

COMPARATOR

PGATE

–

+

6

SLEEP

BURST

DEFEAT

(LTC3801B)

0V

SOFT-START

CLAMP

OVERVOLTAGE

COMPARATOR

SHORT-CIRCUIT

DETECT

0.15V

+

–

ERROR

AMPLIFIER

V

FB

0.225V

0.88V

0.3V

–

+

3

0.8V

1.2V

GND

2

3801 FD

3801f

4

LTC3801/LTC3801B

U

(Refer to the Functional Diagram)

OPERATIO

Main Control Loop (Normal Operation)

and the load will eventually cause the error amplifier out-

puttostartdriftinghigher. Whentheerroramplifieroutput

rises to 0.225V above its zero current level (approximately

0.925V), the sleep comparator will untrip and normal op-

eration will resume. The next oscillator cycle will turn the

external MOSFET on and the switching cycle will repeat.

The LTC3801/LTC3801B are constant frequency current

mode step-down switching regulator controllers. During

normaloperation,anexternalP-channelMOSFETisturned

on each cycle when the oscillator sets the RS latch and

turned off when the current comparator resets the latch.

Thepeakinductorcurrentatwhichthecurrentcomparator

tripsiscontrolledbythevoltageontheI_TH/RUNpin, which

is the output of the error amplifier. The negative input to

the error amplifier is the output feedback voltage V_FB

which is generated by an external resistor divider con-

nected between V_OUTand ground. When the load current

increases, it causes a slight decrease in V_FBrelative to the

0.8V reference, which in turn causes the I_TH/RUN voltage

to increase until the average inductor current matches the

new load current.

Low Load Current Operation (LTC3801B Only)

Under very light load current conditions, the I_TH/RUN pin

voltage will be very close to the zero current level of 0.85V.

As the load current decreases further, an internal offset at

the current comparator input will ensure that the current

comparator remains tripped (even at zero load current)

and the regulator will start to skip cycles, as it must, in

order to maintain regulation. This behavior allows the

regulator to maintain constant frequency down to very

light loads, resulting in less low frequency noise genera-

tion over a wide load current range.

ThemaincontrolloopisshutdownbypullingtheI_TH/RUN

pin to ground. Releasing the I_TH/RUN pin allows an

internal1µAcurrentsource(2µAonLTC3801B)tocharge

up the external compensation network. When the I_TH/

RUN pin voltage reaches approximately 0.6V, the main

control loop is enabled and the I_TH/RUN voltage is pulled

up by a clamp to its zero current level of approximately

onediodevoltagedrop(0.7V). Astheexternalcompensa-

tion network continues to charge up, the corresponding

peak inductor current level follows, allowing normal op-

eration. The maximum peak inductor current attainable is

set by a clamp on the I_TH/RUN pin at 1.2V above the zero

current level (approximately 1.9V).

Figure 1 illustrates this result for the circuit on the front

page of this data sheet using both an LTC3801 (in Burst

Mode operation) and an LTC3801B (with Burst Mode

operation disabled). At an output current of 100mA, the

LTC3801 exhibits an output ripple of 81.6mV_P-P, whereas

the LTC3801B has an output ripple of only 17.6mV_P-P. At

lower output current levels, the improvement is even

greater. This comes at a tradeoff of lower efficiency for the

non Burst Mode part at light load currents (see Figure 2).

Also notice the constant frequency operation of the

LTC3801B, even at 5% of maximum output current.

Dropout Operation

Burst Mode Operation (LTC3801 Only)

When the input supply voltage decreases towards the

output voltage, the rate of change of inductor current

during the on cycle decreases. This reduction means that

at some input-output differential, the external P-channel

MOSFET will remain on for more than one oscillator cycle

(start dropping off-cycles) since the inductor current has

not ramped up to the threshold set by the error amplifier.

Further reduction in input supply voltage will eventually

cause the external P-channel MOSFET to be turned on

100%, i.e., DC. The output voltage will then be determined

by the input voltage minus the voltage drop across the

sense resistor, the MOSFET and the inductor.

The LTC3801 incorporates Burst Mode operation at low

load currents (<25% of I_MAX). In this mode, an internal

clamp sets the peak current of the inductor at a level cor-

responding to an I_TH/RUN voltage 0.3V above its zero

current level (approximately 1V), even though the actual

I_TH/RUN voltage is lower. When the inductor’s average

current is greater than the load requirement, the voltage at

the I_TH/RUN pin will drop. When the I_TH/RUN voltage falls

to0.15Vaboveitszerocurrentlevel(approximately0.85V),

the sleep comparator will trip, turning off the external

MOSFET. In sleep, the input DC supply current to the IC is

reducedto16µAfrom195µAinnormaloperation.Withthe

switch held off, average inductor current will decay to zero

3801f

5

LTC3801/LTC3801B

U

(Refer to the Functional Diagram)

OPERATIO

V_OUTRipple for Front Page Circuit Using the LTC3801

V_OUTRipple for Front Page Circuit Using the LTC3801B

(Burst Mode Operation Disabled)

(with Burst Mode Operation)

20mV_AC/DIV

V_IN= 4.2V

V_OUT= 2.5V

I_OUT= 100mA

5µs/DIV

3801 F01a

V

_IN= 4.2V

5µs/DIV

3801 F01b

V_OUT= 2.5V

I_OUT= 100mA

Figure 1. Output Ripple Waveforms for the Front Page Circuit

100

V

= 2.5V

This lower frequency allows the inductor current to safely

discharge, thereby preventing current runaway. After the

short is removed, the oscillator frequency will gradually

increase back to 550kHz as V_FBrises through 0.3V on its

way back to 0.8V.

OUT

95

90

85

V

= 3.3V

IN

80

75

V

= 4.2V

= 6.6V

IN

70

65

60

55

50

Overvoltage Protection

IN

V

= 8.4V

IN

V

= 9.8V

100

IN

If V_FBexceeds its regulation point of 0.8V by more than

10% for any reason, such as an output short circuit to a

higher voltage, the overvoltage comparator will hold the

external P-channel MOSFET off. This comparator has a

typical hysteresis of 40mV.

0.1

1

10

1000

10000

LOAD CURRENT (mA)

3801 F02

Figure 2. LTC3801B Efficiency vs Load Current

Slope Compensation and Inductor’s Peak Current

Undervoltage Lockout Protection

The switch on duty cycle in normal operation is given by:

To prevent operation of the external P-channel MOSFET

with insufficient gate drive, an undervoltage lockout cir-

cuit is incorporated into the LTC3801/LTC3801B. When

the input supply voltage drops below approximately 1.7V,

the P-channel MOSFET and all internal circuitry other than

the undervoltage block itself are turned off. Input supply

current in undervoltage is approximately 1µA.

V

_OUT+ V_D

Duty Cycle =

V + V_D

IN

where V_Dis the forward voltage drop of the external diode

at the average inductor current. For duty cycles less than

40%, the inductor’s peak current is determined by:

V

_ITH/RUN– 0.7V

10R_SENSE

I_MAX

=

Short-Circuit Protection

If the output is shorted to ground, the frequency of the

oscillator is folded back from 550kHz to approximately

210kHz while maintaining the same minimum on time.

However, for duty cycles greater than 40%, slope com-

pensation begins and effectively reduces the peak

3801f

6

LTC3801/LTC3801B

U

(Refer to the Functional Diagram)

OPERATIO

inductor current. The amount of reduction is given by the

curve in Figure 3.

100

115

105

95

90

LTC3801

80

LTC3801B

Soft-Start

85

70

60

50

40

30

An internal default soft-start circuit is employed at power-

up and/or when coming out of shutdown. The soft-start

circuit works by internally clamping the voltage at the

I_TH/RUN pin to the corresponding zero current level and

graduallyraisingtheclampvoltagesuchthattheminimum

time required for the programmed switch current to reach

its maximum is approximately 0.6ms. After the soft-start

circuit has timed out, it is disabled until the part is put in

shutdown again or the input supply is cycled.

75

65

55

V

A

= 4.2V

45

IN

T

= 25°C

35

60 70

DUTY CYCLE (%)

20 30 40 50

80 90 100

3801 F03

Figure 3. Maximum Current Limit Trip Voltage vs Duty Cycle

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APPLICATIO S I FOR ATIO

ThebasicLTC3801/LTC3801Bapplicationcircuitisshown

on the front page of this data sheet. External component

selectionisdrivenbytheloadrequirementandbeginswith

the selection of the inductor and R_SENSE. Next, the power

MOSFET and the output diode are selected followed by the

input bypass capacitor C_INand output bypass capacitor

However,foroperationthatisabove40%dutycycle,slope

compensation effect has to be taken into consideration to

selecttheappropriatevaluetoprovidetherequiredamount

of current. Using Figure 3, the value of R_SENSEis:

SF

R_SENSE

=

C_OUT

.

(10)(I_OUT)(100)

where SF is the “Slope Factor.”

R_SENSESelection for Output Current

R_SENSEis chosen based on the required output current.

With the current comparator monitoring the voltage

developed across R_SENSE, the threshold of the compara-

tor determines the inductor’s peak current. The output

current the LTC3801 can provide is given by:

Inductor Value Calculation

The operating frequency and inductor selection are inter-

related in that higher operating frequencies permit the use

of a smaller inductor for the same amount of inductor

ripplecurrent. However, thisisattheexpenseofefficiency

due to an increase in MOSFET gate charge losses.

0.117 I_RIPPLE

I_OUT

=

−

R_SENSE

2

The inductance value also has a direct effect on ripple cur-

rent. The ripple current, I_RIPPLE, decreases with higher in-

ductance or frequency and increases with higher V_INor

V_OUT.Theinductor’speak-to-peakripplecurrentisgivenby:

where I_RIPPLEis the inductor peak-to-peak ripple current

(seeInductorValueCalculationsection).FortheLTC3801B

use 104mV in the previous equation and follow through

the analysis using that number.

V − V_OUT

V

_OUT+ V_D

IN

I_RIPPLE

=

A reasonable starting point for setting ripple current is

f(L)

V + V_D

IN

I

_RIPPLE= (0.4)(I_OUT). Rearranging the above equation, it

becomes:

wherefistheoperatingfrequency.Acceptinglargervalues

of I_RIPPLEallows the use of low inductances, but results in

higher output voltage ripple and greater core losses. A

1

R_SENSE

=

for Duty Cycle < 40%

(10)(I_OUT

)

reasonable starting point for setting ripple current is

3801f

7

LTC3801/LTC3801B

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APPLICATIO S I FOR ATIO

I_RIPPLE=0.4(I_OUT(MAX)).Remember,themaximumI_RIPPLE

manufacturerisKoolMµ. Toroidsareveryspaceefficient,

especially when you can use several layers of wire.

Because they generally lack a bobbin, mounting is more

difficult. However, new designs for surface mount that do

not increase the height significantly are available.

occurs at the maximum input voltage.

In Burst Mode operation on the LTC3801, the ripple

current is normally set such that the inductor current is

continuous during the burst periods. Therefore, the peak-

to-peak ripple current must not exceed:

Power MOSFET Selection

0.03

An external P-channel power MOSFET must be selected

for use with the LTC3801/LTC3801B. The main selection

criteria for the power MOSFET are the threshold voltage

I_RIPPLE

≤

R_SENSE

This implies a minimum inductance of:

V

_GS(TH)and the “on” resistance R_DS(ON), reverse transfer

capacitance C_RSSand total gate charge.

V − V_OUT

V

_OUT+ V_D

IN

L_MIN

=

Since the LTC3801/LTC3801B are designed for operation

down to low input voltages, a sublogic level threshold

MOSFET (R_DS(ON)guaranteed at V_GS= 2.5V) is required

for applications that work close to this voltage. When

these MOSFETs are used, make sure that the input supply

to the LTC3801/LTC3801B is less than the absolute maxi-

mum V_GSrating, typically 8V.

V + V_D

0.03

R_SENSE

IN

f

(Use V_IN(MAX)= V_IN)

A smaller value than L_MINcould be used in the circuit;

however, the inductor current will not be continuous

during burst periods.

TherequiredminimumR_DS(ON)oftheMOSFETisgoverned

byitsallowablepowerdissipation.Forapplicationsthatmay

operatetheLTC3801/LTC3801B indropout,i.e.,100%duty

cycle, at its worst case the required R_DS(ON)is given by:

Inductor Core Selection

Once the value for L is known, the type of inductor must be

selected. High efficiency converters generally cannot af-

ford the core loss found in low cost powdered iron cores,

forcing the use of more expensive ferrite, molypermalloy

or Kool Mµ^®cores. Actual core loss is independent of core

size for a fixed inductor value, but it is very dependent on

inductance selected. As inductance increases, core losses

go down. Unfortunately, increased inductance requires

more turns of wire and therefore copper losses will in-

crease. Ferrite designs have very low core losses and are

preferred at high switching frequencies, so design goals

canconcentrateoncopperlossandpreventingsaturation.

Ferrite core material saturates “hard,” which means that

inductance collapses abruptly when the peak design cur-

rent is exceeded. This results in an abrupt increase in

inductor ripple current and consequent output voltage

ripple. Do not allow the core to saturate!

P_P

R_DS(ON)

=

DC=100%

2

I

(

1+ δp

(

)

OUT(MAX)

where P_Pis the allowable power dissipation and δp is the

temperature dependency of R_DS(ON). (1 + δp) is generally

given for a MOSFET in the form of a normalized R_DS(ON)vs

temperature curve, but δp = 0.005/°C can be used as an

approximation for low voltage MOSFETs.

In applications where the maximum duty cycle is less than

100% and the LTC3801/LTC3801B are in continuous

mode, the R_DS(ON)is governed by:

P_P

R_DS(ON)

2

DC I

1+ δp

(

)

(

)

OUT

Molypermalloy (from Magnetics, Inc.) is a very good, low

loss core material for toroids, but it is more expensive

than ferrite. A reasonable compromise from the same

where DC is the maximum operating duty cycle of the

LTC3801/LTC3801B.

Kool Mµ is a registered trademark of Magnetics, Inc.

3801f

8

LTC3801/LTC3801B

W U U

APPLICATIO S I FOR ATIO

U

Output Diode Selection

input capacitor sized for the maximum RMS current must

beused. ThemaximumRMScapacitorcurrentisgivenby:

The catch diode carries load current during the off-time.

The average diode current is therefore dependent on the

P-channel switch duty cycle. At high input voltages the

diode conducts most of the time. As V_INapproaches V_OUT

the diode conducts only a small fraction of the time. The

most stressful condition for the diode is when the output

is short-circuited. Under this condition the diode must

safelyhandleI_PEAKatcloseto100%dutycycle. Therefore,

itisimportanttoadequatelyspecifythediodepeakcurrent

and average power dissipation so as not to exceed the

diode ratings.

1/2

]

V

V − V

OUT

(

)

[

OUT IN

C_INRequired I_RMS≈I_MAX

V

IN

This formula has a maximum value at V_IN= 2V_OUT, where

_RMS= I_OUT/2. This simple worst-case condition is com-

I

monlyusedfordesignbecauseevensignificantdeviations

donotoffermuchrelief.Notethatcapacitormanufacturer’s

ripplecurrentratingsareoftenbasedon2000hoursoflife.

This makes it advisable to further derate the capacitor, or

to choose a capacitor rated at a higher temperature than

required. Several capacitors may be paralleled to meet the

size or height requirements in the design. Due to the high

operating frequency of the LTC3801/LTC3801B, ceramic

capacitors can also be used for C_IN. Always consult the

manufacturer if there is any question.

Under normal load conditions, the average current con-

ducted by the diode is:

V − V_OUT

V + V_D

IN

I_D=

I_OUT

The allowable forward voltage drop in the diode is calcu-

lated from the maximum short-circuit current as:

The selection of C_OUTis driven by the required effective

series resistance (ESR). Typically, once the ESR require-

ment is satisfied, the capacitance is adequate for filtering.

The output ripple (∆V_OUT) is approximated by:

P_D

I_SC(MAX)

V ≈

F

1

∆V_OUT≈I_RIPPLEESR +

8fC_OUT

where P_Dis the allowable power dissipation and will be

determined by efficiency and/or thermal requirements.

where f is the operating frequency, C_OUTis the output

capacitance and I_RIPPLEis the ripple current in the induc-

tor. The output ripple is highest at maximum input voltage

since ∆I_Lincreases with input voltage.

A fast switching diode must also be used to optimize

efficiency. Schottky diodes are a good choice for low

forwarddropandfastswitchingtimes. Remembertokeep

lead length short and observe proper grounding (see

Board Layout Checklist) to avoid ringing and increased

dissipation.

Manufacturers such as Nichicon, United Chemicon and

Sanyoshouldbeconsideredforhighperformancethrough-

hole capacitors. The OS-CON semiconductor dielectric

capacitor available from Sanyo has the lowest ESR (size)

product of any aluminum electrolytic at a somewhat

higher price. Once the ESR requirement for C_OUThas been

met, the RMS current rating generally far exceeds the

An additional consideration in applications where low no-

load quiescent current is critical is the reverse leakage

current of the diode at the regulated output voltage. A

leakage greater than several microamperes can represent

a significant percentage of the total input current.

I_RIPPLE(P-P)requirement.

C_INand C_OUTSelection

In surface mount applications, multiple capacitors may

have to be paralleled to meet the ESR or RMS current

handling requirements of the application. Aluminum elec-

trolytic and dry tantalum capacitors are both available in

In continuous mode, the source current of the P-channel

MOSFET is a square wave of duty cycle (V_OUT+ V_D)/

(V_IN+ V_D). To prevent large voltage transients, a low ESR

3801f

9

LTC3801/LTC3801B

W U U

U

APPLICATIO S I FOR ATIO

surfacemountconfigurations. Inthecaseoftantalum, itis

critical that the capacitors are surge tested for use in

switching power supplies. An excellent choice is the AVX

TPS, AVX TPSV and KEMET T510 series of surface mount

tantalum, available in case heights ranging from 2mm to

4mm. Other capacitor types include Sanyo OS-CON,

Nichicon PL series and Panasonic SP.

Although all dissipative elements in the circuit produce

losses, four main sources usually account for most of the

losses in LTC3801/LTC3801B circuits: 1) LTC3801/

LTC3801B DC bias current, 2) MOSFET gate charge cur-

rent, 3) I²R losses and 4) voltage drop of the output diode.

1. The V_INcurrent is the DC supply current, given in the

electrical characteristics, that excludes MOSFET driver

and control currents. V_INcurrent results in a small loss

which increases with V_IN.

Setting Output Voltage

The LTC3801/LTC3801B develop a 0.8V reference voltage

between the feedback (Pin 3) terminal and ground (see

Figure 4). By selecting resistor R1, a constant current is

caused to flow through R1 and R2 to set the overall output

voltage. The regulated output voltage is determined by:

2. MOSFETgatechargecurrentresultsfromswitchingthe

gate capacitance of the power MOSFET. Each time a

MOSFET gate is switched from low to high to low again,

a packet of charge dQ moves from V_INto ground. The

resulting dQ/dt is a current out of V_INwhich is typically

much larger than the DC supply current. In continuous

mode, I_GATECHG= (f)(dQ).

3. I²R losses are predicted from the DC resistances of the

MOSFET, inductor and current shunt. In continuous

mode the average output current flows through L but is

“chopped” between the P-channel MOSFET (in series

withR_SENSE)andtheoutputdiode.TheMOSFETR_DS(ON)

plusR_SENSEmultipliedbydutycyclecanbesummedwith

the resistances of L and R_SENSEto obtain I²R losses.

R2

V_OUT= 0.8 1+

R1

Formostapplications, an80kresistorissuggestedforR1.

In applications where low no-load quiescent current is

critical, R1 should be made >400k to limit the feedback

dividercurrenttoapproximately10%ofthechipquiescent

current. If R2 then results in a very high impedance, it may

bebeneficialtobypassR2witha5pFto10pFcapacitor. To

prevent stray pickup, locate resistors R1 and R2 close to

LTC3801/LTC3801B.

4. Theoutputdiodeisamajorsourceofpowerlossathigh

currents and gets worse at high input voltages. The

diode loss is calculated by multiplying the forward

voltagetimesthediodedutycyclemultipliedbytheload

current. For example, assuming a duty cycle of 50%

with a Schottky diode forward voltage drop of 0.4V, the

loss increases from 0.5% to 8% as the load current

increases from 0.5A to 2A.

V

OUT

LTC3801/

R2

LTC3801B

3

V

FB

R1

3801 F04

Figure 4. Setting Output Voltage

5. Transition losses apply to the external MOSFET and

increase at higher operating frequencies and input

voltages. Transition losses can be estimated from:

Efficiency Considerations

The efficiency of a switching regulator is equal to the

output power divided by the input power times 100%. It is

oftenusefultoanalyzeindividuallossestodeterminewhat

is limiting the efficiency and which change would produce

the most improvement. Efficiency can be expressed as:

Transition Loss = 2(V_IN)²I_{O(MAX) RSS}

(f)

C

Other losses including C_INand C_OUTESR dissipative

losses, and inductor core losses, generally account for

less than 2% total additional loss.

Efficiency = 100% – (η1 + η2 + η3 + ...)

where η1, η2, etc. are the individual losses as a percent-

age of input power.

3801f

10

LTC3801/LTC3801B

W U U

APPLICATIO S I FOR ATIO

U

Foldback Current Limiting

V

LTC3801/

LTC3801B

OUT

R2

R1

As described in the Output Diode Selection, the worst-

case dissipation occurs with a short-circuited output

when the diode conducts the current limit value almost

continuously. To prevent excessive heating in the diode,

foldback current limiting can be added to reduce the

current in proportion to the severity of the fault.

+

I

/RUN V

FB

TH

D

FB1

FB2

3801 F05

Figure 5. Foldback Current Limiting

Foldback current limiting is implemented by adding di-

odes D_FB1and D_FB2between the output and the I_TH/RUN

pin as shown in Figure 5. In a hard short (V_OUT= 0V), the

current will be reduced to approximately 50% of the

maximum output current.

U

PACKAGE DESCRIPTIO

S6 Package

6-Lead Plastic TSOT-23

(Reference LTC DWG # 05-08-1636)

2.90 BSC

(NOTE 4)

0.62

MAX

0.95

REF

1.22 REF

1.4 MIN

1.50 – 1.75

2.80 BSC

3.85 MAX 2.62 REF

(NOTE 4)

PIN ONE ID

RECOMMENDED SOLDER PAD LAYOUT

PER IPC CALCULATOR

0.30 – 0.45

6 PLCS (NOTE 3)

0.95 BSC

0.80 – 0.90

0.20 BSC

DATUM ‘A’

0.01 – 0.10

1.00 MAX

0.30 – 0.50 REF

1.90 BSC

0.09 – 0.20

(NOTE 3)

S6 TSOT-23 0302

NOTE:

1. DIMENSIONS ARE IN MILLIMETERS

2. DRAWING NOT TO SCALE

4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR

5. MOLD FLASH SHALL NOT EXCEED 0.254mm

6. JEDEC PACKAGE REFERENCE IS MO-193

3. DIMENSIONS ARE INCLUSIVE OF PLATING

3801f

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.

11

LTC3801/LTC3801B

U

TYPICAL APPLICATIO

550kHz Micropower Step-Down DC/DC Controller

LTC3801 No-Load I_Qvs Input Voltage*

220pF

25

V

IN

10k

V

= 2.5V

OUT

2.7V TO 9.8V

I

/RUN

LTC3801/

V

FRONT PAGE APPLICATION

TH

IN

–

10µF

0.025Ω

23

21

19

17

15

LTC3801B

GND

SENSE

V

PGATE

FB

402k

4.7µH

V

2.5V

2A

866k

OUT

+

47µF

3801 TA01

3

4

5

6

7

8

9

10

V

IN

INPUT VOLTAGE (V)

3801 TA04

*SEE THE FRONT PAGE OF THIS DATA SHEET FOR THE EFFICIENCY vs LOAD CURRENT CURVE

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750kHz Optional Burst Mode Operation, 8-Lead MSOP

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550kHz, 2 Phase, Dual Synchronous Controller

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OUT OUT

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LT^®1765

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25V, 2.75A (I ), 1.25MHz Step-Down Converter

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≥ 1.2V, SO-8 and TSSOP16 Packages

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IN

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10µA Supply Current, 93% Efficiency,

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LTC1778/LTC1778-1 No R Current Mode Synchronous Step-Down Controllers

2.5V ≤ V ≤ 9.8V, V

≥ 0.8V, I

≤ 6A

IN

OUT

4V ≤ V ≤ 36V, 0.8V ≤ V

≤ (0.9)(V ), I

Up to 20A

SENSE

IN

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IN OUT

LTC1779

250mA Monolithic Step-Down Converter in ThinSOT

2.5V ≤ V ≤ 9.8V, 550kHz, V

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2.5V ≤ V ≤ 9.8V; 90% Efficiency

IN

LTC3411/LTC3412

LTC3440

1.25/2.5A Monolithic Synchronous Step-Down Converter

600mA (I ), 2MHz Synchronous Buck-Boost DC/DC Converter

95% Efficiency, 2.5V ≤ V ≤ 5.5V, V

TSSOP16 Exposed Pad Package

≥ 0.8V,

OUT

IN

2.5V ≤ V ≤ 5.5V, Single Inductor

OUT

IN

No R

is a trademark of Linear Technology Corporation.

SENSE

3801f

LT/TP 1103 1K • PRINTED IN THE USA

LinearTechnology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

12

●

(408) 432-1900 FAX: (408) 434-0507 www.linear.com

LINEAR TECHNOLOGY CORPORATION 2003


型号：	LTC3801BES6
厂家：	Linear
描述：	Micropower Constant Frequency Step-Down DC/DC Controllers in ThinSOT 微恒频降压型DC / DC采用ThinSOT封装的控制器稳压器开关式稳压器或控制器电源电路开关式控制器光电二极管
文件：	总12页 (文件大小：241K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LTC3801BES6 [Linear]

相关型号：

LTC3801BES6#PBF

LTC3801BES6#TR

LTC3801BES6#TRM

LTC3801BES6#TRMPBF

LTC3801BES6#TRPBF

LTC3801B_15

LTC3801ES6

LTC3801ES6#TR

LTC3801ES6#TRM

LTC3801ES6#TRPBF

LTC3801_1

LTC3801_15