MAX16834 [MAXIM]
High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver; 高功率LED驱动器，集成高边LED电流检测和PWM调光MOSFET驱动器
| 型号： | MAX16834 |
| 厂家： | 
MAXIM INTEGRATED PRODUCTS |
| 描述： | High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver |
| 文件： | 总23页 (文件大小：199K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
19-4235; Rev 3; 1/10
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
General Description
Features
o Wide Input Operating Voltage Range (4.75V to
The MAX16834 is a current-mode high-brightness LED
(HB LED) driver for boost, boost-buck, SEPIC, and high-
side buck topologies. In addition to driving an n-channel
power MOSFET switch controlled by the switching con-
troller, it also drives an n-channel PWM dimming switch to
achieve LED PWM dimming. The MAX16834 integrates
all the building blocks necessary to implement a fixed-fre-
quency HB LED driver with wide-range dimming control.
The MAX16834 features constant-frequency peak cur-
rent-mode control with programmable slope compensa-
tion to control the duty cycle of the PWM controller.
28V)
o Works for Input Voltage > 28V with External
Voltage Clamp on V for Boost Converter
IN
o 3000:1 PWM Dimming/Analog Dimming
o Integrated PWM Dimming MOSFET Driver
o Integrated High-Side Current-Sense Amplifier for
LED Current Sense in Boost-Buck Converter
o 100kHz to 1MHz Programmable High-Frequency
Operation
o External Clock Synchronization Input
o Programmable UVLO
A dimming driver designed to drive an external n-chan-
nel MOSFET in series with the LED string provides
wide-range dimming control up to 20kHz. In addition to
PWM dimming, the MAX16834 provides analog dim-
ming using a DC input at REFI. The programmable
switching frequency (100kHz to 1MHz) allows design
optimization for efficiency and board space reduction.
A single resistor from RT/SYNC to ground sets the
switching frequency from 100kHz to 1MHz while an
external clock signal at RT/SYNC disables the internal
oscillator and allows the MAX16834 to synchronize to
an external clock. The MAX16834’s integrated high-
side current-sense amplifier eliminates the need for a
separate high-side LED current-sense amplifier in
boost-buck applications.
o Internal 7V Low-Dropout Regulator
o Fault Output (FLT) for Overvoltage, Overcurrent,
and Thermal Warning Faults
o Programmable True Differential Overvoltage
Protection
o 20-Pin TQFN-EP and TSSOP-EP Packages
Ordering Information
PART
TEMP RANGE
-40°C to +125°C
-40°C to +125°C
-40°C to +125°C
-40°C to +125°C
PIN-PACKAGE
20 TQFN-EP*
20 TQFN-EP*
20 TSSOP-EP*
20 TSSOP-EP*
MAX16834ATP+
MAX16834ATP/V+
MAX16834AUP+
MAX16834AUP/V+
The MAX16834 operates over a wide supply range of
4.75V to 28V and includes a 3A sink/source gate driver
for driving a power MOSFET in high-power LED driver
applications. It can also operate at input voltages
greater than 28V in boost configuration with an external
voltage clamp. The MAX16834 is also suitable for DC-
DC converter applications such as boost or boost-
buck. Additional features include external enable/
disable input, an on-chip oscillator, fault indicator out-
put (FLT) for LED open/short or overtemperature condi-
tions, and an overvoltage protection sense input
(OVP+) for true overvoltage protection.
+Denotes a lead(Pb)-free/RoHS-compliant package.
*EP = Exposed pad.
/V denotes an automotive qualified part.
Simplified Application Circuit
V
IN
BOOST LED DRIVER
LED+
IN
NDRV
The MAX16834 is available in a thermally enhanced
4mm x 4mm, 20-pin TQFN-EP package and in a thermal-
ly enhanced 20-pin TSSOP-EP package and is specified
over the automotive -40°C to +125°C temperature range.
LEDs
LED-
MAX16834
ON
OFF
PWMDIM
REFI
CS
ANALOG
DIM
DIMOUT
Applications
Single-String LED LCD Backlighting
Automotive Rear and Front Lighting
PGND
SENSE+
Projection System RGB LED Light Sources
Architectural and Decorative Lighting (MR16, M111)
Spot and Ambient Lights
DC-DC Boost/Boost-Buck Converters
Pin Configurations appear at end of data sheet.
________________________________________________________________ Maxim Integrated Products
1
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
ABSOLUTE MAXIMUM RATINGS
IN, HV, LV to SGND................................................-0.3V to +30V
OVP+, SENSE+, DIMOUT, CLV to SGND ..............-0.3V to +30V
SENSE+ to LV........................................................-0.3V to +0.3V
HV, IN to LV............................................................-0.3V to +30V
OVP+, CLV, DIMOUT to LV......................................-0.3V to +6V
PGND to SGND .....................................................-0.3V to +0.3V
20-Pin TSSOP (derate 26.5mW/°C above +70°C)..........2122mW
Junction-to-Ambient Thermal Resistance (θJA) (Note 1)
20-Pin TQFN 4mm x 4mm .................................................39°C/W
20-Pin TSSOP..................................................................37.7°C/W
Junction-to-Case Thermal Resistance (θJC) (Note 1)
20-Pin TQFN 4mm x 4mm...............................................6°C/W
20-Pin TSSOP..................................................................2°C/W
Operating Temperature Range .........................-40°C to +125°C
Junction Temperature......................................................+150°C
Storage Temperature Range.............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
V
CC
to SGND..........................................................-0.3V to +12V
NDRV to PGND...........................................-0.3V to (V
+ 0.3V)
CC
All Other Pins to SGND.............................................-0.3V to +6V
NDRV Continuous Current................................................ 50mA
DIMOUT Continuous Current.............................................. 2mA
MAX16834
V
Short-Circuit Current to SGND Duration ...........................1s
CC
Continuous Power Dissipation (T = +70°C)
A
*As per JEDEC51 standard (multilayer board).
20-Pin TQFN (4mm x 4mm)
(derate 25.6mW/°C* above +70°C) ............................2051mW
Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-
layer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial.
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(V = V
= 12V, V
= 5V, V = V
= V
, C
= 4.7µF, C
= 100nF, C
= 100nF, R
= 0.1Ω,
SGND
IN
HV
UVEN
LV
PWMDIM
VCC
LCV
REF
SENSE+
R
= 10kΩ, T = T = -40°C to +125°C, unless otherwise noted. Typical values are at T = +25°C.)
RT
A
J
A
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
28
UNITS
V
Input Voltage Range
V
4.75
IN
Q
Quiescent Supply Current
Shutdown Supply Current
INTERNAL LINEAR REGULATOR (V
Output Voltage
I
Excluding I
6
10
mA
µA
LED
I
V
= 0
UVEN
30
60
SHDN
)
CC
V
0 ≤ I
≤ 50mA, 9.5V ≤ V ≤ 28V
6.3
80
7
7.7
1
V
V
CC
DO
CC
IN
Dropout Voltage
V
I
= 35mA (Note 2)
0.65
CC
Short-Circuit Current
V
= 0, V = 12V
300
mA
CC
IN
LINEAR REGULATOR (CLV)
0 ≤ I
6V ≤ V
≤ 2mA, 6V ≤ V ≤ 28V,
HV
CLV
Output Voltage
(V
V
)
4.7
5
5.3
V
CLV - LV
≤ 22V
(HV-LV)
Dropout Voltage
V
I
= 2mA, 0 ≤ V ≤ 23.3V (Note 3)
0.5
10
V
DO
CLV
LV
Short-Circuit Current
REFERENCE VOLTAGE (REF)
Output Voltage
V
= 12V, V = 12V, V = 24V
2.2
mA
CLV
IN
HV
V
0 ≤ I
≤ 1mA, 4.75V ≤ V ≤ 28V
3.625
3.70
30
3.775
1.475
V
REF
REF
IN
REF Short-Circuit Current
V
= 0
mA
REF
UNDERVOLTAGE LOCKOUT/ENABLE INPUT (UVEN)
UVEN On Threshold Voltage
V
1.395
1.395
1.435
200
I1I
V
UVEN_THUP
UVEN Threshold Voltage
Hysteresis
mV
µA
Input Leakage Current
PWMDIM
I
V
= 0
UVEN
LEAK
PWMDIM On Threshold Voltage
V
1.435
200
1.475
V
PWMDIM
PWMDIM Threshold Voltage
Hysteresis
mV
2
_______________________________________________________________________________________
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
ELECTRICAL CHARACTERISTICS (continued)
(V = V
= 12V, V
= 5V, V = V
= V
, C
= 4.7µF, C
= 100nF, C
= 100nF, R
= 0.1Ω,
SGND
IN
HV
UVEN
LV
PWMDIM
VCC
LCV
REF
SENSE+
R
= 10kΩ, T = T = -40°C to +125°C, unless otherwise noted. Typical values are at T = +25°C.)
RT
A
J
A
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
Input Leakage Current
V
= 0
I1I
µA
PWMDIM
OSCILLATOR
R
R
= 5kΩ
0.9
180
100
1
1.1
220
MHz
kHz
kHz
RT/SYNC
Oscillator Frequency
f
OSC
= 25kΩ
200
RT/SYNC
Oscillator Frequency Range
(Note 4)
1000
External Sync Input Clock High
Threshold
(Note 4)
2
V
V
External Sync Input Clock Low
Threshold
(Note 4)
(Note 4)
0.4
External Sync Input High Pulse
Width
200
80
ns
µs
Maximum External Sync Period
SLOPE COMPENSATION (SC)
SC Pullup Current
50
I
V
V
= 100mV
100
8
120
µA
SCPU
SC
SC
SC Discharge Resistance
REFI
R
= 100mV
Ω
SCD
REFI Input Bias Current
REFI Input Common-Mode Range
SENSE+
V
= 1V
I1I
µA
V
REFI
(Note 4)
0
2
SENSE+ Input Bias Current
(V
SENSE+
- V ) = 100mV
250
µA
LV
HIGH-SIDE LED CURRENT-SENSE AMPLIFIER (V
- V
)
SENSE+
LV
Input Offset Voltage
Voltage Gain
V
V
> 5V, (V
> 5V, (V
- V ) = 5mV
-2.4
9.7
0
+2.4
10.1
mV
V/V
LV
LV
SENSE+
SENSE+
LV
A
- V ) = 0.2V
9.9
1.8
600
V
LV
(V
- V ) = 0.1V, no load
MHz
kHz
SENSE+
SENSE+
LV
3dB Bandwidth
(V
- V ) = 0.02V, no load
LV
LOW-SIDE LED CURRENT-SENSE AMPLIFIER
Input Offset Voltage
V
V
< 1V, (V
- V ) = 0V
-2
0
+2
mV
V/V
kHz
LV
LV
SENSE+
LV
Voltage Gain
A
< 1V, (V
- V ) = 0.2V
9.7
9.9
600
10.1
V
SENSE+
LV
3dB Bandwidth
CURRENT ERROR AMPLIFIER (TRANSCONDUCTANCE AMPLIFIER)
Transconductance
Open-Loop DC Gain
Input Offset Voltage
COMP Voltage Range
PWM COMPARATOR
Input Offset Voltage
Propagation Delay
g
V
= 2V, V = 5V
PWMDIM
400
500
60
0
600
µS
dB
mV
V
m
COMP
A
V
-10
0.4
+10
2.5
V
(Note 4)
COMP
0.6
0.65
40
0.70
V
t
50mV overdrive
ns
PD
_______________________________________________________________________________________
3
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
ELECTRICAL CHARACTERISTICS (continued)
(V = V
= 12V, V
= 5V, V = V
= V
, C
= 4.7µF, C
= 100nF, C
= 100nF, R
= 0.1Ω,
SGND
IN
HV
UVEN
LV
PWMDIM
VCC
LCV
REF
SENSE+
R
= 10kΩ, T = T = -40°C to +125°C, unless otherwise noted. Typical values are at T = +25°C.)
RT
A
J
A
PARAMETER
SYMBOL
CONDITIONS
On-time includes blanking time
(Note 4)
MIN
TYP
MAX
UNITS
ns
Minimum On-Time
Duty Cycle
t
100
ON(MIN)
90
99.5
0.35
%
CURRENT PEAK LIMIT COMPARATOR
Trip Threshold Voltage
0.25
0.3
40
V
MAX16834
Propagation Delay
50mV overdrive with respect to NDRV
ns
OVERVOLTAGE PROTECTION INPUT (OVP+)
OVP+ On Threshold Voltage
OVP+ Hysteresis
V
1.375
-1
1.435
200
1.495
+1
V
OVP_ON
mV
µA
OVP+ Input Leakage Current
(V
OVP
- V ) = 1.235V
LV
HIGH-SIDE LED SHORT COMPARATOR
Off Threshold
V
V
- V
4.0
4.1
4.3
4.4
256
4.6
4.7
V
V
CLV
LV
On Threshold
- V
LV
CLV
Error Reject Blankout
f
= 500kHz
µs
OSC
LOW-SIDE LED SHORT COMPARATOR
Off Threshold
0.27
0.30
5
0.33
V
Error Reject Blankout
µs
HICCUP TIMER
Hiccup Time
f
= 500kHz
8.2
ms
OSC
GATE-DRIVER OUTPUT (NDRV)
NDRV Peak Pullup Current
NDRV Peak Pulldown Current
V
V
= 7V
= 7V
3
A
A
CC
CC
3
p-Channel MOSFET R
n-Channel MOSFET R
DIMOUT
(V
- V ) = 0.1V
NDRV
1.2
0.9
1.9
1.7
Ω
Ω
DSON
DSON
CC
V
= 0.1V
NDRV
DIMOUT Peak Pullup Current
DIMOUT Peak Pulldown Current
(V
(V
(V
(V
- V ) = 5V
25
25
50
50
31
25
mA
mA
Ω
CLV
CLV
CLV
LV
- V ) = 5V
LV
p-Channel MOSFET R
n-Channel MOSFET R
- V ) = 0.1V
DIMOUT
DSON
DSON
- V ) = 0.1V
Ω
DIMOUT
LV
PWMDIM to DIMOUT
Propagation Delay
200
ns
FAULT FLAG (FLT)
FLT Pulldown Current
FLT Leakage Current
V
V
= 0.2V
2
5
10
mA
µA
°C
FLT
FLT
= 1.0V
I1I
Thermal Warning On Threshold
+140
Thermal Warning Threshold
Hysteresis
20
°C
Note 2: Dropout voltage is defined as V - V , when V
is 100mV below the value of V for V = 9.5V.
CC IN
IN
CC
CC
Note 3: Dropout is defined as V - V
, when V
is 100mV below the value of V for V = 8V.
CLV HV
HV
CLV
CLV
Note 4: Not production tested. Guaranteed by design.
4
_______________________________________________________________________________________
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
Typical Operating Characteristics
(V = V
= 12V, V
A
= 5V, V = V
= V
, C
= 4.7µF, C
= 100nF, C
= 100nF, R
= 0.1Ω,
SENSE+
SGND
IN
HV
UVEN
LV
PWMDIM
VCC
LCV
REF
R
RT
= 10kΩ, T = +25°C, unless otherwise noted.)
V
REF
vs. TEMPERATURE
V
REF
vs. SUPPLY VOLTAGE
V vs. I
REF REF
3.80
3.75
3.70
3.65
3.60
3.55
3.50
3.7020
3.7015
3.7010
3.7005
3.7000
3.6995
3.6990
3.6985
3.74
V
= 12V
IN
3.72
3.70
3.68
3.66
3.64
3.62
V
= 12V
IN
3.60
3.6980
0
-40 -25 -10
5
20 35 50 65 80 95 110 125
4
8
12
16
20
24
28
1
2
3
4
5
6
7
8
9
10
TEMPERATURE (°C)
SUPPLY VOLTAGE (V)
I
(mA)
REF
SUPPLY CURRENT
vs. TEMPERATURE
SUPPLY CURRENT
vs. SUPPLY VOLTAGE
RT vs. SWITCHING FREQUENCY
10
9
8
7
6
5
4
3
2
1
0
100
20
18
16
14
12
10
8
PWMDIM = 0
10
6
4
V
= 12V
IN
2
PWMDIM = 0
V
= 12V
IN
1
0
-40 -25 -10
5
20 35 50 65 80 95 110 125
100
1000
4
8
12
16
20
24
28
TEMPERATURE (°C)
SWITCHING FREQUENCY (kHz)
SUPPLY VOLTAGE (V)
SWITCHING FREQUENCY
vs. TEMPERATURE
V
vs. I
CC
V
vs. I
CC
CC
CC
7.2
7.1
7.0
6.9
6.8
605
604
603
602
601
600
599
598
597
596
595
594
7.16
7.14
7.12
7.10
7.08
7.06
7.04
7.02
7.00
6.98
6.96
6.94
6.92
6.90
V
= 12V
V
= 12V
T = +125°C
A
IN
IN
T
= +100°C
= -40°C
A
T
= +25°C
A
T
A
593
592
591
V
= 12V
IN
590
0
10 20 30 40 50 60 70 80 90 100
(mA)
-40 -25 -10
5
20 35 50 65 80 95 110 125
0
10 20 30 40 50 60 70 80 90 100
(mA)
I
TEMPERATURE (°C)
I
CC
CC
_______________________________________________________________________________________
5
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
Typical Operating Characteristics (continued)
(V = V
= 12V, V
A
= 5V, V = V
= V
, C
= 4.7µF, C
= 100nF, C
= 100nF, R
= 0.1Ω,
SENSE+
SGND
IN
HV
UVEN
LV
PWMDIM
VCC
LCV
REF
R
RT
= 10kΩ, T = +25°C, unless otherwise noted.)
NDRV RISE/FALL TIME
vs. CAPACITANCE
V
vs. V
IN
CC
50
40
30
20
10
0
7.20
7.18
7.16
7.14
7.12
7.10
7.08
7.06
7.04
7.02
7.00
V
= 12V
IN
T
= -40°C
T
A
= +25°C
A
T
= +125°C
A
MAX16834
RISE TIME
FALL TIME
10
22
0
1
2
3
4
5
6
7
8
9
10
6
14
18
26
CAPACITANCE (nF)
V
(V)
IN
V
CLV
vs. I
V
CLV
vs. V
HV
CLV
5.10
5.09
5.08
5.07
5.06
5.05
5.04
5.03
5.02
5.01
5.00
5.50
5.00
4.50
4.00
3.50
3.00
2.50
2.00
1.50
1.00
0.50
0
V
= 12V
IN
V
= 12V
IN
6
10
18
0
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0
(mA)
14
22
26
I
V
(V)
HV
CLV
Pin Description
PIN
TQFN TSSOP
NAME
FUNCTION
LED-String Overvoltage Protection Input. Connect a resistive voltage-divider between the positive
output, OVP+, and LV to set the overvoltage threshold. OVP+ has a 1.435V threshold voltage with a
200mV hysteresis.
1
3
OVP+
2
3
4
5
4
5
6
7
SGND
COMP
REF
Signal Ground
Error-Amplifier Output. Connect an RC network from COMP to SGND for stable operation. See the
Feedback Compensation section.
3.7V Reference Output Voltage. Bypass REF to SGND with a 0.1µF to 0.22µF ceramic capacitor.
Current Reference Input. V
the LED current.
provides a reference voltage for the current-sense amplifier to set
REFI
REFI
Current-Mode Slope Compensation Setting. Connect to an appropriate external capacitor from SC
to SGND to generate a ramp signal for stable operation.
6
8
SC
6
_______________________________________________________________________________________
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
Pin Description (continued)
PIN
NAME
FUNCTION
TQFN TSSOP
7
9
FLT
Active-Low, Open-Drain Fault Indicator Output. See the Fault Indicator (FLT) section.
Resistor-Programmable Switching Frequency Setting/Sync Control Input. Connect a resistor from
8
10
RT/SYNC RT/SYNC to SGND to set the switching frequency. Drive RT/SYNC to synchronize the switching
frequency with an external clock.
Undervoltage-Lockout (UVLO) Threshold/Enable Input. UVEN is a dual-function adjustable UVLO
9
11
UVEN
threshold input with an enable feature. Connect UVEN to V through a resistive voltage-divider to
IN
program the UVLO threshold. Observe the absolute maximum value for this pin.
10
11
12
13
PWMDIM PWM Dimming Input. Connect to an external PWM signal for dimming operation.
Current-Sense Amplifier Positive Input. Connect a resistor from CS to PGND to set the inductor peak
CS
current limit.
12
13
14
15
PGND
NDRV
Power Ground
External n-Channel Gate-Driver Output
7V Low-Dropout Voltage Regulator. Bypass to PGND with at least a 1µF low-ESR ceramic capacitor.
14
16
V
CC
V
provides power to the n-channel gate driver (NDRV).
CC
15
16
17
18
IN
Positive Power-Supply Input. Bypass to PGND with at least a 0.1µF ceramic capacitor.
High-Side Positive Supply Input Referred to LV. HV provides power to high-side linear regulator
HV
5V High-Side Regulator Output. CLV provides power to the dimming MOSFET driver. Connect a
0.1µF to 1µF ceramic capacitor from CLV to LV for stable operation.
17
18
19
19
20
1
CLV
DIMOUT External Dimming MOSFET Gate Driver. DIMOUT is capable of sinking/sourcing 50mA.
High-Side Reference Voltage Input. Connect to SGND for boost configuration. Connect to IN for
boost-buck configuration.
LV
LED Current-Sense Positive Input. Connect a bypass capacitor of at least 0.1µF between SENSE+
and LV close to the IC.
20
—
2
SENSE+
Exposed Pad. Connect EP to a large-area contiguous copper ground plane for effective power
dissipation. Do not use as the main IC ground connection. EP must be connected to SGND.
—
EP
The MAX16834 switching frequency (100kHz to 1MHz)
Detailed Description
The MAX16834 is a current-mode, high-brightness LED
(HB LED) driver designed to control a single-string LED
current regulator with two external n-channel MOSFETs.
is adjustable using a single resistor from RT/SYNC. The
MAX16834 disables the internal oscillator and synchro-
nizes if an external clock is applied to RT/SYNC. The
switching MOSFET driver sinks and sources up to 3A,
making it suitable for high-power MOSFETs driving in
HB LED applications, and the dimming control allows
for wide PWM dimming at frequencies up to 20kHz.
The MAX16834 integrates all the building blocks nec-
essary to implement a fixed-frequency HB LED driver
with wide-range dimming control. The MAX16834
allows implementation of different converter topologies
such as SEPIC, boost, boost-buck, or high-side buck
current regulator.
The MAX16834 is suitable for boost and boost-buck
LED drivers (Figures 2 and 3).
The MAX16834 alone operates over a wide 4.75V to
28V supply range. With a voltage clamp that limits the
IN pin voltage to less than 28V, it can operate in boost
configuration for input voltages greater than 28V.
Additional features include external enable/disable
input, an on-chip oscillator, fault indicator output (FLT)
for LED open/short or overtemperature conditions, and
an overvoltage protection circuit for true differential
overvoltage protection (Figure 1).
The MAX16834 features a constant-frequency, peak-cur-
rent-mode control with programmable slope compensa-
tion to control the duty cycle of the PWM controller. A
dimming driver offers a wide-range dimming control for
the external n-channel MOSFET in series with the LED
string. In addition to PWM dimming, the MAX16834
allows for analog dimming of LED current.
_______________________________________________________________________________________
7
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
IN
REF
TO
INTERNAL
CIRCUITRY
TEMPERATURE
SENSE
REFERENCE
OT
SGND
UVEN
V
CC
7V
LDO
MAX16834
UVLO
V
BG
S
R
Q
NDRV
PGND
RT/SYNC
OSC
RAMP
GENERATOR
SC
CS
PWM
COMP
0.6V
5kΩ
OR
AND
CURRENT-LIMIT
COMPARATOR
NDRVB
BLANK
NDRVB
0.3V
V
REF
FLTB FLTA
REFI
LPF
FLT
ERROR
AMPLIFIER
SENSE+
PWMDIM
OT
A
V
= 9.9
AND
AND
g
m
V
LV
LED CURRENT-
SENSE AMPLIFIERS
CLV
COMP
HV
DIMOUT
HIGH-SIDE
5V
REGULATOR
LV REFERENCE
SWITCH
V
BG
LV
V
LV
REFHI
V
IN
128 TOSC
ERROR
V
BG
4.3V
PWMDIM
REFHI
REJECT
DELAY
FLTB
4096 TOSC
HICCUP
TIMER
AND
FLTB
V
LV
V
REF
V
BG
0.3V
V
HV
5µs ERROR
REJECT
DELAY
OVP+
FLTA
SENSE+
V
BG
MAX16834
V
LV
V
LV
Figure 1. Internal Block Diagram
_______________________________________________________________________________________
8
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
The MAX16834 is also suitable for DC-DC converter
applications such as boost or boost-buck (Figures 6
and 7). Other applications include boost LED drivers
with automotive load dump protection (Figure 4) and
high-side buck LED drivers (Figure 5).
n-Channel MOSFET Switch Driver (NDRV)
The MAX16834 drives an external n-channel switching
MOSFET. NDRV swings between V
and PGND.
CC
NDRV is capable of sinking/sourcing 3A of peak current,
allowing the MAX16834 to switch MOSFETs in high-
power applications. The average current demanded
from the supply to drive the external MOSFET depends
Undervoltage Lockout/Enable
The MAX16834 features an adjustable UVLO using the
on the total gate charge (Q ) and the operating
G
enable input (UVEN). Connect UVEN to V through a
IN
frequency of the converter, f . Use the following equa-
SW
resistive divider to set the UVLO threshold. The
tion to calculate the driver supply current I
required for the switching MOSFET:
NDRV
MAX16834 is enabled when the V
exceeds the
UVEN
1.435V (typ) threshold. See the Setting the UVLO
Threshold section for more information.
I
= Q x f
G SW
NDRV
UVEN also functions as an enable/disable input to the
device. Drive UVEN low to disable the output and high
to enable the output.
Pulse Dimming Inputs (PWMDIM)
The MAX16834 offers a dimming input (PWMDIM) for
pulse-width modulating the output current. PWM dim-
ming can be achieved by driving PWMDIM with a pul-
sating voltage source. When the voltage at PWMDIM is
greater than 1.435V, the PWM dimming MOSFET turns
on and when the voltage on PWMDIM is below 1.235V,
the PWM dimming MOSFET turns off.
Reference Voltage (REF)
The MAX16834 features a 3.7V reference output, REF.
REF provides power to most of the internal circuit blocks
except for the output drivers and is capable of sourcing
1mA to external circuits. Connect a 0.1µF to 0.22µF
ceramic capacitor from REF to SGND. Connect REF to
REFI through a resistive divider to set the LED current.
High-Side Linear Regulator (V
)
CLV
The MAX16834’s 5V high-side regulator (CLV) powers
up the dimming MOSFET driver. V is measured with
CLV
Reference Input (REFI)
The output current is proportional to the voltage at
REFI. Applying an external DC voltage at REFI or using
a potentiometer from REF to SGND allows analog dim-
ming of the output current.
respect to LV and sources up to 2mA of current.
Bypass CLV to LV with a 0.1µF to 1µF low-ESR ceramic
capacitor. The maximum voltage on CLV with respect
to PGND must not exceed 28V. This limits the input volt-
age for boost-buck topology.
High-Side Reference Voltage Input (LV)
LV is a reference input. Connect LV to SGND for boost
and SEPIC topologies. Connect LV to IN for boost-buck
and high-side buck topologies.
Low-Side Linear Regulator (V
)
CC
The MAX16834’s 7V low-side linear regulator (V ) pow-
CC
ers up the switching MOSFET driver with sourcing capa-
bility of up to 50mA. Use at least a 1µF low-ESR ceramic
capacitor from V
to PGND for stable operation.
CC
Dimming Driver Regulator
Input Voltage (HV)
LED Current-Sense Input (SENSE+)
The differential voltage from SENSE+ to LV is fed to an
internal current-sense amplifier. This amplified signal is
then connected to the negative input of the transcon-
ductance error amplifier. The voltage gain factor of this
amplifier is 9.9 (typ).
The voltage at HV provides the input voltage for the
dimming driver regulator. For boost or SEPIC topology,
connect HV either to IN or to V . For boost-buck, con-
CC
nect HV to a voltage higher than IN. The voltage at HV
must not exceed 28V with respect to PGND. For the
high-side buck, connect HV to IN.
Whenever V is greater than 5V, the input impedance
LV
of the LED current-sense amplifier seen at the SENSE+
pin is 1kΩ 30%. In that condition, a bias current of
20µA ( 30%) is pulled from SENSE+, in addition to the
Dimming MOSFET Driver (DIMOUT)
The MAX16834 requires an external n-channel MOSFET
for PWM dimming. Connect the gate of the MOSFET to
the output of the dimming driver, DIMOUT, for normal
operation. The dimming driver is capable of sinking or
sourcing up to 50mA of current.
current due to the 1kΩ resistor. When V is less than
LV
1V, the amplifier input (SENSE+ pin) is in high imped-
ance and the bias current of 20µA ( 30%) is pushed
out of that pin.
Always have a bypass capacitor of at least 0.1µF value
between SENSE+ and LV and close to the IC.
_______________________________________________________________________________________
9
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
grammed by the external capacitor connected at SC.
The current source charging the capacitor is 100µA.
Internal Transconductance Error Amplifier
The MAX16834 has a built-in transconductance amplifi-
er used to amplify the error signal inside the feedback
loop. The amplified current-sense signal is connected
Overvoltage Protection (OVP+)
OVP+ sets the overvoltage threshold limit across the
LEDs. Use a resistive divider between output OVP+
and LV to set the overvoltage threshold limit. An internal
overvoltage protection comparator senses the differen-
tial voltage across OVP+ and LV. If the differential volt-
age is greater than 1.435V, NDRV is disabled and FLT
asserts. When the differential voltage drops by 200mV,
NDRV is enabled and FLT deasserts. The PWM dim-
ming MOSFET is still controlled by the PWMDIM input.
to the negative input of the g amplifier with the current
m
reference connected to REFI. The output of the op amp
is controlled by the input at PWMDIM. When the signal
at PWMDIM is high, the output of the op amp connects
to COMP; when the signal at PWMDIM is low, the out-
put of the op amp disconnects from COMP to preserve
the charge on the compensation capacitor. When the
voltage at PWMDIM goes high, the voltage on the com-
pensation capacitor forces the converter into a steady
state. COMP is connected to the negative input of the
PWM comparator with CMOS inputs, which draw very
little current from the compensation capacitor at COMP
and thus prevent discharge of the compensation
capacitor when the PWMDIM input is low.
MAX16834
Fault Indicator (FLT)
The MAX16834 features an active-low, open-drain fault
indicator (FLT). FLT asserts when one of the following
occurs:
1) Overvoltage across the LED string
2) Short-circuit condition across the LED string, or
3) Overtemperature condition
Internal Oscillator
The internal oscillator of the MAX16834 is programma-
ble from 100kHz to 1MHz using a single resistor at
RT/SYNC. Use the following formula to calculate the
switching frequency:
When the output voltage drops below the overvoltage
set point minus the hysteresis, FLT deasserts. Similarly
during the short-circuit period, the fault signal
deasserts when the dimming MOSFET is on, which
happens every hiccup cycle during short circuit. During
overtemperature fault, the FLT signal is the inverse of
the PWM input.
5000kΩ
RT(kΩ)
f
(kHz) =
× (kHz)
OSC
where RT is the resistor from RT/SYNC to SGND.
The MAX16834 synchronizes to an external clock signal
at RT/SYNC. The application of an external clock dis-
ables the internal oscillator allowing the MAX16834 to
use the external clock for switching operation. The
internal oscillator is enabled if the external clock is
absent for more than 50µs. The synchronizing pulse
minimum width for proper synchronization is 200ns.
Applications Information
Setting the UVLO Threshold
The UVLO threshold is set by resistors R1 and R2 (see
Figure 2). The MAX16834 turns on when the voltage
across R2 exceeds 1.435V, the UVLO threshold. Use
the following equation to set the desired UVLO thresh-
old:
Switching MOSFET
Current-Sense Input (CS)
V
= 1.435V(R1+ R2) R2
UVEN
CS is part of the current-mode control loop. The switch-
In a typical application, use a 10kΩ resistor for R2 and
then calculate R1 based on the desired UVLO threshold.
ing control uses the voltage on CS, set by R , to termi-
CS
nate the on pulse width of the switching cycle, thus
achieving peak current-mode control. Internal leading-
edge blanking is provided to prevent premature turn-off
of the switching MOSFET in each switching cycle.
Setting the Overvoltage Threshold
The overvoltage threshold is set by resistors R4 and R9
(see Figure 2). The overvoltage circuit in the MAX16834
is activated when the voltage on OVP+ with respect to
LV exceeds 1.435V. Use the following equation to set
the desired overvoltage threshold:
Slope Compensation (SC)
The MAX16834 uses an internal-ramp generator for
slope compensation. The ramp signal also resets at the
beginning of each cycle and slews at the rate pro-
V
= 1.435V(R4 + R9) R9
OV
10 ______________________________________________________________________________________
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
V
IN
C1
L1
R1
D1
LED+
LV
FLT
C3
Q1
IN
NDRV
CS
LEDs
UVEN
HV
R4
C2
ON
MAX16834
SC
OFF
LED-
PWMDIM
R3
C5
RT/SYNC
Q2
DIMOUT
R2
V
CC
C4
SENSE+
OVP+
CLV
REF
R6
R5
COMP
PGND
REFI
R9
R10
R8
C8
C7
R7
SGND
C6
Figure 2. Boost LED Driver
Calculate maximum duty cycle using the below equation.
+ V − V
Programming the LED Current
The LED current is programmed using the voltage on
REFI and the LED current-sense resistor R10 (see
Figure 2). The current is given by:
V
LED
D
INMIN
D
=
MAX
V
+ V − V
D FET
LED
V
× R5
REF
where V
is the forward voltage of the LED string in
LED
I
=
A
( )
LED
R10 × (R6 + R5) × 9.9
volts, V is the forward drop of the rectifier diode D1 in
D
volts (approximately 0.6V), V
supply voltage in volts, and V
is the minimum input
is the average drain to
INMIN
where V
is 3.7V and the resistors R5, R6, and R10
REF
FET
are in ohms. The regulation voltage on the LED current-
sense resistor must not exceed 0.3V to prevent activa-
tion of the LED short-circuit protection circuit.
source voltage of the MOSFET Q1 in volts when it is on.
Use an approximate value of 0.2V initially to calculate
D
. A more accurate value of the maximum duty
MAX
cycle can be calculated once the power MOSFET is
selected based on the maximum inductor current.
Boost Configuration
In the boost converter (Figure 2), the average inductor
current varies with the line voltage. The maximum aver-
age current occurs at the lowest line voltage. For the
boost converter, the average inductor current is equal
to the input current.
Use the following equations to calculate the maximum
average inductor current IL
, peak-to-peak inductor
AVG
current ripple ∆I , and the peak inductor current IL in
L
P
amperes:
______________________________________________________________________________________ 11
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
Allowing the peak-to-peak inductor ripple ∆I to be
L
I
30% of the average inductor current:
LED
IL
=
AVG
1− D
∆I = IL
× 0.3 × 2
AVG
MAX
L
∆I
2
L
Allowing the peak-to-peak inductor ripple (∆I ) to be
L
IL = IL
+
AVG
P
30% of the average inductor current:
The inductance value (L) of the inductor L1 in henries is
calculated as:
∆I = IL
× 0.3 × 2
AVG
L
and
(V
− V
) × D
× ∆I
INMIN
FET MAX
MAX16834
∆I
2
L =
L
IL = IL
+
P
AVG
f
SW L
where f
is the switching frequency in hertz, V
INMIN
The inductance value (L) of the inductor L1 in henries
(H) is calculated as:
SW
and V
are in volts, and ∆I is in amperes. Choose an
FET
L
inductor that has a minimum inductance greater than
the calculated value.
(V
− V
) × D
× ∆I
L
INMIN
FET MAX
L =
f
SW
Peak Current-Sense Resistor (R8)
The value of the switch current-sense resistor R8 for the
boost and boost-buck configurations is calculated as
follows:
where f
is the switching frequency in hertz, V
INMIN
SW
and V
are in volts, and ∆I is in amperes.
FET
L
Choose an inductor that has a minimum inductance
greater than the calculated value. The current rating of
the inductor should be higher than IL at the operating
P
temperature.
0.25
R8 =
Ω
(IL ×1.25)
P
Boost-Buck Configuration
In the boost-buck LED driver (Figure 3), the average
inductor current is equal to the input current plus the
LED current.
where 0.25V is the minimum peak current-sense thresh-
old, IL is the peak inductor current in amperes, and
P
the factor 1.25 provides a 25% margin to account for
tolerances. The worst cycle-by-cycle current limiter trig-
gers at 350mV (max). The I
be higher than 0.35V/R8.
of the inductor should
SAT
Calculate maximum duty cycle using the following
equation:
Output Capacitor
V
+ V
D
LED
The function of the output capacitor is to reduce the
output ripple to acceptable levels. The ESR, ESL, and
the bulk capacitance of the output capacitor contribute
to the output ripple. In most applications, the output
ESR and ESL effects can be dramatically reduced by
using low-ESR ceramic capacitors. To reduce the ESL
and ESR effects, connect multiple ceramic capacitors
in parallel to achieve the required bulk capacitance. To
minimize audible noise generated by the ceramic
capacitors during PWM dimming, it may be necessary
to minimize the number of ceramic capacitors on the
output. In these cases an additional electrolytic or tan-
talum capacitor provides most of the bulk capacitance.
D
=
MAX
V
+ V + V
− V
LED
D
INMIN FET
where V
is the forward voltage of the LED string in
volts, V is the forward drop of the rectifier diode D1
(approximately 0.6V) in volts, V
input supply voltage in volts, and V
drain to source voltage of the MOSFET Q1 in volts when
it is on. Use an approximate value of 0.2V initially to cal-
. A more accurate value of maximum duty
cycle can be calculated once the power MOSFET is
selected based on the maximum inductor current.
LED
D
is the minimum
is the average
INMIN
FET
culate D
MAX
Use the below equations to calculate the maximum
average inductor current IL
, peak-to-peak inductor
AVG
Boost and boost-buck configurations: The calcula-
tion of the output capacitance is the same for both
boost and boost-buck configurations. The output ripple
is caused by the ESR and the bulk capacitance of the
output capacitor if the ESL effect is considered negligi-
ble. For simplicity, assume that the contributions from
current ripple ∆I , and the peak inductor current IL in
L
P
amperes:
I
LED
IL
=
AVG
1− D
MAX
12 ______________________________________________________________________________________
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
V
IN
C1
L1
D1
R1
LED+
LV
IN
HV
Q1
NDRV
LEDs
UVEN
SC
CS
C2
R4
Q2
C3
ON
MAX16834
OFF
R3
C5
PWMDIM
LED-
RT/SYNC
DIMOUT
R2
V
CC
C4
REF
SENSE+
OVP+
R6
R5
REFI
CLV
COMP
PGND
FLT
R9
R10
R8
C8
C7
R7
SGND
C6
V
IN
Figure 3. Boost-Buck LED Driver (V
< 28V)
LED+
ESR and the bulk capacitance are equal, allowing 50%
of the ripple for the bulk capacitance. The capacitance
is given by:
Use the below equation to calculate the RMS current
rating of the output capacitor:
2
(IL
× (1 - D
)) × D
AVG
MAX
2
MAX
I
=
COUT(RMS)
I
× 2 × D
LED
MAX
+(IL
× D
)
× (1- D
)
MAX
AVG
MAX
C
≥
OUT
∆V
× f
SW
OUTRIPPLE
Input Capacitor
where I
is in amperes, C
OUTRIPPLE
is in farads, f
is in
LED
hertz, and ∆V
OUT
SW
The input filter capacitor bypasses the ripple current
drawn by the converter and reduces the amplitude of
high-frequency current conducted to the input supply.
The ESR, ESL, and the bulk capacitance of the input
capacitor contribute to the input ripple. Use a low-ESR
input capacitor that can handle the maximum input
RMS ripple current from the converter.
is in volts. The remaining 50%
of allowable ripple is for the ESR of the output capaci-
tor. Based on this, the ESR of the output capacitor is
given by:
∆V
(Ω)
OUTRIPPLE
ESR
<
COUT
(IL × 2)
P
For the boost configuration, the input current is the
same as the inductor current. For boost-buck
where IL is the peak inductor current in amperes.
P
______________________________________________________________________________________ 13
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
configuration, the input current is the inductor current
minus the LED current. But for both configurations, the
ripple current that the input filter capacitor has to sup-
ply is the same as the inductor ripple current with the
condition that the output filter capacitor should be con-
nected to ground for boost-buck configuration. This
reduces the size of the input capacitor, as the inductor
current is continuous with maximum 30% ripple.
Neglecting the effect of LED current ripple, the calcula-
tion of the input capacitor for boost as well as boost-
buck configurations is the same.
source and discharged at the beginning of each switch-
ing cycle to generate the slope compensation ramp.
The value of the slope compensation capacitor C2 is
calculated as shown below:
Boost configuration:
-6
3 × L ×100 ×10
C2 =
(V
- V
) × R8 × 2
LED INMIN
MAX16834
where C2 is in farads, L is the inductance of the induc-
tor L1 in henries, 100µA is the pullup current from SC,
Neglecting the effect of the ESL, the ESR, and the bulk
capacitance at the input contributes to the input voltage
ripple. For simplicity, assume that the contribution from
the ESR and the bulk capacitance is equal. This allows
50% of the ripple for the bulk capacitance. The capaci-
tance is given by:
V
and V
are in volts, and R8 is the switch cur-
INMIN
LED
rent-sense resistor in ohms.
Boost-buck configuration:
-6
3 × L ×100 ×10
C2 =
(V
) × R8 × 2
LED
∆I
L
C
≥
IN
4 × ∆V × f
where C2 is in farads, L is the inductance of the induc-
tor L1 in henries, 100µA is the pullup current from SC,
IN
SW
where ∆I is in amperes, C is in farads, f
is in hertz,
V
is in volts, and R8 is the switch current-sense
LED
L
IN
SW
and ∆V is in volts. The remaining 50% of allowable
resistor in ohms.
IN
ripple is for the ESR of the output capacitor. Based on
this, the ESR of the input capacitor is given by:
Selection of Power Semiconductors
Switching MOSFET
The switching MOSFET (Q1) should have a voltage rat-
ing sufficient to withstand the maximum output voltage
together with the diode drop of the rectifier diode D1
and any possible overshoot due to ringing caused by
parasitic inductances and capacitances. Use a
MOSFET with a drain-to-source voltage rating higher
than that calculated by the following equations:
∆V
IN
ESR
<
CIN
∆I × 2
L
where ∆I is in amperes, ESR
is in ohms, and ∆V
IN
L
CIN
is in volts.
Use the below equation to calculate the RMS current
rating of the input capacitor:
Boost configuration:
∆I
2 3
L
I
=
V
= V
(
+ V ×1.2
)
CIN(RMS)
DS
LED
D
where V
D
is the drain-to-source voltage in volts and
DS
Slope Compensation
V is the forward drop of the rectifier diode D1. The fac-
Slope compensation should be added to converters
with peak current-mode control operating in continuous
conduction mode with more than 50% duty cycle to
avoid current loop instability and subharmonic oscilla-
tions. The minimum amount of slope added to the peak
inductor current to stabilize the current control loop is
half of the falling slope of the inductor.
tor of 1.2 provides a 20% safety margin.
Boost-buck configuration:
V
= V
(
+ V
+ V ×1.2
)
DS
LED
INMAX
D
where V
is the drain-to-source voltage in volts and
DS
V is the forward drop of the rectifier diode D1. The fac-
D
In the MAX16834, the slope compensating ramp is
added to the current-sense signal before it is fed to the
PWM comparator. Connect a capacitor (C2 in the appli-
cation circuit) from SC to ground for slope compensa-
tion. This capacitor is charged with a 100µA current
tor of 1.2 provides a 20% safety margin.
The continuous drain current rating of the selected
MOSFET, when the case temperature is at +70°C,
should be greater than the value calculated by the fol-
14 ______________________________________________________________________________________
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
lowing equation. The MOSFET must be mounted on a
board as per manufacturer specifications to dissipate
the heat.
Rectifier Diode
Use a Schottky diode as the rectifier (D1) for fast
switching and to reduce power dissipation. The select-
ed Schottky diode must have a voltage rating 20%
above the maximum converter output voltage. The max-
The RMS current rating of the switching MOSFET Q1 is
calculated as follows for boost and boost-buck configu-
rations:
imum converter output voltage is V
in boost configu-
LED
ration and V
+ V
in boost-buck configuration.
LED
INMAX
The current rating of the diode should be greater than
I in the following equation:
⎛
2
⎞
ID
=
IL
(
× D
MAX
×1.3
)
⎜
⎝
⎟
⎠
RMS
AVG
D
I
= IL
× (1-D ) ×1.5
MAX
where ID
is the MOSFET Q1’s drain RMS current in
RMS
amperes.
D
AVG
The MOSFET Q1 will dissipate power due to both
switching losses as well as conduction losses. The con-
duction losses in the MOSFET is calculated as follows:
Dimming MOSFET
Select a dimming MOSFET (Q2) with continuous current
rating at +70°C, higher than the LED current by 30%.
The drain-to-source voltage rating of the dimming
2
MOSFET must be higher than V by 20%.
LED
P
= IL
(
× D
× R
DSON
)
COND
AVG
MAX
Feedback Compensation
The LED current control loop comprising of the switch-
ing converter, the LED current amplifier, and the error
amplifier should be compensated for stable control of
the LED current. The switching converter small-signal
transfer function has a right half-plane (RHP) zero for
both boost and boost-buck configurations as the induc-
tor current is in continuous conduction mode. The RHP
zero adds a 20dB/decade gain together with a 90°
phase lag, which is difficult to compensate. The easiest
way to avoid this zero is to roll off the loop gain to 0dB
at a frequency less than one-fifth of the RHP zero fre-
quency with a -20dB/decade slope.
where R
is the on-resistance of Q1 in ohms with
DSON
an assumed junction temperature of +100°C, P
is
COND
in watts, and IL
is in amperes.
AVG
Use the following equations to calculate the switching
losses in the MOSFET:
Boost configuration:
2
⎛
⎜
⎝
⎞
⎟
⎠
IL
× V
× C
× f
GD SW
AVG
LED
P
=
SW
2
⎛
⎝
⎞
⎠
1
1
×
+
⎜
⎟
IG
IG
OFF
ON
The worst-case RHP zero frequency (f
ed as follows:
) is calculat-
ZRHP
Boost-buck configuration:
2
⎛
⎜
⎝
⎞
IL
× (V
+ V
)
× C
× f
Boost configuration:
AVG
LED
INMAX
2
GD
SW
P
=
⎟
⎠
SW
2
V
× (1-D
2π × L ×I
)
LED
MAX
LED
f
=
ZRHP
⎛
⎝
⎞
⎠
1
1
×
+
⎜
⎟
IG
IG
OFF
ON
Boost-buck configuration:
where IG
and IG
are the gate currents of the
OFF
ON
2
MOSFET Q1 in amperes when it is turned on and
turned off, respectively, V and V are in volts,
V
× (1-D
)
MAX
LED
f
=
ZRHP
LED
INMAX
2π × L ×I
× D
MAX
LED
IL
is in amperes, f
is in hertz, and C
is the
GD
AVG
SW
gate-to-drain MOSFET capacitance in farads.
where f
is in hertz, V
is in volts, L is the induc-
ZRHP
LED
Choose a MOSFET that has a higher power rating than
that calculated by the following equation when the
MOSFET case temperature is at +70°C:
tance value of L1 in henries (H), and I
is in amperes.
LED
The switching converter small-signal transfer function
also has an output pole for both boost and boost-buck
configurations. The effective output impedance that
determines the output pole frequency together with the
output filter capacitance is calculated as:
P
(W) = P
(W) + P (W)
TOT
COND SW
______________________________________________________________________________________ 15
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
Boost configuration:
1
C7 =
(R
+ R10) × V
LED
LED
2π × R7 × f
P2
R
=
OUT
(R
+ R10) ×I
+ V
LED
LED LED
where C7 is in farads, f is in hertz, and R7 is in ohms.
P2
Boost-buck configuration:
(R
To minimize switching frequency noise, an additional
capacitor can be added in parallel with the series com-
bination of R7 and C7. The pole from this capacitor and
R7 must be a decade higher than the loop crossover
frequency.
+ R10) × V
LED
LED
R
=
OUT
(R
+ R10) ×I
× D
+ V
LED
LED
MAX LED
MAX16834
where R
is the dynamic impedance (rate of change
LED
of voltage with current) of the LED string at the operat-
ing current, R10 is the LED current-sense resistor in
Short-Circuit Protection
Boost Configuration
In the boost configuration (Figure 2), if the LED string is
shorted then the excess current flowing in the LED cur-
rent-sense resistor will cause NDRV to stop switching.
The input voltage will appear on the output capacitor,
and this causes very high peak currents to flow in the
LED current-sense resistor R10 because the dimming
MOSFET (Q2) is on. Once the voltage across the LED
current-sense resistor exceeds 300mV for more than
5µs, then the dimming MOSFET Q2 turns off and stays
off for 4096 switching clock cycles. At the same time,
NDRV is also off. The MAX16834 goes into the hiccup
mode and recovers from hiccup once the short has
been removed. The power dissipation in the dimming
MOSFET (Q2) is minimized during a short across the
LED string. During the same period, FLT only goes high
when the dimming MOSFET is on.
ohms, V
is in volts, and I
is in amperes.
LED
LED
The output pole frequency for both boost and boost-
buck configurations is calculated as follows:
1
f
=
P2
2π × C
× R
OUT
OUT
where f is in hertz, C
is the output filter capaci-
OUT
P2
tance in farads, R
is the effective output impedance
OUT
in ohms calculated above.
Compensation components R7 and C7 perform two
functions. C7 introduces a low-frequency pole that
introduces a -20dB/decade slope into the loop gain. R7
flattens the gain of the error amplifier for frequencies
above the zero formed by R7 and C7. For compensa-
tion, this zero is placed at the output pole frequency f
P2
such that it provides a -20dB/decade slope for frequen-
Boost-Buck Configuration
In the case of the boost-buck configuration (Figure 3),
once an LED string short occurs then the behavior is
different. A short across the LED string causes a high
current spike due to the external capacitors at the out-
put. The regulation loop will cause NDRV to stop
switching. This causes the voltage on HV to drop if its
voltage is derived from LED+. The voltage on CLV will
drop, and this drop is detected after 128 clock cycles.
The dimming MOSFET and the switching MOSFET will
stop switching. It stays off for 4096 clock cycles, and
the cycle repeats itself. The short across the LED string
will cause the MAX16834 to go into a hiccup mode. At
the same time the FLT signal asserts itself for 4096
clock cycles every hiccup cycle. In the case where the
HV voltage is derived from a source different than
LED+, then the LED current will stay in regulation even
during a short across the LED string. In this case, FLT
does not assert itself during the short.
cies above f for the complete loop gain.
P2
The value of R7 needed to fix the total loop gain at f
P2
such that the total loop gain crosses 0dB at
-20dB/decade at one-fifth of the RHP zero can be cal-
culated as follows:
f
× R8
ZRHP
R7 =
5 × f × (1− D
) × R10 × 9.9 × GM
COMP
P2
MAX
where R7 is the compensation resistor in ohms, f
ZRHP
and f
are in hertz, R8 is the switch current-sense
P2
resistor in ohms, R10 is the LED current-sense resistor
in ohms, factor 9.9 is the gain of the LED current ampli-
fier, and GM
is the transconductance of the error
COMP
amplifier in Siemens.
The value of C7 can be calculated as:
16 ______________________________________________________________________________________
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
V
IN
C1
L1
Q3
R1
D1
C8
LV
FLT
LED+
D2
24V
C3
Q1
IN
NDRV
CS
LEDs
UVEN
HV
R4
C2
ON
MAX16834
SC
OFF
LED-
PWMDIM
R3
C5
RT/SYNC
Q2
DIMOUT
R2
V
CC
C4
SENSE+
OVP+
CLV
REF
R6
R5
COMP
PGND
REFI
R9
R10
C9
R8
C7
R7
SGND
C6
Figure 4. Boost LED Driver with Automotive Load Dump Protection
______________________________________________________________________________________ 17
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
V
IN
LED+
C1
C3
D1
R1
L1
LV
IN
HV
MAX16834
Q1
NDRV
V
LV
LEDs
UVEN
SC
C2
CS
ON
MAX16834
R3
C5
OFF
PWMDIM
DIMOUT
R4
Q2
RT/SYNC
LED-
R2
V
CC
C4
REF
SENSE+
OVP+
R6
R5
REFI
CLV
FLT
COMP
R9
R10
R8
C8
C7
R7
SGND
PGND
C6
V
LV
V
LV
Figure 5. High-Side Buck LED Driver
18 ______________________________________________________________________________________
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
V
IN
C1
L1
V
OUT
R1
D1
FLT
LV
Q1
IN
NDRV
UVEN
C3
R4
HV
SC
C2
CS
MAX16834
V
REF
R3
C5
PWMDIM
DIMOUT
SENSE+
RT/SYNC
R2
V
CC
C4
REF
OVP+
CLV
R6
R5
COMP
PGND
REFI
C7
R10
R9
OPTIONAL
SGND
C6
R7
R8
Figure 6. Boost DC-DC Converter
______________________________________________________________________________________ 19
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
V
IN
C1
L1
R1
MAX16834
D1
HV
LV
V
OUT
Q1
IN
NDRV
UVEN
C2
C3
R4
R11
SC
CS
MAX16834
R3
C5
V
REF
RT/SYNC
PWMDIM
DIMOUT
SENSE+
R2
V
CC
C4
REF
OVP+
CLV
R6
N.C.
R5
REFI
COMP
FLT
C7
R10
R9
C6
R7
SGND
PGND
R8
V
IN
Figure 7. Boost-Buck DC-DC Converter
20 ______________________________________________________________________________________
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
b) The cathode of D1 must be connected very
close to C
Layout Recommendations
Typically, there are two sources of noise emission in a
switching power supply: high di/dt loops and high dv/dt
surfaces. For example, traces that carry the drain cur-
rent often form high di/dt loops. Similarly, the heatsink
of the MOSFET connected to the device drain presents
a dv/dt source; therefore, minimize the surface area of
the heatsink as much as is compatible with the MOS-
FET power dissipation or shield it. Keep all PCB traces
carrying switching currents as short as possible to mini-
mize current loops. Use ground planes for best results.
.
OUT
c) C
and the current-sense resistor R8 must be
OUT
connected directly to the ground plane.
4) Connect PGND and SGND to a star-point configura-
tion.
5) Keep the power traces and load connections short.
This practice is essential for high efficiency. Use
thick copper PCBs (2oz vs.1oz) to enhance full-load
efficiency.
6) Route high-speed switching nodes away from the
sensitive analog areas. Use an internal PCB layer
for the PGND and SGND plane as an EMI shield to
keep radiated noise away from the device, feed-
back dividers, and analog bypass capacitors.
Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. Use a multilayer
board whenever possible for better noise immunity and
power dissipation. Follow these guidelines for good
PCB layout:
7) To prevent discharge of the compensation capaci-
tors during the off-time of the dimming cycle,
ensure that the PCB area close to these compo-
nents has extremely low leakage. Discharge of
these capacitors due to leakage results in reduced
performance of the dimming circuitry.
1) Use a large contiguous copper plane under the
MAX16834 package. Ensure that all heat-dissipat-
ing components have adequate cooling.
2) Isolate the power components and high-current
path from the sensitive analog circuitry.
3) Keep the high-current paths short, especially at the
ground terminals. This practice is essential for sta-
ble, jitter-free operation. Keep switching loops short
such that:
a) The anode of D1 must be connected very close
to the drain of the MOSFET Q1.
______________________________________________________________________________________ 21
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
Pin Configurations
TOP VIEW
TOP VIEW
+
LV
SENSE+
OVP+
SGND
COMP
REF
1
2
3
4
5
6
7
8
9
20 DIMOUT
19 CLV
18 HV
15
14
13
12
11
PWMDIM
UVEN
10
9
HV 16
17 IN
MAX16834
CLV 17
MAX16834
16 V
CC
8
DIMOUT 18
RT/SYNC
FLT
MAX16834
15 NDRV
14 PGND
13 CS
LV
7
19
20
REFI
*EP
6
SC
SC
SENSE+
+
FLT
12 PWMDIM
11 UVEN
1
2
3
4
5
RT/SYNC 10
TSSOP
TQFN
*EP = EXPOSED PAD.
Package Information
Chip Information
For the latest package outline information and land patterns, go
PROCESS: BiCMOS–DMOS
to www.maxim-ic.com/packages.
PACKAGE TYPE PACKAGE CODE DOCUMENT NO.
20-TQFN-EP
20-TSSOP-EP
T2044-3
U20E+1
21-0139
21-0108
22 ______________________________________________________________________________________
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
Revision History
REVISION
NUMBER
REVISION
DATE
PAGES
CHANGED
DESCRIPTION
0
8/08
Initial release
—
Added TSSOP package and automotive version. Also updated Electrical
Characteristics, Pin Description, Detailed Description, and LED Current-
Sense Input (SENSE+) section, Pin Configuration and Package Information
1
2/09
1, 2, 6, 7, 8, 9, 22
1
2
3
5/09
1/10
Added automotive version of TQFN package
1, 2, 7, 9, 11,
13, 17–20
Added requirement for a capacitor on the SENSE+ pin
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 23
© 2010 Maxim Integrated Products
Maxim is a registered trademark of Maxim Integrated Products, Inc.
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