MAX16909RAUE+CFZ [MAXIM]
Switching Regulator,;
| 型号： | MAX16909RAUE+CFZ |
| 厂家： | 
MAXIM INTEGRATED PRODUCTS |
| 描述： | Switching Regulator, 开关 |
| 文件： | 总20页 (文件大小：1219K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
EVALUATION KIT AVAILABLE
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
General Description
Features
The MAX16909 is a 3A, current-mode, step-down con-
verter with an integrated high-side switch. The device is
designed to operate with input voltages from 3.5V to 36V
while using only 30FA quiescent current at no load. The
switching frequency is adjustable from 220kHz to 1MHz
by an external resistor and can be synchronized to an
external clock. The output voltage is pin selectable to
be 5V fixed or adjustable from 1V to 10V. The wide input
voltage range along with its ability to operate at high duty
cycle during undervoltage transients make the device
ideal for automotive and industrial applications.
S Wide 3.5V to 36V Input Voltage Range
S 42V Input Transients Tolerance
S High Duty Cycle During Undervoltage Transients
S 5V Fixed or 1V to 10V Adjustable Output Voltage
S Integrated 3A Internal High-Side (70mI typ)
Switch
S Fast Load-Transient Response and Current-Mode
Architecture
S Adjustable Switching Frequency (220kHz to 1MHz)
S Frequency Synchronization Input
The device operates in skip mode for reduced current
consumption in light-load applications. Protection features
include overcurrent limit, overvoltage, and thermal shut-
down with automatic recovery. The device also features
a power-good monitor to ease power-supply sequencing.
S 30µA Standby Mode Operating Current
S 5µA Typical Shutdown Current
S Overvoltage, Undervoltage, Overtemperature, and
Short-Circuit Protections
The device operates over the -40NC to +125NC automo-
tive temperature range, and is available in 16-pin TSSOP
and TQFN (5mm x 5mm) packages with exposed pads.
Ordering Information appears at end of data sheet.
Applications
Automotive
Industrial
High-Voltage Input DC-DC Converter
Point-of-Load Applications
Typical Application Circuit
VBAT
C
47µF
C
IN2
4.7µF
IN1
C
BST
0.1µF
SUP
SUPSW
BST
L1
10µH
V
EN
OUT
5V AT 3A
LX
FSYNC
V
C
100µF
OUT
OUT
D1
MAX16909
OUT
COMP
C
COMP1
2.7nF
V
BIAS
R
FOSC
C
COMP2
10pF
65kI
V
BIAS
R
COMP
47kI
FOSC
BIAS
FB
R
PGOOD
10kI
PGOOD
POWER GOOD
C
BIAS
1µF
GND
For pricing, delivery, and ordering information, please contact Maxim Direct
at 1-888-629-4642, or visit Maxim’s website at www.maximintegrated.com.
19-5777; Rev 4; 1/17
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
ABSOLUTE MAXIMUM RATINGS
SUP, SUPSW, LX, EN to GND...............................-0.3V to +42V
SUP to SUPSW.....................................................-0.3V to +0.3V
BST to GND...........................................................-0.3V to +47V
BST to LX ...............................................................-0.3V to +6V
OUT to GND..........................................................-0.3V to +12V
FOSC, COMP, BIAS, FSYNC, I.C., PGOOD,
FB to GND............................................................-0.3V to +6V
LX Continuous RMS Current ...................................................4A
Output Short-Circuit Duration....................................Continuous
Continuous Power Dissipation (T = +70NC)
A
o
TSSOP (derate 26.1mW/ C above +70NC) .......... 2088.8mW*
o
TQFN (derate 28.6mW/ C above +70NC) ............ 2285.7mW*
Operating Temperature Range........................ -40NC to +125NC
Junction Temperature .....................................................+150NC
Storage Temperature Range............................ -65NC to +150NC
Lead Temperature (soldering, 10s) ................................+300NC
o
Soldering Temperature (reflow) ..................................... +260 C
*As per the JEDEC 51 standard (multilayer board).
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional opera-
tion of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
PACKAGE THERMAL CHARACTERISTICS (Note 1)
TSSOP
TQFN
Junction-to-Ambient Thermal Resistance (B ) .......38.3NC/W
Junction-to-Ambient Thermal Resistance (B ) ..........35NC/W
JA
JA
Junction-to-Case Thermal Resistance (B ).................3NC/W
Junction-to-Case Thermal Resistance(B ) ...............2.7NC/W
JC
JC
Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer
board. For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.
ELECTRICAL CHARACTERISTICS
(V
= V
= 14V, V
= 14V, R
= 66.5kI, T = T = -40NC to +125NC, unless otherwise noted. Typical values are at
SUP
SUPSW
EN
FOSC
A
J
T
= +25NC.)
A
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
V
,
SUP
Supply Voltage Range
3.5
36
V
V
SUPSW
Load-Dump Event Supply
Voltage
V
t
I
< 1s
42
V
SUP_LD
LD
I
= 1.5A
3.5
30
mA
SUP
LOAD
Standby mode, no load, V
= 5V
60
45
OUT
OUT
Supply Current
I
FA
Standby mode, no load, V
T
= 5V,
SUP_STANDBY
30
= +25°C
A
Shutdown Supply Current
BIAS Regulator Voltage
I
V
V
V
= 0V
5
5
12
5.5
3.3
FA
V
SHDN
EN
V
= V = 6V to 36V
SUPSW
4.7
2.9
BIAS
SUP
BIAS
BIAS Undervoltage Lockout
V
rising
3.1
V
UVBIAS
BIAS Undervoltage-Lockout
Hysteresis
400
+175
15
mV
NC
NC
Thermal-Shutdown Threshold
Thermal-Shutdown Threshold
Hysteresis
OUTPUT VOLTAGE (OUT)
Output Voltage
V
V
= V normal operation
BIAS,
4.925
5
5.075
V
OUT
FB
Maxim Integrated
2
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
ELECTRICAL CHARACTERISTICS (continued)
(V
= V
= 14V, V
= 14V, R
= 66.5kI, T = T = -40NC to +125NC, unless otherwise noted. Typical values are at
FOSC A J
SUP
SUPSW
EN
T
= +25NC.)
A
PARAMETER
SYMBOL
CONDITIONS
No load, V = V
MIN
TYP
MAX
UNITS
Skip-Mode Output Voltage
V
4.925
5
5.15
V
OUT_SKIP
FB
BIAS
Adjustable Output Voltage
Range
V
FB connected to external resistive divider
1
10
V
OUT_ADJ
Load Regulation
Line Regulation
V
V
= V
= V
, 30mA < I < 3A
LOAD
0.5
0.02
1.5
%
%/V
mA
A
FB
FB
BIAS
, 6V < V
< 36V
BIAS
SUPSW
BST Input Current
LX Current Limit
Skip-Mode Threshold
I
High-side on, V
- V = 5V
2.5
6
BST_ON
BST
LX
I
(Note 2)
3.4
4.1
LX
I
300
mA
SKIP_TH
R
measured between SUPSW and LX,
ON
Power-Switch On-Resistance
R
70
150
1
mI
FA
ON
I
= 1A, V
= 5V
LX
BIAS
High-Side Switch Leakage
Current
V
= 36V, V = 0V, T = +25°C
LX A
SUP
TRANSCONDUCTANCE AMPLIFIER (COMP)
FB Input Current
I
10
nA
V
FB
FB connected to an external resistive
0.99
1.0
1.01
divider; 0°C < T < +125°C
FB Regulation Voltage
FB Line Regulation
V
FB
A
-40°C < T < +125°C
0.985
1.0
1.015
A
DV
6V < V
< 36V
0.02
%/V
FS
LINE
SUP
Transconductance (from FB to
COMP)
g
V
= 1V, V = 5V (Note 2)
BIAS
900
m
FB
Minimum On-Time
t
(Note 2)
110
98
ns
ON_MIN
f
f
= 1MHz
SW
SW
Maximum Duty Cycle
DC
%
MAX
= 220kHz
99
OSCILLATOR FREQUENCY
Oscillator Frequency
R
= 66.5kI
360
400
1
444
1
kHz
FOSC
EXTERNAL CLOCK INPUT (FSYNC)
FSYNC Input Current
T
= +25°C
FA
A
External Input Clock Acquisition
Time
t
Cycles
FSYNC
f
+
OSC
10%
External Input Clock Frequency
(Note 2)
Hz
V
External Input Clock
V
V
V
rising
falling
1.4
FSYNC_HI
FSYNC
FSYNC
High Threshold
External Input Clock
Low Threshold
V
0.4
V
FSYNC_LO
Soft-Start Time
t
8.5
ms
SS
Maxim Integrated
3
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
ELECTRICAL CHARACTERISTICS (continued)
(V
= V
= 14V, V
= 14V, R
= 66.5kI, T = T = -40NC to +125NC, unless otherwise noted. Typical values are at
SUP
SUPSW
EN
FOSC
A
J
T
= +25NC.)
A
PARAMETER
ENABLE INPUT (EN)
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
Enable Input-High Threshold
Enable Input-Low Threshold
V
2
V
V
EN_HI
V
0.9
EN_LO
Enable Threshold Voltage
Hysteresis
V
0.2
V
EN,HYS
Enable Input Current
I
T
= +25°C
1
FA
EN
A
RESET
Output Overvoltage Trip
Threshold
V
105
110
115
%V
%V
OUT_OV
FB
V
V
rising, V
= high
93
90
10
95
92.5
35
97
95
60
0.4
1
VTH_RISING
FB
FB
PGOOD
PGOOD Switching Level
FB
V
falling, V
= low
TH_FALLING
PGOOD
PGOOD Debounce
Fs
PGOOD Output Low Voltage
PGOOD Leakage Current
I
= 5mA
V
SINK
V
in regulation, T = +25NC
FA
OUT
A
Note 2: Guaranteed by design; not production tested.
Maxim Integrated
4
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
Typical Operating Characteristics
(V
= V
= V = 14V, V
= 1.8V, R1 = 80.6kΩ, R2 = 100kΩ, T = +25NC (Figure 4), unless otherwise noted.)
SUP
SUPSW
EN
OUT A
STARTUP INTO HEAVY LOAD
(1.8V/400kHz)
STARTUP INTO NO LOAD
(1.8V/400kHz)
MAX16909 toc01
MAX16909 toc02
5V/div
0.6I RESISTIVE LOAD
5V/div
V
IN
V
IN
0V
0V
1V/div
1V/div
V
V
OUT
OUT
0V
0V
2A/div
0A
5V/div
0V
I
LOAD
5V/div
0V
V
V
PGOOD
PGOOD
2ms/div
2ms/div
EFFICIENCY vs. LOAD CURRENT
EFFICIENCY vs. LOAD CURRENT
= 14V
V
= 14V
V
IN
IN
100
90
80
70
60
50
40
30
20
10
0
100
90
80
70
60
50
40
30
20
10
0
5V
3.3V
3.3V
5V
8V
1.8V
1.8V
8V
D1: B360B-13-F, DIODES
L1: DRA125-150-R,
COOPER BUSSMANN
DIODE: B360B-13-F FROM DIODES
INDUCTOR: DRA125-150-R,
COOPER BUSSMANN
0
0.5
1.0
1.5
2.0
2.5
3.0
0
0.0001
0.001
0.01
0.1
LOAD CURRENT (A)
LOAD CURRENT (A)
SWITCHING FREQUENCY vs. I
(1.8V/400kHz)
LOAD
SWITCHING FREQUENCY vs. R
FOSC
400
399
398
397
396
395
394
393
392
391
390
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
V
= 14V
0.5
IN
V
= 14V
IN
0
1.0
1.5
2.0
2.5
3.0
20 30 40 50 60 70 80 90 100 110
(kI)
I
(A)
LOAD
R
FOSC
Maxim Integrated
5
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
Typical Operating Characteristics (continued)
(V
= V
= V = 14V, V
= 1.8V, R1 = 80.6kΩ, R2 = 100kΩ, T = +25NC (Figure 4), unless otherwise noted.)
SUP
SUPSW
EN
OUT A
PWM MODE LOAD-TRANSIENT
RESPONSE (1.8V/400kHz)
SKIP-MODE LOAD-TRANSIENT
RESPONSE (1.8V/400kHz)
MAX16909 toc07
MAX16909 toc08
V
= 14V
V = 14V
IN
IN
V
V
OUT
AC-COUPLED
50mV/div
OUT
200mV/div
1A/div
AC-COUPLED
10mA/div
0A
I
LOAD
I
LOAD
0.5A
0A
100µs/div
100µs/div
FSYNC TRANSITION FROM INTERNAL
TO EXTERNAL FREQUENCY (1.8V/400kHz)
UNDERVOLTAGE PULSE (1.8V/400kHz)
MAX16909 toc09
MAX16909 toc10
f
= 440kHz
FSYNC
5V/div
0V
V
SUPSW
5V/div
SUP = 5V
RESISITIVE LOAD = 0.6I
V
FSNC
0V
2V/div
0V
V
OUT
10V/div
V
LX
20V/div
0V
V
LX
0V
5A/div
0A
I
LOAD
2µs/div
20ms/div
OUTPUT RESPONSE TO SLOW INPUT
(ILOAD = 3A)
LOAD DUMP TEST (1.8V/400kHz)
MAX16909 toc12
MAX16909 toc11
42V
10V/div
0V
V
IN
1.8V/400kHz
0.6I RESISTIVE LOAD
V
IN
2V/div
0V
10V/div
0V
V
OUT
14V
10V/div
0V
V
LX
V
2V/div
0V
OUT
5V/div
0V
V
PGOOD
2s/div
10ms/div
Maxim Integrated
6
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
Typical Operating Characteristics (continued)
(V
= V
= V = 14V, V
= 1.8V, R1 = 80.6kΩ, R2 = 100kΩ, T = +25NC (Figure 4), unless otherwise noted.)
SUP
SUPSW
EN
OUT A
SHORT CIRCUIT TO GROUND TEST
(1.8V/400kHz)
V
LOAD REGULATION (1.8V/400kHz)
OUT
MAX16909 toc13
1.85
1.84
1.83
1.82
1.81
1.80
1.79
1.78
1.77
1.76
1.75
V
= 14V
IN
1V/div
0V
V
OUT
5V/div
0V
V
PGOOD
2A/div
0A
I
LX
10ms/div
0
0.5
1.0
1.5
2.0
2.5
3.0
I
(A)
LOAD
V
OUT
vs. TEMPERATURE
(1.8V/400kHz)
V
OUT
vs. SUPPLY VOLTAGE
(1.8V/400kHz)
V
OUT
vs. SUPPLY VOLTAGE
(1.8V/400kHz)
1.90
1.88
1.86
1.84
1.82
1.80
1.78
1.76
1.74
1.72
1.70
1.85
1.84
1.83
1.82
1.81
1.80
1.79
1.78
1.77
1.76
1.75
1.85
1.84
1.83
1.82
1.81
1.80
1.79
1.78
1.77
1.76
1.75
V
= 14V
I
= 3A
I
= 0A
IN
LOAD
LOAD
I
= 0A
LOAD
I
= 3A
LOAD
-40 -25 -10
5
20 35 50 65 80 95 110 125
TEMPERATURE (°C)
0
6
12
18
24
30
36
0
6
12
18
24
30
36
SUPPLY VOLTAGE (V)
SUPPLY VOLTAGE (V)
Maxim Integrated
7
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
Typical Operating Characteristics (continued)
(V
= V
= V = 14V, V
= 1.8V, R1 = 80.6kΩ, R2 = 100kΩ, T = +25NC (Figure 4), unless otherwise noted.)
SUP
SUPSW
EN
OUT A
V
BIAS
LOAD REGULATION
(1.8V/400kHz)
I
vs. SUPPLY VOLTAGE
I
vs. TEMPERATURE
SHDN
SHDN
5.20
5.18
5.16
5.14
5.12
5.10
5.08
5.06
5.04
5.02
5.00
4.98
20
18
16
14
12
10
8
6
6
6
5
5
5
5
5
4
4
4
V
EN
V
IN
= 0V
= 14V
V
EN
= 0V
T = -40°C
A
T = 125°C
A
T = -40°C
A
T = 25°C
A
6
T = 125°C
A
4.96
4.94
4.92
4.90
4
2
T = 25°C
A
0
0
2
4
6
8
I
10 12 14 16 18 20
(mA)
3
10
17
24
31
38
45
-40 -25 -10
5
20 35 50 65 80 95 110 125
TEMPERATURE (°C)
SUPPLY VOLTAGE (V)
BIAS
DIPS AND DROP TEST
(1.8V/400kHz)
LINE TRANSIENT TEST
(1.8V/400kHz)
MAX16909 toc21
MAX16909 toc22
V
IN
10V/div
0V
10V/div
0V
V
IN
1.8V/400kHz
0.6I LOAD
0.6I RESISTIVE LOAD
2V/div
2V/div
0V
V
OUT
V
OUT
0V
10V/div
10V/div
0V
V
LX
V
LX
0V
5V/div
0V
5V/div
0V
V
PGOOD
V
PGOOD
10ms/div
10ms/div
Maxim Integrated
8
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
Pin Configurations
TOP VIEW
TOP VIEW
12
11
10
9
16 15 14 13 12 11 10
9
SUP
8
7
6
5
EN 13
I.C. 14
BST
MAX16909
MAX16909
FSYNC
GND
BIAS
15
16
EP
FOSC
EP
4
+
+
1
2
3
4
5
6
7
8
1
2
3
TQFN
(5mm × 5mm)
TSSOP
Pin Descriptions
PIN
NAME
FSYNC
FOSC
FUNCTION
TSSOP
TQFN
Synchronization Input. The device synchronizes to an external signal applied to FSYNC.
The external clock frequency must be 10% greater than the internal clock frequency for
proper operation. Connect FSYNC to GND if the internal clock is used.
1
15
Resistor-Programmable Switching-Frequency Setting Control Input. Connect a resistor
from FOSC to GND to set the switching frequency.
2
3
4
5
6
16
1
Open-Drain, Active-Low Output. PGOOD asserts when V
is below the 92.5% regula-
OUT
PGOOD
OUT
tion point. PGOOD deasserts when V
is above the 95% regulation point.
OUT
Switch Regulator Output. OUT also provides power to the internal circuitry when the out-
put voltage of the converter is set between 3V and 5V during standby mode.
2
Feedback Input. Connect an external resistive divider from OUT to FB and GND to set
the output voltage. Connect to BIAS to set the output voltage to 5V.
3
FB
Error-Amplifier Output. Connect an RC network from COMP to GND for stable operation.
See the Compensation Network section for more details.
4
COMP
Linear Regulator Output. BIAS powers up the internal circuitry. Bypass with a 1FF
capacitor to ground.
7
8
9
5
6
7
BIAS
GND
BST
Ground
High-Side Driver Supply. Connect a 0.1FF capacitor between LX and BST for proper
operation.
Maxim Integrated
9
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
Pin Descriptions (continued)
PIN
NAME
FUNCTION
TSSOP
10
TQFN
8
Voltage Supply Input. SUP powers up the internal linear regulator. Connect a minimum
SUP
LX
4.7FF capacitor to ground.
11, 12
13, 14
9, 10
11, 12
Inductor Switching Node. Connect a Schottky diode between LX and GND.
Internal High-Side Switch-Supply Input. SUPSW provides power to the internal switch.
Connect a 0.1FF decoupling capacitor and a 4.7FF ceramic capacitor to ground.
SUPSW
SUP Voltage-Compatible Enable Input. Drive EN low to disable the device. Drive EN
high to enable the device.
15
16
13
14
EN
I.C.
Internally Connected. Connect to ground for proper operation.
Exposed Pad. Connect EP to a large-area contiguous copper ground plane for effective
power dissipation. Do not use as the only IC ground connection. EP must be connected
to GND.
—
—
EP
Internal Block Diagram
OUT
COMP
PGOOD
EN
SUP
BIAS
FB
FBSW
FBOK
AON
HVLDO
SWITCH-
OVER
BST
SUPSW
EAMP
PWM
LOGIC
HSD
REF
LX
CS
SOFT-
START
GND
MAX16909
SLOPE
COMP
OSC
FSYNC FOSC
Maxim Integrated
10
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
the converter needs to maintain a well-regulated output
voltage using an input voltage that varies from 9V to
Detailed Description
18V. Additionally, the device incorporates an innovative
design for fast-loop response that further ensures good
output-voltage regulation during transients.
The MAX16909 is a constant-frequency, current-mode,
automotive buck converter with an integrated high-side
switch. The device operates with input voltages from 3.5V
to 36V and tolerates input transients from 3.5V up to 42V.
During undervoltage events, such as cold-crank condi-
tions, the internal pass device maintains 98% duty cycle.
System Enable (EN)
An enable-control input (EN) activates the device from its
low-power shutdown mode. EN is compatible with inputs
from automotive battery level down to 3.3V. The high-
voltage compatibility allows EN to be connected to SUP,
KEY/KL30, or the INH pin of a CAN transceiver.
The switching frequency is resistor programmable from
220kHz to 1MHz to allow optimization for efficiency, noise,
and board space. A synchronization input FSYNC allows
the device to synchronize to an external clock frequency.
EN turns on the internal regulator. Once V
is above
BIAS
During light-load conditions, the device enters skip mode
for high efficiency. The 5V fixed output voltage eliminates
the need for external resistors and reduces the supply
current to 30FA. See the Internal Block Diagram for more
information.
the internal lockout threshold, V
= 3.15V (typ), the
UVL
controller activates and the output voltage ramps up
within 8.5ms.
A logic-low at EN shuts down the device. During shut-
down, the internal linear regulator and gate drivers turn
off. Shutdown is the lowest power state and reduces the
quiescent current to 5FA (typ). Drive EN high to bring the
device out of shutdown.
Wide Input Voltage Range (3.5V to 36V)
The device includes two separate supply inputs, SUP
and SUPSW, specified for a wide 3.5V to 36V input
voltage range. V
provides power to the device and
SUP
Overvoltage Protection
The device includes overvoltage protection circuitry that
protects the device when there is an overvoltage condi-
tion at the output. If the output voltage increases by more
than 110% of its set voltage, the device stops switching.
The device resumes regulation once the overvoltage
condition is removed.
V
provides power to the internal switch. When
SUPSW
the device is operating with a 3.5V input supply, certain
conditions such as cold crank can cause the voltage at
SUPSW to drop below the programmed output voltage.
As such, the device operates in a high duty-cycle mode
to maintain output regulation.
Linear Regulator Output (BIAS)
The device includes a 5V linear regulator, BIAS, that
provides power to the internal circuitry. Connect a 1FF
ceramic capacitor from BIAS to GND.
Fast Load-Transient Response
Current-mode buck converters include an integrator
architecture and a load-line architecture. The integra-
tor architecture has large loop gain but slow transient
response. The load-line architecture has fast transient
response but low loop gain. The device features an inte-
grator architecture with innovative designs to improve
transient response. Thus, the device delivers high output-
voltage accuracy, plus the output can recover quickly
from a transient overshoot, which could damage other
on-board components during load transients.
External Clock Input (FSYNC)
The device synchronizes to an external clock signal
applied at FSYNC. The signal at FSYNC must have a
10% higher frequency than the internal clock frequency
for proper synchronization.
Soft-Start
The device includes an 8.5ms fixed soft-start time for up
to 500FF capacitive load with a 3A resistive load.
Overload Protection
The overload protection circuitry is triggered when the
Minimum On-Time
The device features a 110ns minimum on-time that
ensures proper operation at 1MHz switching frequency
and high differential voltage between the input and the
output. This feature is extremely beneficial in automo-
tive applications where the board space is limited and
device is in current limit and V
is below the reset
OUT
threshold. Under these conditions the device turns the
high-side FET off for 16ms and re-enters soft-start. If the
overload condition is still present, the device repeats the
cycle.
Maxim Integrated
11
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
down converter, allowing the IC to cool. The thermal
sensor turns on the IC again after the junction tempera-
ture cools by 15NC.
Skip Mode/Standby Mode
During light-load operation the device enters skip mode
operation. Skip mode turns off the majority of circuitry
and allows the output to drop below regulation voltage
before the switch is turned on again. The lower the load
current, the longer it takes for the regulator to initiate
a new cycle. Because the converter skips unneces-
sary cycles and turns off the majority of circuitry, the
converter efficiency increases. When the high-side FET
stops switching for more than 50Fs, most of the internal
Applications Information
Setting the Output Voltage
Connect FB to BIAS for a fixed 5V output voltage. To set
the output to other voltages between 1V and 10V, con-
nect a resistive divider from output (OUT) to FB to GND
(Figure 1). Calculate R
following equation:
(OUT to FB resistor) with the
circuitry, including LDO, draws power from V
(V
FB1
OUT OUT
= 3V to 5.5V), allowing current consumption from the bat-
tery to drop to only 30FA.
V
OUT
R
= R
−1
FB2
FB1
V
Overtemperature Protection
Thermal-overload protection limits the total power dis-
sipation in the device. When the junction temperature
exceeds +175NC (typ), an internal thermal sensor
shuts down the internal bias regulator and the step-
FB
where V = 1V (see the Electrical Characteristics table).
FB
Internal Oscillator
The switching frequency, f , is set by a resistor (R
)
SW
FOSC
connected from FOSC to GND. See Figure 2 to select the
correct R value for the desired switching frequency.
FOSC
V
OUT
For example, a 400kHz switching frequency is set with
= 65kI. Higher frequencies allow designs with
lower inductor values and less output capacitance.
Consequently, peak currents and I2R losses are lower
at higher switching frequencies, but core losses, gate
charge currents, and switching losses increase.
R
FOSC
R
R
FB1
FB2
MAX16909
FB
Inductor Selection
Three key inductor parameters must be specified for
operation with the device: inductance value (L), inductor
Figure 1. Adjustable Output-Voltage Setting
saturation current (I ), and DC resistance (R ). To
SAT DCR
select inductance value, the ratio of inductor peak-to-
peak AC current to DC average current (LIR) must be
selected first. A good compromise between size and loss
is a 30% peak-to-peak ripple current to average-current
ratio (LIR = 0.3). The switching frequency, input voltage,
output voltage, and selected LIR then determine the
inductor value as follows:
SWITCHING FREQUENCY vs. R
FOSC
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
V
(V
− V
)
OUT SUP
OUT
LIR
L =
V
f
I
SUP SW OUT
where V , V , and I are typical values (so that
SUP OUT OUT
efficiency is optimum for typical conditions). The switch-
ing frequency is set by R (see the Internal Oscillator
V
= 14V
IN
FOSC
section). The exact inductor value is not critical and can
be adjusted to make trade-offs among size, cost, efficien-
cy, and transient response requirements. Table 1 shows
a comparison between small and large inductor sizes.
20 30 40 50 60 70 80 90 100 110
(kI)
R
FOSC
Figure 2. Switching Frequency vs. R
FOSC
Maxim Integrated
12
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
I
has a maximum value when the input voltage
RMS
Table 1. Inductor Size Comparison
equals twice the output voltage (V
I
= 2V ), so
OUT
SUP
INDUCTOR SIZE
= I
/2.
RMS(MAX)
LOAD(MAX)
SMALLER
Lower price
LARGER
Choose an input capacitor that exhibits less than +10NC
self-heating temperature rise at the RMS input current for
optimal long-term reliability.
Smaller ripple
Higher efficiency
Smaller form factor
Larger fixed-frequency
range in skip mode
The input-voltage ripple is composed of DV (caused
Faster load response
Q
by the capacitor discharge) and DV
(caused by the
ESR
equivalent series resistance (ESR) of the capacitor). Use
low-ESR ceramic capacitors with high ripple-current
capability at the input. Assume the contribution from the
ESR and capacitor discharge equal to 50%. Calculate
the input capacitance and ESR required for a specified
input-voltage ripple using the following equations:
The inductor value must be chosen so that the maximum
inductor current does not reach the device’s minimum
current limit. The optimum operating point is usually
found between 25% and 35% ripple current. When pulse
skipping (FSYNC low and light loads), the inductor value
also determines the load-current value at which PFM/
PWM switchover occurs.
∆V
ESR
ESR
=
IN
∆I
L
I
+
OUT
Find a low-loss inductor having the lowest possible
DC resistance that fits in the allotted dimensions. Most
inductor manufacturers provide inductors in standard
values, such as 1.0FH, 1.5FH, 2.2FH, 3.3FH, etc. Also
look for nonstandard values, which can provide a bet-
ter compromise in LIR across the input voltage range. If
using a swinging inductor (where the no-load inductance
decreases linearly with increasing current), evaluate
the LIR with properly scaled inductance values. For
the selected inductance value, the actual peak-to-peak
2
where
and
(V
− V
)× V
OUT
×L
SUP
V
OUT
× f
∆I
=
L
SUP SW
I
×D(1− D)
V
OUT
OUT
C
=
and D =
IN
∆V × f
V
Q
SW
SUPSW
where I
duty cycle.
is the maximum output current, and D is the
OUT
inductor ripple current (DI ) is defined by:
INDUCTOR
V
(V
− V
)
OUT SUP
OUT
∆I
=
Output Capacitor
INDUCTOR
V
× f
×L
SUP SW
The output filter capacitor must have low enough ESR to
meet output ripple and load-transient requirements, yet
have high enough ESR to satisfy stability requirements.
The output capacitance must be high enough to absorb
the inductor energy while transitioning from full-load
to no-load conditions without tripping the overvoltage
fault protection. When using high-capacitance, low-ESR
capacitors, the filter capacitor’s ESR dominates the
output-voltage ripple. So the size of the output capaci-
tor depends on the maximum ESR required to meet the
where DI
is in A, L is in H, and f
is in Hz.
INDUCTOR
SW
Ferrite cores are often the best choices, although pow-
dered iron is inexpensive and can work well at 200kHz.
The core must be large enough not to saturate at the
peak inductor current (I
):
PEAK
+
LOAD(MAX)
∆I
INDUCTOR
2
I
= I
PEAK
Input Capacitor
output-voltage ripple (V ) specifications:
RIPPLE(P-P)
The input filter capacitor reduces peak currents drawn
from the power source and reduces noise and voltage
ripple on the input caused by the circuit’s switching.
V
= ESR × I
× LIR
RIPPLE(P-P)
LOAD(MAX)
The actual capacitance value required relates to the
physical size needed to achieve low ESR, as well as
to the chemistry of the capacitor technology. Thus, the
capacitor is usually selected by ESR and voltage rating
rather than by capacitance value.
The input capacitor RMS current requirement (I
defined by the following equation:
) is
RMS
V
(V
− V
)
OUT SUP
OUT
I
= I
RMS LOAD(MAX)
V
SUP
Maxim Integrated
13
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
When using low-capacity filter capacitors, such as
current. Current-mode control eliminates the double pole
in the feedback loop caused by the inductor and output
capacitor, resulting in a smaller phase shift and requiring
less elaborate error-amplifier compensation than voltage-
ceramic capacitors, size is usually determined by the
capacity needed to prevent voltage droop and volt-
age rise from causing problems during load transients.
Generally, once enough capacitance is added to meet
the overshoot requirement, undershoot at the rising load
edge is no longer a problem. However, low-capacity filter
capacitors typically have high-ESR zeros that can affect
the overall stability.
mode control. Only a simple single-series resistor (R )
C
and capacitor (C ) are required to have a stable, high-
C
bandwidth loop in applications where ceramic capacitors
are used for output filtering (Figure 3). For other types of
capacitors, due to the higher capacitance and ESR, the
frequency of the zero created by the capacitance and
ESR is lower than the desired closed-loop crossover fre-
quency. To stabilize a nonceramic output capacitor loop,
Rectifier Selection
The device requires an external Schottky diode recti-
fier as a freewheeling diode. Connect this rectifier close
to the device using short leads and short PCB traces.
Choose a rectifier with a voltage rating greater than the
add another compensation capacitor (C ) from COMP to
F
GND to cancel this ESR zero.
The basic regulator loop is modeled as a power modula-
tor, output feedback divider, and an error amplifier. The
maximum expected input voltage, V
. Use a low
SUPSW
forward-voltage-drop Schottky rectifier to limit the nega-
tive voltage at LX. Avoid higher than necessary reverse-
voltage Schottky rectifiers that have higher forward-
voltage drops.
power modulator has a DC gain set by g
x R
, the output
,
mc
LOAD
with a pole and zero pair set by R
capacitor (C
LOAD
), and its ESR. The following equations
OUT
allow to approximate the value for the gain of the power
modulator (GAIN ), neglecting the effect of the
Compensation Network
The device uses an internal transconductance error
amplifier with its inverting input and its output available
to the user for external frequency compensation. The
output capacitor and compensation network determine
the loop stability. The inductor and the output capaci-
tor are chosen based on performance, size, and cost.
Additionally, the compensation network optimizes the
control-loop stability.
MOD(DC)
ramp stabilization. Ramp stabilization is necessary when
the duty cycle is above 50% and is internally done for
the device.
GAIN
= g
× R
mc LOAD
MOD(DC)
where R
= V
/I
in I, f
is the switch-
SW
LOAD
OUT LOUT(MAX)
ing frequency in MHz, L is the output inductance in H,
and g = 3S.
mc
The controller uses a current-mode control scheme that
regulates the output voltage by forcing the required current
through the external inductor. The device uses the volt-
age drop across the high-side MOSFET to sense inductor
In a current-mode step-down converter, the output
capacitor, its ESR, and the load resistance introduce a
pole at the following frequency:
1
× R
(
f
=
pMOD
2π × C
+ ESR
LOAD
)
V
OUT
OUT
The output capacitor and its ESR also introduce a zero at:
1
R
R
1
2
f
=
zMOD
COMP
2π ×ESR× C
OUT
g
m
V
When C
in parallel, the resulting C
ESR = ESR
is composed of “n” identical capacitors
REF
OUT
R
C
C
F
= n x C
and
OUT
OUT(EACH)
/n. Note that the capacitor zero for a
(EACH)
C
C
parallel combination of alike capacitors is the same as
for an individual capacitor.
Figure 3. Compensation Network
Maxim Integrated
14
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
The feedback voltage-divider has a gain of GAIN
=
Set the error-amplifier compensation zero formed by R
FB
C
C
V
/V
, where V is 1V (typ).
and C (f
as follows:
) at the f
. Calculate the value of C
FB OUT
FB
C
zEA
pMOD
The transconductance error amplifier has a DC gain of
GAIN = g x R , where g is the error-
1
C
=
EA(DC)
m,EA
OUT,EA
m,EA
C
2π × f
×R
C
pMOD
amplifier transconductance, which is 900FS (typ), and
is the output resistance of the error amplifier.
R
OUT,EA
If f
is less than 5 x f , add a second capacitor,
C
zMOD
A dominant pole (f
) is set by the compensa-
dpEA
C , from COMP to GND and set the compensation pole
F
tion capacitor (C ) and the amplifier output resistance
C
formed by R and C (f
) at the f
pEA
. Calculate the
zMOD
C
F
(R
OUT,EA
). A zero (f
) is set by the compensation
zEA
value of C as follows:
F
resistor (R ) and the compensation capacitor (C ).
C
C
1
C
=
There is an optional pole (f
) set by C and R to
F
pEA
F
C
2π × f
×R
zMOD
C
cancel the output capacitor ESR zero if it occurs near
the crossover frequency (f , where the loop gain equals
C
As the load current decreases, the modulator pole
also decreases; however, the modulator gain increases
accordingly and the crossover frequency remains the
same.
1 (0dB)). Thus:
1
f
=
dpEA
2π × C × (R
+ R )
C
C
OUT,EA
For the case where f
is less than f :
C
zMOD
1
f
f
=
=
zEA
The power-modulator gain at f is:
C
2π × C ×R
C
C
f
pMOD
GAIN
= GAIN
×
MOD(dc)
MOD(fC)
1
f
zMOD
pEA
2π × C ×R
F
C
The error-amplifier gain at f is:
C
The loop-gain crossover frequency (f ) should be set
C
below 1/5th of the switching frequency and much higher
than the power-modulator pole (f
f
zMOD
GAIN
= g
×R ×
C
EA(fC)
m,EA
f
):
C
pMOD
Therefore:
GAIN
f
SW
5
f
<< f ≤
C
pMOD
V
f
FB
zMOD
×
×g
×R
×
=1
MOD(fC)
m,EA
C
V
f
The total loop gain as the product of the modulator gain,
the feedback voltage-divider gain, and the error-amplifier
gain at f should be equal to 1. So:
OUT
C
Solving for R :
C
C
V
× f
C
OUT
V
R
=
FB
C
GAIN
×
× GAIN
=1
EA(fC)
g
× V × GAIN × f
MOD(fC) zMOD
MOD(fC)
m,EA
FB
V
OUT
Set the error-amplifier compensation zero formed by R
C
For the case where f
is greater than f :
C
zMOD
and C at the f
(f
= f
) as follows:
C
pMOD zEA
pMOD
1
GAIN
= g × R
m,EA C
EA(fC)
C
=
f
C
pMOD
2π × f
×R
C
GAIN
= GAIN
×
MOD(dc)
pMOD
MOD(fC)
f
C
If f
is less than 5 x f , add a second capacitor C
C
zMOD
F
F
Therefore:
GAIN
from COMP to GND. Set f
as follows:
= f
and calculate C
pEA
zMOD
V
FB
×
×g
×R =1
m,EA C
MOD(fC)
V
OUT
1
C
=
F
2π × f
×R
C
Solving for R :
zMOD
C
V
OUT
R
=
C
g
× V × GAIN
FB MOD(fC)
m,EA
Maxim Integrated
15
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
VBAT
C
3
4.7µF
C
47µF
C
4.7µF
1
2
C
4
C
5
0.1µF
0.1µF
C
BST
0.1µF
SUP
SUPSW
BST
L1
15µH
V
EN
OUT
1.8V AT 3A
LX
FSYNC
COMP
V
OUT
C
OUT
D1
100µF
OUT
MAX16909
C
COMP1
821pF
R
R
1
FOSC
C
COMP2
12pF
80.6kI
62kI
R
COMP
9.1kI
FOSC
BIAS
FB
V
BIAS
R
2
100kI
R
PGOOD
10kI
PGOOD
POWER GOOD
C
BIAS
1µF
GND
Figure 4. 1.8V/3A Configuration
3) Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
jitter-free operation. The high-current path composed
of input capacitor, high-side FET, inductor, and output
capacitor should be as short as possible.
PCB Layout Guidelines
Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. Use a multilayer
board whenever possible for better noise immunity and
power dissipation. Follow these guidelines for good PCB
layout:
4) Keep the power traces and load connections short.
This practice is essential for high efficiency. Use
thick copper PCBs (2oz vs. 1oz) to enhance full-load
efficiency.
1) Use a large contiguous copper plane under the IC
package. Ensure that all heat-dissipating components
have adequate cooling. The bottom pad of the device
must be soldered down to this copper plane for effec-
tive heat dissipation and for getting the full power out
of the IC. Use multiple vias or a single large via in this
plane for heat dissipation.
5) The analog signal lines should be routed away from
the high-frequency planes. This ensures integrity of
sensitive signals feeding back into the IC.
6) The ground connection for the analog and power
section should be close to the IC. This keeps the
ground current loops to a minimum. In cases where
only one ground is used, enough isolation between
analog return signals and high-power signals must
be maintained.
2) Isolate the power components and high-current path
from the sensitive analog circuitry. This is essential to
prevent any noise coupling into the analog signals.
Maxim Integrated
16
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
Package Information
Ordering Information
For the latest package outline information and land patterns (foot-
prints), go to www.maximintegrated.com/packages. Note that a
“+”, “#”, or “-” in the package code indicates RoHS status only.
Package drawings may show a different suffix character, but the
drawing pertains to the package regardless of RoHS status.
PART
TEMP RANGE
PIN-PACKAGE
MAX16909RAUE+
MAX16909RAUE/V+
MAX16909RATE+
MAX16909RATE/V+
-40NC to +125NC 16 TSSOP-EP*
-40NC to +125NC 16 TSSOP-EP*
-40NC to +125NC 16 TQFN-EP*
-40NC to +125NC 16 TQFN-EP*
PACKAGE
TYPE
PACKAGE
CODE
OUTLINE
NO.
LAND
PATTERN NO.
/V denotes an automotive qualified part.
+Denotes a lead(Pb)-free/RoHS-compliant package.
*EP = Exposed pad.
16 TSSOP-EP
16 TQFN-EP
U16E+3
T1655+4
21-0108
21-0140
90-0120
90-0121
Chip Information
PROCESS: BiCMOS
Maxim Integrated
17
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
Revision History
REVISION REVISION
PAGES
CHANGED
DESCRIPTION
NUMBER
DATE
0
3/11
Initial release
Changed R
—
= 120kI to R
=66.5kI in the Electrical Characteristics
FOSC
FOSC
globals and table; changed the min, typ, max values for R
from 190kHz
FOSC
(min), 220kHz (typ), 250kHz (max) to 360kHz (min), 400kHz (typ), 444kHz (max);
2, 3, 4, 11, 14,
17
1
9/11
changed the minimum on-time (t ) from 80ns (typ) to 110ns (typ); updated the
ON_MIN
GAIN
MOD(DC)
and f equations; removed future status from the 16-pin TQFN
pMOD
package in the Ordering Information table
2
3
4
8/12
10/14
1/17
Added two new OPNs in the Ordering Information table
Changed BIAS Regulator Voltage max limit in Electrical Characteristics
Changed Industrial/Military to Industrial in Applications section
17
2
1
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent
licenses are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and
max limits) shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated 160 Rio Robles, San Jose, CA 95134 USA 1-408-601-1000
18
©
2017 Maxim Integrated Products, Inc.
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
Maxim Integrated
19
MAX16909
36V, 220kHz to 1MHz Step-Down Converter
with Low Operating Current
Maxim Integrated
20
相关型号：
MAX16910CASA
200mA, Automotive, Ultra-Low Quiescent Current, Linear Regulator Automotive Qualified
MAXIM
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