MAX16929BGUIV [MAXIM]
Automotive TFT-LCD Power Supply with Boost Converter and Gate Voltage Regulators; 车载TFT -LCD电源与升压转换器和栅极电压稳压器型号: | MAX16929BGUIV |
厂家: | MAXIM INTEGRATED PRODUCTS |
描述: | Automotive TFT-LCD Power Supply with Boost Converter and Gate Voltage Regulators |
文件: | 总25页 (文件大小:2319K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
19-5857; Rev 0; 5/11
E V A L U A T I O N K I T A V A I L A B L E
General Description
Features
The MAX16929 is a highly integrated power supply for
automotive TFT-LCD applications. The device integrates
one buck converter, one boost converter, one 1.8V/3.3V
regulator controller, and two gate voltage regulators.
The device comes in several versions to satisfy com-
mon automotive power-supply requirements (see the
Ordering Information/Selector Guide table).
S Operating Voltage Range of 4V to 28V (Buck) or
3V to 5.5V (Boost)
S Independent 28V Input Buck Converter Powers
TFT Bias Supply Circuitry and External Circuitry
S High-Power (Up to 6W) Boost Output Providing Up
to 18V
S 1.8V or 3.3V Regulator Provides 500mA with
Designed to operate from a single 4V to 28V supply or
5.5V to 28V supply, the device is ideal for automotive
TFT-LCD applications.
External npn Transistor
S One Positive-Gate Voltage Regulator Capable of
Delivering 20mA at 28V
Both the buck and boost converters use spread-spec-
trum modulation to reduce peak interference and to opti-
mize EMI performance.
S One Negative-Gate Voltage Regulator
S High-Frequency Operation
2.1MHz (Buck Converter)
2.2MHz (Boost Converter)
The sequencing input (SEQ) allows flexible sequencing
of the positive-gate and negative-gate voltage regulators.
The power-good indicator (PGOOD) indicates a failure
on any of the converters or regulator outputs. Integrated
thermal shutdown circuitry protects the device from over-
heating.
S Flexible Stand-Alone Sequencing
S True Shutdown™ Boost Converter
S 6µA Low-Current Shutdown Mode (Buck)
S Internal Soft-Start
The MAX16929 is available in a 28-pin TSSOP pack-
age with exposed pad, and operates over the -40NC to
+105NC temperature range.
S Overtemperature Shutdown
S -40NC to +105NC Operation
Applications
Ordering Information/Selector Guide appears at end of data
sheet.
Automotive Dashboards
Automotive Central Information Displays
Automotive Navigation Systems
Typical Application Circuit appears at end of data sheet.
True Shutdown is a trademark of Maxim Integrated Products, Inc.
For related parts and recommended products to use with this part, refer to: www.maxim-ic.com/MAX16929.related
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1
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
ABSOLUTE MAXIMUM RATINGS
INB, ENB to GND..................................................-0.3V to +42V
BST to GND...........................................................-0.3V to +47V
BST to LXB ..............................................................-0.3V to +6V
LXB to GND..............................................................-6V to +42V
AVL, PGOOD to GND .............................................-0.3V to +6V
FBB, ENB to GND .................................................-0.5V to +12V
CP, GH to GND.....................................................-0.3V to +31V
ENP, DR, FB, GATE, COMPI, FBGH,
FBGL, REF, SEQ to GND .....................-0.3V to (V
GND to PGNDP....................................................-0.3V to +0.3V
+ 0.3V)
INA
Continuous Power Dissipation (T = +70NC)
A
TSSOP (derate 27mW/NC above +70NC)...................2162mW
Operating Temperature Range........................ -40NC to +105NC
Junction Temperature Range........................... -40NC to +150NC
Storage Temperature Range............................ -65NC to +150NC
Lead Temperature (soldering, 10s) ................................+300NC
Soldering Temperature (reflow) ......................................+260NC
CP, GH to GND (V
= 3.3V) ..............................-0.3V to +29V
INA
LXP to GND...........................................................-0.3V to +20V
DRVN to GND........................................................-25V to +0.3V
INA, COMPV, FBP to GND......................................-0.3V to +6V
PACKAGE THERMAL CHARACTERISTICS (Note 1)
TSSOP
Junction-to-Ambient Thermal Resistance (B ) ..........37NC/W
JA
Junction-to-Case Thermal Resistance (B ).................2NC/W
JC
Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-
layer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial.
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional opera-
tion of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(V
= 12V, V
= 5V, V
= V
= 0V, T = T = -40NC to +105NC, typical values are at T = +25NC, unless otherwise noted.)
INB
INA
GND
PGNDP A J A
(Note 2)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
BUCK CONVERTER
V
= 5V (Note 3)
5.5
4
28
28
42
9
OUTB
Supply Voltage Range
V
V
= 3.3V (Note 3)
V
INB
OUTB
t < 500ms
V
V
= 0V
6
ENB
Supply Current
I
FA
INB
= V , no load, T = +25NC
70
ENB
INB
A
Undervoltage Lockout
V
AVL rising
3.1
0.5
2.1
3.5
2.3
V
V
INB,UVLO
Undervoltage Lockout Hysteresis
PWM Switching Frequency
Spread-Spectrum Range
f
1.9
MHz
%
SWB
SSR
+6
5
5V, continuous mode
5V, skip mode
P18V,
-3%
-3%
-3%
-3%
+3%
+6%
+3%
+6%
400
5
6V PV
INB
Output-Voltage Accuracy
V
V
OUTB
I
< full load
3.3V, continuous mode
3.3V, skip mode
3.3
3.3
180
LOAD
High-Side DMOS R
R
I = 1A
mI
DS_ON
DS_ON(LXB) LXB
Skip-Current Threshold
I
16
2
%I
SKIP
MAX
I
I
= 1.2A option
= 2.0A option
1.6
2.7
2.4
OUTB
OUTB
Current-Limit Threshold
I
A
MAX
3.4
4.08
����������������������������������������������������������������� Maxim Integrated Products
2
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
ELECTRICAL CHARACTERISTICS (continued)
(V
= 12V, V
= 5V, V
= V
= 0V, T = T = -40NC to +105NC, typical values are at T = +25NC, unless otherwise noted.)
INB
INA
GND
PGNDP A J A
(Note 2)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
3.9
80
MAX
UNITS
ms
%
Soft-Start Ramp Time
Maximum Duty Cycle
Minimum Duty Cycle
Continuous mode
Continuous mode
Dropout
20
%
Maximum Duty Cycle in Dropout
Thermal Shutdown Temperature
Thermal Shutdown Hysteresis
POWER GOOD (PGOOD)
95
%
+175
15
NC
NC
Rising
Falling
94
92
13
PGOOD Threshold
%
90
95
PGOOD Debounce Time
Output High-Leakage Current
Output Low Level
Fs
FA
V
0.2
0.4
LOGIC LEVELS
ENB Threshold
ENB rising
1.4
3
1.8
0.2
5
2.2
9
V
V
ENB Hysteresis
ENB Input Current
FA
BOOST, POSITIVE (GH), NEGATIVE (GL), 1.8V/3.3V CONVERTERS
INA Input Supply Range
V
3
5.5
2.0
V
INA
V
= V
= 1.3V, V
= 0V,
FBP
FBGH
FBGL
INA Supply Current
I
1.5
2.7
mA
INA
LXP not switching
V rising, hysteresis = 200mV,
INA
INA Undervoltage Lockout
Threshold
V
2.5
2.9
V
INA,UVLO
T
= +25NC
A
INA Shutdown Current
I
V
= 0V, T = +25NC
0.5
+165
15
FA
NC
NC
SHDN
ENP
A
Thermal Shutdown Temperature
Thermal Shutdown Hysteresis
T
Temperature rising
SHDN
T
H
V
old
, V
, or V
below its thresh-
FBP FBGH
FBGL
Duration to Trigger Fault Condition
238
1.9
ms
s
Autoretry Time
REFERENCE (REF)
REF Output Voltage
REF Load Regulation
V
No output current
0 < I < 80FA, REF sourcing
1.236
-2
1.25
1.264
+2
V
REF
%
REF
REF Undervoltage Lockout
Threshold
Rising edge, hysteresis = 200mV
1.165
V
OSCILLATOR
Internal Oscillator Frequency
f
T
= +25NC
A
3.96
4.40
4.84
MHz
MHz
OSC
Spread-Spectrum Modulation
Frequency
f
f
/2
OSC
SS
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3
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
ELECTRICAL CHARACTERISTICS (continued)
(V
= 12V, V
= 5V, V
= V
= 0V, T = T = -40NC to +105NC, typical values are at T = +25NC, unless otherwise noted.)
INB
INA
GND
PGNDP A J A
(Note 2)
PARAMETER
SYMBOL
SSR
CONDITIONS
As a percentage of switching frequency,
MIN
TYP
MAX
UNITS
Spread-Spectrum Factor
Q4
%
f
SW
BOOST CONVERTER
Switching Frequency
Maximum Duty Cycle
f
1.98
82
2.20
2.42
93.5
MHz
%
SW
Low boost current-
limit option
0.625
1.25
0.78
1.56
Duty cycle = 70%,
= 220pF
LXP Current Limit
I
A
LIM
C
COMPI
High boost current-
limit option
1.87
LXP On-Resistance
LXP Leakage Current
Soft-Start Time
R
I
= 200mA
110
8.5
30
250
20
mI
FA
ms
V
DS_ON(LXP) LXP
I
V
= 20V, T =+25NC
LK_LXP
LXP A
(Note 4)
Output Voltage Range
V
V
18
SH
INA
T
= +3V to +5.5V,
= +25NC
0.985
0.98
0.74
1.0
1.0
0.85
-1
1.015
1.02
0.96
V
0 < I
A
INA
FBP Regulation Voltage
V
V
FBP
< full load
T =-40NCto+105NC
LOAD
A
PGOOD Threshold
V
Measured at FBP
0 < I < full load
V
%
PG_FBP
FBP Load Regulation
FBP Line Regulation
FBP Input Bias Current
LOAD
V
V
= +3V to +5.5V
0.1
%/V
FA
FS
INA
= +1V, T = +25NC
Q1
FBP
A
FBP to COMPV Transconductance
POSITIVE-GATE VOLTAGE REGULATOR (GH)
DI = Q2.5FA at COMPV, T = +25NC
400
A
With external charge pump, T = +25NC
A
Output Voltage Range
V
5
29
V
GH
(maximum V = 29.5V)
CP
CP Overvoltage Threshold
FBGH Regulation Voltage
PGOOD Threshold
T
= +25NC (Note 6)
29.5
0.98
0.83
30.5
1.0
0.85
2
V
V
A
V
I
= 1mA
1.034
0.87
FBGH
GH
V
Measured at FBGH
V
PG_FBGH
FBGH Load Regulation
I
= 0 to 20mA
%
GH
V
= 12V to 20V at V
= 10mA
= 10V,
GH
CP
FBGH Line Regulation
2
%
I
GH
FBGH Input Bias Current
GH Output Current
GH Current Limit
V
V
= 1V, T = +25NC
Q1
FA
mA
mA
ms
FBGH
A
I
- V = 2V
GH
20
35
GH
CP
I
56
LIM_GH
GH Soft-Start Time
7.45
NEGATIVE-GATE VOLTAGE REGULATOR (GL)
Output Voltage Range
V
-24
-2
V
V
DRVN
FBGL Regulation Voltage
V
I
= 100FA
0.212
0.242
0.271
FBGL
DRVN
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4
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
ELECTRICAL CHARACTERISTICS (continued)
(V
= 12V, V
= 5V, V
= V
= 0V, T = T = -40NC to +105NC, typical values are at T = +25NC, unless otherwise noted.)
INB
INA
GND
PGNDP A J A
(Note 2)
PARAMETER
SYMBOL
CONDITIONS
Measured at FBGL
MIN
TYP
MAX
0.42
Q1
UNITS
V
PGOOD Threshold
V
0.38
0.4
PG_FBGL
FBGL Input Bias Current
DRVN Source Current
V
= +0.25V
FA
FBGL
FBGL
V
= +0.5V, V
= -10V
2
mA
mA
ms
DRVN
DRVN Source Current Limit
GL Soft-Start Time
2.5
4
7.45
1.8V/3.3V REGULATOR CONTROLLER
3.3V regulator option
1.8V regulator option
3.18
3.3
1.8
3.38
Output Voltage
V
V
= V
FB
V
V
FB
DR
1.746
1.854
3.3V regulator option,
FB rising
2.4
2.57
2.7
Measured at FB
(Notes 5, 7)
FB PGOOD Threshold
V
PG_FB
1.8V regulator option
1.364
1.38
2.5
4.5
6
1.396
V
V
V
= 1.8V
= 3.3V
= 1.8V
FB
FB
FB
FB Input Bias Current
FA
DR Drive Current
4.5
33
mA
INPUT SERIES SWITCH CONTROL
p-Channel FET GATE Sink Current
V
= 0.5V
55
75
FA
GATE
Measured at GATE; below this voltage, the
external p-channel FET is considered on
GATE Voltage Threshold
1.25
V
DIGITAL LOGIC
ENP, SEQ Input Pulldown Resistor
Value
R
V
500
kI
PD
ENP, SEQ Input-Voltage Low
ENP, SEQ Input-Voltage High
PGOOD Leakage Current
V
0.3 x V
V
V
IL
INA
0.7 x V
IH
INA
I
T
= +25NC
A
Q1
FA
V
LK_IN
PGOOD Output-Voltage Low
V
2mA sink current, T = +25NC
0.4
OL
A
Note 2: Specifications over temperature are guaranteed by design and not production tested.
Note 3: Operation in light-load conditions or at extreme duty cycles result in skipped cycles, resulting in lower operating frequency
and possibly limited output accuracy and load response.
Note 4: 50% of the soft-start voltage time is due to the soft-start ramp, and the other 50% is due to the settling of the output voltage.
Note 5: Guaranteed by design; not production tested.
Note 6: After the voltage at CP exceeds this overvoltage threshold, the entire circuit switches off and autoretry is started.
Note 7: FB power good is indicated by PGOOD. The condition V < V
does not shutdown/restart the device.
FB
PG_FB
����������������������������������������������������������������� Maxim Integrated Products
5
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
Typical Operating Characteristics
(V
= 5V, V
= 12V, measurements taken on “A” version, unless otherwise noted; V = 12V, V
= 18V, V = -6V, V
= 3.3V,
REG
INA
INB
A
SH
GH
GL
V
= 5V, T = +25NC, unless otherwise noted.)
OUTB
SHUTDOWN SUPPLY CURRENT (BUCK)
EFFICIENCY vs. LOAD CURRENT (BUCK)
20
100
90
80
70
60
50
40
30
20
10
0
18
16
14
12
10
8
V
=12V
INB
V
=18V
INB
V
= 28V
INB
6
4
2
0
4
8
12
16
20
24
28
0
0.4
0.8
1.2
1.6
2.0
SUPPLY VOLTAGE (V)
LOAD CURRENT (A)
LINE REGULATION (BUCK)
LOAD REGULATION (BUCK)
4.0
3.2
6
5
4
I
= 0A
OUTB
2.4
3
1.6
V
= 12V
INB
2
0.8
I
= 1A
OUTB
1
0
0
-1
-2
-3
-4
-5
-6
V
= 18V
INB
-0.8
-1.6
-2.4
-3.2
-4.0
V
= 28V
INB
I
= 2A
OUTB
4
6
8
10 12 14 16 18 20 22 24 26 28
INPUT VOLTAGE (V)
0
0.4
0.8
1.2
1.6
2.0
LOAD CURRENT (A)
STARTUP WAVEFORMS (BUCK)
LOAD-TRANSIENT RESPONSE (BUCK)
MAX16929 toc06
MAX16929 toc05
V
ENB
1.8A
0.2A
5V/div
I
OUTB
1A/div
I
LXB
2A/div
V
OUTB
V
OUTB
2V/div
(AC-COUPLED)
100mV/div
1ms/div
400µs/div
����������������������������������������������������������������� Maxim Integrated Products
6
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
Typical Operating Characteristics (continued)
(V
= 5V, V
= 12V, measurements taken on “A” version, unless otherwise noted; V = 12V, V
= 18V, V = -6V, V = 3.3V,
REG
INA
INB
A
SH
GH
GL
V
= 5V, T = +25NC, unless otherwise noted.)
OUTB
LINE-TRANSIENT RESPONSE (BUCK)
SHORT-CIRCUIT BEHAVIOR (BUCK)
MAX16929 toc08
MAX16929 toc07
V
OUTB
28V
12V
5V/div
V
INB
10V/div
I
LXB
2A/div
V
OUTB
V
PGOOD
(AC-COUPLED)
50mV/div
5V/div
1ms/div
100ms/div
LOAD DUMP RESPONSE
INA SHUTDOWN SUPPLY CURRENT
MAX16929 toc09
10
9
8
7
6
5
4
3
2
1
0
42V
12V
V
INB
20V/div
I
LXB
2A/div
V
OUTB
5V/div
100ms/div
3.0
3.5
4.0
4.5
5.0
5.5
INPUT VOLTAGE (V)
EFFICIENCY vs. LOAD CURRENT
(BOOST)
LOAD REGULATION (BOOST)
LINE REGULATION (BOOST)
100
90
80
70
60
50
40
30
20
10
0
1.0
1.0
0.8
I
= 0mA
LOAD
0.8
0.6
0.6
V
= 5V
INA
0.4
0.2
0.4
V
= 3.3V
INA
0.2
V
= 3.3V
INA
0
0
-0.2
-0.4
-0.6
-0.8
-1.0
-0.2
-0.4
-0.6
-0.8
-1.0
V
= 5V
INA
0
100
200
300
400
500
0
100
200
300
400
500
3.0
3.5
4.0
4.5
5.0
5.5
LOAD CURRENT (mA)
LOAD CURRENT (mA)
INPUT VOLTAGE (V)
����������������������������������������������������������������� Maxim Integrated Products
7
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
Typical Operating Characteristics (continued)
(V
= 5V, V
= 12V, measurements taken on “A” version, unless otherwise noted; V = 12V, V
= 18V, V = -6V, V
= 3.3V,
REG
INA
INB
A
SH
GH
GL
V
= 5V, T = +25NC, unless otherwise noted.)
OUTB
STARTUP WAVEFORMS (BOOST)
LOAD-TRANSIENT RESPONSE (BOOST)
MAX16929 toc14
MAX16929 toc15
V
ENP
5V/div
450mA
50mA
V
LXP
I
SH
10V/div
500mA/div
I
LXP
1A/div
V
SH
100mV/div
V
SH
10V/div
4ms/div
100µs/div
SUPPLY SEQUENCING WAVEFORMS
SUPPLY SEQUENCING WAVEFORMS
(V
SEQ
= 0V)
(V
SEQ
= V
)
INA
MAX16929 toc16
MAX16929 toc17
V
V
ENP
ENP
5V/div
5V/div
V
V
GH
GH
5V/div
5V/div
V
V
SH
SH
5V/div
5V/div
V
REG
V
REG
5V/div
5V/div
V
V
GL
GL
5V/div
5V/div
10ms/div
10ms/div
LOAD REGULATION (GH REGULATOR)
LINE REGULATION (GH REGULATOR)
0
-0.4
-0.8
-1.2
-1.6
-2.0
-2.4
-2.6
-3.2
-3.6
-4.0
1.0
0.8
0.6
0.4
I
= 10mA
LOAD
0.2
0
-0.2
-0.4
-0.6
-0.8
-1.0
I
= 20mA
LOAD
0
2
4
6
8
10 12 14 16 18 20
18 19 20 21 22 23 24 25 26 27 28 29 30
VOLTAGE (V)
LOAD CURRENT (mA)
V
CP
����������������������������������������������������������������� Maxim Integrated Products
8
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
Typical Operating Characteristics (continued)
(V
= 5V, V
= 12V, measurements taken on “A” version, unless otherwise noted; V = 12V, V
= 18V, V = -6V, V
= 3.3V,
REG
INA
INB
A
SH
GH
GL
V
= 5V, T = +25NC, unless otherwise noted.)
OUTB
LINE REGULATION
(GL REGULATOR)
LOAD REGULATION
(GL REGULATOR)
1.0
0.8
3.0
2.7
2.4
2.1
1.8
1.5
1.2
0.9
0.6
0.3
0
0.6
0.4
0.2
I
= 20mA
= 10mA
LOAD
0
-0.2
-0.4
-0.6
-0.8
-1.0
I
LOAD
-24 -22 -20 -18 -16 -14 -12 -10 -8 -6
VOLTAGE (V)
0
2
4
6
8
10 12 14 16 18 20
V
LOAD CURRENT (mA)
CN
LOAD REGULATION
(3.3V LINEAR REGULATOR)
LOAD-TRANSIENT RESPONSE
(3.3V LINEAR REGULATOR)
MAX16929 toc23
0
-0.02
-0.04
-0.06
-0.08
-0.10
-0.12
-0.14
-0.16
-0.18
-0.20
450mA
50mA
I
OUTB
500mA/div
V
REG
(AC-COUPLED)
100mV/div
0
50 100 150 200 250 300 350 400 450 500
LOAD CURRENT (mA)
100µs/div
����������������������������������������������������������������� Maxim Integrated Products
9
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
Pin Configuration
TOP VIEW
1
2
28
27
26
25
24
23
22
21
20
19
18
17
16
15
ENP
DR
SEQ
REF
3
FB
FBGL
FBGH
COMPI
GND
DRVN
GH
4
GATE
PGNDP
LXP
5
MAX16929
6
7
INA
8
COMPV
FBP
9
CP
10
11
12
13
14
FBB
PGOOD
GND
ENB
AVL
BST
LXB
INB
EP
LXB
INB
TSSOP
Pin Description
PIN
NAME
FUNCTION
Boost Circuitry and 1.8V/3.3V Regulator Controller Enable Input. ENP has an internal 500kI pulldown
resistor. Drive high for normal operation and drive low to place the device (except buck converter) in
shutdown.
1
ENP
1.8V or 3.3V Regulator Output. DR has a 4.5mA (min) drive capability. For greater output current capa-
bility, use an external npn bipolar transistor whose base is connected to DR.
2
3
DR
FB
1.8V or 3.3V Regulator Feedback Input. FB is regulated to 1.8V or 3.3V. Connect FB to DR when power-
ing loads demanding less than 4.5mA. For greater output current capability, use an external npn bipo-
lar transistor whose emitter is connected to FB.
External p-Channel FET Gate Drive. GATE is an open-drain driver connected to the gate of the external
input series p-channel FET. Connect a pullup resistor between GATE and INA. During a fault condition,
the gate driver turns off and the pullup resistor turns off the FET.
4
GATE
5
6
7
PGNDP
LXP
Boost Converter Power Ground
Boost Converter Switching Node. Connect LXP to the inductor and catch diode of the boost converter.
Boost Circuitry and 1.8V/3.3V Regulator Controller Power Input. Connect INA to a 3V to 5.5V supply.
INA
Boost Error Amplifier Compensation Connection. Connect a compensation network between COMPV to
GND.
8
9
COMPV
FBP
Boost Converter Feedback Input. FBP is regulated to 1V. Connect FBP to the center of a resistive divid-
er connected between the boost output and GND.
���������������������������������������������������������������� Maxim Integrated Products 10
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
Pin Description (continued)
PIN
NAME
FUNCTION
Buck Converter Feedback Input. FBB is regulated to either 3.3V or 5V. Connect FBB to the output-volt-
age node, OUTB, as shown in the Typical Application Circuit.
10
FBB
Buck Converter Internal 5V Regulator. Connect a 1FF capacitor between AVL and the analog ground
plane. Do not use AVL to power external circuitry.
11
12
AVL
BST
LXB
Buck Converter Bootstrap Capacitor Connection. Connect a 0.1FF capacitor between BST and LXB.
Buck Converter Inductor Connection. Connect the inductor, boost capacitor, and catch diode at this
node.
13, 14
Buck Converter Power Input. Connect to a 4V to 28V supply. Connect a 1FF or larger ceramic capaci-
tor in parallel with a 47FF bulk capacitor from INB to the power ground plane. Connect both INB power
inputs together.
15, 16
INB
Buck Converter Enable Input. ENB is a high-voltage compatible input. Connect to INB for normal opera-
tion and connect to ground to disable the buck converter.
17
ENB
18, 23
19
GND
Analog Ground
PGOOD Open-Drain Power-Good Output. Connect PGOOD to INA through an external pullup resistor.
Positive-Gate Voltage Regulator Power Input. Connect CP to the positive output of the external charge
20
CP
pump. Ensure that V does not exceed the CP overvoltage threshold as given in the Electrical
CP
Characteristics table.
21
22
GH
Positive-Gate Voltage Regulator Output
Negative-Gate Voltage Regulator Driver Output. DRVN is the open drain of an internal p-channel FET.
Connect DRVN to the base of an external npn pass transistor.
DRVN
Boost Slope Compensation Connection. Connect a capacitor between COMPI and GND to set the
slope compensation.
24
25
COMPI
FBGH
Positive-Gate Voltage Regulator Feedback Input. FBGH is regulated to 1V. Connect FBGH to the center
of a resistive divider connected between GH and GND.
Negative-Gate Voltage Regulator Feedback Input. FBGL is regulated to 0.25V. Connect FBGL to the
center of a resistive divider connected between REF and the output of the negative-gate voltage
regulator.
26
FBGL
27
28
REF
SEQ
1.25V Reference Output. Bypass REF to GND with a 0.1FF ceramic capacitor.
Sequencing Input. SEQ has an internal 500kI pulldown resistor. SEQ determines the sequence in
which V
and V power-up. See Table 1 for supply sequencing options.
GH
GL
Exposed Pad. Connect to a large contiguous copper ground plane for optimal heat dissipation. Do not
use EP as the only electrical ground connection.
—
EP
���������������������������������������������������������������� Maxim Integrated Products 11
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
enabling the buck converter, V
begins to rise. Once
AVL
Detailed Description
V
exceeds the undervoltage lockout voltage of 3.5V
AVL
(max), LXB starts switching. Bypass AVL to GND with a
The MAX16929 is a highly integrated power supply for
automotive TFT-LCD applications. The device integrates
one buck converter, one boost converter, one 1.8V/3.3V
regulator controller, one positive-gate voltage regulator,
and one negative-gate voltage regulator.
1FF ceramic capacitor.
Spread-Spectrum Modulation
The buck converter features spread-spectrum operation
that varies the internal operating frequency of the buck
converter by +6% relative to the switching frequency of
2.1MHz (typ).
The device achieves enhanced EMI performance through
spread-spectrum modulation. Digital input control allows
the device to be placed in a low-current shutdown mode
and provides flexible sequencing of the gate voltage
regulators.
Soft-Start
The buck converter features an internal soft-start timer.
The output voltage takes 3.9ms to ramp up to its set
voltage. If a short circuit or undervoltage is encountered
after the soft-start timer has expired, the device enters
hiccup mode, during which soft-start is reattempted
every 16ms. This process repeats until the short circuit
has been removed.
Internal thermal shutdown circuitry protects the device
from overheating. The buck converter is designed to
shut down when its die temperature reaches +175NC
(typ), while the boost circuitry does so at +165NC (typ).
Each resumes normal operation once its die temperature
has fallen 15NC below its respective thermal shutdown
temperature.
Overcurrent Protection
The device enters hiccup mode in one of three ways. If
eight consecutive current limits are detected and the out-
put is below 77% of its nominal value, the buck converter
enters hiccup mode. The converter enters hiccup mode
immediately if the output is short circuited to ground
(output below 1V). Additionally, the device enters hiccup
mode if 256 consecutive overcurrent events are detected
when the output is greater than 77% of its nominal value.
In hiccup mode, the buck controller idles for 16ms before
reattempting soft-start.
The device is factory-trimmed to provide a variety of
power options to meet the most common automotive
TFT-LCD display power requirements, as outlined in the
Ordering Information/Selector Guide table.
Buck Converter
The device features a current-mode buck converter with
an integrated high-side FET, which requires no external
compensation network. The buck converter regulates the
output voltage to within Q3% in continuous mode over
line and load conditions. The high 2.1MHz (typ) switching
frequency allows for small external components, reduced
output ripple, and guarantees no AM interference.
Power Good (PGOOD)
When an overcurrent condition causes the buck output to
fall below 92% of its set voltage, the open-drain power-
good indicator output (PGOOD) asserts low. PGOOD
deasserts once the output voltage has risen above 95%
of its set voltage.
A power-good (PGOOD) indicator is available to moni-
tor output-voltage quality. The enable input allows the
device to be placed in shutdown, reducing supply cur-
rent to 70FA.
PGOOD serves as a general fault indicator for all the
converters and regulators. Besides indicating an under-
voltage on the buck output, it also indicates any of the
faults listed in the Fault Conditions and PGOOD section.
The buck converter comes with a preset output voltage
of either 3.3V or 5V, and can deliver either 1.2A or 2A to
the output.
Enable (ENB)
Connect ENB to INB for always-on operation. ENB is also
compatible with 3.3V logic systems and can be con-
trolled through a microcontroller or by automotive KEY or
CAN inhibit signals.
Boost Converter
The boost converter employs a current-mode, fixed-
frequency PWM architecture to maximize loop bandwidth
and provide fast transient response to pulsed loads typical
of TFT-LCD panel source drivers. The 2.2MHz switching
frequency allows the use of low-profile inductors and
ceramic capacitors to minimize the thickness of LCD panel
designs. The integrated low on-resistance MOSFET and
Internal 5V Regulator (AVL)
AVL is an internal 5V regulator that supplies power to the
buck controller and charges the boost capacitor. After
���������������������������������������������������������������� Maxim Integrated Products 12
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
the device’s built-in digital soft-start functions reduce the
number of external components required while control-
ling inrush currents. The output voltage can be set from
to 1V and changes the COMPV output. The voltage at
COMPV sets the peak inductor current. As the load var-
ies, the error amplifier sources or sinks current to the
COMPV output accordingly to produce the peak induc-
tor current necessary to service the load. To maintain
stability at high duty cycles, a slope-compensation sig-
nal (set by the capacitor at COMPI) is summed with the
current-sense signal. On the rising edge of the internal
clock, the controller turns on the n-channel MOSFET and
applies the input voltage across the inductor. The current
through the inductor ramps up linearly, storing energy in
its magnetic field. Once the sum of the current feedback
signal and the slope compensation exceeds the COMPV
voltage, the controller turns off the MOSFET. The inductor
current then flows through the diode to the output. The
MOSFET remains off for the rest of the clock cycle.
V
INA
to 18V with an external resistive voltage-divider.
The regulator controls the output voltage by modulat-
ing the duty cycle (D) of the internal power MOSFET in
each switching cycle. The duty cycle of the MOSFET is
approximated by:
ηV
IN
D=1−
V
O
where V is the voltage at INA, V = V (the boost
SH
IN
O
output voltage), and E is the efficiency of the boost con-
verter, as shown in the Typical Operating Characteristics.
Figure 1 shows the functional diagram of the boost
regulator. An error amplifier compares the signal at FBP
LXP
CLOCK
LOGIC AND
DRIVER
PGNDP
I
LIM
COMPARATOR
SOFT-
START
V
LIMIT
PWM
COMPARATOR
CURRENT
SENSE
Σ
2.2MHz
OSCILLATOR
SLOPE
COMP
COMPI
FBP
ERROR
AMP
TO FAULT LOGIC
0.85V
FAULT
COMPARATOR
1V
MAX16929
COMPV
Figure 1. Boost Converter Functional Diagram
���������������������������������������������������������������� Maxim Integrated Products 13
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
The external p-channel FET controlled by GATE protects
the output during fault conditions and provides True
Shutdown of the converter. Connect a pullup resistor
between GATE and INA (see the Boost Converter section
to select the value for the pullup resistor). Under normal
operation, GATE turns on the p-channel FET, connecting
the supply to the boost input. During a fault condition or
in shutdown, GATE is off and the pullup resistor turns off
the p-channel FET, disconnecting the supply from the
boost input.
4) The LXP voltage is greater than 21V (typ).
5) The positive charge-pump voltage (V ) is greater
CP
than 30.5V (typ).
6) The 1.8V/3.3V regulator output voltage falls below
85% of its nominal value.
7) The buck output voltage falls below 92% of its nominal
value.
If any of the first three fault conditions persists for longer
than the 238ms fault blanking period, the device pulls
PGOOD low, turns off all outputs, and starts the autoretry
timer.
Spread-Spectrum Modulation
The high-frequency 2.2MHz operation of the boost con-
verter keeps switching noise outside of the AM band. The
device achieves enhanced EMI performance by modu-
lating the switching frequency by Q4%. The modulating
signal is pseudorandom and changes each switching
If either condition 4 or 5 occurs, the device pulls PGOOD
low and turns off all outputs immediately. The device initi-
ates startup only after the fault has cleared.
If condition 6 occurs, the device pulls PGOOD low, but
does not turn off any of the outputs.
period (i.e., f = 2.2MHz).
SS
Startup
During startup, PGOOD is masked and goes high as
soon as the 1.8V/3.3V regulator controller turns on. This
Immediately after power-up, coming out of shutdown,
or going into autoretry, the boost converter performs a
short-circuit detection test on the output by connecting
the input (INA) to the switching node (LXP) through an
internal 50I resistor.
regulator turns on as soon as V
undervoltage lockout threshold.
exceeds the INA
INA
Autoretry
When the autoretry counter finishes incrementing after
1.9s, the device attempts to turn on the boost converter
and gate voltage regulators in the order shown in
Table 1. The device continues to autoretry as long as the
fault condition persists. A fault on the 1.8V/3.3V regulator
output causes PGOOD to go low, but does not result in
the device shutting down and going into autoretry.
If the resulting voltage on LXP exceeds 1.2V, the device
turns on the external pMOS switch by pulling GATE low.
The boost output ramps to its final value in 15ms.
An overloaded or shorted output is detected if the result-
ing voltage on LXP is below 1.2V. The external pMOS
switch remains off and the converter does not switch.
After the fault blanking period of 238ms, the device pulls
PGOOD low and starts the autoretry timer.
Current Limit
The effective current limit of the boost converter is
reduced by the internally injected slope compensation by
an amount dependent on the duty cycle of the converter.
The effective current limit is given by:
The short-circuit detection feature places a lower limit
on the output load of approximately 46I when the input
voltage is 3V.
Fault Conditions and PGOOD
PGOOD signals whether all the regulators and the boost
converter are operating normally. PGOOD is an open-
drain output that pulls low if any of the following faults
occur:
D
-12
I
=192 ×10
×I
×
LIM(EFF)
LIM_DC_0
C
COMPI
where I
is the effective current limit, I
=
LIM(EFF)
LIM_DC_0
1.1A or 2.2A depending on the boost converter current-
limit option, D is the duty cycle of the boost converter,
1) The boost output voltage falls below 85% of its set
value.
and C
is the value of the capacitor at the COMPI
input. Estimate the duty cycle of the converter using the
formulas shown in the Design Procedure section.
COMPI
2) The positive-gate voltage regulator output (V ) falls
GH
below 85% of its set value.
3) The negative-gate voltage regulator output (V ) falls
GL
below 85% of its set value.
���������������������������������������������������������������� Maxim Integrated Products 14
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
dependent bypassing requirements. Connect a ceramic
capacitor between the collector and ground with the
value shown in Table 3.
1.8V/3.3V Regulator Controller
The 1.8V/3.3V regulator controller delivers 4.5mA (min)
to an external load. Connect FB to DR for a regulated
1.8V/3.3V output.
The regulator derives its negative supply voltage from an
inverting charge pump, a single-stage example of which
is shown in the Typical Application Circuit. A more nega-
tive voltage using a multistage charge pump is possible
as described in the Charge Pumps section.
For higher output capability, use an external npn transis-
tor as shown in the Typical Application Circuit. The drive
capability of the regulator is then increased by the cur-
rent gain of the transistor (h ). When using an external
FE
transistor, use DR as the base drive and connect FB to
the transistor’s emitter. Bypass the base to ground with a
0.1FF ceramic capacitor.
The external npn transistor is not short-circuit protected.
To maintain proper pulldown capability of external npn
transistor and optimal regulation, a minimum load of at
least 500µA is recommended on the output of the GL
regulator.
If the boost output current is greater than 300mA, con-
nect a 30kI resistor between DR and GND.
Enable (ENP)
Use the enable input (ENP) to enable and disable the
boost section of the device. Connect ENP to INA for
normal operation and to GND to place the device in shut-
down. In shutdown, the INA supply current is reduced to
0.5FA.
Positive-Gate Voltage Regulator (GH)
The positive-gate voltage regulator includes a p-channel
FET output stage to generate a regulated output between
+5V and V - 2V. The regulator maintains accuracy over
CP
wide line and load conditions. It is capable of at least
20mA of output current and includes current-limit protec-
tion. V
drivers’ gate-on voltage.
is typically used to provide the TFT-LCD gate
GH
Soft-Start and Supply Sequencing (SEQ)
When enabled, the boost output ramps up from V
to
INA
The regulator derives its positive supply voltage from a
noninverting charge pump, a single-stage example of
which is shown in the Typical Application Circuit. A high-
er voltage using a multistage charge pump is possible,
as described in the Charge Pumps section.
its set voltage. Once the boost output reaches 85% of the
set voltage and the soft-start timer expires, the gate volt-
age regulators turn on in the order shown in Table 1. The
1.8V/3.3V regulator controller is enabled at the beginning
of the boost converter’s soft-start.
Both gate voltage regulators have a 7.45ms soft-start
time. The second one turns on as soon as the output of
the first reaches 85% of its set voltage.
Negative-Gate Voltage Regulator (GL)
The negative-gate voltage regulator is an analog gain
block with an open-drain p-channel output. It drives an
external npn pass transistor with a 6.8kIbase-to-emitter
resistor (see the Pass Transistor Selection section). Its
guaranteed base drive source current is at least 2mA.
Thermal Shutdown
Internal thermal shutdown circuitry shuts down the
device immediately when the die temperature exceeds
+165NC. A 15NC thermal shutdown hysteresis prevents
the device from resuming normal operation until the die
temperature falls below +150NC.
V
is typically used to provide the TFT-LCD gate driv-
GL
ers’ gate-off voltage.
The output of the negative-gate voltage regulator (i.e.,
the collector of the external npn pass transistor) has load-
Table 1. Supply Sequencing
CONTROL INPUTS
SUPPLY SEQUENCING
ENP
SEQ
FIRST
SECOND
THIRD
0
1
1
X
0
1
Device is in shutdown
V
V
V
V
GL
SH
GH
V
V
GH
SH
GL
���������������������������������������������������������������� Maxim Integrated Products 15
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
Table 2. Minimum Buck Inductor Value
Required for Normal Operation During
Load Dump
Design Procedure
Buck Converter
Inductor Selection
Three key inductor parameters must be specified for
operation with the device: inductance value (L), induc-
BUCK V
(V)
BUCK I
(A)
L
(µH)
MIN
OUTB
OUTB
3.3
1.2
3.3
tor saturation current (I
), and DC resistance (R ).
SAT
DC
3.3
5
2
6.8
3.3
4.7
To determine the inductance value, select the ratio of
inductor peak-to-peak ripple current to average output
current (LIR) first. For LIR values that are too high, the
RMS currents are high, and therefore I2R losses are high.
Use high-valued inductors to achieve low LIR values.
Typically, inductance is proportional to resistance for a
given package type, which again makes I2R losses high
for very low LIR values. A good compromise between
size and loss is to select a 30%-to-60% peak-to-peak
ripple current to average-current ratio. If extremely thin
high-resistance inductors are used, as is common for
LCD-panel applications, the best LIR can increase
between 0.5 and 1.0. The size of the inductor is deter-
mined as follows:
1.2
2
5
Capacitor Selection
The input and output filter capacitors should be of a low-
ESR type (tantalum, ceramic, or low-ESR electrolytic) and
should have I
ratings greater than:
RMS
2
LIR
12
I
= I D× (1-D +
)
for the input capacitor
INB(RMS)
O
LIR×I
O
I
=
for the output capacitor
OUTB(RMS)
12
where D is the duty cycle given above.
(V
-V )×D
O
INB
L =
and
The output voltage contains a ripple component whose
peak-to-peak value depends on the value of the ESR
and capacitance of the output capacitor, and is approxi-
mately given by:
LIR×I × f
O
SWB
V
O
D =
η× V
INB
DV
= DV
+ DV
RIPPLE
ESR CAP
where V
is the input voltage, V is the output volt-
O
INB
DV
= LIR x I x R
O ESR
ESR
age, I is the output current, E is the efficiency of the
O
LIR×I
O
buck converter, D is the duty cycle, and f
is 2.1MHz
SWB
∆V
=
CAP
8 × C× f
(the switching frequency of the buck converter). The
efficiency of the buck converter can be estimated from
the Typical Operating Characteristics and accounts for
SWB
Diode Selection
The catch diode should be a Schottky type to minimize
its voltage drop and maximize efficiency. The diode must
be capable of withstanding a reverse voltage of at least
the maximum input voltage in the application. The diode
should have an average forward current rating greater
than:
losses in the internal switch, catch diode, inductor R
and capacitor ESR.
,
DC
To ensure the buck converter does not shut down
during load dump input-voltage transients to 42V, an
inductor value larger than calculated above should be
used. Table 2 lists the minimum inductance that should
be used for proper operation during load dump. The
I
= I × (1-D)
O
D
saturation current rating (I
) must be high enough to
where D is the duty cycle given above. In addition, ensure
that the peak current rating of the diode is greater than:
SAT
ensure that saturation can occur only above the maxi-
mum current-limit value. Find a low-loss inductor having
the lowest possible DC resistance that fits in the allotted
dimensions.
LIR
2
I
× 1+
OUTB
���������������������������������������������������������������� Maxim Integrated Products 16
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
The output voltage contains a ripple component whose
peak-to-peak value depends on the value of the ESR and
capacitance of the output capacitor and is approximately
given by:
Boost Converter
Inductor Selection
Three key inductor parameters must be specified for
operation with the device: inductance value (L), induc-
tor saturation current (I
), and DC resistance (R ).
DV
= DV + DV
ESR CAP
SAT
DC
RIPPLE
To determine the inductance value, select the ratio of
inductor peak-to-peak ripple current to average input
current (LIR) first. For LIR values that are too high, the
RMS currents are high, and therefore I2R losses are high.
Use high-valued inductors to achieve low LIR values.
Typically, inductance is proportional to resistance for a
given package type, which again makes I2R losses high
for very low LIR values. A good compromise between
size and loss is to select a 30%-to-60% peak-to-peak
ripple current to average-current ratio. If extremely thin
high-resistance inductors are used, as is common for
LCD-panel applications, the best LIR can increase
between 0.5 and 1.0. The size of the inductor is deter-
mined as follows:
LIR
2
∆V
=I
× (1+
)×R
ESR
ESR INP
I
×D
O
∆V
=
CAP
C
×f
OUT SW
where I
given above.
and D are the input current and duty cycle
INP
Rectifier Diode
The catch diode should be a Schottky type to minimize
its voltage drop and maximize efficiency. The diode must
be capable of withstanding a reverse voltage of at least
V
SH
. The diode should have an average forward current
rating greater than:
V
×D
V ×I
O O
INA
L =
and I
=
I
= I
× (1-D)
D
INP
INP
LIR×I
× f
ηV
INP SW
INA
where I
and D are the input current and duty cycle
INP
given above. In addition ensure that the peak current rat-
ing of the diode is greater than:
ηV
V
INA
D=1−
O
LIR
2
I
× 1+
INP
where V
is the input voltage, V is the output voltage,
O
INA
I
is the output current, I
is the average boost input
O
INP
current, Eis the efficiency of the boost converter, D is the
duty cycle, and f is 2.2MHz (the switching frequency
of the boost converter). The efficiency of the boost
converter can be estimated from the Typical Operating
Characteristics and accounts for losses in the internal
Output-Voltage Selection
The output voltage of the boost converter can be adjust-
ed by using a resistive voltage-divider formed by R
SW
TOP
and R
FBP and connect R
Select R
. Connect R
between the output and
between FBP and GND.
BOTTOM
TOP
BOTTOM
switch, catch diode, inductor R , and capacitor ESR.
DC
in the 10kI to 50kI range. Calculate
BOTTOM
R
with the following equation:
TOP
Capacitor Selection
The input and output filter capacitors should be of a low-
ESR type (tantalum, ceramic, or low-ESR electrolytic) and
V
O
R
=R
× (
BOTTOM
−1)
TOP
V
FBP
should have I
ratings greater than:
RMS
where V
, the boost converter’s feedback set point, is
FBP
LIR×I
INP
1V. Place both resistors as close as possible to the device
and connect R to the analog ground plane.
I
=
for the input capacitor
RMS
12
BOTTOM
Loop Compensation
to set the high-frequency integrator
COMPV
2
LIR
Choose R
D+
12
gain for fast transient response. Choose C
to set
I
=I
O
COMPV
for the output capacitor
RMS
1− D
the integrator zero to maintain loop stability. For low-ESR
output capacitors, use Table 3 to select the initial values
where I
given above.
and D are the input current and duty cycle
INP
for R
lel with R
and C
. Use a 22pF capacitor in paral-
COMPV
COMPV
+ C
.
COMPV
COMPV
���������������������������������������������������������������� Maxim Integrated Products 17
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
normal operation, R
55FA and the resulting gate source voltage (V ) turns
on the FET. When the gate drive is removed under a fault
condition or in shutdown, R
carries a gate drive current of
SG
Table 3. Compensation Component Values
GS
V
(V)
8
18
200
5
SH
bleeds off charge to turn
I
(mA)
200
3.3
1.75
5
SG
SH
off the FET. Size R
on the FET.
to produce the V
needed to turn
SG
GS
V
(V)
INA
P
(W)
3.75
5
IN
1.8V/3.3V Regulator Controller
L (µH)
(kI)
npn Bipolar Transistor Selection
R
33
39
COMPV
There are two important considerations in selecting the
pass npn bipolar transistor: current gain (h ) and power
dissipation. Select a transistor with an h high enough to
C
(pF)
(pF)
220
820
180
330
COMPV
FE
C
COMPI
FE
ensure adequate drive capability. This condition is satis-
fied when I
x (h + 1) is greater than the maximum
DR
FE
load current. The regulator can source I = 4.5mA (min).
The transistor should be capable of dissipating:
DR
V
SH
V
CP
P
= (V
- V
) × I
NPN_REG
INA
REG_OUT LOAD(MAX)
LXP
where V
= 1.8V or 3.3V. Bypass DR to ground
REG_OUT
with a 0.1FF ceramic capacitor. For applications in which
the boost output current exceeds 300mA, connect a
30kI resistor from DR to ground.
Figure 2. Multistage Charge Pump for Positive Output Voltage
Supply Considerations
INA needs to be at least 4.5V for the 3.3V regulator to
operate properly.
V
CN
Charge Pumps
Selecting the Number of Charge-Pump Stages
For most applications, a single charge-pump stage is
sufficient, as shown in the Typical Application Circuit.
Connect the flying capacitors to LXP. The output voltages
generated on the storage capacitors are given by:
LXP
Figure 3. Multistage Charge Pump for Negative Output Voltage
V
= 2 x V + V
- 2 x V
SCHOTTKY D
CP
SH
V
= -(V + V
- 2 x V )
CN
SH
SCHOTTKY D
To further optimize transient response, vary R
COMPV
where V is the positive supply for the positive-gate volt-
in 20% steps and C
in 50% steps while observ-
CP
COMPV
age regulator, and V
is the negative supply for the neg-
ing transient-response waveforms. The ideal transient
response is achieved when the output settles quickly with
little or no overshoot. Connect the compensation network
to the analog ground plane.
CN
ative-gate voltage regulator. Where larger output voltages
are needed, use multistage charge pumps (however, the
maximum charge-pump voltage is limited by the absolute
maximum ratings of CP and DRVN). Figure 2 and Figure 3
show the configuration of a multistage charge pump for
both positive and negative output voltages.
Use the following formula to calculate the value for C
-6
:
COMPI
C
COMPI
≤ 550 × 10 × L/f
× (V + V
- V
)
SW
SH
SCHOTTKY
INA
where f
= 2.2MHz.
SW
For mutistage charge pumps the output voltages are:
p-Channel FET Selection
The p-channel FET used to gate the boost converter’s
input should have low on-resistance. Connect a resistor
V
CP
V
= V + n × (V + V
- 2 x V )
SH
SH
SCHOTTKY D
= -n × (V + V
- 2 x V )
D
CN
SH
SCHOTTKY
(R ) between the source and gate of the FET. Under
SG
���������������������������������������������������������������� Maxim Integrated Products 18
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
For highest efficiency, choose the lowest number of
charge-pump stages that meets the output requirement.
The number of positive charge-pump stages needed is
given by:
where C
is the output capacitor of the charge
OUT_CP
pump, D is the duty cycle of the boost converter, I
is the load current of the charge pump, f
ing frequency of the boost converter, and V
the peak-to-peak value of the output ripple.
LOAD_CP
is the switch-
SW
is
RIPPLE_CP
V
+V
− V
− 2 × V
GH DROPOUT SH
n
=
CP
For the inverting charge pump connected to CN, use the
following equation to approximate the required output
capacitance:
V
+V
SH SCHOTTKY D
and the number of negative charge-pump stages is
given by:
(1-D)×I
LOAD_CN
C
≥
OUT_CN
|V |+V
f
× V
RIPPLE_CN
GL
DROPOUT
SW
n
=
CN
V
+ V
− 2 × V
SH
SCHOTTKY D
where C
is the output capacitor of the charge
OUT_CN
where n
is the number of positive charge-pump stag-
is the number of negative charge-pump stages,
pump, D is the duty cycle of the boost converter,
CP
es, n
I
is the load current of the charge pump, f
CN
LOAD_CN SW
V
GH
is the positive-gate voltage regulator output volt-
is the switching frequency of the boost converter, and
age, V
is the negative-gate voltage regulator output
V
is the peak-to-peak value of the output
GL
RIPPLE_CN
voltage, V
is the boost converter’s output voltage, V
ripple.
SH
D
is the forward-voltage drop of the charge-pump diode,
is the forward drop of the Schottky diode
Charge-Pump Rectifier Diodes
V
SCHOTTKY
Use high-speed silicon switching diodes with a current
rating equal to or greater than two times the average
charge-pump input current. If it helps avoid an extra
stage, some or all of the diodes can be replaced with
Schottky diodes with an equivalent current rating.
of the boost converter, and V
margin for the regulator. Use V
negative voltage regulator and V
is the dropout
= 0.3V for the
= 2V at 20mA
DROPOUT
DROPOUT
DROPOUT
for the positive-gate voltage regulator.
Flying Capacitors
Increasing the flying capacitor (C ) value lowers the
Positive-Gate Voltage Regulator
X
Output-Voltage Selection
The output voltage of the positive-gate voltage regula-
tor can be adjusted by using a resistive voltage-divider
effective source impedance and increases the output
current capability. Increasing the capacitance indefi-
nitely has a negligible effect on output current capability
because the internal switch resistance and the diode
impedance place a lower limit on the source impedance.
A 0.1FF ceramic capacitor works well in most low-current
applications. The voltage rating of the flying capacitors
formed by R
and R
. Connect R
between
between
TOP
BOTTOM
TOP
the output and FBGH, and connect R
FBGH and GND. Select R
BOTTOM
in the 10kI to 50kI
BOTTOM
range. Calculate R
with the following equation:
TOP
for the positive charge pump should exceed V , and
that for the negative charge pump should exceed the
CP
V
GH
R
= R
× (
BOTTOM
−1)
TOP
V
FBGH
magnitude of V
.
CN
where V
is the desired output voltage and V
= 1V
FBGH
Charge-Pump Output Capacitor
GH
(the regulated feedback voltage for the regulator). Place
both resistors as close as possible to the device.
Increasing the output capacitance or decreasing the ESR
reduces the output-ripple voltage and the peak-to-peak
transient voltage. With ceramic capacitors, the output-
voltage ripple is dominated by the capacitance value.
Use the following equation to approximate the required
output capacitance for the noninverting charge pump
connected to CP:
Avoid excessive power dissipation within the internal
pMOS device of the regulator by paying attention to the
voltage drop across the drain and source. The amount of
power dissipation is given by:
P
GL
= (V - V ) × I
CP GH LOAD(MAX)
D×I
LOAD_CP
where V
is the noninverting charge-pump output volt-
CP
C
≥
OUT_CP
f
× V
RIPPLE_CP
age applied to the drain, V is the regulated output
voltage, and I
SW
GH
is the maximum load current.
LOAD(MAX)
���������������������������������������������������������������� Maxim Integrated Products 19
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
Stability Requirements
The positive-gate voltage regulator (GH) requires a mini-
mum output capacitance for stability. For an output volt-
P
= (V - V ) × I
NPN_GL
GL CN LOAD(MAX)_GL
where V
is the regulated output voltage on the collec-
GL
tor of the transistor, V
is the inverting charge-pump
CN
age of 5V to (V - 2V) and an output current of 10mA to
CP
output voltage applied to the emitter of the transistor,
and I is the maximum load current. Note
15mA, use a minimum capacitance of 0.47FF.
LOAD(MAX)_GL
that the external transistor is not short-circuit protected.
Negative-Gate Voltage Regulator
Output-Voltage Selection
The output voltage of the negative-gate voltage regula-
tor can be adjusted by using a resistive voltage-divider
Stability Requirements
The device’s negative-gate voltage regulator uses an
internal transconductance amplifier to drive an external
pass transistor. The transconductance amplifier, the
pass transistor, the base-emitter resistor, and the output
capacitor determine the loop stability.
formed by R
REF and FBGL, and connect R
and R
. Connect R
between
TOP
BOTTOM
TOP
between FBGL
BOTTOM
and the collector of the external npn transistor. Select
greater than 20kI to avoid loading down the ref-
R
TOP
The transconductance amplifier regulates the output volt-
age by controlling the pass transistor’s base current. The
total DC loop gain is approximately:
erence output. Calculate R
equation:
with the following
BOTTOM
V
− V
GL
FBGL
R
= R
×
TOP
BOTTOM
I
×h
I
LOAD
4
V
− V
BIAS
FE
REF
FBGL
A
≅ ( )× (1+
)× V
REF
V_GL
V
T
where V
and V
the regulator).
is the desired output voltage, V
= 0.25V (the regulated feedback voltage of
= 1.25V,
GL
FBGL
REF
where V is 26mV at room temperature, and I
current through the base-to-emitter resistor (R ). For
is the
BIAS
BE
T
the device, the bias current for the negative-gate voltage
regulator is 0.1mA. Therefore, the base-to-emitter resistor
should be chosen to set 0.1mA bias current:
Pass Transistor Selection
The pass transistor must meet specifications for current
gain (h ), input capacitance, collector-emitter saturation
FE
V
0.7V
voltage, and power dissipation. The transistor’s current
gain limits the guaranteed maximum output current to:
BE
R
=
=
= 7kΩ
BE
0.1mA 0.1mA
V
Use the closest standard resistor value of 6.8kI. The
output capacitor and the load resistance create the
dominant pole in the system. However, the internal
amplifier delay, pass transistor’s input capacitance,
and the stray capacitance at the feedback node create
additional poles in the system, and the output capacitor’s
ESR generates a zero. For proper operation, use the fol-
lowing equations to verify that the regulator is properly
compensated:
BE
I
= (I
−
)×h
FE(MIN)
LOAD(MAX)
DRVN
R
BE
where I
rent, V
is the minimum guaranteed base-drive cur-
is the transistor’s base-to-emitter forward volt-
DRVN
BE
age drop, and R
is the pulldown resistor connected
BE
between the transistor’s base and emitter. Furthermore,
the transistor’s current gain increases the regulator’s DC
loop gain (see the Stability Requirements section), so
excessive gain destabilizes the output.
1) First, determine the dominant pole set by the regula-
tor’s output capacitor and the load resistor:
The transistor’s saturation voltage at the maximum output
current determines the minimum input-to-output volt-
age differential that the regulator can support. Also, the
package’s power dissipation limits the usable maximum
input-to-output voltage differential. The maximum power-
dissipation capability of the transistor’s package and
mounting must exceed the actual power dissipated in
the device. The power dissipated equals the maximum
I
LOAD(MAX)_GL
f
=
POLE_GL
2π × C
× V
OUT_GL
OUT_GL
The unity-gain crossover frequency of the regulator is:
= A × f
f
CROSSOVER
V_GL
POLE_GL
2) The pole created by the internal amplifier delay is
approximately 1MHz:
load current (I
) multiplied by the maximum
input-to-output voltage differential:
LOAD(MAX)_GL
f
= 1MHz
POLE_AMP
���������������������������������������������������������������� Maxim Integrated Products 20
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
3) Next, calculate the pole set by the transistor’s input
Table 4. Minimum Output Capacitance vs.
capacitance, the transistor’s input resistance, and the
Output Voltage Range for Negative-Gate
base-to-emitter pullup resistor:
Voltage Regulator (I
= 10mA to 15mA)
OUT
1
f
=
=
POLE_IN
OUTPUT VOLTAGE
RANGE
MINIMUM OUTPUT
CAPACITANCE (µF)
2π × C × (R /R
)
IN
BE IN
where:
-2V R V R -4V
2.2
1.5
1
GL
g
h
FE
m
C
, R
=
IN
-5V R V R -7V
GL
IN
2πf
g
T
m
-8V R V R -13V
GL
g
is the transconductance of the pass transistor, and
m
f is the transition frequency. Both parameters can be
T
Table 4 is a list of recommended minimum output capaci-
tance for the negative-gate voltage regulator and are
applicable for output currents in the 10mA to 15mA range.
found in the transistor’s data sheet. Because R
is
BE
much greater than R , the above equation can be
simplified:
IN
Applications Information
1
2π × C ×R
f
=
POLE_IN
IN
IN
Power Dissipation
An IC’s maximum power dissipation depends on the ther-
mal resistance from the die to the ambient environment
and the ambient temperature. The thermal resistance
depends on the IC package, PCB copper area, other
thermal mass, and airflow. More PCB copper, cooler
ambient air, and more airflow increase the possible dis-
sipation, while less copper or warmer air decreases the
IC’s dissipation capability. The major components of
power dissipation are the power dissipated in the buck
converter, boost converter, positive-gate voltage regula-
tor, negative-gate voltage regulator, and the 1.8V/3.3V
regulator controller.
Substituting for C and R yields:
IN
IN
f
T
f
=
POLE
h
FE
4) Next, calculate the pole set by the regulator’s feed-
back resistance and the capacitance between FBGL
and GND (including stray capacitance):
1
× (R
f
=
POLE_FBGL
2π × C
/R
)
FBGL
TOP BOTTOM
where C
GND and is equal to 30pF, R
is the capacitance between FBGL and
FBGL
is the upper resistor
TOP
of the regulator’s feedback divider, and R
the lower resistor of the divider.
is
BOTTOM
Buck Converter
In the buck converter, conduction and switching losses
in the internal MOSFET are dominant. Estimate these
losses using the following formula:
5) Next, calculate the zero caused by the output capaci-
tor’s ESR:
2
1
P
LXB
≈ [(I
× √D) × R
] + [0.5 × V
×
OUTB
DS_ON(LXB)
INB
f
=
ZERO_ESR
I
× (t + t ) × f
]
2π × C
×R
OUTB
R
F
SWB
OUT_LR
ESR
where I
of the buck converter, R
of the internal high-side FET, V
is the output current, D is the duty cycle
OUTB
where R
is the equivalent series resistance of
ESR
is the on-resistance
is the input voltage,
DS_ON(LXB)
C
. To ensure stability, make C
large
OUT_LR
OUT_LR
INB
enough so the crossover occurs well before the poles
and zero calculated in steps 2 to 5. The poles in steps
3 and 4 generally occur at several MHz and using
ceramic capacitors ensures the ESR zero also occurs
at several MHz. Placing the crossover frequency below
500kHz is sufficient to avoid the amplifier delay pole
and generally works well, unless unusual component
choices or extra capacitances move one of the other
poles or the zero below 1MHz.
(t + t is the time is takes for the switch current and
R
F)
voltage to settle to their final values during the rising and
falling transitions, and f is the switching frequency
SWB
DS_ON(LXB)
of the buck converter. R
is 180mI (typ) and
(t + t is 4.4ns + 4.6ns = 9ns at V = 12V.
R
F)
INB
���������������������������������������������������������������� Maxim Integrated Products 21
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
Boost Converter
Power dissipation in the boost converter is primarily due
to conduction and switching losses in the low-side FET.
Conduction loss is produced by the inductor current
flowing through the on-resistance of the FET during the
on-time. Switching loss occurs during switching transi-
tions and is a result of the finite time needed to fully turn
on and off the FET. Power dissipation in the boost con-
verter can be estimated with the following formula:
Positive-Gate Voltage Regulator
Use the lowest number of charge-pump stages possible
in supplying power to the positive-gate voltage regulator.
Doing so minimizes the drain-source voltage of the inte-
grated pMOS switch and power dissipation. The power
dissipated in the switch is given as:
P
GH
= (V - V ) × I
CP GH LOAD(MAX)_GH
Ensure that the voltage on CP does not exceed the
CP overvoltage threshold as given in the Electrical
Characteristics table.
2
P
≈ [(I
IN(DC,MAX)
× √D) × R
] + V
×
LXP
IN(DC,MAX)
DS_ON(LXP)
SH
I
× f
× [(t
+ t ) + (t + t )]
SW
R-V F-I R-I F-V
Negative-Gate Voltage Regulator
Use the lowest number of charge-pump stages possible
to provide the negative voltage to the negative-gate
voltage regulator. Estimate the power dissipated in the
negative-gate voltage regulator using the following:
where I
is the maximum expected average
IN(DC,MAX)
input (i.e., inductor) current, D is the duty cycle of the
boost converter, R
the internal low-side FET, V
is the on-resistance of
is the output voltage, and
DS_ON(LXP)
SH
f
R
is the switching frequency of the boost converter.
SW
P
GL
= (V + |V | - V ) × I
INA CN BE DRVN
is 110mI (typ) and f
is 2.2MHz.
DS_ON(LXP)
SW
where V is the base-emitter voltage of the external npn
BE
The voltage and current rise and fall times at the LXP
node are equal to t (voltage rise time), t (voltage fall
bipolar transistor, and I
is the current sourced from
DRVN
R-V
F-V
DRVN to the R
bias resistor and to the base of the
BE
time), t
(current rise time), and t (current fall time),
R-I
F-I
transistor, which is given by:
and are determined as follows:
V
I
GL
BE
I
=
+
DRVN
V
+ V
SCHOTTKY
SH
R
h
+1
t
t
=
=
BE
FE
R-V
F-V
K
R-V
1.8V/3.3V Regulator Controller
The power dissipated in the 1.8V/3.3V regulator controller
is given by:
V
+ V
K
SH
SCHOTTKY
F-V
P
= (V
- V
- V ) × I
REG
INA
OUT_REG BE DR
I
IN(DC,MAX)
where V
= 1.8V or 3.3V, V is the base-emitter
BE
t
=
=
OUT_REG
R-I
F-I
K
voltage of the external npn bipolar transistor, and I
is
R-I
DR
the current sourced from DR to the base of the transistor.
is given by:
I
IN(DC,MAX)
I
DR
t
K
I
F-I
LOAD
I
=
DR
h
+1
FE
K
R-V
, K , K , and K
are the voltage and current
F-I
slew rates of the LXP node and are supply dependent.
Use Table 5 to determine their values.
F-V
R-I
where I
is load current of the 1.8V/3.3V regulator
FE
LOAD
controller, and h is the current gain of the transistor.
Table 5. LXP Voltage and Current Slew Rates vs. Supply Voltage
LXP VOLTAGE AND CURRENT SLEW RATES
RISING VOLTAGE
SLEW RATE
FALLING VOLTAGE
SLEW RATE
RISING CURRENT
SLEW RATE
FALLING CURRENT
SLEW RATE
V
(V)
INA
K
(V/ns)
K
(V/ns)
K
(A/ns)
K
(A/ns)
R-V
F-V
R-I
F-I
3.3
0.52
1.35
1.7
0.13
0.3
0.38
0.44
5
2
���������������������������������������������������������������� Maxim Integrated Products 22
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
Total Power Dissipation
The total power dissipated in the package is the sum of
the losses previously calculated. Therefore, total power
dissipation can be estimated as follows:
2) Connect input and output capacitors to the power
ground planes; connect all other capacitors to the
analog ground plane.
3) Keep the high-current paths as short and wide as
possible. Keep the path of switching currents short.
P = P
+ P
+ P + P + P
GH GL REG
T
LXB
LXP
Achieve maximum heat transfer by connecting the exposed
pad to a thermal landing pad and connecting the thermal
landing pad to a large ground plane through thermal vias.
4) Place the feedback resistors as close to the IC as
possible. Connect the negative end of the resistive
divider and the compensation network to the analog
ground plane.
Layout Considerations
Careful PCB layout is critical in achieving stable and
optimized performance. Follow the following guidelines
for good PCB layout:
5) Route the high-speed switching node LXB and LXP
away from sensitive analog nodes (FB, FBP, FBGH,
FBGL, FBB, and REF).
Refer to the MAX16929 Evaluation Kit data sheet for a
recommended PCB layout.
1) Place decoupling capacitors as close as possible to
the device. Connect the power ground planes and the
analog ground plane together at one point close to the
device.
Ordering Information/Selector Guide
REGULATOR
BUCK
BUCK
BOOST
(A)
PART
PIN-PACKAGE
V
(V)
V
(V)
I
(A)
I
LIM
REG
3.3
1.8
1.8
3.3
3.3
1.8
1.8
1.8
1.8
OUTB
5
OUTB
2
MAX16929AGUI/V+
MAX16929BGUI/V+
MAX16929CGUI/V+
MAX16929DGUI/V+
MAX16929EGUI/V+
MAX16929FGUI/V+
MAX16929GGUI/V+
MAX16929HGUI/V+
MAX16929IGUI/V+
1.5
28 TSSOP-EP*
28 TSSOP-EP*
28 TSSOP-EP*
28 TSSOP-EP*
28 TSSOP-EP*
28 TSSOP-EP*
28 TSSOP-EP*
28 TSSOP-EP*
28 TSSOP-EP*
5
2
1.5
3.3
5
2
1.5
2
0.75
0.75
0.75
0.75
0.75
0.75
5
1.2
2
5
5
1.2
2
3.3
3.3
1.2
Note: All devices are specified over the -40°C to +105°C operating temperature range.
/V denotes an automotive qualified part.
+Denotes a lead(Pb)-free/RoHS-compliant package.
*EP = Exposed pad.
Chip Information
Package Information
For the latest package outline information and land patterns
(footprints), go to www.maxim-ic.com/packages. Note that a
“+”, “#”, or “-” in the package code indicates RoHS status only.
Package drawings may show a different suffix character, but
the drawing pertains to the package regardless of RoHS status.
PROCESS: BiCMOS
PACKAGE
TYPE
PACKAGE
CODE
OUTLINE
NO.
LAND
PATTERN NO.
28 TSSOP-EP
U28ME+1
21-0108
90-0147
���������������������������������������������������������������� Maxim Integrated Products 23
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
Typical Application Circuit
OUTB
R
COMPV
C
COMPI
C
COMPV
COMPI
COMPV
GATE
INA
L
P
OPTIONAL
LXP
V
SH
DR
FB
V TO 18V
INA
BOOST
1.8V/3.3V
REGULATOR
CONTROLLER
PGNDP
FBP
V
REG
1.8V/3.3V
LXP
V
OSCILLATOR
CN
V
CN
CP
DRVN
FBGL
V
SH
POSITIVE
GATE
VOLTAGE
REGULATOR
NEGATIVE
GATE
VOLTAGE
REGULATOR
GH
V
GH
V
GL
FBGH
BST
INB
4V TO 28V
REF
BANDGAP
REFERENCE
3.3V/5V
BUCK
GND
LXB
OUTB
ENP
SEQ
INA
CONTROL
FBB
ENB
AVL
GND
PGOOD
MAX16929
���������������������������������������������������������������� Maxim Integrated Products 24
MAX16929
Automotive TFT-LCD Power Supply with Boost
Converter and Gate Voltage Regulators
Revision History
REVISION REVISION
PAGES
CHANGED
DESCRIPTION
NUMBER
DATE
0
5/11
Initial release
—
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied.
Maxim reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits) shown in the Electrical
Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
25
©
2011 Maxim Integrated Products
Maxim is a registered trademark of Maxim Integrated Products, Inc.
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