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MAX1846EUB+T

更新时间：2024-09-18 19:05:02
品牌：MAXIM
描述：Switching Controller, Current-mode, 500kHz Switching Freq-Max, BICMOS, PDSO10, MICRO MAX PACKAGE-10

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MAX1846EUB+T 概述

Switching Controller, Current-mode, 500kHz Switching Freq-Max, BICMOS, PDSO10, MICRO MAX PACKAGE-10 稳压芯片 开关式稳压器或控制器

MAX1846EUB+T 规格参数

是否无铅： 不含铅是否Rohs认证： 符合
生命周期：Active零件包装代码：SOIC
包装说明：TSSOP, TSSOP10,.19,20针数：10
Reach Compliance Code：compliantECCN代码：EAR99
HTS代码：8542.31.00.01Factory Lead Time：6 weeks
风险等级：1.46模拟集成电路 - 其他类型：SWITCHING CONTROLLER
控制模式：CURRENT-MODE控制技术：PULSE WIDTH MODULATION
最大输入电压：16.5 V最小输入电压：3 V
标称输入电压：12 VJESD-30 代码：S-PDSO-G10
JESD-609代码：e3长度：3 mm
湿度敏感等级：1功能数量：1
端子数量：10最高工作温度：85 °C
最低工作温度：-40 °C封装主体材料：PLASTIC/EPOXY
封装代码：TSSOP封装等效代码：TSSOP10,.19,20
封装形状：SQUARE封装形式：SMALL OUTLINE, THIN PROFILE, SHRINK PITCH
峰值回流温度（摄氏度）：260认证状态：Not Qualified
座面最大高度：1.1 mm子类别：Switching Regulator or Controllers
表面贴装：YES切换器配置：SINGLE
最大切换频率：500 kHz技术：BICMOS
温度等级：INDUSTRIAL端子面层：Matte Tin (Sn)
端子形式：GULL WING端子节距：0.5 mm
端子位置：DUAL处于峰值回流温度下的最长时间：30
宽度：3 mmBase Number Matches：1

MAX1846EUB+T 数据手册

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EVALUATION KIT AVAILABLE  
MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
General Description  
Features  
● 90% Efficiency  
MAX1846/MAX1847 high-efficiency PWM inverting  
controllers allow designers to implement compact, low-  
noise, negative-output DC-DC converters for telecom  
and networking applications. Both devices operate  
from +3V to +16.5V input and generate -500mV to  
-200V output. To minimize switching noise, both devices  
feature a current-mode, constant-frequency PWM control  
scheme. The operating frequency can be set from 100kHz  
to 500kHz through a resistor.  
● +3.0V to +16.5V Input Range  
● -500mV to -200V Output  
● Drives High-Side P-Channel MOSFET  
● 100kHz to 500kHz Switching Frequency  
● Current-Mode, PWM Control  
● Internal Soft-Start  
● Electrolytic or Ceramic Output Capacitor  
The MAX1846 is available in an ultra-compact 10-pin  
● The MAX1847 also offers:  
®
µMAX package. Operation at high frequency, com-  
Synchronization to External Clock Shutdown  
N-Channel Inverting Flyback Option  
patibility with ceramic capacitors, and inverting topol-  
ogy without transformers allow for a compact design.  
Compatibility with electrolytic capacitors and flexibility to  
operate down to 100kHz allow users to minimize the cost  
of external components. The high-current output drivers  
are designed to drive a P-channel MOSFET and allow the  
converter to deliver up to 30W.  
Ordering Information  
PART  
TEMP RANGE  
-40°C to +85°C  
-40°C to +105°C  
-40°C to +85°C  
-40°C to +85°C  
PIN-PACKAGE  
10 µMAX  
MAX1846EUB  
MAX1846EUB+  
MAX1847EEE  
MAX1847EEE+  
10 µMAX  
16 QSOP  
The MAX1847 features clock synchronization and shut-  
down functions. The MAX1847 can also be configured to  
operate as an inverting flyback controller with an N-channel  
MOSFET and a transformer to deliver up to 70W. The  
MAX1847 is available in a 16-pin QSOP package.  
16 QSOP  
+Denotes a lead(Pb)-free/RoHS-compliant package.  
Typical Operating Circuit  
Current-mode control simplifies compensation and  
provides good transient response. Accurate current-mode  
control and over current protection are achieved through  
low-side current sensing.  
POSITIVE  
V
IN  
P
NEGATIVE  
OUT  
Applications  
V
● Cellular Base Stations  
● Networking Equipment  
● Optical Networking Equipment  
● SLIC Supplies  
VL  
IN  
EXT  
MAX1846  
MAX1847  
● CO DSL Line Driver Supplies  
● Industrial Power Supplies  
● Servers  
COMP  
FREQ  
REF  
CS  
PGND  
● VOIP Supplies  
FB  
GND  
Pin Configurations appear at end of data sheet.  
µMAX is a registered trademark of Maxim Integrated Products, Inc.  
19-2091; Rev 4; 7/16  
MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Absolute Maximum Ratings  
IN, SHDN to GND .................................................-0.3V to +20V  
PGND to GND......................................................-0.3V to +0.3V  
Continuous Power Dissipation (T = +70°C)  
10-Pin µMAX (derate 5.6mW/°C above +70°C)..........444mW  
A
VL to PGND for VIN ≤ 5.7V........................-0.3V to (V + 0.3V)  
VL to PGND for VIN > 5.7V.....................................-0.3V to +6V  
16-Pin QSOP (derate 8.3mW/°C above +70°C)..........696mW  
Operating Temperature Range......................... -40°C to +105°C  
Junction Temperature......................................................+150°C  
Storage Temperature Range............................ -65°C to +150°C  
Lead Temperature (soldering, 10s) .................................+300°C  
Soldering Temperature (reflow)  
IN  
EXT to PGND.............................................-0.3V to (V + 0.3V)  
IN  
REF, COMP to GND....................................-0.3V to (VL + 0.3V)  
CS, FB, FREQ, POL, SYNC to GND......................-0.3V to +6V  
Lead(Pb)-free...............................................................+260°C  
Containing lead(Pb).....................................................+240°C  
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these  
or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect  
device reliability.  
Electrical Characteristics  
(V  
= V = +12V, SYNC = GND, PGND = GND, R  
= 147kΩ ±1%, C  
= 0.47µF, C  
= 0.1µF, T = 0°C to +85°C,  
SHDN  
IN  
FREQ  
VL  
REF A  
unless otherwise noted.)  
PARAMETER  
PWM CONTROLLER  
CONDITIONS  
MIN  
3.0  
TYP  
MAX  
UNITS  
Operating Input Voltage Range  
16.5  
2.95  
2.96  
V
V
-40°C to ~+85°C  
2.8  
2.8  
2.74  
2.74  
60  
V
V
rising  
falling  
IN  
-40°C to ~+105°C  
-40°C to ~+85°C  
-40°C to ~+105°C  
UVLO Threshold  
2.6  
IN  
2.59  
UVLO Hysteresis  
FB Threshold  
mV  
mV  
nA  
No load  
= -0.1V  
-12  
-50  
0
12  
50  
FB Input Current  
V
-6  
FB  
C
= 0.068µF, V  
= -48V,  
COMP  
OUT  
Load Regulation  
Line Regulation  
-1  
0
%
%
I
= 20mA to 200mA (Note 1)  
OUT  
C
= 0.068µF, V  
= -48V,  
COMP  
OUT  
0.04  
V
= +8V to +16.5V, I  
= 100mA  
IN  
OUT  
Current-Limit Threshold  
CS Input Current  
Supply Current  
85  
100  
10  
115  
20  
mV  
µA  
CS = GND  
= -0.1V, V = +3.0V to +16.5V  
V
0.75  
1.2  
mA  
FB  
IN  
SHDN = GND, V = +3.0V to +16.5V  
IN  
Shutdown Supply Current  
10  
25  
µA  
V
= +3.0V to +16.5V  
IN  
REFERENCE AND VL REGULATOR  
REF Output Voltage  
REF Load Regulation  
VL Output Voltage  
VL Load Regulation  
I
= 50µA  
1.236  
3.85  
1.25  
-2  
1.264  
-15  
V
mV  
V
REF  
I
I
I
= 0 to 500µA  
REF  
= 100µA  
4.25  
-20  
4.65  
-60  
VL  
VL  
= 0.1mA to 2.0mA  
mV  
Maxim Integrated  
│ 2  
www.maximintegrated.com  
MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Electrical Characteristics (continued)  
(V  
= V = +12V, SYNC = GND, PGND = GND, R  
= 147kΩ ±1%, C  
= 0.47µF, C  
= 0.1µF, T = 0°C to +85°C,  
SHDN  
IN  
FREQ  
VL  
REF A  
unless otherwise noted.)  
PARAMETER  
OSCILLATOR  
CONDITIONS  
= 500kΩ ±1%  
MIN  
TYP  
MAX  
UNITS  
R
R
R
R
R
R
88  
100  
300  
500  
96  
112  
345  
FREQ  
FREQ  
FREQ  
FREQ  
FREQ  
FREQ  
Oscillator Frequency  
Maximum Duty Cycle  
= 147kΩ ±1%  
= 76.8kΩ ±1%  
= 500kΩ ±1%  
= 147kΩ ±1%  
= 76.8kΩ ±1%  
255  
kHz  
93  
85  
98  
92  
88  
%
80  
SYNC Input Signal Duty-Cycle  
Range  
7
93  
%
Minimum SYNC Input Logic-Low  
Pulse Width  
50  
200  
ns  
SYNC Input Rise/Fall Time  
SYNC Input Frequency Range  
DIGITAL INPUTS  
(Note 2)  
200  
550  
ns  
100  
2.0  
kHz  
POL, SYNC, SHDN Input High  
Voltage  
V
POL, SYNC, SHDN Input Low  
Voltage  
0.45  
V
POL, SYNC Input Current  
POL, SYNC = GND or VL  
20  
-4  
40  
0
µA  
µA  
V
V
= +5V or GND  
= +16.5V  
-12  
SHDN  
SHDN  
SHDN Input Current  
1.5  
6
SOFT-START  
Soft-Start Clock Cycles  
Soft-Start Levels  
1024  
64  
EXT OUTPUT  
EXT Sink/Source Current  
V
= +5V, V  
forced to +2.5V  
1
3
5
A
IN  
EXT  
EXT high or low, tested with 100mA load, V = +5V  
7.5  
12  
IN  
EXT On-Resistance  
Ω
EXT high or low, tested with 100mA load, V = +3V  
IN  
Note 1: Production test correlates to operating conditions.  
Note 2: Guaranteed by design and characterization.  
Maxim Integrated  
│ 3  
www.maximintegrated.com  
MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Electrical Characteristics  
(V  
= V = +12V, SYNC = GND, PGND = GND, R  
= 147kΩ ±1%, C = 0.47µF, C  
= 0.1µF, T = -40°C to +85°C,  
SHDN  
IN  
FREQ  
VL  
REF A  
unless otherwise noted.) (Note 3)  
PARAMETER  
PWM CONTROLLER  
CONDITIONS  
MIN  
MAX  
UNITS  
Operating Input Voltage Range  
3.0  
16.5  
2.95  
V
V
V
V
rising  
falling  
IN  
IN  
UVLO Threshold  
FB Threshold  
2.6  
-20  
-50  
No load  
+20  
+50  
mV  
-40°C to ~+85°C  
-40°C to ~+105°C  
= -48V,  
FB Input Current  
Load Regulation  
V
= -0.1V  
nA  
%
FB  
-150  
+150  
C
= 0.068µF, V  
OUT  
COMP  
-2  
0
I
= 20mA to 200mA (Note 1)  
OUT  
Current Limit Threshold  
CS Input Current  
85  
115  
20  
mV  
µA  
CS = GND  
= -0.1V, V = +3.0V to +16.5V  
Supply Current  
V
1.2  
25  
mA  
µA  
FB  
IN  
Shutdown Supply Current  
SHDN = GND, V = +3.0V to +16.5V  
IN  
REFERENCE AND VL REGULATOR  
REF Output Voltage  
REF Load Regulation  
VL Output Voltage  
VL Load Regulation  
OSCILLATOR  
I
I
I
I
= 50µA  
1.225  
3.85  
1.275  
-15  
V
mV  
V
REF  
REF  
= 0 to 500µA  
= 100µA  
4.65  
-60  
VL  
VL  
= 0.1mA to 2.0mA  
mV  
R
R
R
R
= 500kΩ ±1%  
= 147kΩ ±1%  
= 500kΩ ±1%  
= 147kΩ ±1%  
84  
255  
93  
116  
345  
98  
FREQ  
FREQ  
FREQ  
FREQ  
Oscillator Frequency  
Maximum Duty Cycle  
kHz  
%
84  
93  
SYNC Input Signal Duty-Cycle  
Range  
7
93  
%
Minimum SYNC Input Logic Low  
Pulse Width  
200  
ns  
SYNC Input Rise/Fall Time  
SYNC Input Frequency Range  
DIGITAL INPUTS  
(Note 2)  
200  
550  
ns  
100  
2.0  
kHz  
POL, SYNC, SHDN Input High  
Voltage  
V
V
POL, SYNC, SHDN Input Low  
Voltage  
0.45  
Maxim Integrated  
│ 4  
www.maximintegrated.com  
MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Electrical Characteristics (continued)  
(V  
= V = +12V, SYNC = GND, PGND = GND, R  
= 147kΩ ±1%, C = 0.47µF, C  
= 0.1µF, T = -40°C to +85°C,  
SHDN  
IN  
FREQ  
VL  
REF A  
unless otherwise noted.) (Note 3)  
PARAMETER  
CONDITIONS  
POL, SYNC = GND or VL  
MIN  
MAX  
40  
0
UNITS  
POL, SYNC Input Current  
µA  
V
V
= +5V or GND  
-12  
SHDN  
SHDN  
SHDN Input Current  
µA  
= +16.5V  
6
EXT OUTPUT  
-40°C to ~+85°C  
-40°C to ~+105°C  
7.5  
8.75  
12  
EXT high or low, I  
100mA, V = +5V  
IN  
=
EXT  
EXT On-Resistance  
Ω
EXT high or low, I  
= 100mA, V = +3V  
IN  
EXT  
Note 3: Parameters to -40°C are guaranteed by design and characterization.  
Typical Operating Characteristics  
(Circuit references are from Table 1 in the Main Application Circuits section, C = 0.47µF, C  
= 0.1°F, T = +25°C, unless otherwise  
VL  
REF A  
noted.)  
EFFICIENCY vs. LOAD CURRENT  
EFFICIENCY vs. LOAD CURRENT  
EFFICIENCY vs. LOAD CURRENT  
100  
100  
100  
V
IN  
= 5V  
90  
80  
90  
80  
70  
60  
50  
40  
30  
20  
90  
80  
70  
60  
50  
40  
30  
20  
V
IN  
= 12V  
V
IN  
= 5V  
70  
60  
V
IN  
= 3.3V  
V
IN  
= 16.5V  
V
IN  
= 3V  
V
IN  
= 16.5V  
50  
40  
30  
20  
10  
0
10  
0
10  
0
V
= -48V  
OUT  
V
= -5V  
V
= -12V  
APPLICATION CIRCUIT A  
10 100  
APPLICATION CIRCUIT B  
APPLICATION CIRCUIT C  
10  
LOAD CURRENT (mA)  
OUT  
OUT  
1
1000  
10,000  
1
10  
100  
1000  
10,000  
1
100  
1000  
LOAD CURRENT (mA)  
LOAD CURRENT (mA)  
SUPPLY CURRENT  
REFERENCE VOLTAGE  
vs. TEMPERATURE  
OUTPUT VOLTAGE LOAD REGULATION  
vs. SUPPLY VOLTAGE  
1.6  
-11.90  
-11.92  
-11.94  
-11.96  
-11.98  
-12.00  
-12.02  
-12.04  
-12.06  
-12.08  
-12.10  
1.262  
1.258  
1.254  
1.250  
1.246  
1.242  
1.238  
1.4  
1.2  
1.0  
0.8  
0.6  
0.4  
0.2  
0
V
= -0.1V  
FB  
APPLICATION CIRCUIT B  
100 200 300  
LOAD CURRENT (mA)  
V = 5V  
IN  
0
400  
500  
600  
0
2
4
6
8
10 12 14 16  
-40 -20  
0
20  
40  
60  
80 100  
V
IN  
(V)  
TEMPERATURE (C)  
Maxim Integrated  
│ 5  
www.maximintegrated.com  
MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Typical Operating Characteristics (continued)  
(Circuit references are from Table 1 in the Main Application Circuits section, C = 0.47µF, C  
= 0.1°F, T = +25°C, unless otherwise  
VL  
REF A  
noted.)  
VL VOLTAGE  
vs. TEMPERATURE  
REFERENCE LOAD REGULATION  
VL LOAD REGULATION  
4.27  
1.260  
1.255  
1.250  
1.245  
1.240  
4.340  
4.300  
4.260  
4.220  
4.180  
4.140  
4.26  
4.25  
4.24  
4.23  
4.22  
I
= 0  
VL  
4.100  
0
100  
200  
I
300  
(A)  
400  
500  
0
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0  
(mA)  
-40 -20  
0
20  
40  
60  
80 100  
I
TEMPERATURE (C)  
REF  
VL  
OPERATING CURRENT  
vs. TEMPERATURE  
SWITCHING FREQUENCY  
SHUTDOWN SUPPLY CURRENT  
vs. TEMPERATURE  
vs. R  
FREQ  
16  
14  
12  
10  
8
14  
12  
10  
8
500  
400  
300  
200  
100  
0
A
A: V = 3V, V  
= -12V  
IN  
OUT  
V
IN  
= 10V  
V
IN  
= 16.5V  
APPLICATION CIRCUIT A  
B: V = 5V, V = -5V  
V
IN  
= 3V  
IN  
IN  
OUT  
6
C: V = 16.5V, V  
= -5V  
OUT  
6
4
B
4
2
2
0
C
0
-40 -20  
0
20  
60  
80 100  
0
100  
200  
300  
400  
500  
600  
40  
-40 -20  
0
20  
40  
60  
80 100  
TEMPERATURE (C)  
R
FREQ  
(k)  
TEMPERATURE (C)  
SWITCHING FREQUENCY  
vs. TEMPERATURE  
EXT RISE/FALL TIME  
vs. CAPACITANCE  
EXITING SHUTDOWN  
MAX1846/7 toc15  
302  
301  
300  
299  
298  
160  
140  
120  
100  
80  
5V/di  
SHDN  
0
FALL TIME  
V
OUT  
5V/di  
1A/di  
60  
297  
296  
40  
RISE TIME  
20  
I
295  
294  
L
R
= 147k 1%  
V
= 12V  
FREQ  
IN  
0
-40 -20  
0
20  
60  
80 100  
0
2000  
4000  
6000  
8000 10,000  
40  
APPLICATION CIRCUIT B  
1ms/div  
TEMPERATURE (C)  
CAPACITANCE (pF)  
Maxim Integrated  
│ 6  
www.maximintegrated.com  
MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Typical Operating Characteristics (continued)  
(Circuit references are from Table 1 in the Main Application Circuits section, C = 0.47µF, C  
= 0.1°F, T = +25°C, unless otherwise  
A
VL  
REF  
noted.)  
HEAVY-LOAD SWITCHING  
WAVEFORM  
ENTERING SHUTDOWN  
MAX1846/7 toc17  
MAX1846/7 toc16  
SHDN  
0
V
OUT  
100mV/div  
1A/div  
5V/div  
5V/div  
I
L
V
OUT  
LX  
10V/div  
1A/div  
I
L
APPLICATION CIRCUIT B  
1s/div  
APPLICATION CIRCUIT B  
1ms/div  
I
= 600mA  
LOAD  
LIGHT-LOAD SWITCHING  
WAVEFORM  
MAX1846/7 toc18  
OUT  
100mV/d  
1A/div  
I
L
LX  
10V/div  
APPLICATION CIRCUIT B  
1s/div  
I
= 50mA  
LOAD  
LOAD-TRANSIENT RESPONSE  
LOAD-TRANSIENT RESPONSE  
MAX1846/7 toc20  
MAX1846/7 toc19  
I
LOAD  
LOAD  
V
V
OUT  
OUT  
200mV/div  
500mA/div  
500mV/  
1A/div  
I
I
L
L
APPLICATION CIRCUIT B  
2ms/div  
APPLICATION CIRCUIT C  
400s/div  
I
= 4mA to 100mA  
LOAD  
I
= 10mA to 400mA  
LOAD  
Maxim Integrated  
│ 7  
www.maximintegrated.com  
MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Pin Description  
PIN  
NAME  
FUNCTION  
MAX1846  
MAX1847  
Sets polarity of the EXT pin. Connect POL to GND to set EXT for use with an external  
PMOS high-side FET. Connect POL to VL to set EXT for use with an external NMOS low-  
side FET in transformer-based applications.  
—
1
1
2
POL  
VL  
VL Low-Dropout Regulator. Connect 0.47µF ceramic capacitor from VL to GND.  
Oscillator Frequency Set Input. Connect a resistor (R  
) from FREQ to GND to set the  
FREQ  
internal oscillator frequency from 100kHz (R  
= 500kW) to 500kHz (R  
= 76.8kW).  
FREQ  
FREQ  
2
3
FREQ  
R
is still required if an external clock is used at SYNC. See Setting the Operating  
FREQ  
Frequency section.  
Compensation Node for Error Amp/Integrator. Connect a series resistor/capacitor network  
from COMP to GND for loop compensation. See Design Procedure.  
3
4
4
5
COMP  
REF  
1.25V Reference Output. REF can source up to 500µA. Bypass with a 0.1µF ceramic  
capacitor from REF to GND.  
Feedback Input. Connect FB to the center of a resistor-divider connected between the  
output and REF. The FB threshold is 0.  
5
6
7, 9  
8
FB  
—
—
N.C.  
No Connection  
Shutdown Control. Drive SHDN low to turn off the DC-DC controller. Drive high or connect  
to IN for normal operation.  
SHDN  
6
7
10, 11  
12  
GND  
Analog Ground. Connect to PGND.  
PGND  
Negative Rail for EXT Driver and Negative Current-Sense Input. Connect to GND.  
Positive Current-Sense Input. Connect a current-sense resistor (R ) between CS and  
CS  
PGND.  
8
13  
CS  
9
14  
15  
EXT  
IN  
External MOSFET Gate-Driver Output. EXT swings from IN to PGND.  
Power-Supply Input  
10  
Operating Frequency Synchronization Control. Drive SYNC low or connect to GND to set  
the internal oscillator frequency with R  
. Drive SYNC with a logic-level clock input  
FREQ  
—
16  
SYNC  
signal to externally set the converter’s operating frequency. DC-DC conversion cycles  
initiate on the rising edge of the input clock signal. Note that when driving SYNC with an  
external signal, R  
must still be connected to FREQ.  
FREQ  
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MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Typical Application Circuit  
3 x 22µF  
10V  
V
IN  
+3V to +5.5V  
22kΩ  
FDS6375  
CMSH5-40  
V
OUT  
2
VL  
15  
0.47µF  
-12V AT 400mA  
IN  
47µF  
47µF  
16V  
8
10µH  
16V  
SHDN  
SYNC  
14  
13  
DO5022P-103  
EXT  
16  
CS  
N.C.  
SANYO  
16TPB47M  
220pF  
MAX1847  
7, 9  
0.02Ω  
1W  
R1  
4
3
5
95.3kΩ  
COMP  
FREQ  
REF  
12  
6
1%  
PGND  
10kΩ  
0.22µF  
R2  
10.0kΩ  
1%  
FB  
150kΩ  
POL  
1
GND  
1200pF  
10, 11  
0.1µF  
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MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Functional Diagram  
IN  
EXT  
STARTUP  
CIRCUITRY  
SHDN  
MAX1847 ONLY  
PGND  
EXT DRIVER  
VL  
VL  
REGULATOR  
UNDER-  
VOLTAGE  
LOCK OUT  
MAX1846  
MAX1847  
CONTROL  
CIRCUITRY  
POL  
SYNC  
MAX1847 ONLY  
OSCILLATOR  
FREQ  
ERROR  
COMPARATOR  
COMP  
CS  
FB  
G
M
CURRENT-  
SENSE  
AMPLIFIER  
ERROR  
AMPLIFIER  
PGND  
SOFT-START  
REFERENCE  
X3.3  
SLOPE  
COMP  
REF  
GND  
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MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
2) when exiting shutdown with power already applied, and  
3) when exiting undervoltage lockout.  
Detailed Description  
The MAX1846/MAX1847 current-mode PWM controllers  
use an inverting topology that is ideal for generating  
output voltages from -500mV to -200V. Features include  
shutdown, adjustable internal operating frequency or  
synchronization to an external clock, soft-start, adjustable  
current limit, and a wide (+3V to +16.5V) input range.  
Shutdown (MAX1847 only)  
The MAX1847 shuts down to reduce the supply current  
to 10µA when SHDN is low. In this mode, the internal  
reference, error amplifier, comparators, and biasing  
circuitry turn off. The EXT output becomes high imped-  
ance and the external pullup resistor connected to EXT  
PWM Controller  
pulls V  
to V , turning off the P-channel MOSFET.  
EXT  
IN  
The architecture of the MAX1846/MAX1847 current-mode  
PWM controller is a BiCMOS multi-input system that  
simultaneously processes the output-error signal, the  
current-sense signal, and a slope-compensation ramp  
(Functional Diagram). Slope compensation prevents sub-  
harmonic oscillation, a potential result in current-mode  
regulators operating at greater than 50% duty cycle. The  
controller uses fixed-frequency, current-mode operation  
where the duty ratio is set by the input-to-output voltage  
ratio. The current-mode feedback loop regulates peak  
inductor current as a function of the output error signal.  
When in shutdown mode, the converter's output goes to 0.  
Frequency Synchronization  
(MAX1847 only)  
The MAX1847 is capable of synchronizing its switching  
frequency with an external clock source. Drive SYNC  
with a logic-level clock input signal to synchronize the  
MAX1847. A switching cycle starts on the rising edge  
of the signal applied to SYNC. Note that the frequen-  
cy of the signal applied to SYNC must be higher than  
the default frequency set by R  
. This frequency  
FREQ  
is required so that the internal clock does not start a  
switching cycle prematurely. If SYNC is inactive for an  
entire clock cycle of the internal oscillator, the internal  
oscillator takes over the switching operation. Choose  
Internal Regulator  
The MAX1846/MAX1847 incorporate an internal low-  
dropout regulator (LDO). This LDO has a 4.25V output  
and powers all MAX1846/MAX1847 functions (excluding  
EXT) for the primary purpose of stabilizing the perfor-  
mance of the IC over a wide input voltage range (+3V to  
+16.5V). The input to this regulator is connected to IN,  
and the dropout voltage is typically 100mV, so that when  
R
such that f  
= 0.9 5 f  
.
FREQ  
OSC  
SYNC  
EXT Polarity (MAX1847 only)  
The MAX1847 features an option to utilize an N-channel  
MOSFET configuration, rather than the typical p-channel  
MOSFET configuration (Figure 1). In order to drive the  
different polarities of these MOSFETs, the MAX1847  
is capable of reversing the phase of EXT by 180  
degrees. When driving a P-channel MOSFET, connect  
POL to GND. When driving an n-channel MOSFET,  
connect POL to VL. These POL connections ensure the  
proper polarity for EXT. For design guidance in regard to  
this application, refer to the MAX1856 data sheet.  
V
IN  
is less than 4.35V, VL is typically V minus 100mV.  
IN  
When the LDO is in dropout, the MAX1846/MAX1847 still  
operate with V as low as 3V. For best performance, it is  
IN  
recommended to connect VL to IN when the input supply  
is less than 4.5V.  
Undervoltage Lockout  
The MAX1846/MAX1847 have an undervoltage lockout  
circuit that monitors the voltage at VL. If VL falls below  
the UVLO threshold (2.8V typ), the control logic turns  
the P-channel FET off (EXT high impedance). The rest  
of the IC circuitry is still powered and operating. When  
VL increases to 60mV above the UVLO threshold, the IC  
resumes operation from a start up condition (soft-start).  
Design Procedure  
Initial Specifications  
In order to start the design procedure, a few parameters  
must be identified: the minimum input voltage expect-  
ed (V  
), the maximum input voltage expected  
IN(MIN)  
Soft-Start  
(V  
), the desired output voltage (V  
), and the  
IN(MAX)  
OUT  
The MAX1846/MAX1847 feature a “digital” soft-start  
that is preset and requires no external capacitor. Upon  
startup, the FB threshold decrements from the refer-  
expected maximum load current (I  
).  
LOAD  
Calculate the Equivalent Load Resistance  
This is a simple calculation used to shorten the verifica-  
tion equations:  
ence voltage to 0 in 64 steps over 1024 cycles of f  
OSC  
or f  
. See the Typical Operating Characteristics for  
SYNC  
a scope picture of the soft-start operation. Soft-start is  
implemented: 1) when power is first applied to the IC,  
R
LOAD  
= V  
/ I  
OUT LOAD  
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MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
V
IN  
+12V  
12µF  
25V  
VP1-0190  
12.2µH  
1:4  
CMR1U-02  
V
OUT  
1
POL  
2
VL  
15  
IN  
0.47µF  
-48V AT 100mA  
IRLL2705  
8
12µF  
100V  
470Ω  
14  
SHDN  
SYNC  
EXT  
CS  
16  
100pF  
100V  
13  
MAX1847  
7, 9  
0.05Ω  
0.5W  
383kΩ  
N.C.  
1%  
4
3
COMP  
FREQ  
12  
6
PGND  
270kΩ  
0.033µF  
5
REF  
FB  
150kΩ  
10.0kΩ  
1%  
GND  
10, 11  
1800pF  
0.1µF  
Figure 1. Using an N-Channel MOSFET (MAX1847 only)  
1.25V and the regulation voltage for FB is nominally 0.  
The load presented to the reference by the feedback  
resistors must be less than 500µA to guarantee that  
Calculate the Duty Cycle  
The duty cycle is the ratio of the on-time of the MOSFET  
switch to the oscillator period. It is determined by the  
ratio of the input voltage to the output voltage. Since  
the input voltage typically has a range of operation, a  
V
is in regulation (see Electrical Characteristics  
REF  
Table). Conversely, the current through the feedback  
resistors must be large enough so that the leakage  
current of the FB input (50nA) is insignificant. Therefore,  
minimum (D  
) and maximum (D  
MIN  
) duty cycle is  
MAX  
calculated by:  
select R2 so that I is between 50µA and 250µA.  
R2  
−V  
+ V  
D
OUT  
− V  
I
= V  
/ R2  
=
R2  
REF  
MIN  
V
− V  
− V  
+ V  
OUT D  
IN(MAX)  
SW  
LIM  
where V  
= 1.25V. A typical value for R2 is 10kW.  
REF  
Once R2 is selected, calculate R1 with the following  
equation:  
−V  
+ V  
D
OUT  
− V  
D
=
MAX  
V
− V  
− V  
+ V  
OUT D  
IN(MIN)  
SW  
LIM  
R1 = R2 x (-V  
/ V  
)
OUT  
REF  
where V is the forward drop across the output diode,  
D
Setting the Operating Frequency  
V
SW  
is the drop across the external FET when on,  
The MAX1846/MAX1847 are capable of operating at  
switching frequencies from 100kHz to 500kHz. Choice  
of operating frequency depends on a number of factors:  
and V  
is the current-limit threshold. To begin with,  
LIM  
assume V = 0.5V for a Schottky diode, V  
= 100mV,  
D
SW  
and V  
= 100mV. Remember that V  
is negative  
LIM  
OUT  
when using this formula.  
1) Noise considerations may dictate setting (or  
synchronizing) f  
frequency or band of frequencies, particularly in RF  
applications.  
above or below a certain  
OSC  
Setting the Output Voltage  
The output voltage is set using two external resistors to  
form a resistive-divider to FB between the output and  
REF (refer to R1 and R2 in Figure 1). V  
is nominally  
REF  
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High-Efficiency, Current-Mode,  
Inverting PWM Controller  
2) Higher frequencies allow the use of smaller value  
(hence smaller size) inductors and capacitors.  
to the rate set by R  
. Choose R  
such that  
FREQ  
FREQ  
f
= 0.9 x f  
.
OSC  
SYNC  
3) Higher frequencies consume more operating power  
both to operate the IC and to charge and discharge  
the gate at the external FET, which tends to reduce the  
efficiency at light loads.  
Choosing Inductance Value  
The inductance value determines the operation of the  
current-mode regulator. Except for low-current applica-  
tions, most circuits are more efficient and economical  
operating in continuous mode, which refers to continu-  
ous current in the inductor. In continuous mode there is  
a trade-off between efficiency and transient response.  
Higher inductance means lower inductor ripple current,  
lower peak current, lower switching losses, and, there-  
fore, higher efficiency. Lower inductance means higher  
inductor ripple current and faster transient response. A  
reasonable compromise is to choose the ratio of inductor  
ripple current to average continuous current at mini-  
mum duty cycle to be 0.4. Calculate the inductor ripple  
with the following formula:  
4) Higher frequencies may exhibit lower overall efficiency  
due to more transition losses in the FET; however, this  
shortcoming can often be nullified by trading some  
of the inductor and capacitor size benefits for lower-  
resistance components.  
5) High-duty-cycle applications may require lower fre-  
quencies to accommodate the controller minimum  
off-time of 0.4µs. Calculate the maximum oscillator  
frequency with the following formula:  
V
− V  
− V  
LIM  
IN(MIN)  
− V  
SW  
− V  
f
=
OSC(MAX)  
V
− V  
+ V  
OUT D  
IN(MIN)  
1
SW  
LIM  
I
=
RIPPLE  
×
0.4 ×I  
× V  
− V  
− V  
− V  
+ V  
OUT D  
(
)
LOAD(MAX)  
IN(MAX)  
SW  
LIM  
t
OFF(MIN)  
V
− V  
− V  
(
)
IN(MAX)  
SW  
LIM  
Remember that V  
is negative when using this formula.  
OUT  
Then calculate an inductance value:  
When running at the maximum oscillator frequency  
(f ) and maximum duty cycle (D ), do  
L = (V / I ) x (D  
/ f )  
MIN OSC  
IN(MAX) RIPPLE  
OSCILLATOR  
MAX  
not exceed the minimum value of D  
stated in the  
Choose the closest standard value. Once again, remem-  
ber that V is negative when using this formula.  
MAX  
Electrical Characteristics table. For designs that exceed  
the D and f , an autotransformer can reduce  
OUT  
MAX  
OSC(MAX)  
Determining Peak Inductor Current  
the duty cycle and allow higher operating frequencies.  
The peak inductor current required for a particular output  
is:  
The oscillator frequency is set by a resistor, RFREQ,  
which is connected from FREQ to GND. The relation-  
ship between fOSC (in Hz) and RFREQ (in W) is slightly  
nonlinear, as illustrated in the Typical Operating  
Characteristics. Choose the resistor value from the graph  
and check the oscillator frequency using the following  
formula:  
I
= I  
+ (I  
/ 2)  
LPEAK  
LDC  
LPP  
where I  
is the average DC inductor current and I  
LDC  
LPP  
and  
is the inductor peak-to-peak ripple current. The I  
LDC  
I
terms are determined as follows:  
LPP  
I
LOAD  
I
I
=
=
LDC  
LPP  
1 −D  
(
)
1
MAX  
f
=
OSC  
2  
−7  
5.21×10  
+ 1.92×10−11 × R  
− 4.86×10−19 × R  
(
)
(
)
(
)
(
)
FREQ  
FREQ  
V
− V  
− V  
x D  
MAX  
(
SW  
LIM  
)
IN MIN  
(
)
L x f  
OSC  
External Synchronization (MAX1847 only)  
The SYNC input provides external-clock synchroniza-  
tion (if desired). When SYNC is driven with an exter-  
nal clock, the frequency of the clock directly sets the  
MAX1847's switching frequency. A rising clock edge on  
SYNC is interpreted as a synchronization input. If the  
sync signal is lost, the internal oscillator takes over at  
the end of the last cycle, and the frequency is returned  
where L is the selected inductance value. The  
saturation rating of the selected inductor should meet  
or exceed the calculated value for I  
, although  
LPEAK  
most coil types can be operated up to 20% over their  
saturation rating without difficulty. In addition to the sat-  
uration criteria, the inductor should have as low a series  
resistance as possible. For continuous inductor current,  
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the power loss in the inductor resistance (PLR) is  
approximated by:  
Power MOSFET Selection  
The MAX1846/MAX1847 drive a wide variety of P-channel  
power MOSFETs (PFETs). The best performance,  
especially with input voltages below 5V, is achieved with  
low-threshold PFETs that specify on-resistance with  
2
I
LOAD  
P
~R x  
L
LR  
I − D  
MAX   
a gate-to-source voltage (V ) of 2.7V or less. When  
GS  
where R is the inductor series resistance.  
L
selecting a PFET, key parameters include:  
Once the peak inductor current is calculated, the  
● Total gate charge (Q )  
G
current sense resistor, R , is determined by:  
CS  
● Reverse transfer capacitance (C  
)
RSS  
R
= 85mV / I  
LPEAK  
CS  
● On-resistance (  
)
RDS(ON)  
For high peak inductor currents (>1A), Kelvin-sensing  
connections should be used to connect CS and PGND  
to RCS. Connect PGND and GND together at the ground  
● Maximum drain-to-source voltage (V  
)
DS(MAX)  
● Minimum threshold voltage (V  
)
TH(MIN)  
side of R . A lowpass filter between R  
be required to prevent switching noise from tripping the  
current-sense comparator at heavy loads. Connect a  
100W resistor between CS and the high side of R , and  
connect a 1000pF capacitor between CS and GND.  
and CS may  
CS  
CS  
At high-switching rates, dynamic characteristics (para-  
meters 1 and 2 above) that predict switching losses  
may have more impact on efficiency than R  
,
DS(ON)  
CS  
which predicts DC losses. Q includes all capacitance  
G
associated with charging the gate. In addition, this  
parameter helps predict the current needed to drive the  
gate at the selected operating frequency. The power  
MOSFET in an inverting converter must have a high  
enough voltage rating to handle the input voltage plus  
the magnitude of the output voltage and any spikes  
induced by leakage inductance and ringing.  
Checking Slope-Compensation Stability  
In  
a
current-mode regulator, the cycle-by-cycle  
stability is dependent on slope compensation to prevent  
subharmonic oscillation at duty cycles greater than  
50%. For the MAX1846/MAX1847, the internal slope  
compensation is optimized for a minimum inductor value  
An RC snubber circuit across the drain to ground might be  
required to reduce the peak ringing and noise.  
(L  
) with respect to duty cycle. For duty cycles greater  
MIN  
then 50%, check stability by calculating LMIN using the  
following equation:  
Choose R  
specified at V  
< V to be  
IN(MIN)  
DS(ON)(MAX)  
GS  
one to two times R . Verify that V  
< V  
CS  
IN(MAX)  
GS(MAX)  
= V  
(
/M  
S
L
xR  
)
MIN  
IN(MIN)  
CS  
and V  
> V  
- V  
+ V . Choose the rise-  
OUT D  
DS(MAX)  
IN(MAX)  
and-fall times (t , t ) to be less than 50ns.  
R
F
(
)
x
2 xD  
− 1 / 1− D  
) (  
MAX  
MAX  
Output Capacitor Selection  
where V  
is the minimum expected input voltage,  
IN(MIN)  
The output capacitor (C  
) does all the filtering in an  
OUT  
M
is the Slope Compensation Ramp (41 mV/µs) and  
s
inverting converter. The output ripple is created by the  
variations in the charge stored in the output capacitor with  
each pulse and the voltage drop across the capacitor’s  
equivalent series resistance (ESR) caused by the current  
into and out of the capacitor. There are two properties  
of the output capacitor that affect ripple voltage: the  
capacitance value, and the capacitor’s ESR. The output  
ripple due to the output capacitor’s value is given by:  
D
is the maximum expected duty cycle. If L  
is  
MAX  
MIN  
larger than L, increase the value of L to the next standard  
value that is larger than L  
tion stability.  
to ensure slope compensa-  
MIN  
Choosing the Inductor Core  
Choosing the most cost-effective inductor usually requires  
optimizing the field and flux with size. With higher output  
voltages the inductor may require many turns, and this  
V
= (I  
× D  
× T  
) / C  
OSC OUT  
RIPPLE-C  
LOAD  
MAX  
can drive the cost up. Choosing an inductor value at L  
MIN  
The output ripple due to the output capacitor’s ESR is  
given by:  
can provide a good solution if discontinuous inductor  
current can be tolerated. Powdered iron cores can pro-  
vide the most economical solution but are larger in size  
than ferrite.  
V
= I × R  
LPP ESR  
RIPPLE-R  
These two ripple voltages are additive and the total output  
ripple is:  
V
= V  
+ V  
RIPPLE-T  
RIPPLE-C RIPPLE-R  
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The ESR-induced ripple usually dominates this last  
equation, so typically output capacitor selection is based  
mostly upon the capacitor's ESR, voltage rating, and  
ripple current rating. Use the following formula to deter-  
mine the maximum ESR for a desired output ripple volt-  
2
− 1−D  
x V  
−V  
×L  
xR  
LOAD  
(
)
(
)
MAX  
IN(MIN)  
OUT  
Z
=
RHP  
2πxV  
(
)
OUT  
The calculations for p  
applications where V  
are very complex. For most  
does not exceed -48V (in a  
OUT2  
OUT  
age (V  
):  
RIPPLE-D  
negative sense), the p  
will not be lower than 1/8th  
OUT2  
R
= V  
/ I  
ESR  
RIPPLE-D LPP  
of the oscillator frequency and is generally at a higher  
frequency than z . Therefore:  
Select a capacitor with ESR rating less than R  
. The  
ESR  
RHP  
value of this capacitor is highly dependent on dielectric  
type, package size, and voltage rating. In general, when  
choosing a capacitor, it is recommended to use low-ESR  
capacitor types such as ceramic, organic, or tanta-  
lum capacitors. Ensure that the selected capacitor has  
sufficient margin to safely handle the maximum RMS  
ripple current.  
p
≥ 0.125 × f  
OSC  
OUT2  
A pole is created by the output capacitor and the load  
resistance. This pole must also be compensated and its  
center frequency is given by the formula:  
p
= 1 / (2π × R  
× C  
)
OUT1  
LOAD  
OUT  
Finally, there is a zero introduced by the ESR of the  
output capacitor. This zero is determined from the follow-  
ing equation:  
For continuous inductor current the maximum RMS ripple  
current in the output filter capacitor is:  
I
2
LOAD  
zESR = 1 / (2π × C  
× R  
)
OUT  
ESR  
I
=
x D  
−D  
MAX MAX  
RMS  
I−D  
MAX  
Calculating the Required Pole Frequency  
To ensure stability of the MAX1846/MAX1847, the gain  
of the error amplifier must roll-off the total loop gain to  
Choosing Compensation Components  
The MAX1846/MAX1847 are externally loop-compensat-  
ed devices. This feature provides flexibility in designs to  
accommodate a variety of applications. Proper compen-  
sation of the control loop is important to prevent excessive  
output ripple and poor efficiency caused by instability. The  
goal of compensation is to cancel unwanted poles and  
zeros in the DC-DC converter’s transfer function created  
by the power-switching and filter elements. More precise-  
ly, the objective of compensation is to ensure stability by  
ensuring that the DC-DC converter’s phase shift is less  
than 180° by a safe margin, at the frequency where the  
loop gain falls below unity. One method for ensuring ade-  
quate phase margin is to introduce corresponding zeros  
and poles in the feedback network to approximate a sin-  
gle-pole response with a -20dB/decade slope all the way  
to unity-gain crossover.  
1 before Z  
or P  
occurs. First, calculate the DC  
RHP  
OUT2  
open-loop gain A  
:
DC  
BxG xR x(1−D  
)R  
M
O
A
MAX LOAD  
A
=
DC  
xR  
CS  
CS  
where:  
A
is the current-sense amplifier gain = 3.3  
CS  
B is the feedback-divider attenuation factor =  
R2  
R1+ R2  
G
is the error-amplifier transconductance =  
M
400 µA/V  
Calculating Poles and Zeros  
R
R
is the error-amplifier output resistance = 3 MW  
O
The MAX1846/MAX1847 current-mode architecture takes  
the double pole caused by the inductor and output  
capacitor and shifts one of these poles to a much higher  
frequency to make loop compensation easier. To compen-  
sate these devices, we must know the center frequencies  
is the selected current-sense resistor  
CS  
Determining the Compensation Component Values  
Select a unity-gain crossover frequency (f ), which is  
CROS  
lower than z  
and p  
and higher than p  
. Using  
OUT1  
RHP  
OUT2  
of the right-half plane zero (z  
) and the higher frequen-  
RHP  
f
, calculate the compensation resistor (R  
).  
COMP  
CROS  
cy pole (p  
). Calculate the z  
OUT2  
frequency with the  
RHP  
following formula:  
f
xR  
O
CROS  
R
=
COMP  
A
xP  
− f  
DC  
OUT1 CROS  
Maxim Integrated  
│ 15  
www.maximintegrated.com  
MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Select the next smaller standard value of resistor and  
then calculate the compensation capacitor required to  
Applications Information  
Maximum Output Power  
cancel out the output-capacitor-induced pole (P  
determined previously.  
)
OUT1  
The maximum output power that the MAX1846/MAX1847  
can provide depends on the maximum input power  
available and the circuit's efficiency:  
1
C
=
COMP  
6.28 xP  
xR  
COMP  
P
= Efficiency × P  
IN(MAX)  
OUT1  
OUT(MAX)  
Furthermore, the efficiency and input power are both  
functions of component selection. Efficiency losses can  
be divided into three categories: 1) resistive losses across  
the inductor, MOSFET on-resistance, current-sense resis-  
tor, rectification diode, and the ESR of the input and out-  
put capacitors; 2) switching losses due to the MOSFET's  
transition region, and charging the MOSFET's gate  
capacitance; and 3) inductor core losses. Typically,  
80% efficiency can be assumed for initial calculations.  
The required input power depends on the inductor  
current limit, input voltage, output voltage, output current,  
inductor value, and the switching frequency. The max-  
imum output power is approximated by the following  
formula:  
Choose the next larger standard value of capacitor.  
In order for p to compensate the loop, the open-  
COMP  
loop gain must reach unity at a lower frequency than the  
right-half-plane zero or the second output pole, whichever  
is lower in frequency. If the second output pole and the  
right-half-plane zero are close together in frequency, the  
higher resulting phase shift at unity gain may require  
a lower crossover frequency. For duty cycles greater  
than 50%, slope compensation reduces A , reducing  
DC  
the actual crossover frequency from f  
. It is also a  
CROS  
good practice to reduce noise on COMP with a capacitor  
(C ) to ground. To avoid adding extra phase margin  
COMP2  
at the crossover, the capacitor (C  
) should roll-off  
COMP2  
noise at five times the crossover frequency. The value for  
P
= [V - (V  
+ I  
x R  
)] x I  
LIM  
- V  
SW LIM  
x
MAX  
IN  
LIM  
LIM  
DS(ON)  
IN  
C
can be found using:  
COMP2  
[1 - (LIR / 2)] x [(-V  
+ V ) / (V - V  
OUT  
D
- V  
+ V )]  
OUT  
D
R
+ R  
COMP  
O
where I  
is the peak current limit and LIR is the inductor  
current-ripple ratio and is calculated by:  
C
=
LIM  
COMP2  
5 x 6.28 x f  
x R x R  
O COMP  
CROS  
LIR = I / I  
LPP LDC  
It might require a couple iterations to obtain a suitable  
combination of compensation components.  
Again, remember that V  
for the MAX1846/MAX1847  
OUT  
is negative.  
Finally, the zero introduced by the output capacitor's  
ESR must be compensated. This compensation is  
accomplished by placing a capacitor between REF  
and FB creating a pole directly in the feedback loop.  
Calculate the value of this capacitor using the frequency  
Diode Selection  
The MAX1846/MAX1847's high-switching frequency  
demands a high-speed rectifier. Schottky diodes are  
recommended for most applications because of their  
fast recovery time and low forward voltage. Ensure that  
the diode's average current rating exceeds the peak  
inductor current by using the diode manufacturer's data.  
Additionally, the diode's reverse breakdown voltage must  
of z  
and the selected feedback resistor values with  
ESR  
the formula:  
R +R  
1
2
2
C
=R  
xC  
x
OUT  
FB  
ESR  
R xR  
1
exceed the potential difference between V  
and the  
OUT  
input voltage plus the leakage-inductance spikes. For  
high output voltages (-50V or more), Schottky diodes  
may not be practical because of this voltage requirement.  
In these cases, use an ultrafast recovery diode with  
adequate reverse-breakdown voltage.  
When using low-ESR, ceramic chip capacitors (MLCCs)  
at the output, calculate the value of C as follows:  
FB  
R +R  
1
2
C
=
FB  
2×3.14×f  
×R ×R  
1 2  
OSC  
Maxim Integrated  
│ 16  
www.maximintegrated.com  
MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Input Filter Capacitor  
PC Board Layout Guidelines  
The input capacitor (C ) must provide the peak current  
into the inverter. This capacitor is selected the same way  
Good PC board layout and routing are required in high-  
frequency-switching power supplies to achieve good  
regulation, high efficiency, and stability. It is strongly  
recommended that the evaluation kit PC board layouts  
be followed as closely as possible. Place power components  
as close together as possible, keeping their traces short,  
direct, and wide. Avoid interconnecting the ground pins  
of the power components using vias through an internal  
ground plane. Instead, keep the power components close  
together and route them in a “star” ground configuration  
using component-side copper, then connect the star  
ground to internal ground using multiple vias.  
IN  
as the output capacitor (C ). Under ideal conditions,  
OUT  
the RMS current for the input capacitor is the same as the  
output capacitor. The capacitor value and ESR must be  
selected to reduce noise to an acceptable value and also  
handle the ripple current (I  
where:  
NRMS  
1.2 xIO  
2
I
=
x D  
−D  
MAX MAX  
NRMS  
(I − D  
)
MAX  
Bypass Capacitor  
In addition to C  
capacitors are required with the MAX1846/MAX1847.  
Bypass REF to GND with a 0.1µF or larger capacitor.  
and C  
, other ceramic bypass  
OUT  
Main Application Circuits  
IN  
The MAX1846/MAX1847 are extremely versatile devices.  
Figure 2 shows a generic schematic of the MAX1846.  
Table 1 lists component values for several typical  
applications. These component values also apply to the  
MAX1847. The first two applications are featured in the  
MAX1846/MAX1847 EV kit.  
Bypass V to GND with a 0.47µF or larger capacitor. All  
L
bypass capacitors should be located as close to their  
respective pins as possible.  
V
IN  
APPLICATION B  
ONLY  
22k  
C
IN  
P
D1  
V
OUT  
1
VL  
10  
IN  
0.47µF  
C
OUT  
L1  
9
8
EXT  
CS  
MAX1846  
R
CS  
R1  
R2  
C
COMP2  
3
COMP  
FREQ  
7
5
PGND  
FB  
C
R
COMP  
COMP  
2
4
REF  
R
FREQ  
GND  
C
FB  
6
0.1µF  
NOTE: APPLICATIONS A & B USE POS CAPACITORS. APPLICATIONS C & D USE ALUMINUM ELECTROLYTIC CAPACITORS.  
Figure 2. MAX1846 Main Application Circuit  
Maxim Integrated  
│ 17  
www.maximintegrated.com  
MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Table 1. Component List for Main Application Circuits  
CIRCUIT ID  
Input (V)  
A
12  
B
3 to 5.5  
-12  
C
12  
D
12  
Output (V)  
Output (A)  
-5  
-48  
-72  
2
0.4  
0.1  
0.1  
C
C
C
C
(µF)  
0.047  
3 x 10  
2 x 100  
390  
0.22  
0.1  
0.068  
10  
COMP  
(µF)  
3 x 22  
2 x 47  
1200  
95.3  
10  
IN  
OUT  
(pF)  
(µF)  
39  
39  
1000  
383  
1000  
576  
FB  
R1 (kW) (1%)  
R2 (kW) (1%)  
40.2  
10  
10  
10  
10  
R
R
R
(kΩ)  
8.2  
10  
220  
470  
COMP  
(W)  
0.02  
150  
0.02  
0.05  
150  
0.05  
150  
CS  
FREQ  
(kW)  
150  
D1  
CMSH5-40  
10  
CMSH5-40  
10  
CMR1U-02  
47  
CMR1U-02  
82  
L1 (µH)  
P1  
FDS6685  
220  
FDS6375  
220  
IRFR5410  
22  
IRFR5410  
12  
C
(pF)  
COMP2  
Component Suppliers  
SUPPLIER  
COMPONENT  
Capacitors  
Diodes  
PHONE  
WEBSITE  
AVX  
803-946-0690  
516-435-1110  
847-639-6400  
402-564-3131  
408-721-2181  
310-322-3331  
512-992-7900  
864-963-6300  
602-303-5454  
201-348-7522  
619-661-6835  
408-988-8000  
603-224-1961  
847-956-0666  
203-268-6261  
www.avxcorp.com  
www.centralsemi.com  
www.coilcraft.com  
Central Semiconductor  
Coilcraft  
Inductors  
Dale  
Resistors  
www.vishay.com/company/brands/dale/  
www.fairchildsemi.com  
www.irf.com  
Fairchild  
MOSFETs  
International Rectifier  
IRC  
MOSFETs  
Resistors  
www.irctt.com  
Kemet  
Capacitors  
MOSFETs, Diodes  
Capacitors, resistors  
Capacitors  
MOSFETs  
www.kemet.com  
On Semiconductor  
Panasonic  
Sanyo  
www.onsemi.com  
www.panasonic.com  
www.secc.co.jp  
Siliconix  
www.siliconix.com  
Sprague  
Capacitors  
Inductors  
www.vishay.com/company/brands/sprague/  
www.remtechcorp.com  
www.vishay.com/company/brands/vitramon/  
Sumida  
Vitramon  
Resistors  
Note: Indicate that you are using the MAX1846/MAX1847 when contacting these component suppliers.  
Maxim Integrated  
│ 18  
www.maximintegrated.com  
MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Chip Information  
PROCESS: BiCMOS  
Pin Configurations  
TOP VIEW  
Package Information  
+
+
POL  
VL  
1
2
3
4
5
6
7
8
16 SYNC  
15 IN  
VL  
FREQ  
COMP  
REF  
1
2
3
4
5
10 IN  
For the latest package outline information and land patterns  
(footprints), go to www.maximintegrated.com/packages. Note  
that a “+”, “#”, or “-” in the package code indicates RoHS status  
only. Package drawings may show a different suffix character, but  
the drawing pertains to the package regardless of RoHS status.  
9
8
7
6
EXT  
MAX1846  
FREQ  
COMP  
REF  
14 EXT  
13 CS  
CS  
MAX1847  
PGND  
GND  
12 PGND  
11 GND  
10 GND  
FB  
FB  
µMAX  
LAND  
PATTERN NO.  
PACKAGE PACKAGE OUTLINE  
N.C.  
TYPE  
CODE  
NO.  
SHDN  
9
N.C.  
90-0330  
90-0167  
10 µMAX  
16 QSOP  
U10+2  
E16+1  
21-0061  
21-0055  
QSOP  
Maxim Integrated  
│ 19  
www.maximintegrated.com  
 
MAX1846/MAX1847  
High-Efficiency, Current-Mode,  
Inverting PWM Controller  
Revision History  
REVISION  
NUMBER  
REVISION  
DATE  
PAGES  
CHANGED  
DESCRIPTION  
2
3
4
9/10  
3/14  
7/16  
Added equation in the Determining the Compensation Component Values section  
Removed automotive application from the Applications section  
16  
1
Extended maximum operating temperature from +85°C to +105°C  
1, 2, 4  
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated’s website at www.maximintegrated.com.  
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses  
are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits)  
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.  
©
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.  
2016 Maxim Integrated Products, Inc.  
│ 20  

MAX1846EUB+T CAD模型

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    • MAX1846EUB+T 替代型号

    型号 制造商 描述 替代类型 文档
    MAX1846EUB-T MAXIM High-Efficiency, Current-Mode, Inverting PWM Controller 完全替代
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