MAX20034ATIRVY [MAXIM]
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck Controller with 17μA Quiescent Current;
| 型号： | MAX20034ATIRVY |
| 厂家： | 
MAXIM INTEGRATED PRODUCTS |
| 描述： | Automotive High-Efficiency 2.2MHz, 36V, Dual Buck Controller with 17μA Quiescent Current |
| 文件： | 总24页 (文件大小：2434K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
EVALUATION KIT AVAILABLE
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual
Buck Controller with 17μA Quiescent Current
General Description
Benefits and Features
● Meets Stringent Automotive OEM Module Power-
Consumption and Performance Specifications
• 17µA Quiescent Current in Skip Mode
The MAX20034 is an automotive 2.2MHz, dual synchro-
nous step-down controller IC that provides two high-
voltage, synchronous step-down controllers that operate
180° out-of-phase. The IC operates with a 3.5V to 42V
input-voltage supply and can function in dropout condition
by running at 99% duty cycle. It is intended for applications
with mid- to high-power requirements that perform at a
wide input voltage range, such as during automotive cold-
crank or engine stop-start conditions.
• ±1.5% Output-Voltage Accuracy: 5.0V/3.3V Fixed,
or Adjustable Between 1V and 10V
● Enables Crank-Ready Designs
• Wide 3.5V to 36V Input Supply Range
● EMI Reduction Features Reduce Interference with
Sensitive Radio Bands without Sacrificing Wide Input
Voltage Range
The IC’s step-down controllers operate at up to 2.2MHz
frequency to allow small external components, reduced
output ripple, and to guarantee no AM band interference.
The switching frequency is resistor adjustable (220kHz
to 2.2MHz). SYNC input programmability enables three
frequency modes for optimized performance: forced
fixed-frequency operation, skip mode with ultra-low
quiescent current, and synchronization to an external
clock. The IC is also available with spread-spectrum
frequency modulation to minimize EMI interference.
• 50ns (typ) Minimum On-Time Guarantees Skip-
Free Operation for 3.3V Output from a Car Battery
at 2.2MHz
• Spread-Spectrum Option
• Frequency-Synchronization Input
• Resistor-Programmable Frequency Between
220kHz and 2.2MHz
● Integration and Thermally Enhanced Packages Save
Board Space and Cost
The IC features
a
power-OK monitor, overvolt-
• Dual, Up to 2.2MHz Step-Down Controllers
• 180° Out-of-Phase Operation
• Current-Mode Controllers with Forced-PWM
(FPWM) and Skip Modes
age lockout, and undervoltage lockout. Protection
features include cycle-by-cycle current limit and
thermal shutdown. The MAX20034 is specified
for operation over the -40°C to +125°C automotive
temperature range.
• Thermally Enhanced, 28-Pin TQFN-EP Package
● Protection Features Improve System Reliability
• Supply Overvoltage and Undervoltage Lockout
• Overtemperature and Short-Circuit Protection
Applications
● POL Applications for Automotive Power
● Distributed DC Power Systems
● Navigation and Radio Head Units
Ordering Information appears at end of data sheet.
19-100159; Rev 3; 8/19
MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Simplified Block Diagram
PGOOD1 COMP1
DC-DC1
CONTROL LOGIC
PGOOD LOW LEVEL
PGOOD HIGH LEVEL
PGOOD
COMP
MAX20034
FB1
FEEDBACK-
SELECT LOGIC
EAMP1
INTERNAL
SOFT-START
EN1
BST1
DH1
LX1
V
REF
= 1V
PWM1
CLK1
ZX1
OUT1
CS1
PWM1
80mV (TYP) MAX
DIFFERENTIAL INPUT
STEP-DOWN DC-DC1
CSA1
GATE-DRIVE
LOGIC
DL1
CL
ZERO-
CROSS
COMP
SLOPE-
COMP LOGIC
CURRENT-LIMIT
THRESHOLD
EN1
PGND1
LX1
LX1
CLK1
FOSC
OSCILLATOR
IN
SPREAD-SPECTRUM
OPTION AVAILABLE WITH
INTERNAL CLOCK ONLY
EXTERNAL
CLOCK INPUT
BIAS
BIAS
INTERNAL LINEAR
REGULATOR
CONNECTED HIGH (PWM MODE)
CONNECTED LOW (SKIP MODE)
FSYNC
AGND
FSYNC-SELECT LOGIC
IF 3.25V <
< 5.2V
V
EXTVCC
EXTVCC
SWITCHOVER
CLK 180°
OUT-OF-PHASE
CLK2
EN2
PWM2
CLK2
ZX2
BST2
DH2
COMP2
FB2
OUT2
CS2
STEP-DOWN DC-DC2
LX2
DC-DC2 CONTROL LOGIC
SAME AS DC-DC1 ABOVE
GATE-DRIVE
LOGIC
EN2
DL2
PGOOD2
PGND2
LX2
LX2
EP
Maxim Integrated
│ 2
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Absolute Maximum Ratings
IN, EN1, EN2, LX_ to PGND.................................-0.3V to +42V
OUT1, OUT2 to AGND..........................................-0.3V to +12V
CS1 to OUT1........................................................-0.3V to +0.3V
CS2 to OUT2........................................................-0.3V to +0.3V
BIAS, FSYNC, PGOOD_, FB_,
DH_ to LX_ (Note 1)................................ -0.3V to V
PGND_ to AGND..................................................-0.3V to +0.3V
+ 0.3V
BST_
Continous Power Dissipation (T = +70°C)
A
28 TQFN (derate 37mW/°C above +70°C)................2285mW
Operating Temperature Range......................... -40°C to +125°C
Junction Temperature......................................................+150°C
Storage Temperature Range............................ -65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Soldering Temperature (reflow).......................................+260°C
EXTVCC to AGND...............................................-0.3V to +6V
COMP_, FOSC to PGND_.......................-0.3V to V
DL_ to PGND_ (Note 1)...........................-0.3V to V
+ 0.3V
+ 0.3V
BIAS
BIAS
BST_ to LX_ (Note 1)..............................................-0.3V to +6V
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these
or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect
device reliability.
Note 1: Self-protected against transient voltages exceeding these limits for ≤ 50ns under normal operation and loads up to the
maximum rated output current.
Package Information
PACKAGE TYPE: 28-PIN TQFN
Package Code
T2855Y-5C
21-100130
90-0027
Outline Number
Land Pattern Number
THERMAL RESISTANCE, FOUR-LAYER BOARD:
Junction to Ambient (θ
)
35°C/W
3°C/W
JA
Junction to Case (θ
)
JC
For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a “+”,
“#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing
pertains to the package regardless of RoHS status.
Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer board.
For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.
Electrical Characteristics
(V
= 14V, C
= 6.8μF, R
= 12kΩ, T = T = -40°C to +125°C, unless otherwise noted. Typical values are at
IN
BIAS
FOSC A J
T
= +25°C. Limits are 100% tested at T = +25°C and T = +125°C. Limits over the operating temperature range and relevant supply
A
A A
voltage range are guaranteed by design and characterization.)
PARAMETER
SYMBOL
CONDITIONS
Normal operation
t < 1s
MIN
TYP
MAX
36
UNITS
3.5
Supply Voltage Range
V
V
IN
42
V
= V
= 0V
6.5
25
10
EN1
EN2
V
V
= 5V, V
= 5V, V
= 0V,
EN1
OUT1
= 5V (no switching)
EN2
40
28
Supply Current
I
µA
IN
EXTVCC
V
V
= 5V, V
= 3.3V, V
= 0V,
EN2
OUT2
= 3.3V (no switching)
EN1
17
5
EXTVCC
Buck 1 Fixed-Output
Voltage
V
V
V
= V , V = 5V, PWM mode
BIAS OUT1
4.925
3.25
5.075
3.35
V
V
OUT1
FB1
FB2
Buck 2 Fixed-Output
Voltage
V
= V
, V
= 3.3V, PWM mode
3.3
OUT2
BIAS OUT2
Maxim Integrated
│ 3
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Electrical Characteristics (continued)
(V
= 14V, C
= 6.8μF, R
= 12kΩ, T = T = -40°C to +125°C, unless otherwise noted. Typical values are at
IN
BIAS
FOSC A J
T
= +25°C. Limits are 100% tested at T = +25°C and T = +125°C. Limits over the operating temperature range and relevant supply
A
A A
voltage range are guaranteed by design and characterization.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
Output Voltage-Adjustable
Range
Buck 1, Buck 2
PWM
1
10
V
Regulated Feedback
Voltage
V
I
V
0.995
1.005
0.01
0.01
470
1.015
1
V
FB1, FB2
Feedback Leakage
Current
I
T
= +25°C
µA
FB1, FB2
A
Feedback Line-
Regulation Error
V
V
= 3.5V to 36V, V
= 1V
%/V
µS
IN
FB_
Transconductance (from
FB1, 2 to COMP1, 2)
g
= 1V, V = 5V
BIAS
300
97
700
m
FB_
DL_ low to DH_ high
DH_ low to DL_ high
Buck 1, Buck 2
15
15
99
50
Dead Time
ns
Maximum Duty Cycle
Minimum On-Time
%
t
Buck 1, Buck 2
ns
ON,MIN
PWM Switching-
Frequency Range
f
Programmable
0.22
2
2.2
2.4
MHz
MHz
SW
Switching-Frequency
Accuracy
R
= 12kΩ, V
= 5V, 3.3V
2.2
FOSC
BIAS
CS_ Current-Limit
Voltage Threshold
V
V
LIMIT1,
LIMIT2
V
- V
; V
= 5V, V ≥ 2.5V
OUT
68
3
80
5
92
8
mV
ms
deg
µA
Ω
CS_
OUT BIAS
Soft-Start Ramp Time
Buck 1 and Buck 2
Phase Shift Between
Buck 1 and Buck 2
PWM operation (Note 2)
180
0.001
3
LX1, LX2 Leakage Current
V
= V
or V , T = +25°C
1
6
LX_
PGND_
IN
A
DH1, DH2 Pullup
Resistance
V
= 5V, l
= -100mA
= 100mA
BIAS
DH
DH1, DH2 Pulldown
Resistance
V
V
V
= 5V, l
1.5
3
3
6
3
Ω
Ω
Ω
BIAS
BIAS
BIAS
DH
DL1,2 Pullup Resistance
= 5V, l = -100mA
DL
DL1, DL2 Pulldown
Resistance
= 5V, l = 100mA
1.5
DL
Output Overvoltage
Threshold
Detected with respect to V
rising
105
108
3
112
%
%
FB_
Output Overvoltage
Hysteresis
Maxim Integrated
│ 4
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Electrical Characteristics (continued)
(V
= 14V, C
= 6.8μF, R
= 12kΩ, T = T = -40°C to +125°C, unless otherwise noted. Typical values are at
IN
BIAS
FOSC A J
T
= +25°C. Limits are 100% tested at T = +25°C and T = +125°C. Limits over the operating temperature range and relevant supply
A
A A
voltage range are guaranteed by design and characterization.)
PARAMETER
SYMBOL
PGOOD_R
PGOOD_F
CONDITIONS
MIN
92
TYP
94
MAX
97
UNITS
Percentage of V
Percentage of V
, rising
OUT_
OUT_
PGOOD Threshold
%
, falling
90
92
95
Leakage Current
Output Low Voltage
Debounce Time
FSYNC INPUT
V
= 5V, T = +25°C
0.01
1
µA
V
PGOOD_
A
I
= 1mA
0.2
SINK
Fault detection, rising and falling
20
µs
Minimum sync pulse > (1/FSYNC - 1/FOSC)
1.8
2.6
MHz
kHz
R
= 12kΩ
FOSC
FSYNC Frequency Range
Minimum sync pulse > (1/FSYNC - 1/FOSC)
= 70kΩ
250
1.4
550
R
FOSC
High threshold
Low threshold
FSYNC Switching
Thresholds
V
0.4
INTERNAL LDO BIAS
Internal BIAS Voltage
V
V
V
> 6V, no load
5
V
V
IN
rising
falling
3.1
2.6
3.25
BIAS
BIAS
BIAS UVLO Threshold
2.35
3.25
2.85
EXTVCC Operating
Range
5.5
V
V
EXTVCC Threshold
V
EXTVCC rising, hysteresis = 110mV
3
3.25
TH,EXTVCC
THERMAL OVERLOAD
Thermal-Shutdown
Temperature
(Note 2)
(Note 2)
170
20
°C
°C
Thermal-Shutdown
Hysteresis
ENABLE LOGIC INPUT
High Threshold
EN1, EN2
EN1, EN2
1.8
V
V
Low Threshold
0.8
1
EN_ Input Bias Current
EN1, EN2 logic inputs only, T = +25°C
0.01
µA
A
Note 2: Guaranteed by design, not production tested.
Maxim Integrated
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Typical Operating Characteristics
(T = +25°C, unless otherwise noted.)
A
SHUTDOWN SUPPLY CURRENT
vs. SUPPLY VOLTAGE
STARTUP INTO LOAD
STARTUP INTO NO-LOAD
toc02
toc01
toc03
15
12
9
5V/div
5V/div
2V/div
V
VEN_
EN_
EN1 = EN2
FSYNC = 0
2V/div
5V/div
VEXTVCC = VOUT1
IOUT1 = IOUT2 = 0A
V
VOUT1
OUT1
5V/div
EN1 = EN2
FSYNC = 0
V
VPGOOD1
PGOOD1
I
V
= I
= 3A
OUT1
OUT1 OUT2
= V
EXTVCC
6
2V/div
5V/div
2V/div
5V/div
V
VOUT2
OUT2
3
V
VPGOOD2
PGOOD2
2ms/div
2ms/div
0
6
9
12 15 18 21 24 27 30 33 36
SUPPLY VOLTAGE (V)
QUIESCENT CURRENT
vs. SUPPLY VOLTAGE
BUCK1 EFFICIENCY vs OUTPUT CURRENT
(fSW = 2.2MHz)
toc04
toc05
70
60
50
40
30
20
10
0
100
90
80
70
60
50
40
30
20
10
0
ONLY BUCK 1 ON, VEXTVCC = VOUT1
ONLY BUCK 2 ON, VEXTVCC = VOUT2
BOTH BUCKS ON, VEXTVCC = VOUT1
SKIP ENABLED
FPWM
EN1 = HIGH
EN2 = LOW
EXTVCC = VOUT1
VOUT1 = 5V
0.001
0.01
0.1
1
10
OUTPUT CURRENT (A)
6
12
18
24
30
36
SUPPLY VOLTAGE (V)
BUCK 2 EFFICIENCY vs. OUTPUT CURRENT
(fSW = 400kHz)
BUCK 1 EFFICIENCY vs. OUTPUT CURRENT
(fSW = 400kHz)
toc07
toc06
100
90
80
70
60
50
40
30
20
10
0
100
90
80
70
60
50
40
30
20
10
0
SKIP ENABLED
FPWM
SKIP ENABLED
FPWM
EN1 = HIGH
EN2 = HIGH
VEXTVCC = VOUT1
VOUT2 = 3.3V
EN1 = HIGH
EN2 = LOW
VEXTVCC = VOUT1
VOUT1 = 5V
0.001
0.01
0.1
1
10
0.001
0.01
0.1
1
10
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
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│ 6
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Typical Operating Characteristics (continued)
(T = +25°C, unless otherwise noted.)
A
SWITCHING FREQUENCY
vs. LOAD CURRENT
SWITCHING FREQUENCY
vs. AMBIENT TEMPERATURE
SWITCHING FREQUENCY vs. RFOSC
toc08
toc10
2.22
toc09
2.22
2.21
2.20
2.19
2.18
2.17
2.16
2.15
2.14
2.13
2.12
2.6
2.4
2.2
2.0
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0.0
2.21
2.20
2.19
2.18
2.17
2.16
2.15
BUCK1
BUCK2
FSYNC = 1
2.14
2.13
2.12
0
1
2
3
4
5
LOAD CURRENT (A)
0
20
40
60
80 100 120 140 160
-40 -25 -10
5
20 35 50 65 80 95 110 125
AMBIENT TEMPERATURE (°C)
RFOSC (kΩ)
FSYNC SYNCHRONIZATION
OUT-OF-PHASE OPERATION
toc11
toc12
5V/div
5V/div
VFSYNC
V
FSYNC
1.8MHz, 50% DUTY-CYCLE SIGNAL ON FSYNC
IOUT1 = IOUT2 = 2.5A
2.2MHz, 50% DUTY-CYCLE SIGNAL ON FSYNC
= I = 2.5A
I
OUT1 OUT2
VLX1
10V/div
V
10V/div
LX1
VLX2
10V/div
V
10V/div
LX2
1µs/div
1µs/div
LOAD TRANSIENT RESPONSE (BUCK 2)
LOAD-TRANSIENT RESPONSE (BUCK 1)
toc14
toc13
IOUT1
5A
5A/div
5A
0A
IOUT2
5A/div
5A/div
0A
ILX
ILX
5A/div
10V/div
10V/div
VLX_
VLX
VOUT2 = 3.3V
FPWM
EXTVCC = VOUT2
VOUT
VOUT_
200mV/
div
200mV/div
VOUT1 = 5V
FPWM
VEXTVCC = VOUT1
200µs/div
200µs/div
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│ 7
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Typical Operating Characteristics (continued)
(T = +25°C, unless otherwise noted.)
A
OUTPUT LOAD REGULATION
(BUCK 2)
OUTPUT LOAD REGULATION
(BUCK 1)
OUTPUT LINE REGULATION (BUCK 1)
toc15
toc16
toc17
5.10
5.02
5.01
5.00
4.99
4.98
4.97
4.96
3.366
3.333
3.300
3.267
3.234
SKIP
FPWM
IOUT = 0A
IOUT = 0.1A
IOUT = 0.5A
SKIP
FPWM
VEXTVCC = VOUT1
5.05
VEXTVCC = VOUT1
5.00
4.95
4.90
IOUT = 1A
IOUT = 2A
IOUT = 3A
IOUT = 4A
FPWM
VEXTVCC = VOUT1
0.001
0.01
0.1
1
10
6
9
12 15 18 21 24 27 30 33 36
0.001
0.01
0.1
1
10
INPUT VOLTAGE (V)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
COLD CRANK (BUCK 1)
OUTPUT LINE REGULATION (BUCK 2)
toc19
toc18
3.300
3.295
3.290
3.285
3.280
3V/div
1V/div
VIN
VOUT
IOUT = 0A
IOUT = 0.1A
IOUT = 0.5A
IOUT = 1A
IOUT = 2A
IOUT = 3A
IOUT = 4A
ILX
2A/div
5V/div
VIN = 14V TO 3.5V TO 5V TO 14V
FPWM, IOUT = 4A, VEXTVCC = VOUT1
FPWM
VEXTVCC = VOUT2
VPGOOD1
6
9
12 15 18 21 24 27 30 33 36
INPUT VOLTAGE (V)
100ms/div
COLD CRANK (BUCK 2)
LOAD DUMP (BUCK 1)
toc20
toc21
3V/div
10V/div
1V/div
VIN
VIN
VOUT
VOUT
1V/div
ILX
ILX
2A/div
5V/div
2A/div
5V/div
VIN = 14V TO 3.5V TO 5V TO 14V
FPWM, IOUT = 4A, VEXTVCC = VOUT2
FPWM, IOUT = 4A, VEXTVCC = VOUT1
VPGOOD
VPGOOD
100ms/div
50ms/div
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Typical Operating Characteristics (continued)
(T = +25°C, unless otherwise noted.)
A
SLOW INPUT-VOLTAGE RISING (BUCK 2)
LOAD DUMP (BUCK 2)
SLOW INPUT-VOLTAGE RISING (BUCK 1)
toc24
toc22
toc23
V
10V/div
1V/div
10V/div
10V/div
1V/div
VIN
VIN
IN
VOUT
VOUT
V
= 0 TO 14V
IN
1V/div
FPWM, I
V
= 4A
OUT
= V
OUT1
EXTVCC
ILX
ILX
2A/div
5V/div
V
VIN = 0 TO 14V
FPWM, IOUT = 4A
VEXTVCC = VOUT1
OUT
2A/div
5V/div
2A/div
5V/div
FPWM, IOUT = 4A, VEXTVCC = VOUT1
I
LX
VPGOOD
VPGOOD
V
PGOOD
5s/div
50ms/div
5s/div
LINE TRANSIENT OUT OF DROPOUT
(BUCK 1)
LINE TRANSIENT OUT OF DROPOUT
(BUCK 2)
toc25
toc26
3V/div
3V/div
3.42V
VIN = 5V TO 14V
FPWM, NO LOAD
5.13V
300mV/div
100mV/div
VLX
VLX
5V/div
1A/div
5V/div
1A/div
VIN = 5V TO 14V
FPWM, NO LOAD
ILX
ILX
50µs/div
50µs/div
SHORT CIRCUIT AFTER REGULATION
(BUCK 1)
SHORT CIRCUIT AFTER REGULATION
(BUCK 2)
toc27
toc28
2V/div
VOUT1
VOUT1
2V/div
2A/div
5V/div
2A/div
5V/div
VPGOOD
VPGOOD
ILX
ILX
10V/div
VLX
VLX
10V/div
400µs/div
400µs/div
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Pin Configuration
TOP VIEW
21 20 19 18 17 16 15
14
13
LX2 22
DH2 23
FSYNC
PGOOD2
12 PGOOD1
24
25
26
27
28
BST2
EN2
MAX20034
IN
11
10
9
EN1
EXTVCC
AGND
BIAS
BST1
DH1
EP
6
+
8
1
2
3
4
5
7
TQFN
(5mm x 5mm)
Pin Description
PIN
NAME
FUNCTION
Inductor Connection for Buck 1. Connect LX1 to the switched side of the inductor. LX1 serves as
the lower supply rail for the DH1 high-side gate driver.
1
LX1
2
3
DL1
Low-Side Gate-Driver Output for Buck 1. DL1 output voltage swings from V
Power Ground for Buck 1
to V
.
PGND1
BIAS
PGND1
Positive Current-Sense Input for Buck 1. Connect CS1 to the positive terminal of the current-sense
element. See the Current Limiting and Current-Sense Inputs (OUT_ and CS_) and Current-Sense
Measurement sections.
4
CS1
Output Sense and Negative Current-Sense Input for Buck 1. When using the internal preset 5V
feedback-divider (FB1 = BIAS), the controller uses OUT1 to sense the output voltage. Connect
OUT1 to the negative terminal of the current-sense element. See the Current Limiting and Current-
Sense Inputs (OUT_ and CS_) and Current-Sense Measurement sections.
5
OUT1
Feedback Input for Buck 1. Connect FB1 to BIAS for the 5V fixed output or to a resistive divider
between OUT1 and AGND to adjust the output voltage between 1V and 10V. In adjustable version,
FB1 regulates to 1V (typ). See the Setting the Output Voltage in Buck Converters section.
6
7
8
FB1
COMP1
BIAS
Buck 1 Error-Amplifier Output. Connect an RC network to COMP1 to compensate.
5V Internal Linear Regulator Output. Bypass BIAS to PGND with a low-ESR ceramic capacitor of
6.8µF minimum value. BIAS provides the power to the internal circuitry and external loads. See the
Fixed 5V Linear Regulator (BIAS) section.
9
AGND
Signal Ground for IC
Switchover Comparator Input. Connect a voltage between 3.25V and 5.5V to EXTVCC to power the
IC and bypass the internal bias LDO.
10
EXTVCC
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Pin Description (continued)
PIN
NAME
FUNCTION
11
IN
Supply Input. Bypass IN with enough capacitors to supply the two out-of-phase buck converters.
Open-Drain Power-Good Output for Buck 1. PGOOD1 is low if OUT1 is more than 92% (typ)
below the normal regulation point. PGOOD1 asserts low during soft-start and in shutdown.
PGOOD1 becomes high impedance when OUT1 is in regulation. To obtain a logic signal, pull
PGOOD1 up with an external resistor connected to a positive voltage lower than 5.5V.
12
13
14
PGOOD1
PGOOD2
FSYNC
Open-Drain Power-Good Output for Buck 2. PGOOD2 is low if OUT2 is more than 92% (typ)
below the normal regulation point. PGOOD2 asserts low during soft-start and in shutdown.
PGOOD2 becomes high impedance when OUT2 is in regulation. To obtain a logic signal, pull
PGOOD2 up with an external resistor connected to a positive voltage lower than 5.5V.
External Clock-Synchronization Input. Synchronization to the controller operating-frequency ratio
is 1. See the Switching Frequency/External Synchronization section. For FSYNC high, and T
<
ON
T
, ensure there is at least 50μA (including the resistor-divider current on V
) of load
OUT1,2
ON,MIN
current if V
- V
> 1.3V.
BIAS
OUT
Frequency-Setting Input. Connect a resistor from FOSC to AGND to set the switching frequency of
the DC-DC converters.
15
16
FOSC
COMP2
Buck 2 Error-Amplifier Output. Connect an RC network to COMP2 to compensate buck converter 2.
Feedback Input for Buck 2. Connect FB2 to BIAS for the 3.3V fixed output or to a resistive divider
between OUT2 and AGND to adjust the output voltage between 1V and 10V. In adjustable version,
FB2 regulates to 1V (typ). See the Setting the Output Voltage in Buck Converters section.
17
18
19
FB2
OUT2
CS2
Output Sense and Negative Current-Sense Input for Buck 2. When using the internal preset 3.3V
feedback-divider (FB2 = BIAS), the buck uses OUT2 to sense the output voltage. Connect OUT2
to the negative terminal of the current-sense element. See the Current Limiting and Current-Sense
Inputs (OUT_ and CS_) and Current-Sense Measurement sections.
Positive Current-Sense Input for Buck 2. Connect CS2 to the positive terminal of the current-sense
element. See Current Limiting and Current-Sense Inputs (OUT_ and CS_) and Current-Sense
Measurement sections.
20
21
PGND2
DL2
Power Ground for Buck 2
Low-Side Gate-Driver Output for Buck 2. DL2 output voltage swings from V
to V
.
PGND2
BIAS
Inductor Connection for Buck 2. Connect LX2 to the switched side of the inductor. LX2 serves as
the lower supply rail for the DH2 high-side gate driver.
22
23
LX2
DH2
High-Side Gate-Driver Output for Buck 2. DH2 output voltage swings from V
to V
.
LX2
BST2
Boost Flying-Capacitor Connection for High-Side Gate Voltage of Buck 2. Connect a high-voltage
diode between BIAS and BST2. Connect a ceramic capacitor between BST2 and LX2. See the
High-Side Gate-Driver Supply (BST_) section.
24
BST2
25
26
EN2
EN1
High-Voltage-Tolerant, Active-High Digital Enable Input for Buck 2. Driving EN2 high enables Buck 2.
High-Voltage-Tolerant, Active-High Digital Enable Input for Buck 1. Driving EN1 high enables Buck 1.
Boost Flying-Capacitor Connection for High-Side Gate Voltage of Buck 1. Connect a high-voltage
diode between BIAS and BST2. Connect a ceramic capacitor between BST1 and LX1. See the
High-Side Gate-Driver Supply (BST_) section.
27
28
BST1
DH1
High-Side Gate-Driver Output for Buck 2. DH1 output voltage swings from V
to V
.
LX1
BST1
Exposed Pad. Connect EP to ground. Connecting the exposed pad to ground does not remove the
requirement for proper ground connections to PGND1, PGND2, and AGND. The exposed pad is
attached with epoxy to the substrate of the die, making it an excellent path to remove heat from the
IC.
—
EP
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Undervoltage Lockout (UVLO)
Detailed Description
The BIAS input undervoltage-lockout (UVLO) circuitry
inhibits switching if the 5V bias supply (BIAS) is below its
2.6V (typ) UVLO falling threshold. Once BIAS rises above
its UVLO rising threshold and EN1 and EN2 enable the
buck controllers, the controllers start switching and the
output voltages begin to ramp up using soft-start.
The MAX20034 is an automotive-rated dual-out-
put switching power-supply IC. The IC integrates two
synchronous step-down controllers and can provide two
independent-controlled power rails as follows:
● Buck controller with a fixed 5V output voltage, or an
adjustable 1V to 10V output voltage.
Buck Controllers
● Buck controller with a fixed 3.3V output voltage, or an
adjustable 1V to 10V output voltage.
The IC provides two buck controllers with synchronous
rectification. The step-down controllers use a pulse-width
modulation (PWM) current-mode control scheme. External
MOSFETs allow for optimized load-current design. Fixed-
frequency operation with optimal interleaving minimizes
input ripple current from the minimum to the maximum
input voltages. Output-current sensing provides an accu-
rate current limit with a sense resistor, or power dissipation
can be reduced using lossless current sensing across the
inductor.
EN1 and EN2 enable the respective buck controllers.
Connect EN1 and EN2 directly to V
supply sequencing logic.
, or to power-
BAT
In skip mode, the total supply current is reduced to 17μA
(typ) with Buck 1 disabled and Buck 2 enabled. When
both controllers are disabled, the total current drawn is
further reduced to 6.5µA (typ).
Fixed 5V Linear Regulator (BIAS)
The internal circuitry of the IC requires a 5V bias supply.
An internal 5V linear regulator (BIAS) generates this bias
supply. Bypass BIAS with a ≥ 6.8µF ceramic capacitor to
guarantee stability under the full-load condition.
Soft-Start
Once a buck converter is enabled by driving the corre-
sponding EN_ high, the soft-start circuitry gradually ramps
up the reference voltage during soft-start time (t
SSTART
The internal linear regulator can source up to 100mA
(150mA under EXTVCC switchover; see the EXTVCC
Switchover section). Use the following equation to
estimate the internal current requirements for the IC:
= 5ms (typ)) to reduce the input surge currents during
startup. Before the IC can begin the soft-start, the follow-
ing conditions must be met:
1) V
exceeds the 3.25V (max) undervoltage-lockout
BIAS
I
= I
+ f
(Q
+ Q
+
threshold.
BIAS
CC
SW
G_DH1
G_DL1
Q
+ Q
) = 10mA to 50mA (typ)
G_DH2
G_DL2
2) V
is logic-high.
EN_
where I
is the internal 5mA (typ) supply current, f
SW
CC
Switching Frequency/External Synchronization
is the switching frequency, and Q
is the MOSFET’s
G_
The IC provides an internal oscillator, adjustable from
220kHz to 2.2MHz. High-frequency operation optimizes
the application for the smallest component size, trading off
efficiency to higher switching losses. Low-frequency opera-
tion offers the best overall efficiency at the expense of
component size and board space. To set the switching fre-
total gate charge (specification limits at V
minimize the internal power dissipation, bypass BIAS to
an external 5V rail.
= 5V). To
GS
EXTVCC Switchover
The internal linear regulator can be bypassed by connect-
ing an external supply (3.25V to 5.2V) or one of the buck
converter outputs to EXTVCC. BIAS internally switches to
EXTVCC and the internal linear regulator turns off. This
configuration has several advantages:
quency, connect a resistor (R ) from FOSC to AGND:
FOSC
R_FOSC
6
25.5 +
f_SW =
● It reduces the internal power dissipation of the device.
R_FOSC
● The low-load efficiency improves as the internal supply
current is scaled down proportionally to the duty cycle.
See the Typical Operating Characteristics to determine
the relationship between switching frequency and R
.
If V
drops below 3.25V, the internal regulator is
FOSC
EXTVCC
enabled and BIAS switches back to the internal regulator.
The IC can be synchronized to an external clock by con-
necting the external clock signal to FSYNC. A rising edge
on FSYNC resets the internal clock. The FSYNC clock
should have a minimum 150ns high pulse width.
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Light-Load Efficiency Skip Mode (V
= 0V)
MOSFET Gate Drivers (DH_ and DL_)
FSYNC
Drive FSYNC low to enable skip mode. In skip mode, the
IC stops switching until the FB_ voltage drops below the
reference voltage. Once the FB_ voltage has dropped
below the reference voltage, the IC begins switching until
the inductor current reaches 30% (skip threshold) of the
maximum current defined by the inductor DCR or output
shunt resistor.
The DH_ high-side n-channel MOSFET drivers are
powered from capacitors at BST_, while the low-side
drivers (DL_) are powered by the 5V linear regulator
(BIAS). On each channel, a shoot-through protection
circuit monitors the gate-to-source voltage of the external
MOSFETs to prevent a MOSFET from turning on until the
complementary switch is fully off. There must be a low-
resistance, low-inductance path from the DL_ and DH_
drivers to the MOSFET gates for the protection circuits to
work properly. Follow the instructions listed to provide the
necessary low-resistance and low-inductance path:
Forced-PWM Mode (V
= High)
FSYNC
Driving FSYNC high prevents the IC from entering skip
mode by disabling the zero-crossing detection of the induc-
tor current. This forces the low-side gate-driver waveform
to constantly be the complement of the high-side gate-
driver waveform, so the inductor current reverses at light
loads and discharges the output capacitor. The benefit
of forced-PWM (FPWM) mode is to keep the switching
frequency constant under all load conditions; however,
forced-frequency operation diverts a considerable amount
of the output current to PGND, reducing the efficiency
under light-load conditions.
● Use very short, wide traces (50 mils to 100 mils wide
if the MOSFET is 1in from the driver).
● It may be necessary to decrease the slew rate for the
gate drivers to reduce switching noise or to compen-
sate for low-gate-charge capacitors. For the low-side
drivers, use 1nF to 5nF gate capacitors from DL_ to
PGND, and for the high-side drivers, connect a small
5Ω to 10Ω resistor between BST_ and the bootstrap
capacitor.
FPWM mode is useful for improving load-transient
response and eliminating unknown frequency harmonics
that can interfere with AM radio bands.
Note: Gate drivers must be protected during shutdown,
at the absence of the supply voltage (V
= 0V) when
BIAS
the gate is pulled high either capacitively or by the leak-
age path on the PCB; therefore, external-gate pulldown
resistors are needed to prevent making a direct path from
Maximum Duty-Cycle Operation
The IC has a maximum duty cycle of 97% (min). The inter-
nal logic of the IC looks for approximately 10 consecutive
high-side FET-on pulses and decides to turn on the low-
side FET for 150ns (typ) every 12μs. The input voltage at
which the IC enters dropout changes depending on the
input voltage, output voltage, switching frequency, load
current, and the efficiency of the design. The input voltage
at which the IC enters dropout can be approximated as:
V
BAT
to PGND.
High-Side Gate-Driver Supply (BST_)
The high-side MOSFET is turned on by closing an inter-
nal switch between BST_ and DH_ and transferring the
bootstrap capacitor’s (at BST_) charge to the gate of the
high-side MOSFET. This charge refreshes when the high-
side MOSFET turns off and the LX_ voltage drops down to
ground potential, taking the negative terminal of the capaci-
tor to the same potential. At this time, the bootstrap diode
recharges the positive terminal of the bootstrap capacitor.
V
OUT_
= [V
+ (I
x R
)]/0.97
OUT_
OUT_
ON_H
Note: The above equation does not take into account
the efficiency and switching frequency, but is a good first-
order approximation. Use the R
the data sheet of the high-side MOSFET used.
(max) number from
ON_H
The selected n-channel high-side MOSFET determines the
appropriate boost capacitance values (C
in the Typical
BST_
Operating Circuit) according to the following equation:
Spread Spectrum
Q
G
The IC features enhanced EMI performance, which per-
forms ±6% dithering of the switching frequency to reduce
peak emission noise at the clock frequency and its har-
monics, making it easier to meet stringent emission limits.
C
=
BST_
∆V
BST_
where Q is the total gate charge of the high-side
G
MOSFET and ΔV
on the high-side MOSFET driver after turn-on. Choose
is the voltage variation allowed
BST_
When using an external clock source (i.e., driving the
FSYNC input with an external clock), spread spectrum is
disabled.
ΔV such that the available gate-drive voltage is not
BST_
significantly degraded (e.g., ΔV
= 100mV to 300mV)
BST_
when determining C
.
BST_
The boost capacitor should be a low-ESR ceramic
capacitor. A minimum value of 100nF works in most cases.
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Current Limiting and
Overcurrent Protection
Current-Sense Inputs (OUT_ and CS_)
If the inductor current in the IC exceeds the maximum
current limit programmed at CS_ and OUT_, the respec-
tive driver turns off. In an overcurrent mode, this results in
shorter and shorter high-side pulses.
The current-limit circuit uses differential current-sense
inputs (OUT_ and CS_) to limit the peak inductor current.
If the magnitude of the current-sense signal exceeds the
current-limit threshold (V
= 80mV (typ)), the PWM
LIMIT1,2
A hard short results in a minimum on-time pulse every
clock cycle. Choose the components so they can with-
stand the short-circuit current if required.
controller turns off the high-side MOSFET. The actual
maximum load current is less than the peak current-
limit threshold by an amount equal to half the inductor
ripple current; therefore, the maximum load capability is
a function of the current-sense resistance, inductor value,
Overvoltage Protection
The IC limits the output voltage of the buck converters
by turning off the high-side gate driver at approximately
109% of the regulated output voltage. The output voltage
needs to come back in regulation before the high-side
gate driver starts switching again.
switching frequency, and duty cycle (V
/V ).
OUT_ IN
For the most accurate current sensing, use a current-
sense shunt resistor (R ) between the inductor and the
SH
output capacitor. Connect CS_ to the inductor side of R
SH
and OUT_ to the capacitor side. Dimension R such that
SH
LOAD,MAX
Design Procedure
the maximum inductor current (I
= I
+ 1/2
across R
L,MAX
I ) induces a voltage of V
RIPPLE,PP
,
SH
LIMIT1,2
Buck Converter Design Procedure
Effective Input Voltage Range in Buck Converters
Although the MAX20034 can operate from input supplies
up to 36V (42V transients) and regulate down to 1V, the
including all tolerances.
For higher efficiency, the current can also be measured
directly across the inductor. This method could cause
up to 30% error over the entire temperature range and
requires a filter network in the current-sense circuit. See
the Current-Sense Measurement section.
minimum voltage conversion ratio (V
/V ) might be
OUT IN
limited by the minimum controllable on-time. For proper
fixed-frequency PWM operation and optimal efficiency,
Buck 1 and Buck 2 should operate in continuous conduc-
tion during normal operating conditions. For continuous
conduction, set the voltage-conversion ratio as follows:
Voltage Monitoring (PGOOD_)
The IC includes several power-monitoring signals to facili-
tate power-supply sequencing and supervision. PGOOD_
can be used to enable circuits that are supplied by the cor-
responding voltage rail, or to turn on subsequent supplies.
V
OUT
> t
× f
ON(MIN)
SW
V
IN
Each PGOOD_ goes high (high impedance) when the cor-
responding regulator output voltage is in regulation. Each
PGOOD_ goes low when the corresponding regulator
output voltage drops below 92% (typ) or rises above 95%
(typ) of its nominal regulated voltage. Connect a 10kΩ (typ)
pullup resistor from PGOOD_ to the relevant logic rail to
level-shift the signal.
where t
is 50ns (typ) and f
is the switching
ON(MIN)
SW
frequency in Hz. If the desired voltage conversion does not
meet the above condition, pulse skipping occurs to
decrease the effective duty cycle. Decrease the switching
frequency if constant switching frequency is required. The
same is true for the maximum voltage conversion ratio.
PGOOD_ asserts low during soft-start, soft-discharge, and
when either buck converter is disabled (either EN1 or EN2
is low).
The maximum voltage conversion ratio is limited by the
maximum duty cycle (97%).
V
OUT
< 0.97
Thermal-Overload, Overcurrent, and
Overvoltage/Undervoltage Behavior
V
− V
DROP
IN
where V
= I
(R
+ R
) is the sum of
DCR
Thermal-Overload Protection
DROP
OUT
ON,HS
the parasitic voltage drops in the high-side path, and f
is the programmed switching frequency. During low-drop
operation, the device reduces f to 80kHz. In practice,
the above condition should be met with adequate margin
SW
Thermal-overload protection limits total power dissipation in
the IC. When the junction temperature exceeds +170°C, an
internal thermal sensor shuts down the device, allowing it to
cool. The thermal sensor turns the device on again after the
junction temperature cools by 20°C.
SW
for good load-transient response.
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
The ratio of the inductor peak-to-peak AC current to DC
average current (LIR) must be selected first. A good initial
value is a 30% peak-to-peak ripple current to average-
current ratio (LIR = 0.3). The switching frequency, input
voltage, output voltage, and selected LIR then determine
the inductor value as follows:
Setting the Output Voltage in Buck Converters
Connect FB1 and FB2 to BIAS to enable the fixed buck
controller output voltages (5V and 3.3V) set by a preset
internal resistive voltage-divider connected between
the output (OUT_) and AGND. To externally adjust the
output voltage between 1V and 10V, connect a resistive
divider from the output (OUT_) to FB_ to AGND (see
(V − V
) x D
OUT
IN
the Typical Operating Circuit). Calculate R
and
L
=
MIN1
FB_1
f
x I
x LIR
R
with the following equation:
SW OUT
, and I are typical values (so that
OUT
FB_2
where V , V
IN
OUT
V
OUT_
efficiency is optimum for typical conditions).
FB_2
−1
R
= R
FB_1
V
FB_
The next equation ensures that the inductor current
downslope is less than the internal slope compensation:
where V
table).
= 1V (typ) (see the Electrical Characteristics
FB_
m2
m ≥
DC output accuracy specifications in the Electrical
Characteristics table refer to the error comparator’s thresh-
2
V
OUT
L
m2 =
× A
× R
VCS CS
old, V
= 1V (typ). When the inductor conducts continu-
FB_
ously, the device regulates the peak of the output ripple,
so the actual DC output voltage is lower than the slope-
compensated trip level by 50% of the output ripple voltage.
where m is the internal slope compensation, m2 is the
inductor current downslope, A is current-sense gain
VCS
In discontinuous-conduction mode (skip or STDBY active
(11V/V), and R
is current-sense resistor.
CS
and I
< I ), the device regulates the valley of
LOAD(SKIP)
OUT
Solving for L and adding 1.5 multiplier to account for
tolerances in the system:
the output ripple, so the output voltage has a DC regulation
level higher than the error-comparator threshold.
Inductor Selection in Buck Converters
Three key inductor parameters must be specified
for operation with the MAX20034: inductance value (L),
( V
× A
× R
×1.5)
CS
OUT
VCS
L
=
MIN2
2m
inductor saturation current (I
), and DC resistance
SAT
Select the larger of L
and L
as L .
MIN
MIN1
MIN2
(R
). To determine the optimum inductance, knowing
DCR
the typical duty cycle (D) is important.
The maximum inductor value is recommended to:
= 1.6 x L
L
MAX
MIN
V
V
OUT
OUT
D =
or D =
For example, in a typical-use case, 5V output voltage and
5A output current, R is 15mΩ, m is 0.4V/µs for 2.2MHz
V
V
−I
(R
+ R )
DCR
IN
IN OUT DS(ON)
CS
if the R
of the inductor and R
of the MOSFET
and 0.08V/µs for 400kHz. For 2.2MHz:
DCR
DS(ON)
are available with V = (V
- V ). All values should
DIODE
IN
BAT
be typical to optimize the design for normal operation.
(5×11× 0.015×1.5)
L
=
= 1.5µH
MIN
2× 0.4
Inductance
(
)
The exact inductor value is not critical and can be adjust-
ed to make trade-offs among size, cost, efficiency, and
transient-response requirements.
L
= 1.6×1.5 = 2.4µH
MAX
Therefore, a 2.2µH inductor is chosen for 2.2MHz.
For 400kHz, L is calculated as 7.7µH and L
● Lower inductor values increase LIR, which minimizes
size and cost, and improves transient response at the
cost of reduced efficiency due to higher peak currents.
is
MIN
MAX
12.3µH; therefore, a 10µH inductor is chosen.
● Higher inductance values decrease LIR, which
increases efficiency by reducing the RMS current at
the cost of requiring larger output capacitors to meet
load-transient specifications.
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Current-Sense Measurement
Peak Inductor Current
For the best current-sense accuracy and overcurrent
protection, use a ±1% tolerance current-sense resistor
between the inductor and output, as shown in Figure
1A. This configuration constantly monitors the inductor
current, allowing accurate current-limit protection. Use low-
inductance current-sense resistors for accurate measure-
ment.
Inductors are rated for maximum saturation current. The
maximum inductor current equals the maximum load
current, in addition to half the peak-to-peak ripple
current:
∆I
INDUCTOR
I
= I
+
LOAD(MAX)
PEAK
2
Alternatively, high-power applications that do not require
highly accurate current-limit protection can reduce the
overall power dissipation by connecting a series RC circuit
across the inductor (see Figure 1B) with an equivalent time
constant:
For the selected inductance value, the actual peak-to-peak
inductor ripple current (ΔI ) is calculated as:
INDUCTOR
V
(V − V
)
OUT IN
OUT
x L
SW
∆I
=
INDUCTOR
V
x f
IN
R2
where ΔI
is in mA, L is in µH, and f
is in kHz.
R
=
R
DCR
INDUCTOR
SW
CSHL
R1+ R2
Once the peak current and inductance are known, the
inductor can be selected. The saturation current should
and:
be larger than I
, or at least in a range where the
PEAK
L
1
1
inductance does not degrade significantly. The MOSFETs
are required to handle the same range of current without
dissipating too much power.
R
=
+
DCR
C
R1 R2
EQ
where R
is the required current-sense resistor and
CSHL
MOSFET Selection in Buck Converters
R
is the inductor’s series DC resistor. Use the
DCR
Each step-down controller drives two external logic-level
n-channel MOSFETs as the circuit switch elements. The
key selection parameters to choose these MOSFETs
include the items in the following sections.
inductance and R
values provided by the inductor
DCR
manufacturer. If DCR sense is the preferred current-
sense method, the recommended resistor value for R1
(Figure 1B) should be less than 1kΩ.
Threshold Voltage
All four n-channel MOSFETs must be a logic-level type,
with guaranteed on-resistance specifications at V
4.5V. If the internal regulator is bypassed (for example:
Carefully observe the PCB layout guidelines to ensure
the noise and DC errors do no corrupt the differential
current-sense signals seen by CS_ and OUT_. Place the
sense resistor close to the device with short, direct traces,
making a Kelvin-sense connection to the current-sense
resistor.
=
GS
V
= 3.3V), the n-channel MOSFETs should be
EXTVCC
chosen to have guaranteed on-resistance at that gate-to-
source voltage.
Input Capacitor in Buck Converters
Maximum Drain-to-Source Voltage (V
DS(MAX)
All MOSFETs must be chosen with an appropriate V
rating to handle all V voltage conditions.
)
The discontinuous input current of the buck converters
cause large input ripple currents and therefore the input
capacitor must be carefully chosen to withstand the input
ripple current and keep the input-voltage ripple within
design requirements. The 180° ripple phase operation
increases the frequency of the input-capacitor ripple
current to twice the individual converter switching
frequency. When using ripple phasing, the worst-case
input-capacitor ripple current is when the converter with
the highest output current is on.
DS
IN
Current Capability
The n-channel MOSFETs must deliver the average
current to the load and the peak current during switching.
Choose MOSFETs with the appropriate average current
at V
= 4.5V, or V
= V
when the internal
GS
GS
EXTVCC
linear regulator is bypassed. For load currents below
approximately 3A, dual MOSFETs in a single package
can be an economical solution. To reduce switching noise
for smaller MOSFETs, use a series resistor in the BST_
path and additional gate capacitance. Contact the factory
for guidance using gate resistors.
The input-voltage ripple is composed of ΔV (caused by
Q
the capacitor discharge) and ΔV
(caused by the ESR
ESR
of the input capacitor). The total voltage ripple is the sum
of ΔV and ΔV that peaks at the end of an on-cycle.
Q
ESR
Maxim Integrated
│ 16
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
INPUT (V )
IN
C
IN
MAX20034
NH
NL
DH_
LX_
DL_
R
L
SENSE
C
OUT
GND
CS_
OUT_
A) OUTPUT SERIES-RESISTOR SENSING
INPUT (V )
IN
C
IN
MAX20034
NH
INDUCTOR
DH_
L
DCR
LX_
C
OUT
R1
R2
NL
DL_
R2
C
EQ
R
=
R
DCR
CSHL
GND
(R1 + R2)
L
1
1
R2
+
CS_
R
=
DCR
[R1 ]
C
EQ
OUT_
B) LOSSLESS INDUCTOR SENSING
Figure 1. Current-Sense Configurations
Maxim Integrated
│ 17
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Calculate the input capacitance and ESR required for a
specific ripple using the following equation:
The amount of overshoot (V
no-load transient due to stored inductor energy can be
calculated as:
) during a full-load to
SOAR
∆V
ESR
2
ESR[Ω] =
(∆I
) L
LOAD(MAX)
∆I
P− P
2
V
≈
SOAR
I
+
LOAD(MAX)
2C
V
OUT OUT
V
ESR Considerations
OUT
V
I
x
LOAD(MAX)
The output-filter capacitor must have low enough equiva-
lent series resistance (ESR) to meet output ripple and
load-transient requirements, yet have high enough ESR
to satisfy stability requirements. When using high-capac-
itance, low-ESR capacitors, the filter capacitor’s ESR
dominates the output-voltage ripple, so the output capaci-
tor’s size depends on the maximum ESR required to meet
IN
C
[µF] =
IN
∆V x f
(
)
Q
SW
where:
V
− V
x V
OUT
(
=
)
IN
OUT
x f
∆I
P−P
V
x L
IN SW
the output-voltage ripple (V
) specifications:
x LIR
LOAD(MAX)
RIPPLE(P-P)
I
is the maximum output current in A, ΔI
is
P-P
LOAD(MAX)
V
= ESR x I
the peak-to-peak inductor current in A, f
is the switch-
RIPPLE(P−P)
SW
ing frequency in MHz, and L is the inductor value in µH.
In standby mode, the inductor current becomes
discontinuous, with peak currents set by the idle-mode
The internal 5V linear regulator (BIAS) includes an output
UVLO with hysteresis to avoid unintentional chattering
during turn-on. Use additional bulk capacitance if the
input source impedance is high. At lower input voltage,
additional input capacitance helps avoid possible under-
shoot below the undervoltage-lockout threshold during
transient loading.
current-sense threshold (V
= 26mV (typ)).
CS,SKIP
Transient Considerations
The output capacitor must be large enough to absorb
the inductor energy while transitioning from no-load to
full-load condition without tripping the overvoltage fault
protection. The total output voltage sag is the sum of the
voltage sag while the inductor is ramping up, and the volt-
age sag before the next pulse can occur. Therefore:
Output Capacitor in Buck Converters
The actual capacitance value required relates to the
physical size needed to achieve low ESR, as well as to
the chemistry of the capacitor technology. The capacitor
is usually selected by ESR and the voltage rating rather
than by capacitance value.
2
L ∆I
(
)
LOAD(MAX)
C
=
OUT
2V
(V x D
− V
)
OUT
SAG IN
MAX
When using low-capacity filter capacitors, such as
ceramic capacitors, size is usually determined by the
∆I
t − ∆t
(
)
LOAD(MAX)
+
V
SAG
capacity needed to prevent V
and V
from
SAG
SOAR
causing problems during load transients. Generally, once
enough capacitance is added to meet the overshoot
requirement, undershoot at the rising load edge is no
longer a problem (see the Transient Considerations
section). However, low-capacity filter capacitors typically
have high-ESR zeros that can affect the overall stability.
where D
is the maximum duty factor (approximately
MAX
95%), L is the inductor value in µH, C
capacitor value in µF, t is the switching period (1/f ) in
µs, and Δt equals (V
is the output
OUT
SW
/V ) x t.
OUT IN
The IC uses a peak current-mode-control scheme that
regulates the output voltage by forcing the required
current through the external inductor, so the controller
uses the voltage drop across the DC resistance of the
inductor or the alternate series current-sense resistor
to measure the inductor current. Current-mode control
eliminates the double pole in the feedback loop caused
by the inductor and output capacitor, resulting in a smaller
phase shift and requiring less elaborate error-amplifier
The total voltage sag (V
) can be calculated as follows:
SAG
L(∆I
2
)
LOAD(MAX)
((V ×D
V
=
SAG
2C
) − V
)
OUT
OUT
IN
MAX
∆I
(t − ∆t)
LOAD(MAX)
+
C
OUT
Maxim Integrated
│ 18
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
compensation than voltage-mode control. A single series
resistor (R ) and capacitor (C ) is all that is required
(typ), and R
is the output resistance of the error
OUT,EA
amplifier, which is 2.2MΩ (typ) (see the Electrical
Characteristics table.)
C
C
to have a stable, high-bandwidth loop in applications
where ceramic capacitors are used for output filter-
ing (see Figure 2). For other types of capacitors, due
to the higher capacitance and ESR, the frequency of
the zero created by the capacitance and ESR is lower
than the desired closed-loop crossover frequency. To
stabilize a nonceramic output capacitor loop, add another
A dominant pole (f
capacitor (C ) and the amplifier output resistance
) is set by the compensation
dpEA
C
(R
). A zero (f
OUT,EA
) is set by the compensation
ZEA
resistor (R ) and the compensation capacitor (C ). There
C
C
is an optional pole (f
output capacitor ESR zero if it occurs near the crossover
) set by C and R to cancel the
PEA
F C
compensation capacitor (C ) from COMP to AGND to
F
frequency (f , where the loop gain equals 1 (0dB)). Thus:
C
cancel this ESR zero.
1
The basic regulator loop is modeled as a power
modulator, output feedback-divider, and an error amplifier,
as shown in Figure 2. The power modulator has a DC
f
=
dpEA
2π × C × (R
+ R )
C
C
OUT,EA
1
gain set by g
x R , with a pole and zero pair set
LOAD
f
=
mc
zEA
pEA
2π × C × R
C
C
C
by R , the output capacitor (C ), and its ESR. The
LOAD OUT
loop response is set by the following equations:
1
GAIN = g ×R
f
=
MOD(dc)
mc
LOAD
2π × C × R
F
where R
= V
/I
in Ω and g =1/(A
LOAD
OUT LOUT(MAX) mc V_CS
The loop-gain crossover frequency (f ) should be set
C
below 1/15th the switching frequency and much higher
x R ) in S. A
is the voltage gain of the current-sense
DC
V_CS
amplifier and is typically 11V/V. R
the inductor or the current-sense resistor in Ω.
is the DC resistance of
DC
than the power-modulator pole (f ). Select a value
pMOD
for f in the range:
C
In a current-mode step-down converter, the output
capacitor and the load resistance introduce a pole at the
following frequency:
f
SW
15
f
<< f
≤
C
pMOD
1
At the crossover frequency, the total loop gain must be
equal to 1. Therefore:
f
=
pMOD
2π × C
× R
LOAD
OUT
The unity-gain frequency of the power stage is set by
and g
V
FB
C
:
OUT
mc
GAIN
×
× GAIN
= 1
)
C
MOD(f
)
EA(f
C
C
V
OUT
gmc
2π × C
f
=
UGAINpMOD
OUT
GAIN
= g
×R
EA(f
)
m,EA
C
The output capacitor and its ESR also introduce a zero at:
1
f
pMOD
GAIN
= GAIN
×
f
=
MOD(f
MOD(f
)
MOD(dc)
zMOD
C
C
f
2π × ESR× C
C
OUT
Therefore:
GAIN
Solving for R :
When C
is composed of “n” identical capacitors in
OUT
V
FB
×
parallel, the resulting C
= n x C
, and ESR
OUT
OUT(EACH)
× GAIN
= 1
)
)
EA(f
C
V
= ESR
/n. Note that the capacitor zero for a parallel
(EACH)
OUT
combination of alike capacitors is the same as for an
individual capacitor.
C
V
The feedback voltage-divider has a gain of GAIN
=
FB
OUT
R
=
C
V
/V
, where V is 1V (typ).
g
× V ×GAIN
FB MOD(f
FB OUT FB
m,EA
)
C
The transconductance error amplifier has a DC gain
of GAIN = g x R , where g is
EA(DC)
m,EA
OUT,EA
m,EA
the error-amplifier transconductance, which is 470µS
Maxim Integrated
│ 19
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Set the error-amplifier compensation zero formed by R
C
and C at the f
. Calculate the value of C as follows:
C
pMOD
C
g
= 1/(A
x R
)
DC
VCS
mc
1
CS_
C
=
C
CURRENT-MODE
POWER
2π × f
× R
C
pMOD
OUT_
MODULATION
If f
is less than 5 x f , add a second capacitor (C )
C F
zMOD
from COMP to AGND. The value of C is:
F
R1
R2
g
= 470µS
mea
R
ESR
1
FB_
C
=
F
2π × f
× R
C
COMP_
ERROR
AMP
zMOD
C
OUT
As the load current decreases, the modulator pole also
decreases; however, the modulator gain increases accord-
ingly and the crossover frequency remains the same.
R
V
REF
C
2.2MΩ
C
F
C
C
The following is a numerical example to calculate the
compensation network component values of Figure 2:
● A
= 11V/V
V_CS
DCR
Figure 2. Compensation Network
● R
= 15mΩ
● g = 1/(A
x R ) = 1/(11 x 0.015) = 6.06
DC
mc
V_CS
Applications Information
● V
= 5V
OUT
Layout Recommendations
● I
= 5.33A
OUT(MAX)
Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. The switching power
stage requires particular attention (see Figure 3). If pos-
sible, mount all the power components on the top side of
the board, with their ground terminals flush against one
another. Follow these guidelines for good PCB layout:
● R
● C
= V
/I
= 5V/5.33A = 0.9375Ω
LOAD
OUT OUT(MAX)
= 2x47µF = 94µF
OUT
● ESR = 9mΩ/2 = 4.5mΩ
● f = 0.403MHz
SW
● Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
jitter-free operation.
GAIN
= 6.06×0.9375 = 5.68
MOD(dc)
1
f
=
≈ 1.8kHz
pMOD
● Keep the power traces and load connections short.
This practice is essential for high efficiency. Using
thick copper PCBs (2oz vs. 1oz) can enhance full load
efficiency by 1% or more.
2π × 94µF× 0.9375
f
SW
15
f
<< f
≤
C
pMOD
● Minimize current-sensing errors by connecting CS_ and
1.8kHz << f ≤ 80.6kHz
C
OUT_. Use kelvin sensing directly across the current-
sense resistor (R
). A high-frequency filter is
SENSE_
select f = 25kHz:
C
required if operating above 1.8MHz. The recommended
RC filter values are 20Ω/100pF. Refer to the MAX20034
EV kit data sheet schematic for details.
1
f
=
≈ 376kHz
zMOD
2π × 4.5mΩ × 94µF
● Route high-speed switching nodes (BST_, LX_, DH_,
and DL_) away from sensitive analog areas (FB_, CS_,
and OUT_).
since f
> f :
C
zMOD
● R ≈ 25kΩ
C
● C ≈ 3.3nF
C
● C ≈ 18pF
F
Maxim Integrated
│ 20
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
from DH_ to the gate of the external HS switch and
from the LX_ pin to the inductor. Up to 100mA of
current flow from the BIAS capacitor through the
bootstrap diode to the bootstrap capacitor. Dimension
those traces accordingly.
Layout Procedure
1) Place the power components first, with ground
terminals adjacent (low-side FET, C , C , and
IN
OUT_
Schottky). If possible, make all these connections on
the top layer with wide, copper-filled areas.
4) Make the DC-DC controller ground connections as
shown in Figure 3. This diagram can be viewed as
having two separate ground planes: power ground,
where all the high-power components go; and an
analog ground plane for sensitive analog components.
The analog ground plane and power ground plane
must meet only at a single point directly under the IC.
2) Mount the controller IC adjacent to the low-side
MOSFET, preferably on the back side opposite DL_
and DH_ to keep LX_, PGND, DH_, and the DL_ gate
drive lines short and wide. The DL_ and DH_ gate
traces must be short and wide (50 mils to 100 mils
wide if the MOSFET is 1in from the controller IC) to
keep the driver impedance low and for proper adaptive
dead-time sensing.
5) Connect the output-power planes directly to the output-
filter capacitor positive and negative terminals with
multiple vias. Place the entire DC-DC converter circuit
as close as possible to the load.
3) Group the gate-drive components (BST_ diode and
capacitor and LDO bypass capacitor BIAS) together
near the controller IC. Be aware that gate currents of
up to 1A flow from the bootstrap capacitor to BST_,
KELVIN-SENSE VIAS
UNDER THE SENSE RESISTOR
(REFER TO THE MAX20034 EVALUATION KIT)
INDUCTOR
LOW-SIDE
n-CHANNEL
MOSFET (NH)
C
C
OUT
OUT
OUTPUT
GROUND
HIGH-SIDE
n-CHANNEL
MOSFET (NL)
INPUT
Figure 3. Layout Example
Maxim Integrated
│ 21
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Typical Operating Circuit
R
PGOOD1
FB1
PGOOD1
DL1
CS1
COUT1
OUT1
CS1
OUT1
CS1
R
CS1
*
OUT1
LX1
L1
BST1
EN1
MAX20034
DH1
PGND2
FSYNC
FOSC
COMP1
AGND
COMP2
EN2
BIAS
PGND1
EXTVCC
OUT1
V
BAT
IN
C
IN
DH2
BST2
LX2
R
*
CS2
L2
OUT2
CS2
CS2
OUT2
OUT2
C
OUT2
CS2
DL2
PGOOD2
FB2
PGND2
*DCR SENSE IS ALSO AN OPTION.
Maxim Integrated
│ 22
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Ordering Information
V
ADJUSTABLE
1V to 10V
OUT
SPREAD
SPECTRUM
PART
TEMP RANGE
PIN-PACKAGE
FIXED
5V/3.3V
5V/3.3V
MAX20034ATIR/VY+
MAX20034ATIS/VY+
-40°C to +125°C
-40°C to +125°C
28 TQFN-EP**
28 TQFN-EP**
Off
1V to 10V
On
/V denotes an automotive qualified part.
+Denotes a lead(Pb)-free/RoHS-compliant package.
**EP = Exposed pad.
Chip Information
PROCESS: BiCMOS
Maxim Integrated
│ 23
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MAX20034
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck
Controller with 17μA Quiescent Current
Revision History
REVISION REVISION
PAGES
DESCRIPTION
CHANGED
NUMBER
DATE
0
1
9/17
Initial release
Removed future product status from MAX20034ATIR/VY+ in Ordering Information
—
2/18
23
Updated the Simplified Block Diagram, TOC01–TOC02, TOC08, TOC11–TOC12,
TOC23, Pin Description table, and the Inductance and Transient Considerations section.
2, 6–7, 9, 11,
15, 19–20
2
3
4/18
8/19
Updated title to indicate automotive part; updated Benefits and Features, TOC5 inTypi-
cal Operating Characteristics, and Pin Description
1, 6, 10
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated’s website at www.maximintegrated.com.
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses
are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits)
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
©
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
2018 Maxim Integrated Products, Inc.
│ 24
相关型号：
MAX20034ATISVY
Automotive High-Efficiency 2.2MHz, 36V, Dual Buck Controller with 17μA Quiescent Current
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