MAX25611CAUD/V+T [MAXIM]
LED Driver,;
| 型号： | MAX25611CAUD/V+T |
| 厂家： | 
MAXIM INTEGRATED PRODUCTS |
| 描述： | LED Driver, |
| 文件： | 总25页 (文件大小：2124K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
EVALUATION KIT AVAILABLE
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
General Description
The MAX25611A/MAX25611B/MAX25611C/MAX25611D
Benefits and Features
● Automotive Ready: AEC-Q100 Qualified
are single-channel HBLED drivers for automo-
tive front light applications such as high beam,
low beam, daytime running light (DRL), turn
indicator, fog light, and other LED lights. It can take an
input voltage from 5V to 36V and can drive a string of
LEDs with a maximum output voltage of 65V.
● Integration Minimizes BOM for High-Brightness
LED Driver
• Integrated pMOS Dimming FET Gate Driver Allows
Single-Wire Connection to LED String
• PWM, Analog-to-PWM and Analog Dimming
• Integrated High-Side, Current-Sense Amplifier
• 12-Pin SWTQFN-EP Package
The MAX25611A/B/C/D sense output current at
the high side of the LED string. High-side current
sensing is required to protect for shorts from the output
to the ground or battery input. It is also the most flex-
ible scheme for driving LEDs, allowing boost, high-side
buck, SEPIC mode, or buck-boost mode configura-
tions. The PWM input provides LED dimming ratios of
up to 5000:1, and the REFI input provides additional
analog dimming capability in the MAX25611A/B/C/D.
The MAX25611A/B/C/D have built-in spread-spectrum
modulation for improved electromagnetic compat-
ibility performance. The MAX25611A/B/C/D can also
be used in zeta and Cuk converter configurations if it is
necessary in some applications.
● Flexible Application Configurations
• +5V to +36V Wide Input Voltage Range with a
Maximum +65V Boost Output
• Boost, Buck-Boost, High-Side Buck, SEPIC, Zeta,
and Cuk Single-Channel LED Drivers
● Protection Features and Wide Temperature Range
Increase System Reliability
• Short-Circuit, Overvoltage, and Thermal Protection
• -40°C to +125°C Operating Temperature Range
Ordering Information appears at end of data sheet.
Simplified Application Circuit
The MAX25611A/B/C/D are available in a space-saving
12-pin SWTQFN-EP package. They are specified to oper-
ate over the -40°C to +125°C automotive temperature
range. The switching frequency is internally set at 350kHz
for the MAX25611A/MAX25611C and 2.2MHz for the
MAX25611B/MAX25611D.
Applications
● Automotive Exterior Lighting
● High Beam/Low Beam/Signal/Position Lights
● Daytime Running Lights (DRLs)
● Fog Lights and Adaptive Front-Light Assemblies
● Head-Up Displays
● Commercial, Industrial, and Architectural Lighting
19-100429; Rev 2; 5/19
MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Absolute Maximum Ratings
IN to GND (MAX25611A/B)...................................-0.3V to +40V
IN to GND (MAX25611C/D)...................................-0.3V to +52V
ISENSEP, ISENSEN, DIMOUT to GND................-0.3V to +70V
DIMOUT to ISENSEP..............................................-6V to +0.3V
ISENSEP to ISENSEN.........................................-0.3V to +0.6V
Continuous Current on NDRV..........................................+50mA
Short-Circuit Duration on V ...................................Continuous
CC
Continuous Power Dissipation (T = +70°C)
A
Multilayer Board
TQFN (derate 25mW/°C above +70°C).....................1951mW
TSSOP (derate 10mW/°C above +70°C) ...............796.80mW
Operating Temperature Range......................... -40°C to +125°C
Junction Temperature......................................................+150°C
Soldering Temperature (reflow).......................................+260°C
Lead Temperature (soldering, 10s) .................................+300ºC
Storage Temperature Range............................ -65°C to +150°C
V
to GND ............................................................-0.3V to +6V
CC
NDRV to GND .............................................-0.3V to V
PWMDIM, REFI, OVP to GND................................-0.3V to +6V
COMP, CS to GND......................................-0.3V to V + 0.3V
Continuous Current on IN ................................................100mA
Peak Current on NDRV.........................................................±1A
+ 0.3V
CC
CC
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these
or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect
device reliability.
Package Information
12-Pin, SWTQFN-EP
Package Code
T1244Y+4C
21-100312
90-0068
Outline Number
Land Pattern Number
Thermal Resistance, Single-Layer Board:
Junction to Ambient (θ
)
59.3°C/W
6°C/W
JA
Junction to Case (θ
)
JC
Thermal Resistance, Four-Layer Board:
Junction to Ambient (θ
)
24.4°C/W
41°C/W
JA
Junction to Case (θ
)
JC
14-Pin, TSSOP
Package Code
Outline Number
U14+5C
21-0066
Land Pattern Number
Thermal Resistance, Single-Layer Board:
Junction to Ambient (θ
)
110°C/W
30°C/W
JA
Junction to Case (θ
)
JC
Thermal Resistance, Four-Layer Board:
Junction to Ambient (θ
)
100.4°C/W
30°C/W
JA
Junction to Case (θ
)
JC
For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a “+”,
“#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing
pertains to the package regardless of RoHS status.
Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer board.
For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Electrical Characteristics
(V = 12V, C = C
= 1μF, NDRV = COMP = DIMOUT = PWMDIM = unconnected, V
= V
= V
= 0V, V
=
IN
IN
VCC
CS
OVP
GND
ISENSEP
V
= 45V, V
= 1.20V. Limits are 100% tested at T = +25°C and T = +125°C. Limits over the operating temperature range
ISENSEN
REFI A A
and relevant supply voltage range are guaranteed by design and characterization.) (Note 1)
PARAMETER
SYMBOL
CONDITIONS
MIN
5
TYP
MAX UNITS
SUPPLY VOLTAGES AND SUPPLY CURRENT
V
MAX25611A/B
36
IN
V
V
t < 1s, MAX25611A/B
MAX25611C/D
40
IN_MAX
Input Voltage Range
V
V
IN
5
48
t < 1s, MAX25611C/D
52
IN_MAX
Quiescient Supply Current
I
V
= 1.5V, no switching, T = +25°C
1.8
3.5
mA
INQ
OVP
A
V
CC
REGULATOR
Output Voltage
V
Load = 0.1mA to 15mA
4.875
5
5.125
V
V
CC
VCC UVLO Rising
VCC
Rising, 1V (typ) hysteresis
4.3
50
UVLOR
Short-Circuit Current Limit
SWITCHING FREQUENCY
I
V
shorted to GND
mA
VCC_SC
CC
MAX25611A
MAX25611B
Dither enable
315
350
2200
±6
385
Switching Frequency
f
kHz
%
SW
1980
2420
Frequency Dither
f
SW_DITH
SLOPE COMPENSATION
Slope Compensation Current
Ramp Height
Peak current ramp out from CS pin
per switching period
I
42.5
50
57.5
μA
SLOPE
ANALOG DIMMING
REFI Input Control Voltage Range
REFI Zero Current Threshold
REFI Internal Clamp Voltage
REFI Input Bias Current
V
0.2
1.2
0.20
1.35
500
V
V
REFI_RNG
V
(V
- V ) < 5mV
ISENSEN
0.16
1.25
0.18
1.3
20
REFI_ZTH
ISENSEP
V
REFI sink = 1μA
= 0V to 5.5V
V
REFI_CLMP
I
V
nA
REFI
REFI
LED CURRENT SENSE AMP
Common-Mode Input Range
Differential Signal Range
-0.2
0
+65
200
V
mV
(V
V
- V
= 60V
) = 200mV,
) = 200mV,
ISENSEP
ISENSEP
ISENSEN
ISENSEP Input Bias Current
ISENSEN Input Bias Current
I
350
22
550
60
μA
μA
B_ISENSEP
(V
- V
ISENSEP
ISENSEN
ISENSEN
I
B_ISENSEN
V
= 60V
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Electrical Characteristics (continued)
(V = 12V, C = C
= 1μF, NDRV = COMP = DIMOUT = PWMDIM = unconnected, V
= V
= V
= 0V, V
=
IN
IN
VCC
CS
OVP
GND
ISENSEP
V
= 45V, V
= 1.20V. Limits are 100% tested at T = +25°C and T = +125°C. Limits over the operating temperature range
ISENSEN
REFI A A
and relevant supply voltage range are guaranteed by design and characterization.) (Note 1)
PARAMETER
Voltage Gain
SYMBOL
CONDITIONS
- V ) = 200mV,
MIN
TYP
MAX UNITS
(V
3V < (V
ISENSEP
ISENSEN
4.9
5
5.1
226
206
44
V/V
, V
) < 60V
) < 60V
) < 60V
) < 60V
< 3V
ISENSEP ISENSEN
V
= 1.3V,
REFI
214
194
36
220
200
40
3V < (V
, V
ISENSEP ISENSEN
LED Current-Sense
Regulation Voltage
V
REFI
= 1.2V,
V
mV
SENSE
3V < (V
, V
ISENSEP ISENSEN
V
REFI
= 0.4V,
3V < (V
, V
ISENSEP ISENSEN
V
= 1.2V,
REFI
193
35
200
40
207
45
0V < V
, V
ISENSEP ISENSEN
LED Current-Sense
Regulation Voltage (Low Range)
V
mV
V
SENSE_LOW
V
= 0.4V,
REFI
0V < V
, V
< 3V
ISENSEP ISENSEN
RNG
V
V
rising
falling
2.72
2.48
2.85
2.6
2.98
2.72
SEL
ISENSEP
Common-Mode Input
Range Selector
RNGSEL
ISENSEP
ERROR AMP
Transconductance
g
(V
- V ) = 200mV
ISENSEN
1170
1800
300
2430
μS
μA
μA
M
ISENSEP
COMP Sink Current
COMP
V
V
= 5V
ISINK
ISRC
COMP
COMP Source Current
PWM COMPARATOR
Input Offset Voltage
COMP
= 0V
300
COMP
1
V
PWM to NDRV Propagation Delay
CURRENT LIMIT COMPARATOR
Current Limit Threshold
GATE DRIVER (NDRV)
RDSon Pullup pMOS
RDSon Pulldown nMOS
Rise Time
Includes leading edge blanking time
90
ns
V
388
418
448
mV
CS_LIMIT
R
1.5
1.5
Ω
Ω
NDRV_HIGH
R
V
= 0V, I
= 100mA
NDRV_LOW
COMP
SINK
t
C
= 10nF
= 10nF
100
100
ns
ns
R
NDRV
NDRV
Fall Time
t
F
C
PWM DIMMING
Internal Ramp Frequency
External Sync Frequency Range
External Sync Low-Level Voltage
External Sync High-Level Voltage
f
160
60
200
240
2000
0.4
Hz
Hz
V
RAMP
f
DIM
V
PWMDIM_L
PWMDIM_H
V
2
V
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Electrical Characteristics (continued)
(V = 12V, C = C
= 1μF, NDRV = COMP = DIMOUT = PWMDIM = unconnected, V
= V
= V
= 0V, V
=
IN
IN
VCC
CS
OVP
GND
ISENSEP
V
= 45V, V
= 1.20V. Limits are 100% tested at T = +25°C and T = +125°C. Limits over the operating temperature range
ISENSEN
REFI A A
and relevant supply voltage range are guaranteed by design and characterization.) (Note 1)
PARAMETER
SYMBOL
CONDITIONS
MIN
170
3.1
TYP
MAX UNITS
DIM Comparator Offset Voltage
DIM Voltage for 100% Duty Cycle
PWMDIM Low to NDRV Low Delay
V
200
230
mV
V
PWMDIM_OFS
120
88
ns
PWMDIM High to NDRV
High Delay
ns
μs
μs
PWMDIM falling edge to rising edge
PWMDIM to LED Turn-Off Time
4.2
3.9
on DIMOUT, C
= 7nF
DIMOUT
PWMDIM rising edge to falling edge
PWMDIM to LED Turn-On Time
on DIMOUT, C
= 7nF
DIMOUT
DIMMING MOSFET GATE DRIVER (DIMOUT)
PWMDIM = low,
(V - V
Peak Pullup Current
25
10
50
25
-5
80
50
mA
mA
V
I
DIMOUT_PU
) = 5V (Note 2)
) = 0V (Note 2)
ISENSEP
DIMOUT
PWMDIM = high,
Peak Pulldown Current
I
DIMOUT_PD
(V
- V
ISENSEP
DIMOUT
DIMOUT Low Voltage with
Respect to ISENSEP
-5.4
-4.6
SHORT-CIRCUIT HICCUP MODE
Short-Circuit Current Threshold
V
(V
(V
- V
)
369
398
427
mV
V
IOUT_SHRT
ISENSEP
ISENSEP
ISENSEN
Short-Circuit Voltage Detect
Threshold
V
- V ) falling, V = 12V
1.15
1.55
1.95
VOUT_SHRT
IN
IN
After (V
and
) detected
Clock
Cycles
IOUT_SHRT
Hiccup Time
t
8192
HICCUP
V
VOUT_SHRT
THERMAL SHUTDOWN
Thermal Shutdown Threshold
Thermal Shutdown Hysteresis
T
Temperature rising
165
15
°C
°C
SHDN
T
HYS
OVERVOLTAGE PROTECTION (OVP)
OVP Threshold Rising
V
Output rising, 70mV hysteresis
= 1.235V
1.17
-500
1.23
1.29
V
OVP_TH
OVP Input Bias Current
I
V
+500
nA
OVP
OVP
Note 1: Limits are 100% tested at T = +25°C and T = +125°C. Limits over the operating temperature range and relevant supply
A
A
voltage range are guaranteed by design and characterization.
Note 2: Guaranteed by design. Not production tested.
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Typical Operating Characteristics
(Typical Operating Circuit, V = 12V, 8x LEDs, T = +25°C, unless otherwise noted.)
IN
A
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Typical Operating Characteristics (continued)
(Typical Operating Circuit, V = 12V, 8x LEDs, T = +25°C, unless otherwise noted.)
IN
A
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Pin Configuration
TOP VIEW
+
1
2
3
4
5
6
7
14 ISENSEP
PWMDIM
OVP
13
12
11
10
9
NC
C OM P
NC
ISENSEN
DIMOUTB
IN
MAX25611A/B/C/D
REFI
CS
VCC
8
GND
NDRV
TSSOP
Pin Description
TSSOP SW-TQFN
NAME
FUNCTION
14-PIN
12-PIN
Positive LED Current-Sense Input. Place a 0.1μF common-mode filter capacitor from ISENSEP
to
14
1
ISENSEP GND near the IC. Place a 100pF differential mode filter capacitor across ISENSEP and IS-
ENSEN
near the IC.
Dimming Control Input. Connect PWMDIM to an external 3.3V or 5V PWM signal for PWM dim-
ming.
For analog voltage controlled PWM dimming, connect PWMDIM to V
through a resistive
CC
voltage-divider with voltage between 0.2V and 3V. The dimming frequency is 200Hz under
1
2
PWMDIM
these conditions, and the duty cycle is (V - 0.2)/2.8V.
PWMDIM
Connect PWMDIM to GND to turn off the LEDs. Connect PWMDIM to V
for 100% duty cycle.
CC
Bypass PWMDIM to GND with a 0.1µF ceramic capacitor when using analog PWMDIM.
Overvoltage-Protection Input for the LED String. Connect a resistor-divider between the boost
output, OVP, and GND. When the voltage on OVP exceeds 1.23V, a fast-acting comparator im-
mediately stops PWM switching and pulls DIMOUT high to disconnect the LED string from the
boost output.
2
3
OVP
R
+ R
OVP2
(
)
OVP1
R
V
= 1.23
OVP
OVP2
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Pin Description (continued)
TSSOP SW-TQFN
NAME
FUNCTION
14-PIN
12-PIN
Compensation Network Connection. For proper compensation, connect a suitable RC network
3
4
COMP
from
COMP to GND.
Analog Dimming Control Input. The voltage at REFI sets the LED current level when V
1.3V.
<
REFI
This voltage reference can be set using a voltage divider from V
the internal reference sets the LED current.
to GND. For V
> 1.3V,
CC
REFI
5
6
5
6
REFI
V
− 0.2V
(
)
REFI
I
=
LED
5 × R
CS_LED
Bypass REFI to GND with at least a 10nF ceramic capacitor for noise filter. Not needed if V
REFI
> 1.3V.
Current-Sense Amplifier Positive Input for the Switching Regulator. Add a resistor from CS to
CS
the
switching MOSFET current-sense resistor terminal to program the slope compensation.
7
8
7
8
GND
Power and Analog Ground. Star point connection for power ground and analog ground.
External n-Channel Gate Driver Output
NDRV
5V Low-Dropout Voltage Regulator Output. V
supplies the bias for the gate drive and internal
CC
9
9
V
CC
control logic. Bypass V
to GND with a 4.7µF and 0.1µF ceramic capacitor.
CC
10
10
IN
Positive Power-Supply Input. Bypass IN to GND with at least a 1µF ceramic capacitor.
External Dimming p-Channel MOSFET Gate Driver. DIMOUT drives the gate of the external
p-Channel MOSFET based on the signal at PWMDIM.
PWMDIM
APPLICATION FUNCTION
DIMOUT
11
11
DIMOUT
External PFET off. LEDs disabled
Low
High, pulled up to ISENSEP
Low, pulled down to ISENSEP-5V
(dimmed).
High
External PFET on. LEDs enabled.
Negative LED Current-Sense Input. A 100Ω series resistor protects against large differential
voltages across ISENSEP and ISENSEN that might occur during an output short.v
12
12
ISENSEN
13
14
–
–
NC
NC
No Connection
No Connection
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Functional Diagram
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
When the external MOSFET is turned off, the inductor
current is transferred to the output. When the next switch-
ing cycle starts and the external MOSFET is turned on,
the inductor current starts ramping back up. Through this
repetitive action, the PWM-control algorithm establishes a
switching duty cycle to regulate current to the LED load.
Detailed Description
Functional Operation of the
MAX25611A/B/C/D
The MAX25611A/B/C/D is a constant-frequency, current-
mode controller with a low-side nMOS gate driver. The
operation of the devices is best understood by seeing the
Functional Diagram. The devices are enabled when the
5V regulator is above its UVLO limit of 4.5V (typ), before
switching on NDRV can begin. The nMOS gate-drive volt-
age and the control circuitry inside the device uses the
5V supply.
The external pMOS is turned on when PWMDIM is high
and is turned off when PWMDIM is low. This external dim-
ming MOSFET is a p-channel MOSFET and is connected
on the high side. The source of this pMOS is connected
to ISENSEN and the gate is connected to DIMOUT.
The drain of this MOSFET is connected to the anode of
the external LED string. In certain applications, it is not
necessary to use this dimming MOSFET and in these
cases, the DIMOUT pin is left open. During normal
operation when PWMDIM is high, the voltage across the
resistor from ISENSEP to ISENSEN is regulated to a
programmed voltage set by REFI.
When PWMDIM goes high, switching is initiated. The
internal oscillator runs at either 350kHz (MAX25611A) or
2.2MHz (MAX25611B). Additional spread-spectrum dith-
ering is added to the oscillator to alleviate EMI problems
in the LED driver. The internal oscillator is synchronized
to the positive going edge of the PWMDIM pulse. This
means that the NDRV pulse goes high at the same instant
as the positive-going pulse on PWMDIM. This guarantees
that the switching frequency variation over a period of a
PWMDIM pulse is the same from one PWMDIM pulse to
the next. This prevents flicker during PWM dimming espe-
cially for short PWMDIM pulse widths.
The external pMOS switch is also used for fault protection
as well. Once a fault condition is detected, the DIMOUT
pin is pulled high to turn off the pMOS switch. This iso-
lates the LED string from the fault condition and prevents
excessive voltage or current from damaging the LEDs.
Input Voltage (IN)
Once PWMDIM goes high, the external switching MOSFET
is turned on. Current flows through the external switching
MOSFET and this current is sensed by the voltage across
the current-sense resistor from the source of the external
MOSFET to GND. The source of the external MOSFET
is connected to the CS pin of the device through a slope-
The input supply pin (IN) must be locally bypassed with a
minimum of 1μF capacitance close to the pin. All the input
current drawn by the device goes through this pin. The
positive terminal of the bypass capacitor must be placed
as close as possible to this pin and the negative terminal
of the bypass capacitor must be placed as close as pos-
sible to the GND pin.
compensation resistor (R
). See the Typical Boost
Application Circuit. The slope-compensation current flows
SLOPE
out of the CS pin into the R resistor. The voltage
on CS is the voltage across the current-sense resistor
SLOPE
V
Linear Regulator
CC
The devices feature a 5V linear regulator (V ) with IN
as its source. Use a 4.7μF and a 0.1μF low-ESR ceramic
capacitor from V
regulator provides power to all the internal logic, control cir-
CC
(R ) + slope-compensation current x R . The
CS_FET
SLOPE
slope compensation prevents subharmonic oscillation
when duty cycles exceed 50%.
to GND for stable operation. The V
CC
CC
The voltage on CS is fed to a current-limit compara-
tor. This current-limit comparator is used to protect the
external switch from overcurrents and causes switch-
ing to stop for that particular cycle if the CS voltage
exceeds 0.418V (typ). An offset of 1.0V is added to the CS
voltage, and this voltage is fed to the positive input of a
PWM comparator. The negative input of this comparator
is a control voltage from the error amplifier that regulates
the LED current. When the positive input of the PWM
comparator exceeds the control voltage from the error
amplifier, the switching is stopped for that particular
cycle and the external nMOS stays off until the next
switching cycle.
cuitry and the MOSFET gate drive. The devices are enabled
when V
is above its UVLO threshold of 4.3V (typ).
CC
The overcurrent limit on the V
regulator is 150mA (typ)
CC
and the foldback short to GND current limit is 50mA (typ).
It is also possible to apply an external voltage on the
V
regulator output and save its power dissipation. The
CC
maximum externally applied voltage on V
should not
CC
exceed its absolute maximum rating.
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Dimming MOSFET Driver (DIMOUT)
n-Channel Switching-MOSFET Driver (NDRV)
The device drives an external n-channel switching
The devices require an external p-channel MOSFET
for PWM dimming. For normal operation, connect the
gate of the MOSFET to the output of the dimming driver
(DIMOUT). The dimming driver can sink up to 25mA
or source up to 50mA of peak current for fast charging
and discharging of the pMOS gate. When the PWMDIM
signal is high, this driver pulls the pMOS gate to 5V below
the ISENSEP pin to completely turn on the p-channel
dimming MOSFET. The DIMOUT pin inverts and level
shifts the signal on PWMDIM to drive the gate of the
external pMOS. In some applications, the pMOS dimming
MOSFET is not required, and the DIMOUT pin can be left
open.
MOSFET (NDRV). NDRV swings between V
and
CC
GND. NDRV can sink/source 1A of peak current, allowing
the ICs to switch MOSFETs in high-power applications.
The average current demanded from the supply to drive
the external MOSFET depends on the total gate charge
(Qg) and the operating frequency of the converter (f ).
SW
Use the following equation to calculate the driver supply
current (INDRV) required for the switching MOSFET:
I
= Qg x f
SW
NDRV
Switching-MOSFET Current-Sense Input (CS)
CS is part of the current-mode-control loop. The switching
control uses the voltage on CS, set by R
and R
CS_FET
SLOPE
LED Current-Sense Inputs
(ISENSEP, ISENSEN)
to terminate the on-pulse width of the switching cycle, thus
achieving peak current-mode control. Internal leading-edge
blanking of 66ns is provided to prevent premature turn-off
of the switching MOSFET in each switching cycle. Resistor
The differential voltage from ISENSEP to ISENSEN is
fed to an internal current-sense amplifier. This ampli-
fied signal is then connected to the negative input of the
transconductance error amplifier. The voltage-gain factor
of this amplifier is 5. The resistor connected between
ISENSEP and ISENSEN to programs the maximum LED
current. The full-scale signal is 220mV when the REFI
voltage is 1.3V or higher.
R
is connected between the source of the n-channel
CS_FET
switching MOSFET and GND. During switching, a current
ramp with a slope of 50μA x f is sourced from the CS pin.
SW
This current ramp, along with resistor R , programs
SLOPE
the amount of slope compensation.
Overvoltage Protection (OVP)
OVP sets the overvoltage-threshold limit across the LEDs.
Use a resistor-divider between ISENSEP to OVP and
GND to set the overvoltage-threshold limit. An internal
overvoltage-protection comparator senses the differential
voltage across OVP and GND. If the differential voltage
is greater than 1.23V, the device stops switching, NDRV
goes low, and DIMOUT goes high. When the differential
voltage drops by 70mV, NDRV is enabled if PWMDIM is
high and DIMOUT goes low.
Switching Frequency
The internal oscillator runs at either 350kHz (MAX25611A/
MAX25611C) or 2.2MHz (MAX25611B/MAX25611D). The
devices have built-in frequency dithering of ±6% of the
programmed frequency to alleviate EMI problems.
The internal oscillator is synchronized to the positive
going edge of the PWMDIM pulse. This means that
the NDRV pulse goes high at the same instant as the
positive-going pulse on PWMDIM. This guarantees that
the switching frequency variation over a period of a
PWMDIM pulse is the same from one PWMDIM pulse
to the next. This prevents flicker during PWM dimming
especially for short PWMDIM pulse widths.
Output Short-Circuit Protection
The MAX25611A/B/C/D feature output short-circuit pro-
tection. This feature is most useful where the LEDs are
connected over long cables and there is possibility of
shorts occurring when connectors are exposed.
A short circuit is detected when the following two condi-
tions are met:
Spread Spectrum
The device has an internal spread-spectrum option to
optimize EMI performance. The switching frequency is
varied ±6%, centered on the oscillator frequency (f
The modulation signal is a triangular wave with a period
● V
is lower than V by the V
ISENSEP
IN OUT_SHRT
).
OSC
threshold, -1.55V (typ)
● The current sense voltage across V
-
ISENSEP
of 418 clocks. Therefore, f ramps down 6% and back
OSC
V
exceeds the V
threshold,
ISENSEN
IOUT_SHRT
to the set frequency in 418 clocks, and also ramps up 6%
and back to the set frequency in another 418 clocks. The
total modulation period is 2.4ms for 350kHz and 380μs
for 2.2MHz.
398mV (typ)
The MAX25611A/B/C/D respond by stopping NDRV and
pulling DIMOUT high to ISENSEP to turn off the DIM
FET, disconnecting the output capacitors from the shorted
output.
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MAX25611A/MAX25611B/
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Automotive High-Voltage HB LED Controller
successive edges of a PWM signal with a frequency
between 60Hz and 2kHz, the device synchronizes to
the external signal and pulse-width modulates the LED
Current Limited Short-Circuit Protection
Faster current limited output short-circuit protection can be
achieved by adding a small-signal PNP transistor across
current at the external PWMDIM input frequency, with
the same duty cycle as the PWMDIM input. If an analog
control signal is applied to PWMDIM, the device com-
pares the DC input to an internally generated 200Hz ramp
R
as shown in Figure 1. The current is limited to
CS_LED
V
/R
, which is roughly three times the maximum
BE CS_LED
programmed current. When this limit is reached, the PNP
pulls up on the gate of the DIM FET P1, reducing the gate
voltage that increases the drain-source resistance to limit
the current. A 1kΩ resistor on DIMOUT allows the PNP
to drive the DIM FET gate high while DIMOUT is still low.
to pulse-width-modulate the LED current (f
= 200Hz).
DIM
The output-current duty cycle is linearly adjustable from
0% to 100% (0.2V < V < 3V). Use the following
PWMDIM
formula to calculate the voltage (V
), necessary
PWMDIM
Internal Transconductance Amplifier
for a given output-current duty cycle (D):
The devices have a built-in transconductance amplifier
used to amplify the error signal inside the feedback loop.
The typical transconductance is 1800µS.
V
= (D x 2.8V) + 0.2V
PWMDIM
where V
is the voltage applied to the PWMDIM pin.
PWMDIM
Rearranged to calculate duty cycle:
Analog Dimming
The device offers an analog dimming-control input pin
(REFI). The voltage at REFI sets the LED current level
V
− 0.2V
(
)
PWMDIM
Duty Cycle =
2.8V
linearly from zero with up to maximum when V
=
REFI
1.3V. For V
> 1.3V, an internal reference sets the
REFI
LED current. The maximum withstand voltage of this input
is 6V. The LED current is guaranteed to be at zero when
the REFI voltage is at or below the zero current threshold
of 0.18V (typ). The LED current can be linearly adjusted
from zero to full scale for the REFI voltage in the range
of 0.2V to 1.3V.
Ground (GND)
This pin is both the power ground and the analog ground.
Place the negative terminal of the IN and V
bypass
CC
capacitors as close as possible to the GND pin. The nega-
tive terminals for other decoupling capacitors on REFI,
PWMDIM, and COMP should be connected together and
connected to the GND pin through a path that does not
carry any high switching current.
Pulsed-Dimming Input (PWMDIM)
PWMDIM functions with either analog or PWM control
signals. Once the internal pulse detector detects three
Figure 1. Current Limited Short-Circuit Protection
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Automotive High-Voltage HB LED Controller
Thermal Shutdown
Setting the Overvoltage Threshold
Internal thermal-shutdown circuitry is provided to protect
the device in the event the maximum junction tempera-
ture is exceeded. The threshold for thermal shutdown is
+165°C (typ) with 15°C (typ) hysteresis. The part returns
to regulation mode once the junction temperature goes
below +150°C (typ). This results in a cycled output during
continuous thermal-overload conditions.
The overvoltage threshold is set by resistors ROVP1 and
ROVP2. See the Simplified Application Circuit. The over-
voltage circuit in the device is activated when the voltage
on OVP with respect to GND exceeds 1.23V. Use the fol-
lowing equation to set the desired overvoltage threshold:
R
+ R
OVP2
(
)
OVP1
R
V
= 1.23
OVP
OVP2
Fault Protection
The device shuts down when one of the following condi-
tions occur:
Inductor Selection
Boost and Buck-Boost Configurations
● Overvoltage or open across the LED string. The
device restarts after the output voltage drops below
the OVP hysteresis (70mV (typ) at the OVP pin).
Boost and buck-boost configurations are similar in that
the total output voltage seen by the inductor is always
higher than the input voltage. The difference being that for
the boost configuration, the total output voltage is depen-
dent on the total LED voltage, while for the buck-boost
configuration, the total output voltage is dependent on the
sum of the LED voltage and the input voltage.
● Short-circuit condition across the LED string. The
device enters hiccup mode and restarts after the
hiccup timer has expired. The hiccup timer is 8192
clock cycles.
●
Overtemperature condition. The device restarts after the
die temperature falls below the 15°C (typ) hysteresis.
In the boost converter, the average inductor current var-
ies with the line voltage. The maximum average current
occurs at the lowest line voltage.
Exposed Pad
For the boost converter, the average inductor current is
equal to the input current. Calculate maximum duty cycle
using the following equation:
The MAX25611A/B/C/D package features an exposed
thermal pad on its underside that should be used as a
heat sink. This pad lowers the package's thermal resis-
tance by providing a direct heat-conduction path from the
die to the PCB. Connect the exposed pad and GND to
the system ground using a large pad or ground plane, or
multiple vias to the ground plane layer.
V
+ V + V
+ V
− V
(
)
LED
D
RCS_LED
+ V
PFET
− V
INMIN
D
=
MAX
V
+ V + V
− V
(
)
LED
D
RCS_LED
PFET
NFET
RCS_FET
where:
● V
Applications Information
is the forward voltage of the LED string
LED
Programming the LED Current
● V is the forward drop of rectifier diode D1 (approxi-
D
Normal sensing of the LED current should be done on
the high side where the LED current-sense resistor is
connected to the anode of the LED string. The LED
current is programmed using resistor R
Simplified Application Circuit.
mately 0.6V)
● V
is the voltage across the LED current
RCS_LED
sense resistor R
(use 0.2V)
CS_LED
. See the
CS_LED
● V
is the average drain-to source voltage of
PFET
MOSFET P1 when it is on (use 0.2V initially)
The LED current can also be programmed adjusting the
● V
is the minimum input supply voltage
voltage on REFI when V
The current is given by:
≤ 1.3V (analog dimming).
INMIN
NFET
REFI
● V
is the average drain-to source voltage of
MOSFET N1 when it is on (use 0.2V initially)
V
− 0.2V
)
(
REFI
● V is the voltage across the NFET current
RCS_FET
I
=
LED
5 × R
sense resistor R
(use 0.3V initially)
CS_FET
CS_LED
Actual voltages for the above can be determined once
component selection is completed.
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Automotive High-Voltage HB LED Controller
In the buck-boost LED driver, the average inductor current
is equal to the input current plus the LED current. Calculate
the maximum duty cycle using the following equation:
● V
is the voltage across the LED current
RCS_LED
sense resistor R
(use 0.2V)
CS_LED
● V
is the maximum input supply voltage
INMAX
● V
is the average drain-to source voltage of
NFET
MOSFET N1 when it is on (use 0.2V initially)
● V is the voltage across the NFET current
RCS_FET
sense resistor R
(use 0.3V initially)
CS_FET
The maximum peak-to-peak inductor ripple (∆IL) occurs
at the maximum input line. The peak inductor current is
given by:
with the variables being the same as defined in the calcu-
lation of the boost configuration.
For both boost and buck-boost configurations, use the
following equations to calculate the maximum aver-
IL = I
+ 0.5 x ∆IL
PK
LED
The inductance value of inductor L
is calculated as:
age inductor current (IL
), peak-to-peak inductor
BUCK
DC_MAX
current ripple (∆IL), and the peak inductor current (IL ):
PK
IL
= I
/(1 - D
)
DC_MAX
LED
MAX
V
− V
− V
× D
MAX
(
INMIN
NFET
RCS_FET
L
=
Allowing the peak-to-peak inductor ripple to be ∆IL, the
peak inductor current is given by:
BUCK
f
× ∆ IL
where f
is the switching frequency, V
, V
INMAX
,
NFET
SW
IL
= IL
+ 0.5 x ∆IL
PK
DC_MAX
V
, V
, V
and ∆IL are defined above.
RCS_FET LED RCS_LED
The inductance value of inductor
L
or
BOOST
Choose an inductor that has a minimum inductance
greater than the calculated value.
L
is calculated as:
BUCK-BOOST
SEPIC, Zeta, and Cuk Configurations
V
− V
− V
× D
INMIN
NFET
RCS_FET MAX
(
In the SEPIC, zeta, and Cuk converters, there are sepa-
rate inductors for L1 and L2. Neglecting the drops in the
switching MOSFET and diode, the maximum duty cycle
L =
f
× ∆ IL
SW
where f
V
is the switching frequency, V
and ∆IL are defined above. Choose an induc-
tor that has a minimum inductance greater than the cal-
culated value. The current rating of the inductor should be
, V
,
SW
RCS_FET
INMIN
NFET
(D
) occurs at low line and is given by:
MAX
V
LED
D
=
MAX
V
+ V
INMIN
LED
higher than IL at the operating temperature.
PK
High-Side Buck Configuration
where V
LED
is the LED string voltage and V is the
INMIN
In the high-side buck LED driver, the average inductor
current is the same as the LED current. The peak inductor
current occurs at the maximum input line voltage where
the duty cycle is at the minimum:
minimum input voltage. If the desired maximum input cur-
rent ripple is ∆IL , then the inductor value of L1 is given by:
IN
V
× D
MAX
INMIN
× ∆ IL
L1 =
f
SW
IN
V
+ V + V
(
)
LED
− V
D
RCS_LED
− V
The peak inductor current in L1 is IL
and is given by:
D
=
INPK
MIN
V
+ V
(
)
INMAX
NFET
RCS_FET
D
D
MAX
IL
= I
+ 0.5 × ∆ IL
IN
INPK
LED
where:
1 − D
(
)
MAX
● V
is the forward voltage of the LED string
LED
● V is the forward drop of rectifier diode D1
To account for current transients, the peak saturation
rating of the inductor should be 1.2 times the calculated
value above.
D
(approximately 0.6V)
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Automotive High-Voltage HB LED Controller
The average output current in inductor L2 is the same
as the LED current. The desired maximum peak-to-peak
High-side buck configuration:
output current ripple is ∆IL
L2 is given by:
. The value of the inductor
OUT
2 × V
− V
× R
)
(
)
LED
INMIN
CS_FET
V
= D
× 1.5
SLOPE
MAX
2 × L × f
(
SW
V
× D
INMIN
SW
MAX
L2 =
f
× ∆ IL
OUT
SEPIC configuration:
The peak inductor current in L2 is IL
by:
and is given
OUTPK
V
− V
× R
(
)
LED
INMIN
CS_FET
IL
OUTPK
= I
+ 0.5 x ∆IL
OUT
V
= D
× 1.5
LED
SLOPE
MAX
2 × L
× f
(
)
SEPIC SW
Slope Compensation
Slope compensation should be added to converters
with peak current-mode-control operating in continuous-
conduction mode with more than 50% duty cycle to avoid
current-loop instability and subharmonic oscillations. The
minimum amount of slope compensation required for
stability is:
where L
= SQRT (L1 x L2) where L1 and L2 are the
SEPIC
two inductors in the SEPIC configuration.
MOSFET Current-Sense Resistor
The minimum value of the peak current-limit comparator
is 0.388V. The current-sense resistor value is given by:
V
= 0.5 x (inductor current downslope -
SLOPE(MIN)
R
= (0.388 - D
x V
)/IL
CS_FET
MAX
SLOPE PK
inductor current upslope) x R
CS_FET
where IL is the peak inductor current that occurs at low
PK
line in the boost, SEPIC, and buck-boost configurations.
In the MAX25611A/B/C/D, the slope-compensating ramp
is added to the current-sense signal before it is fed to the
PWM comparator. Connect a resistor (R
For boost configuration:
) from CS
SLOPE
to the switch current-sense resistor terminal for program-
ming the amount of slope compensation.
0.388
R
=
CS_FET
V
−2V
(
)
LED
L × f
INMIN
SW
The device generates a current ramp with a slope of
IL
+ 0.75D
MAX
PK
50μA/t
for slope compensation. The current-ramp sig-
OSC
nal is forced into an external resistor (R
) connected
SLOPE
For buck-boost configuration:
between CS and the source of the external MOSFET,
thereby adding a programmable slope-compensating volt-
0.388
R
=
age (V
) at the current-sense input CS. Therefore:
SLOPE
CS_FET
V
(
−V
)
LED
L × f
INMIN
SW
dV
)/dt = (R x 50μA)/t
SLOPE SLOPE OSC
IL
+ 0.75D
MAX
PK
The slope-compensation voltage that needs to be added
to the current signal at minimum line voltage, with margin
of 1.5x, is:
For SEPIC configuration:
0.388
Boost configuration:
R
=
CS_FET
V
f
−V
(
)
LED
INMIN
IL1
PK
+ IL2
PK
+ 0.75D
MAX
V
− 2 × V
× R
)
(
)
LED
INMIN
CS_FET
L1 × L2
(
√
SW
V
= D
× 1.5
)
SLOPE
MAX
2 × L × f
(
SW
Buck-boost configuration:
V
− V
× R
(
)
LED
INMIN
CS_FET
V
= D
× 1.5
SLOPE
MAX
2 × L × f
(
)
SW
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Automotive High-Voltage HB LED Controller
the required bulk capacitance. To minimize audible noise
generated by the ceramic capacitors during PWM dim-
ming, it may be necessary to minimize the number of
ceramic capacitors on the output. In these cases, an
additional electrolytic or tantalum capacitor provides most
of the bulk capacitance.
Input Capacitor
The input-filter capacitor bypasses the ripple current
drawn by the converter and reduces the amplitude of
high-frequency current conducted to the input supply.
The ESR, ESL, and bulk capacitance of the input
capacitor contribute to the input ripple. Use a low-ESR
input capacitor that can handle the maximum input
RMS ripple current from the converter. For the boost
configuration, the input current is the same as the induc-
tor current. For buck-boost configuration, the input current
is the inductor current minus the LED current. However,
for both configurations, the ripple current that the input
filter capacitor has to supply is the same as the induc-
tor ripple current with the condition that the output filter
capacitor should be connected to ground for buck-boost
configuration. Neglecting the effect of LED current ripple,
the calculation of the input capacitor for boost, as well
as buck-boost configurations is the same. Neglecting
the effect of the ESL, ESR, and bulk capacitance at the
input contributes to the input-voltage ripple. For simplicity,
assume that the contribution from the ESR and the bulk
capacitance is equal. This allows 50% of the ripple for the
bulk capacitance. The capacitance is given by:
Boost and Buck-Boost Configurations
The calculation of the output capacitance is the same
for both boost and buck-boost configurations. The output
ripple is caused by the ESR and bulk capacitance of the
output capacitor if the ESL effect is considered negligible.
For simplicity, assume that the contributions from ESR
and bulk capacitance are equal, allowing 50% of the rip-
ple for the bulk capacitance. The capacitance is given by:
I
× 2 × D
LED
=
VOUT
MAX
× f
C
OUT
RIPPLE SW
The remaining 50% of allowable ripple is for the ESR of
the output capacitor.
Based on this, the ESR of the output capacitor is given by:
VOUT
RIPPLE
ESR
=
COUT
IL
× 2
PK
∆ IL
C
=
4 × f
IN
× ∆ V
SW
IN
Rectifier Diode Selection
Use a Schottky diode as the rectifier (D1) for fast switch-
ing and to reduce power dissipation. Select a Schottky
diode with a voltage rating higher than that calculated by
the following equations:
The remaining 50% of allowable ripple is for the ESR of
the output capacitor.
Use X7R ceramic capacitors for optimal performance.
The selected capacitor should have the minimum required
capacitance at the operating voltage.
Boost configuration:
In the buck mode, the input capacitor has large pulsed
currents due to the current flowing in the freewheel-
ing diode when the switching MOSFET is off. It is very
important to consider the ripple-current rating of the input
capacitor in this application.
V
≥ (V
+ V + V
+ V
) x 1.2
PFET
D(KA)
LED
D
RCS_LED
Buck-boost configuration:
V
≥ (V
+V
+ V +
D(KA)
LED
INMAX D
V
+ V
) x 1.2
RCS_LED
PFET
where V
is the diode cathode to anode voltage rat-
D(KA)
Output Capacitor Selection
ing. The factor 1.2 provides 20% safety margin.
The function of the output capacitor is to reduce the out-
put ripple to acceptable levels. The ESR, ESL, and bulk
capacitance of the output capacitor contribute to the out-
put ripple. In most applications, the output ESR and ESL
effects can be dramatically reduced by using low ESR
ceramic capacitors. To reduce the ESL and ESR effects,
connect multiple ceramic capacitors in parallel to achieve
The current rating of the diode should be greater than I
in the following equation:
D
I
D
≥ IL
(1 - D ) x 1.5
MAX
DCMAX
where IL
is the average inductor current at V
.
DCMAX
INMIN
The factor 1.5 provides 50% safety margin.
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Automotive High-Voltage HB LED Controller
The switching converter small-signal transfer function
also has an output pole for both boost and buck-boost
configurations. The effective output impedance that deter-
mines the output pole frequency together with the output
filter capacitance is calculated as:
Switching MOSFET Selection
The switching MOSFET (N1) should have a voltage
rating sufficient to withstand the maximum output voltage
together with the diode drop of rectifier diode D1, and
any possible overshoot due to ringing caused by parasitic
inductances and capacitances. Use a MOSFET with a
drain-to-source voltage rating higher than that calculated
by the following equations:
Boost configuration:
R
+ R
× V
(
)
LED
CS_LED
LED
+ V
R
=
OUT
Boost configuration:
R
+
R
× I
(
)
LED
CS_LED
LED
LED
V
= (V
+ V + V
+ V ) x 1.2
PFET
DS
LED
D
RCS_LED
Buck-boost configuration:
Buck-boost configuration:
= (V +V + V + V
V
+ V ) x 1.2
PFET
DS
LED
INMAX
D
RCS_LED
R
+ R
× V
(
)
LED
CS_LED
LED
The factor 1.2 provides 20% safety margin.
R
=
OUT
R
+
R
× I
)
× D
MAX
+ V
(
LED
CS_LED
LED
LED
Dimming MOSFET Selection
Select a dimming MOSFET (P1) with continuous current
rating at the operating temperature higher than the LED
current by 30%. The drain-to-source voltage rating of the
where R
at the operating current.
is the dynamic impedance of the LED string
LED
dimming MOSFET must be higher than V
by 20%.
LED
The output pole frequency for both boost and buck-boost
configurations is calculated as follows:
Feedback Compensation
The LED current-control loop comprising the switching
converter, LED current amplifier, and the error amplifier
should be compensated for stable control of the LED
current. The switching converter small-signal transfer
function has a right half-plane (RHP) zero for both boost
and buck-boost configurations, as the inductor current is
in continuous-conduction mode. The RHP zero adds a
20dB/decade gain together with a 90° phase lag, which
is difficult to compensate. The easiest way to avoid this
zero is to roll off the loop gain to 0dB at a frequency less
than 1/5 of the RHP zero frequency with a -20dB/decade
slope.
1
f
=
2πR
P
C
OUT OUT
The feedback-loop compensation is done by connecting
a resistor (R ) and capacitor (C ) in series from
is chosen to set the highfre-
quency integrator gain for fast transient response, while
is chosen to set the integrator zero to maintain
loop stability. For optimum performance, choose the com-
ponents using the following equations:
COMP
COMP
COMP to GND. R
COMP
C
COMP
f
= 0.2× f
ZRHP
C
The value of R
and C
can be calculated as:
COMP
COMP
The worst-case RHP zero frequency (f
as follows:
) is calculated
ZRHP
Boost configuration:
2
)
V
× 1 − D
(
LED
MAX
25
f
=
ZRHP
C
=
2π × L × I
COMP
LED
π
x
f
x R
ZRHP
COMP
Buck-boost configuration:
2
)
V
+ V
× 1 − D
(
(
)
LED
INMIN
MAX
f
=
ZRHP
2π × L × I
LED
Maxim Integrated
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
b) The cathode of D1 must be connected very close
PCB Layout
to C
.
OUT
Typically, there are two sources of noise emission in a
switching power supply: high di/dt loops and high dV/dt
surfaces. For example, traces that carry the drain cur-
rent often form high di/dt loops. Similarly, the heatsink
of the MOSFET connected to the device drain presents
a dV/dt source; therefore, minimize the surface area of
the heatsink as much as is compatible with the MOSFET
power dissipation, or shield it. Keep all PCB traces car-
rying switching currents as short as possible to minimize
current loops. Use ground planes for best results.
c) C
and current-sense resistor R
must be
OUT
CS_FET
connected directly to the ground plane.
4) Connect the power GND of the high current switching
components to a star-point configuration.
5) Keep the power traces and load connections short.
This practice is essential for high efficiency. Use thick
copper PCBs (2oz vs. 1oz) to enhance full-load ef-
ficiency.
6) Route high-speed switching nodes away from the
sensitive analog areas. Use an internal PCB GND
plane as an EMI shield to keep radiated noise away
from the device, feedback dividers, and analog
bypass capacitors.
Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. Use a multilayer board
whenever possible for better noise immunity and power
dissipation. Follow these guidelines for good PCB layout:
1) Use a large contiguous copper plane under the IC
package. Ensure that all heat-dissipating components
have adequate cooling.
Voltage Regulator Configuration
The MAX25611A/B/C/D can be configured as voltage
regulators by using the voltage across ISENSEP and
ISENSEN as the feedback input for the output voltage
feedback divider.
2) Isolate the power components and high-current path
from the sensitive analog circuitry.
3) Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
jitter-free operation. Keep switching loops short:
V
− 0.2
R
+ R
VOUT2
(
)
(
)
REFI
5
VOUT1
R
V
=
×
OUT
VOUT1
a) The anode of D1 must be connected very close to
the drain of MOSFET N1.
Setting V
= 1.2V selects a large feedback signal that
REFI
improves accuracy and noise immunity.
Maxim Integrated
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Typical Application Circuits
Typical Operating Circuit
Typical Boost Application Circuit
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Typical Application Circuits (continued)
Typical Buck-Boost Application Circuit
Typical High-Side Buck Application Circuit
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Typical Application Circuits (continued)
Typical SEPIC Application Circuit
Typical Zeta Application Circuit
Maxim Integrated
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Typical Application Circuits (continued)
Typical Cuk Application Circuit
Typical Voltage Regulator Application Circuit
Maxim Integrated
│ 23
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Ordering Information
PART
PIN-PACKAGE
12 SWTQFN-EP*
14 TSSOP
FEATURE
350kHz
350kHz
2.2MHz
2.2MHz
350kHz
350kHz
2.2MHz
2.2MHz
MAXIMUM V
IN
MAX25611AATC/VY+
MAX25611AAUD/V+**
MAX25611BATC/VY+
MAX25611BAUD/V+**
MAX25611CATC/VY+
MAX25611CAUD/V+**
MAX25611DATC/VY+
MAX25611DAUD/V+**
36
36
36
36
48
48
48
48
12 SWTQFN-EP*
14 TSSOP
12 SWTQFN-EP*
14 TSSOP
12 SWTQFN-EP*
14 TSSOP
Note: All parts operate over the -40°C to +125°C automotive temperature range.
/V Denotes an automotive-qualified part.
Y Denotes side-wettable package.
+ Denotes a lead(Pb)-free/RoHS-compliant package.
*EP = Exposed pad.
** Future product – contact factory for details
Maxim Integrated
│ 24
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MAX25611A/MAX25611B/
MAX25611C/MAX25611D
Automotive High-Voltage HB LED Controller
Revision History
REVISION REVISION
PAGES
DESCRIPTION
CHANGED
NUMBER
DATE
0
12/18
Initial release
—
Updated Simplified Application Circuit, Pin Configuration, Pin Description, Output
Short-Circuit Protection, Figure 1, Typical Boost Application Circuit, Typical Buck-
Boost Application Circuit, Typical High-Side Buck Application Circuit, Typical SEPIC
Application Circuit, Typical Zeta Application Circuit, Typical Cuk Application Circuit,
Typical Voltage Regulator Application Circuit, Ordering Information, and added
Functional Diagram
1, 8, 10, 12,
13, 20–24
1
2
1/19
5/19
Updated title to include MAX25611C and MAX25611D; updated Absolute Maximum
Ratings, Package Information, Electrical Characteristics, Pin Configuration, Pin
Description, Detailed Description, and Ordering Information
1–24
For pricing, delivery, and ordering information, please visit Maxim Integrated’s online storefront at https://www.maximintegrated.com/en/storefront/storefront.html.
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses
are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits)
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
©
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
2019 Maxim Integrated Products, Inc.
│ 25
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