MAX8505 [MAXIM]
3A. 1MHz. 1% Accurate. Internal Switch Step-Down Regulator with Power-OK ; 3A 。 1MHz的。 1 ％准确。内置开关的降压型稳压器，带有电源就绪\n
| 型号： | MAX8505 |
| 厂家： | 
MAXIM INTEGRATED PRODUCTS |
| 描述： | 3A. 1MHz. 1% Accurate. Internal Switch Step-Down Regulator with Power-OK
|
| 文件： | 总15页 (文件大小：291K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
19-2992; Rev 0; 10/03
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
General Description
Features
The MAX8505 step-down regulator operates from a 2.6V
to 5.5V input and generates an adjustable output voltage
from 0.8V to 0.85 ✕ VIN at up to 3A. With a 2.6V to 5.5V
bias supply, the input voltage can be as low as 2.25V.
ꢀ Saves Space—4.9mm x 6mm Footprint, 1µH
Inductor, 47µF Ceramic Output Capacitor
ꢀ Input Voltage Range
2.6V to 5.5V
The MAX8505 integrates power MOSFETs and
operates at 1MHz/500kHz switching frequency to
provide a compact design. Current-mode pulse-width-
modulated (PWM) control simplifies compensation with
ceramic or polymer output capacitors and provides
excellent transient response.
Down to 2.25V with Bias Supply
ꢀ 0.8V to 0.85 ✕ V , 3A Output
IN
ꢀ Ceramic or Polymer Capacitors
ꢀ
1ꢀ Output Accuracy Over ꢁoad, ꢁine, and
Temperature
The MAX8505 features 1% accurate output over load,
line, and temperature variations. Adjustable soft-start is
achieved with an external capacitor. During the
soft-start period, the voltage-regulation loop is active.
This limits the voltage dip when the active devices,
such as microprocessors or ASICs connected to the
MAX8505’s output, apply a sudden load current step
upon passing their undervoltage thresholds.
ꢀ Fast Transient Response
ꢀ Adjustable Soft-Start
ꢀ In-Regulation Soft-Start ꢁimits Output-Voltage
Dips at Power-On
ꢀ POK Monitors Output Voltage
The MAX8505 features current-limit, short-circuit, and
thermal-overload protection and enables a rugged
design. Open-drain power-OK (POK) monitors the
output voltage.
Ordering Information
PART
TEMP RANGE
PIN-PACKAGE
Applications
µP/ASIC/DSP/FPGA Core and I/O Supplies
MAX8505EEE
-40°C to +85°C
16 QSOP
Chipset Supplies
Functional Diagram appears at end of data sheet.
Server, RAID, and Storage Systems
Network and Telecom Equipment
Pin Configuration
Typical Operating Circuit
TOP VIEW
LX
1
2
3
4
5
6
7
8
16 LX
BST
OUTPUT
0.8V TO
INPUT
2.6V TO 5.5V
IN
LX
15 PGND
14 LX
IN
LX
0.85 x V
3A
IN
MAX8505
IN
MAX8505
13 PGND
12 GND
11 REF
10 FB
PGND
V
CC
BST
FB
V
CC
COMP
REF
ENABLE
POWER-OK
CTL
POK
POK
CTL
9
COMP
GND
QSOP
________________________________________________________________ Maxim Integrated Products
1
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
ABSOLUTE MAXIMUM RATINGS
CTL, FB, IN, V
to GND .........................................-0.3V to +6V
Operating Temperature Range
CC
COMP, REF, POK to GND..........................-0.3V to (V
BST to LX..................................................................-0.3V to +6V
PGND to GND .......................................................-0.3V to +0.3V
+ 0.3V)
MAX8505EEE...................................................-40°C to +85°C
Storage Temperature Range.............................-65°C to +150°C
Junction Temperature......................................................+150°C
Lead Temperature (soldering, 10s) .................................+300°C
CC
Continuous Power Dissipation (T = +70°C)
A
16-Pin QSOP (derate 12.5mW/°C above +70°C).......1000mW
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(V = V
= V
= +3.3V, V = 0.8V, V
= 1.25V, C = 0.01µF, T = 0°C to +85°C, unless otherwise noted.)
REF A
IN
CC
CTL
FB
COMP
PARAMETER
CC
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
IN AND V
IN Voltage Range
Voltage Range
V
2.25
2.6
V
V
V
IN
CC
V
V
5.5
10
CC
CC
V
V
V
V
= 3.3V
= 5.5V
6
10
3
IN
IN Supply Current
Supply Current
I
IN
Switching with no load
Switching with no load
mA
mA
µA
V
IN
10
= 3.3V
= 5.5V
CC
CC
V
I
CC
CC
6
Total Shutdown Current into IN
and V
V
V
= V
= 0
= V
- V = 5.5V, V
= 0V,
IN
CC
BST
LX
CTL
I
20
50
SHDN
CC
LX
V
V
rising
falling
2.40
2.35
2.55
CC
CC
V
Undervoltage Lockout
When LX starts/stops
switching
CC
UVLO
th
Threshold
2.2
0.792
20
REF
REF Voltage
V
I
= 0µA, V = V = 2.6V to 5.5V
CC
0.800
13
0.808
100
30
V
Ω
REF
REF
IN
REF Shutdown Resistance
REF Soft-Start Current
From REF to GND, V
= 0V
CTL
V
= 0.4V
25
µA
REF
Output from 0% to 100%, C
1µF
= 0.01µF to
REF
Soft-Start Ramp Time
32
ms/µF
FB
FB Regulation Voltage
FB Input Bias Current
V
V
V
= 2.6V to 5.5V
= 0.7V
0.792
3
0.800
0.01
0.808
0.1
V
IN
µA
FB
= V
= 3.3V, V = 1.2V,
OUT
IN
CC
Maximum Output Current
I
A
OUT_MAX
L = 1µH/5.9mΩ (Note 1)
10.5
12
-12
50
13.5
FB high
FB low
FB Threshold for POK Transition
FB rising or falling
%
-13.5
-10.5
FB to POK Delay
COMP
FB rising or falling
µs
COMP Transconductance
Gain from FB to COMP
From FB to COMP
60
100
80
160
µS
dB
V
= 1.25V to 1.75V
COMP
2
_______________________________________________________________________________________
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
ELECTRICAL CHARACTERISTICS (continued)
(V = V
= V
= +3.3V, V = 0.8V, V
= 1.25V, C = 0.01µF, T = 0°C to +85°C, unless otherwise noted.)
REF A
IN
CC
CTL
FB
COMP
PARAMETER
SYMBOL
CONDITIONS
= 2.6V, 3.3V, 5.5V, V = 0.9V
MIN
0.45
1.7
TYP
0.75
1.9
MAX
1.00
2.1
UNITS
COMP Clamp Voltage, Low
COMP Clamp Voltage, High
COMP Shutdown Resistance
V
V
= V
= V
V
V
IN
IN
CC
CC
FB
= 2.6V, 3.3V, 5.5V, V = 0.7V
FB
From COMP to GND, V
= 0V
13
100
Ω
CTL
LX (All LX Outputs Connected Together)
V
= V
- V = 3.3V
BST LX
38
42
74
74
IN
LX On-Resistance, High
mΩ
V
V
= V
= V
- V = 2.6V
LX
IN
IN
BST
BST
- V = 3.3V
LX
38
LX On-Resistance, Low
mΩ
Ω
V
= V
- V = 2.6V
LX
42
IN
BST
LX Current-Sense Transresistance
LX Current-Limit Threshold
R
T
From LX to COMP
Sourcing, Typical Application Circuit
Sinking, V = V = 2.6V to 5.5V
0.068
4.6
0.086
5.6
0.104
6.6
A
-4.3
-2.6
-1.0
100
IN
CC
LX = 5.5V
LX = 0V
V
V
= V
= 5.5V,
CC
IN
CTL
LX Leakage Current
µA
= 0
-100
0.85
0.44
95
CTL = V
1
0.5
110
94
89
5
1.15
0.56
135
CC
V
5.5V
= V
= 2.6V, 3.3V,
IN
CC
LX Switching Frequency
LX Minimum Off-Time
LX Maximum Duty Cycle
MHz
ns
CTL = 2/3V
CC
V
V
5.5V
= V = 2.6V, 3.3V, 5.5V
CC
IN
500kHz
90
= V = 2.6V, 3.3V,
IN
CC
%
1MHz
500kHz
1MHz
84
8
V
5.5V
= V
= 2.6V, 3.3V,
CC
IN
LX Minimum Duty Cycle
%
10
15
SLOPE COMPENSATION
Slope Compensation
BST
Extrapolated to 100% duty cycle
245
300
400
mV
V
V
= 5.5V
= 0V
10
10
10
LX
LX
(V
- V ) = V
IN
=
BST
LX
BST Shutdown Supply Current
CTL
µA
V
= 5.5V, V
= 0
CTL
CC
LX open
For 1MHz
80
55
V
= V
= 2.6V,
CC
% of
V
CC
IN
CTL Input Threshold
For 500kHz
70
45
+1
3.3V, 5.5V
For shutdown
CTL Input Current
V
= 0V or 5.5V, V = V = 5.5V
CC
-1
µA
CTL
IN
POK (Power-OK)
POK Output Voltage, Low
POK Leakage Current
POK Fault Delay Time
THERMAL SHUTDOWN
Thermal-Shutdown Threshold
Thermal-Shutdown Hysteresis
V
V
= 0.6V or 1.0V, I
= 2mA
25
0.001
50
100
1
mV
µA
µs
FB
POK
= 5.5V
POK
From FB to POK, any threshold
25
100
When LX stops switching T rising
+170
20
°C
°C
J
_______________________________________________________________________________________
3
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
ELECTRICAL CHARACTERISTICS
(V = V
= V
= +3.3V, V = 0.8V, V
= 1.25V, C = 0.01µF, T = -40°C to +85°C, unless otherwise noted.) (Note 2)
REF A
IN
CC
CTL
FB
COMP
PARAMETER
CC
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
IN AND V
IN Voltage Range
Voltage Range
V
2.25
2.6
V
V
IN
CC
V
5.5
10
10
V
CC
mA
mA
IN Supply Current
Supply Current
I
IN
Switching with no load
Switching with no load
V
V
= 3.3V
IN
V
I
= 3.3V
CC
CC
CC
Total Shutdown Current into IN
and V
V
V
= V
= 0
= V
- V = 5.5V, V
= 0V,
IN
CC
BST
LX
CTL
I
50
µA
V
SHDN
CC
LX
V
V
rising
falling
2.55
CC
CC
V
Undervoltage Lockout
When LX starts/stops
switching
CC
UVLO
th
Threshold
2.2
REF
REF Voltage
V
I
= 0µA, V = V = 2.6V to 5.5V
CC
0.791
0.808
100
30
V
Ω
REF
REF
IN
REF Shutdown Resistance
REF Soft-Start Current
FB
From REF to GND, V
= 0V
CTL
V
= 0.4V
20
µA
REF
FB Regulation Voltage
FB Input Bias Current
V
V
V
= 2.6V to 5.5V
= 0.7V
0.791
0.808
0.1
V
FB
IN
µA
FB
V
= V = 3.3V, V
= 1.2V,
IN
CC
OUT
Maximum Output Current
I
3
A
OUT_MAX
L = 1µH/5.9mΩ (Note 1)
FB high
FB low
10.5
13.5
FB Threshold for POK Transition
FB rising or falling
%
-13.5
-10.5
COMP
COMP Transconductance
COMP Clamp Voltage, Low
COMP Clamp Voltage, High
COMP Shutdown Resistance
From FB to COMP
60
0.45
1.7
160
1.00
2.1
µS
V
V
V
= V
= V
= 2.6V, 3.3V, 5.5V, V = 0.9V
FB
IN
IN
CC
CC
= 2.6V, 3.3V, 5.5V, V = 0.7V
V
FB
From COMP to GND, V
= 0V
100
Ω
CTL
LX (All LX Outputs Connected Together)
LX On-Resistance, High
V
V
= V
= V
- V = 3.3V
LX
74
74
mΩ
mΩ
Ω
IN
IN
BST
BST
- V = 3.3V
LX
LX On-Resistance, Low
LX Current-Sense Transresistance
R
From LX to COMP
Sourcing, Typical Application Circuit
Sinking, V = V = 2.6V to 5.5V
0.068
4.6
0.104
5.6
T
LX Current-Limit Threshold
A
-4.3
-1.0
100
IN
CC
LX = 5.5V
V
V
= V
= 5.5V,
CC
IN
CTL
LX Leakage Current
µA
= 0
LX = 0V
CTL = V
-100
0.85
0.44
95
1.15
0.56
135
CC
V
= V = 2.6V,
IN
CC
LX Switching Frequency
LX Minimum Off-Time
MHz
ns
3.3V, 5.5V
CTL = 2/3 ✕ V
CC
V
= V
= 2.6V, 3.3V, 5.5V
CC
IN
4
_______________________________________________________________________________________
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
ELECTRICAL CHARACTERISTICS (continued)
(V = V
= V
= +3.3V, V = 0.8V, V
= 1.25V, C = 0.01µF, T = -40°C to +85°C, unless otherwise noted.) (Note 2)
REF A
IN
CC
CTL
FB
COMP
PARAMETER
SYMBOL
CONDITIONS
500kHz
= 2.6V, 3.3V,
MIN
90
TYP
MAX
UNITS
V
5.5V
= V
IN
CC
CC
LX Maximum Duty Cycle
LX Minimum Duty Cycle
%
1MHz
500kHz
1MHz
84
8
V
5.5V
= V
= 2.6V, 3.3V,
IN
%
15
SLOPE COMPENSATION
Slope Compensation
BST
Extrapolated to 100% duty cycle
245
406
mV
V
V
= 5.5V
10
10
10
LX
LX
(V
- V ) = V
=
= 0
BST
LX
IN
BST Shutdown Supply Current
CTL
µA
= 0V
V
= 5.5V, V
CC
CTL
LX open
For 1MHz
80
55
V
= V
= 2.6V,
CC
% of
V
CC
IN
CTL Input Threshold
For 500kHz
For shutdown
70
45
+1
3.3V, 5.5V
CTL Input Current
V
= 0V or 5.5V, V = V = 5.5V
CC
-1
µA
CTL
IN
POK (Power-OK)
POK Output Voltage, Low
POK Leakage Current
POK Fault Delay Time
V
V
= 0.6V or 1.0V, I
= 2mA
100
1
mV
µA
µs
FB
POK
= 5.5V
POK
From FB to POK, any threshold
25
100
Note 1: Under normal operating conditions, COMP moves between 1.25V and 2.15V as the duty cycle changes from 10% to 90%
and peak inductor current changes from 0 to 3A. Maximum output current is related to peak inductor current, inductor value
input voltage, and output voltage by the following equations:
I
−(1−D)× t × V
/2L
+R )/2L
L
LIM
S
OUT
I
=
OUT_MAX
1+(1−D)× t ×(R
S
NLS
where V
= output voltage; I
= current limit of high-side switch; t = switching period; R = ESR of inductor; R
=
OUT
LIM
S
L
NLS
on-resistance of low-side switch; L = inductor. Equations for I
and D are shown as follows:
LIM
1−D
I
=I
+ V
LIM LIM_DC100
SW
R
T
where I
= current limit at D = 100%; R = transresistance from LX to COMP; V
= slope compensation (310mV
T
LIM_DC100
SW
20%); D = duty cycle:
V
+I (R
+R )
OUT
O
NLS L
D =
V
+I (R
−R
)
IN
O
NLS
NHS
where V
= output voltage; V = input voltage; I = output current; R = ESR of inductor; R
= on-resistance of high-
OUT
IN
O
L
NHS
side switch; R
= on-resistance of low-side switch. See the Typical Application Circuit for external components.
NLS
Note 2: Specifications to -40°C are guaranteed by design and not production tested.
Note 3: LX has internal clamp diodes to PGND and IN pins 2 and 4. Applications that forward bias these diodes should take care
not to exceed the IC’s package power dissipation limits.
Note 4: When connected together, the LX output is designed to provide 3.5A
current.
RMS
_______________________________________________________________________________________
5
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
Typical Operating Characteristics
(Typical values are at V = V
= V
= 5V, V
= 1.2V, I
= 3A, and T = +25°C, unless otherwise noted.)
OUT A
IN
CC
CTL
OUT
EFFICIENCY vs. OUTPUT CURRENT
EFFICIENCY vs. OUTPUT CURRENT
EFFICIENCY vs. OUTPUT CURRENT
(V = V = 5V, f = 1MHz)
(V = V = 3.3V, f = 500kHz)
(V = V = 3.3V, f = 1MHz)
IN
CC
SW
IN
CC
SW
IN
CC
SW
100
90
80
70
60
50
100
90
80
70
60
50
100
90
80
70
60
50
A
A
A
B
B
B
D
C
C
D
C
D
A: V
OUT
B: V
OUT
C: V
OUT
D: V
OUT
= 3.3V
A: V
OUT
B: V
OUT
C: V
OUT
D: V
OUT
= 2.5V
= 1.8V
= 1.2V
= 0.8V
A: V
OUT
B: V
OUT
C: V
OUT
D: V
OUT
= 2.5V
= 2.5V
= 1.2V
= 0.8V
= 1.8V
= 1.2V
= 0.8V
0
1
2
3
4
0
1
2
3
4
0
1
2
3
4
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
EFFICIENCY vs. OUTPUT CURRENT
(V = 2.5V, V = 5V, f = 1MHz)
FREQUENCY vs. INPUT VOLTAGE AND
TEMPERATURE
IN
CC
SW
100
90
80
70
60
50
1.05
1.03
1.01
0.99
0.97
0.95
A
B
+85°C
C
+25°C
A: V
B: V
C: V
= 1.8V
= 1.2V
= 0.8V
OUT
OUT
OUT
-40°C
0
1
2
3
4
2.5
3.0
3.5
4.0
4.5
5.0
5.5
OUTPUT CURRENT (A)
INPUT VOLTAGE (V)
FREQUENCY vs. INPUT VOLTAGE AND
TEMPERATURE
OUTPUT LOAD REGULATION
530
520
510
500
490
480
470
6
5
4
3
2
1
0
A: V
B: V
C: V
D: V
= 0.8V
= 1.2V
= 1.8V
= 2.5V
OUT
OUT
OUT
OUT
+85°C
+25°C
C
D
B
A
-40°C
V
IN
= V = 3.3V
CC
2.5
3.0
3.5
4.0
4.5
5.0
5.5
0
1
2
3
4
INPUT VOLTAGE (V)
OUTPUT CURRENT (A)
6
_______________________________________________________________________________________
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
Typical Operating Characteristics (continued)
(Typical values are at V = V
= V
= 5V, V
= 1.2V, I
= 3A, and T = +25°C, unless otherwise noted.)
IN
CC
CTL
OUT
OUT
A
SHUTDOWN SUPPLY CURRENT
vs. INPUT VOLTAGE
CURRENT LIMIT vs. OUTPUT VOLTAGE
0.4
0.3
0.2
0.1
0
5.5
f
= 1MHz
SW
5.4
5.3
5.2
5.1
5.0
4.9
4.8
4.7
4.6
4.5
f
= 1MHz
5.0
SW
2.5
3.0
3.5
4.0
4.5
5.5
0.8
1.3
1.8
2.3
2.8
3.3
INPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
OUTPUT SHORT-CIRCUIT CURRENT
vs. INPUT VOLTAGE
GND-MEASURED TEMPERATURE
vs. OUTPUT CURRENT
5.5
4.5
3.5
2.5
120
100
80
60
40
20
0
f
= 1MHz
SW
T
= +85°C
A
T
= +25°C
A
V
OUT
= 5V,
= 1.5V
IN
V
T
= -40°C
A
2.5
3.0
3.5
4.0
4.5
5.0
5.5
3.00
3.25
3.50
3.75
4.00
INPUT VOLTAGE (V)
OUTPUT CURRENT (A)
TRANSIENT RESPONSE
REFERENCE VOLTAGE
vs. TEMPERATURE
(V = 5V, V
= 1.2V)
IN
OUT
MAX8505 toc13
0.810
0.805
0.800
0.795
0.790
OUTPUT VOLTAGE
AC-COUPLED
100mV/div
2.25A
OUTPUT
CURRENT
1A/div
0.75A
0
f
= 1MHz
110
SW
40µs/div
-40
-15
10
35
60
85
TEMPERATURE (°C)
_______________________________________________________________________________________
7
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
Typical Operating Characteristics (continued)
(Typical values are at V = V
= V
= 5V, V
= 1.2V, I
= 3A, and T = +25°C, unless otherwise noted.)
OUT A
IN
CC
CTL
OUT
TRANSIENT RESPONSE
(V = 3.3V, V = 1.2V)
SWITCHING WAVEFORM
(V = 5V, V = 1.2V, I = 2.5A)
IN
OUT
IN
OUT
OUT
MAX8505 toc15
MAX8505 toc14
V
LX
2V/div
OUTPUT VOLTAGE
AC-COUPLED
100mV/div
INDUCTOR CURRENT
AC-COUPLED
2A/div
2.25A
OUTPUT
CURRENT
1A/div
0.75A
0
V
OUT
AC-COUPLED
20mV/div
40µs/div
200ns/div
SOFT-START/SHUTDOWN WAVEFORM
(V = 3.3V, V = 1.2V, I = 3A, C = 0.068µF)
TRANSIENT RESPONSE DURING SOFT-START
IN
OUT
OUT
REF
MAX8505 toc16
MAX8505 toc17
V
OUT
100mV/div
V
OUT
500mV/div
V
CTRL
5V/div
INPUT CURRENT
1A/div
I
OUT
2A/div
V
POK
5V/div
400µs/div
100µs/div
8
_______________________________________________________________________________________
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
Pin Description
PIN
NAME
FUNCTION
Inductor Connection. Connect an inductor between these pins and the regulator output. All LX pins must
be connected together externally. Connect a 3300pF ceramic capacitor from LX to PGND.
1, 3, 14, 16
LX
IN
Power-Supply Inputs. Ranges from 2.6V to 5.5V. Bypass with two ceramic 22µF capacitors to GND. All IN
pins must be connected together externally.
2, 4
5
Bootstrapped Voltage Input. High-side driver supply pin. Bypass to LX with a 0.1µF capacitor. Charged
from IN with an external Schottky diode.
BST
Supply Voltage and Gate-Drive Supply for Low-Side Driver. Decouple with a 10Ω resistor and bypass to
GND with 0.1µF.
6
V
CC
Power-OK Output. Open-drain output of a window comparator that pulls POK low when the FB pin is
outside the 0.8V 12% range.
7
POK
CTL
Output Control. When at GND, the regulator is off. When at V , the regulator is operating at 1MHz. For a
CC
8
500kHz application, raise the pin to 2/3 V
.
CC
Regulator Loop Compensation. Connect a series RC network to GND. This pin is pulled to GND when the
output is shut down, or in UVLO or thermal shutdown.
9
COMP
FB
Feedback Input. This pin regulates to 0.8V. Use an external resistive-divider from the output to set the
output voltage.
10
11
12
REF
GND
Place a capacitor at this pin to set the soft-start time. This pin goes to 0V when the part is shut down.
Ground
13, 15
PGND
Power Ground. Connect this pin to GND at a single point.
compensation ramp is summed into the main PWM com-
Detailed Description
parator. During the second half of the cycle, the internal
The MAX8505 is a high-efficiency synchronous buck
regulator capable of delivering up to 3A of output
current. It operates in PWM mode at a high fixed
frequency of 500kHz or 1MHz, thereby reducing
external component size. The MAX8505 operates from
a 2.6V to 5.5V input voltage and can produce an output
high-side N-channel MOSFET turns off, and the internal
low-side N-channel MOSFET turns on. The inductor
releases the stored energy as its current ramps down
while still providing current to the output. The output
capacitor stores charge when the inductor current
exceeds the load current, and discharges when the
inductor current is lower, smoothing the voltage across
the load. Under overload conditions, when the inductor
current exceeds the current limit (see the Current Limit
section), the high-side MOSFET does not turn on at the
rising edge of the clock and the low-side MOSFET
remains on to let the inductor current ramp down.
voltage from 0.8V to 0.85 ✕ V .
IN
Controller Block Function
The MAX8505 step-down converter uses a PWM
current-mode control scheme. An open-loop comparator
compares the voltage-feedback error signal against the
sum of the amplified current-sense signal and the slope
compensation ramp. At each rising edge of the internal
clock, the internal high-side MOSFET turns on until the
PWM comparator trips. During this on-time, current ramps
up through the inductor, sourcing current to the output
and storing energy in the inductor. The current-mode
feedback system regulates the peak inductor current as a
function of the output-voltage error signal. Since the aver-
age inductor current is nearly the same as the peak
inductor current, the circuit acts as a switch-mode
transconductance amplifier. To preserve inner-loop
stability and eliminate inductor staircasing, a slope-
Current Sense
An internal current-sense amplifier produces a current
signal proportional to the voltage generated by the high-
side MOSFET on-resistance and the inductor current
(R
DS(ON)
✕ I ). The amplified current-sense signal and
LX
the internal slope-compensation signal are summed
together into the comparator’s inverting input. The PWM
comparator turns off the internal high-side MOSFET
when this sum exceeds the feedback voltage from the
voltage-error amplifier.
_______________________________________________________________________________________
9
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
Current Limit
Soft-Start
The MAX8505 offers both high-side and low-side
current limits. The high-side current limit monitors the
inductor peak current and the low-side current limit
monitors the inductor valley current. Current-limit thresh-
olds are 6A (typ) for high side and 3.8A (typ) for low
side. If the output inductor current exceeds the high-
side current limit during its on-time, the high-side MOS-
FET turns off and the synchronous rectifier turns on. The
inductor current is continuously monitored during the
on-time of the low-side MOSFET. If the inductor current
is still above the low-side current limit at the moment of
the next clock cycle, the high-side MOSFET is not
turned on and the low-side MOSFET is kept on to contin-
ue discharging the output inductor current. Once the
inductor current is below the low-side current limit, the
high-side MOSFET is turned on at the next clock cycle.
If the inductor current stays less than the high-side cur-
rent limit during the minimum on duty ratio, the normal
operation resumes at the next clock cycle. Otherwise,
the current-limit operation continues.
To reduce input transient currents during startup, a pro-
grammable soft-start is provided. The soft-start time is
given by:
0.8V
25µA
t
= C
×
SOFT_START
REF
A minimum capacitance of 0.01µF at REF is recom-
mended to reduce the susceptibility to switching noise.
Power-OK (POK)
The MAX8505 also includes an open-drain POK output
that indicates when the regulator output is within 12%
of its nominal output. If the output voltage moves
outside this range, the POK output is pulled to ground.
Since this comparator has no hysteresis on either
threshold, a 50µs delay time is added to prevent the
POK output from chattering between states. The POK
should be pulled to V or another supply voltage less
IN
than 5.5V through a resistor.
V
Decoupling
UVLO
CC
Due to the high switching frequency and tight output
tolerance (1%), decouple V from IN with a 10Ω
If V
drops below +2.25V, the UVLO circuit inhibits
CC
switching. Once V
rises above +2.35V, the UVLO
CC
CC
resistor and bypass to GND with a 0.1µF capacitor.
Place the capacitor as close to V as possible.
clears, and the soft-start sequence activates.
CC
Thermal Protection
Bootstrap (BST)
Thermal-overload protection limits total power dissipa-
tion in the device. When the junction temperature
Gate-drive voltage for the high-side N-channel switch is
generated by a bootstrapped capacitor boost circuit.
The bootstrapped capacitor is connected between the
BST pin and LX. When the low-side N-channel MOSFET
is on, it forces LX to ground and charges the capacitor
exceeds T = +170°C, a thermal sensor forces the
J
device into shutdown, allowing the die to cool. The ther-
mal sensor turns the device on again after the junction
temperature cools by 20°C, resulting in a pulsed output
during continuous overload conditions. Following a
thermal-shutdown condition, the soft-start sequence
begins anew.
to V through diode D1. When the low-side N-channel
IN
MOSFET turns off and the high-side N-channel MOSFET
turns on, LX is pulled to V . D1 prevents the capacitor
IN
from discharging into V and the voltage on the boot-
IN
Design Procedure
strapped capacitor is boosted above V . This provides
IN
the necessary voltage for the high driver. A Schottky
diode should be used for D1.
Duty Cycle
The equation below shows how to calculate the result-
ing duty cycle when series losses from the inductor and
internal switches are accounted for:
Frequency Selection/Enable (CTL)
The MAX8505 includes a frequency selection circuit to
allow it to run at 500kHz or 1MHz. The operating fre-
quency is selected through a control input, CTL, which
has three input threshold ranges that are ratiometric to
the input supply voltage. When CTL is driven to GND, it
acts like an enable pin, switching the output off. When
V
V
+I
(R
+ R )
V
+I
(R
+ R )
OUT OUT NLS
L
OUT OUT NLS L
D =
=
+I
(R
− R
)
V
IN
IN OUT NLS
NHS
if R
= R
NLS
NHS
the CTL input is driven to >0.8 ✕ V , the MAX8505 is
CC
where V
= output voltage; V = input voltage;
IN
OUT
enabled with 1MHz switching. When the CTL input is
I
= output current (3A maximum); R = ESR of the
OUT
L
between 0.55 ✕ V
and 0.7 ✕ V , the part operates
CC
CC
inductor; R
= on-resistance of the high-side switch;
NHS
at 500kHz. When the CTL input is <0.45 x V , the
CC
and R
= on-resistance of the low-side switch.
NLS
device is in shutdown.
10 ______________________________________________________________________________________
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
Output Voltage Selection
The output voltage of the MAX8505 can be adjusted
from 0.8V to 85% of the input voltage at 500kHz or up
to 80% of the input voltage at 1MHz. This is done by
connecting a resistive-divider (R2 and R3) between the
output and the FB pin (see the Typical Operating
Circuit). For best results, keep R3 below 50kΩ and
select R2 using the following equation:
2
2
2
V
= V
+ V
+ V
RIPPLE
RIPPLE(C)
RIPPLE(ESR) RIPPLE(ESL)
where the output ripples due to output capacitance,
ESR, and ESL are:
I
P−P
V
=
RIPPLE(C)
8 × C
× f
S
OUT
V
V
OUT
R2 =R3×
−1
REF
V
=I
×ESR
RIPPLE(ESR) P−P
where V
= 0.8V.
REF
I
t
I
P−P
P−P
Inductor Design
When choosing the inductor, the key parameters are
inductor value (L) and peak current (I ). The
following equation includes a constant, denoted as LIR,
which is the ratio of peak-to-peak inductor AC current
(ripple current) to maximum DC load current. A higher
value of LIR allows smaller inductance but results in
higher losses and ripple. A good compromise between
size and losses is found at approximately 20% to 30%
ripple-current to load-current ratio (LIR = 0.20 to 0.30):
V
=
× ESL or
× ESL,
RIPPLE(ESL)
t
ON
OFF
PEAK
or, whichever is greater.
The ESR is the main contribution to the output voltage
ripple.
I
, the peak-to-peak inductor current, is:
P-P
(V − V
)
V
OUT
V
IN
IN
OUT
I
=
×
P−P
f × L
S
V
× (1− D)
× LIR × f
S
OUT
Use these equations for initial capacitor selection,
but determine final values by testing a prototype or
evaluation circuit. As a rule, a smaller ripple current
results in less output voltage ripple. Since the inductor
ripple current is a factor of the inductor value, the
output voltage ripple decreases with larger inductance.
Use ceramic capacitors for their low ESR and ESL at the
switching frequency of the converter. The low ESL of
ceramic capacitors makes ripple voltages negligible.
Load-transient response depends on the selected
output capacitor. During a load transient, the output
L =
I
OUT
where f is the switching frequency and
S
(I
−I
)
PEAK OUT
LIR = 2×
I
OUT
Choose an inductor with a saturation current at least as
high as the peak inductor current. Additionally, verify
the peak inductor current does not exceed the current
limit. The inductor selected should exhibit low losses at
the chosen operating frequency.
instantly changes by ESR ✕ I
. Before the controller
LOAD
can respond, the output deviates further, depending on
the inductor and output capacitor values. After a short
time (see Transient Response in the Typical Operating
Characteristics), the controller responds by regulating the
output voltage back to its nominal state. The controller
response time depends on the closed-loop bandwidth,
the inductor value, and the slew rate of the transconduc-
tance amplifier. A higher bandwidth yields a faster
response time, thus preventing the output from deviating
further from its regulating value.
Output Capacitor Design and Output Ripple
The key selection parameters for the output capacitor
are capacitance, ESR, ESL, and the voltage rating
requirements. These affect the overall stability, output
ripple voltage, and transient response of the DC-DC
converter. The output ripple occurs due to variations in
the charge stored in the output capacitor, the voltage
drop due to the capacitor’s ESR, and the voltage drop
due to the capacitor’s ESL. Calculate the output voltage
ripple due to the output capacitance, ESR, and ESL as:
______________________________________________________________________________________ 11
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
Input Capacitor Design
The input capacitor reduces the current peaks drawn
from the input power supply and reduces switching
noise in the IC. The impedance of the input capacitor at
the switching frequency should be less than that of the
input source so high-frequency switching currents do
not pass through the input source but instead are
shunted through the input capacitor. A high source
impedance requires larger input capacitance. The
input capacitor must meet the ripple current require-
ment imposed by the switching currents. The RMS
input ripple current is given by:
For customized compensation networks that increase
stability or transient response, the simplified loop gain
can be described by the equation:
V
FB
A
=
× gm
× R
×
OERR
VOL
ERR
V
OUT
s × C
× R
+1)× (s × C
+1
COMP
COMP
×
(s × C
× R
× R
+1)
COMP
COMP
COMP
PARA
+1
ESR
R
R
s × C
s × C
× R
L
T
OUT
×
× R +1
OUT L
where:
V
× (V − V
)
OUT
IN
2
OUT
I
= I
×
RIPPLE
LOAD
gm
(COMP transconductance) = 100µmho
(output resistance of transconductance
amplifier) = 20MΩ
C (compensation capacitor at COMP pin)
COMP
ERR
V
IN
R
OERR
where I
is the input RMS ripple current.
RIPPLE
Use sufficient input bypass capacitance to ensure that
the absolute maximum voltage rating of the MAX8505 is
not exceeded in any condition. When input supply is
not located close to the MAX8505, a bulk bypass input
capacitor may be needed.
R (current-sense transresistance) = 0.086Ω
T
C
PARA
(parasitic capacitance at COMP pin) = 10pF
R (load resistor)
L
C
(output capacitor)
OUT
Compensation Design
The double pole formed by the inductor and output
capacitor of most voltage-mode controllers introduces
a large phase shift, which requires an elaborate
compensation network to stabilize the control loop.
The MAX8505 controller utilizes a current-mode control
scheme that regulates the output voltage by forcing
the required current through the external inductor,
eliminating the double pole caused by the inductor
and output capacitor, and greatly simplifying the
compensation network. A simple type 1 compensation
with single compensation resistor (R1) and compensa-
tion capacitor (C8) create a stable and high-bandwidth
loop (see the Typical Operating Circuit).
R
(series resistance of C
)
ESR
OUT
s = j2πf
In designing the compensation circuit, select an appro-
priate converter bandwidth (f ) to stabilize the system
C
while maximizing transient response. This bandwidth
should not exceed 1/10 of the switching frequency. Use
100kHz as a reasonable starting point. Calculate
C
based on this bandwidth using the following
COMP
equation:
I
×R ×(R3+R2)×2π × f ×C
T C OUT
OUT
R
=
COMP
V
×gm
×R2
OUT
ERR
An internal transconductance error amplifier compen-
sates the control loop. Connect a series resistor and
capacitor between COMP (the output of the error amplifi-
er) and GND to form a pole-zero pair. The external
inductor, internal current-sensing circuitry, output capaci-
tor, and external compensation circuit determine the loop
stability. Choose the inductor and output capacitor based
on performance, size, and cost. Additionally, select the
compensation resistor and capacitor to optimize control-
loop stability. The component values shown in the Typical
Operating Circuit yield stable operation over a broad
range of input-to-output voltages.
where R2 and R3 are the feedback resistors.
Calculate C
to cancel out the pole created by R
L
COMP
and C
using the following equation;
OUT
C
OUT
C
=R ×
L
COMP
R
COMP
12 ______________________________________________________________________________________
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
3) Keep the high-current paths as short and wide as
Applications Information
possible. Keep the path of switching current short
PC Board Layout Considerations
Careful PC board layout is critical to achieve clean and
stable operation. The switching power stage requires
particular attention. Follow these guidelines for good
PC board layout:
and minimize the loop area formed by the high-side
MOSFET, the low-side MOSFET, and the input
capacitors. Avoid vias in the switching paths.
4) If possible, connect IN, LX, and PGND separately to
a large copper area to help cool the IC to further
improve efficiency and long-term reliability.
1) Place decoupling capacitors as close to the IC as
possible. Keep power ground plane (connected to
PGND) and signal ground plane (connected to
GND) separate.
5) Ensure all feedback connections are short and
direct. Place the feedback resistors as close to the
IC as possible.
2) Connect input and output capacitors to the power
ground plane; connect all other capacitors to the
signal ground plane.
6) Route high-speed switching nodes away from sen-
sitive analog areas (FB, COMP).
Chip Information
TRANSISTOR COUNT: 3352
PROCESS: BiCMOS
Typical Application Circuit
D1
C7
0.1µF
(CENTRAL CMOSH-3)
L1
1µH
(FDV3H-IRON)
BST
V
OUT
1.2V
V
IN
IN
LX
2.6V TO 5.5V
C9
C
OUT
47µF
C2
3300pF
R7
2 x 22µF
(10V CERAMIC)
R2
MAX8505
10Ω
(6.3V CERAMIC)
11.3kΩ
PGND
FB
V
CC
C5
0.1µF
R1
51kΩ
R6
20kΩ
COMP
REF
CTL
POK
R3
22.6kΩ
C6
0.01µF
C8
220pF
POWER-OK
GND
______________________________________________________________________________________ 13
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
Functional Diagram
REFERENCE
1.25V
POK
MAX8505
N
50µs
V
CC
FB
UVLO
BST
IN
GND
25µA
25µA
REF
LX
PWM
FB
PGND
GND
COMP
CTL
14 ______________________________________________________________________________________
3A, 1MHz, 1% Accurate, Internal Switch
Step-Down Regulator with Power-OK
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)
PACKAGE OUTLINE, QSOP .150", .025" LEAD PITCH
1
21-0055
E
1
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 15
© 2003 Maxim Integrated Products
Printed USA
is a registered trademark of Maxim Integrated Products.
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