MIC2182-3.3YSM_技术文档

MIC2182

Micrel

MIC2182

High-Efficiency Synchronous Buck Controller

General Description

Features

Micrel’s MIC2182 is a synchronous buck (step-down) switch-

ing regulator controller. An all N-channel synchronous archi-

tecture and powerful output drivers allow up to a 20A output

current capabilty. The PWM and skip-mode control scheme

allows efficiency to exceed 95% over a wide range of load

current, making it ideal for battery powered applications, as

well as high current distributed power supplies.

• 4.5V to 32V Input voltage range

• 1.25V to 6V Output voltage range

• 95% efficiency

• 300kHz oscillator frequency

• Current sense blanking

• 5Ω impedance MOSFET Drivers

• Drives N-channel MOSFETs

• 600µA typical quiescent current (skip-mode)

The MIC2182 operates from a 4.5V to 32V input and can

operate with a maximum duty cycle of 86% for use in low-

dropout conditions. It also features a shutdown mode that

reduces quiescent current to 0.1µA.

• Logic controlled micropower shutdown (I < 0.1µA)

• Current-mode control

Q

• Cycle-by-cycle current limiting

• Built-in undervoltage protection

• Adjustable undervoltage lockout

• Easily synchronizable

• Precision 1.245V reference output

• 0.6% total regulation

• 16-pin SOP and SSOP packages

• Frequency foldback overcurrent protection

• Sustained short-circuit protection at any input voltage

• 20A output current capability

The MIC2182 achieves high efficiency over a wide output

current range by automatically switching between PWM and

skip mode. Skip-mode operation enables the converter to

maintain high efficiency at light loads by turning off circuitry

pertaining to PWM operation, reducing the no-load supply

current from 1.6mA to 600µA. The operating mode is inter-

nally selected according to the output load conditions. Skip

mode can be defeated by pulling the PWM pin low which

reduces noise and RF interference.

Applications

The MIC2182 is available in a 16-pin SOP (small-outline

package) and SSOP (shrink small-outline package) with an

operating range from –40°C to +85°C.

• DC power distribution systems

• Notebook and subnotebook computers

• PDAs and mobile communicators

• Wireless modems

• Battery-operated equipment

Typical Application

C11

22uf

35V

V_IN

D2

MIC2182-3.3BSM

4.5V to 30V*

₁₁SD103BWS

10

VIN

VDD

C9

4.7µF

16V

x2

C5

0.1µF

R7

14

BST

C6

0.1µF

100k

6

16

L1

EN/UVLO HSD

Q2*

V_OUT

3.3V/4A

R2

0.02Ω

10µH

Si4884

2

15

13

12

8

PWM

VSW

LSD

C4

C7

D1

B140

1nF

Q1*

220uf

Si4884

10V ×2

1

3

5

SS

PGND

CSH

GND

COMP

SYNC

C13, 1nF

R1

2k

9

C3

0.1µF

VOUT

VREF

7

* 30V maximum input voltage limit is due

to standard 30V MOSFET selection.

C2

2.2nF

SGND

C1

0.1µF

See “Application Information” section for

5V to 3.3V/10A and other circuits.

4

GND

4.5V–30V* to 3.3V/4A Converter

Micrel, Inc. • 1849 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com

M9999-042204

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MIC2182

Micrel

Ordering Information

Part Number

Voltage

Adjustable

3.3V

Temperature Range

–40°C to +85°C

Package

Lead Finish

Standard

Pb-Free

MIC2182BM

16-pin narrow SOP

16-pin narrow SSOP

16-pin narrow SOP

16-pin narrow SSOP

MIC2182-3.3BM

MIC2182-5.0BM

MIC2182BSM

5.0V

Adjustable

3.3V

MIC2182-3.3BSM

MIC2182-5.0BSM

MIC2182YM

5.0V

Adjustable

3.3V

MIC2182-3.3YM

MIC2182-5.0YM

MIC2182YSM

5.0V

Adjustable

3.3V

MIC2182-3.3YSM

MIC2182-5.0YSM

5.0V

Pin Configuration

MIC2182

MIC2182-x.x

SS

PWM

1

2

3

4

5

6

7

8

16 HSD

SS

1

16 HSD

15 VSW

14 BST

13 LSD

12 PGND

11 VDD

10 VIN

15 VSW

14 BST

13 LSD

12 PGND

11 VDD

10 VIN

PWM 2

COMP 3

SGND 4

COMP

SGND

SYNC

5

6

EN/UVLO

VREF

EN/UVLO

FB 7

CSH 8

9

VOUT

CSH

9 VOUT

Adjustable

16-pin SOP (M)

16-Pin SSOP (SM)

Fixed

16-pin SOP (M)

16-Pin SSOP (SM)

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MIC2182

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Pin Description

Pin Number

Pin Name

Pin Function

1

SS

Soft-Start (External Component): Connect external capacitor to ground to

reduce inrush current by delaying and slowing the output voltage rise time.

Rise time is controlled by an internal 5µA current source that charges an

external capacitor to V_DD

.

2

3

4

5

PWM

COMP

SGND

SYNC

PWM/Skip-Mode Select (Input): Low sets PWM-mode operation. 1nF

capacitor to ground sets automatic PWM/skip-mode selection.

Compensation (Output): Internal error amplifier output. Connect to capacitor

or series RC network to compensate the regulator control loop.

Small Signal Ground (Return): Route separately from other ground traces to

the (–) terminal of C_OUT

.

Frequency Synchronization (Input): Optional. Connect to external clock

signal to synchronize the oscillator. Leading edge of signal above the

threshold terminates the switching cycle. Connect to SGND if unused.

6

EN/UVLO

Enable/Undervoltage Lockout (Input): Low-level signal powers down the

controller. Input below the 2.5V threshold disables switching and functions

as an accurate undervoltage lockout (UVLO). Input below the threshold

forces complete micropower (< 0.1µA) shutdown.

7 (fixed)

7 (adj)

8

VREF

FB

Reference Voltage (Output): 1.245V output. Requires 0.1µf capacitor to

ground.

Feedback (Input): Regulates FB pin to 1.245V. See “Application Information”

for resistor divider calculations.

CSH

Current-Sense High (Input): Current-limit comparator noninverting input. A

built-in offset of 100mV between CSH and V_OUTpins in conjunction with the

current-sense resistor set the current-limit threshold level. This is also the

positive input to the current sense amplifier.

9

VOUT

Current-Sense Low (Input): Output voltage feedback input and inverting

input for the current limit comparator and the current sense amplifier.

10

11

VIN

[Battery] Unregulated Input (Input): +4.5V to +32V supply input.

VDD

5V Internal Linear-Regulator (Output): V_DDis the external MOSFET gate

drive supply voltage and an internal supply bus for the IC. Bypass to SGND

with 4.7µF. V_DDcan supply up to 5mA for external loads.

12

13

14

PGND

LSD

MOSFET Driver Power Ground (Return): Connects to source of synchro-

nous MOSFET and the (–) terminal of C_IN

Low-Side Drive (Output): High-current driver output for external synchronous

MOSFET. Voltage swing is between ground and V_DD

.

BST

Boost (Input): Provides drive voltage for the high-side MOSFET driver. The

drive voltage is higher than the input voltage by V_DDminus a diode drop.

15

16

VSW

HSD

Switch (Return): High-side MOSFET driver return.

High-Side Drive (Output): High-current driver output for high-side MOSFET.

This node voltage swing is between ground and V_IN+ 5V – V_{diode drop}

.

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MIC2182

Micrel

Absolute Maximum Ratings (Note 1)

Operating Ratings (Note 2)

Analog Supply Voltage (V ) .......................................+34V

Analog Supply Voltage (V ) ........................ +4.5V to +32V

IN

Digital Supply Voltage (V ) .........................................+7V

Ambient Temperature (T ) ......................... –40°C to +85°C

DD

A

Driver Supply Voltage (B ) ....................................V +7V

Junction Temperature (T ) ....................... –40°C to +125°C

ST

IN

J

Sense Voltage (V

, C ) ............................. 7V to –0.3V

Package Thermal Resistance

OUT

SH

SOP (θ ) ..........................................................100°C/W

Sync Pin Voltage (V

) ................................ 7V to –0.3V

JA

SYNC

SSOP (θ )........................................................150°C/W

JA

Enable Pin Voltage (V

) ...................................... V

IN

EN/UVLO

Power Dissipation (P )

D

SOP................................................ 400mW @ T = 85°C

A

SSOP ............................................. 270mW @ T = 85°C

A

Ambient Storage Temperature (T ) ......... –65°C to +150°C

S

ESD, Note 3

Electrical Characteristics

V_IN= 15V; SS = open; V_PWM= 0V; V_SHDN= 5V; I_LOAD= 0.1A; T_A= 25°C, bold values indicate –40°C ≤ T_A≤ +85°C; Note 4; unless

noted

Parameter

Condition

Min

Typ

Max

Units

MIC2182 [Adjustable], (Note 5)

Feedback Voltage Reference

Feedback Bias Current

Output Voltage Range

Output Voltage Line Regulation

Output Voltage Load Regulation

Output Voltage Total Regulation

MIC2182-3.3

1.233

1.220

1.208

1.245

10

1.257

1.270

1.282

V

4.5V < V_IN< 32V, 0 < V_CSH– V_OUT< 75mV

V

nA

V

1.25

6

V_IN= 4.5V to 32V, V_CSH– V_OUT= 50mV

0.03

0.5

%/V

%

25mV < (V_CSH– V_OUT) < 75mV (PWM mode only)

0mV < (V

– V

) < 75mV (full load range) 4.5V < V < 32V

0.6

%

CSH

OUT

IN

Output Voltage

3.267

3.234

3.201

3.3

3.333

3.366

3.399

V

Output Voltage

4.5V < V_IN< 32V, 0 < V_CSH– V_OUT< 75mV

V_IN= 4.5V to 32V, V_CSH– V_OUT= 50mV

3.3

V

Output Voltage Line Regulation

Output Voltage Load Regulation

Output Voltage Total Regulation

MIC2182-5.0

0.03

0.5

0.8

%/V

%

25mV < (V_CSH– V_OUT) < 75mV (PWM mode only)

0mV < (V

– V

) < 75mV (full load range) 4.5V < V < 32V

%

CSH

OUT

IN

Output Voltage

4.95

4.90

4.85

5.0

5.05

5.10

V

Output Voltage

6.5V < V_IN< 32V, 0 < V_CSH– V_OUT< 75mV

V_IN= 6.5V to 32V, V_CSH– V_OUT= 50mV

5.0

5.150

V

Output Voltage Line Regulation

Output Voltage Load Regulation

Output Voltage Total Regulation

Input and VDD Supply

PWM Mode

0.03

0.5

0.8

%/V

%

25mV < (V_CSH– V_OUT) < 75mV (PWM mode only)

0mV < (V_CSH– V_OUT) < 75mV (full load range) 6.5V < V < 32V

%

IN

V_PWM= 0V, excluding external MOSFET gate drive current

I_L= 0mA, V_PWMfloating (1nF capacitor to ground)

V_EN/UVLO= 0V

1.6

600

0.1

2.5

1500

5

mA

µA

V

Skip Mode

Shutdown Quiescent Current

Digital Supply Voltage (V_DD

Undervoltage Lockout

)

I_L= 0mA to 5mA

4.7

5.3

V_DDupper threshold (turn on threshold)

V_DDlower threshold (turn off threshold)

4.2

4.1

V

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MIC2182

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Parameter

Condition

Min

Typ

Max

Units

Reference Output (Fixed Versions Only)

Reference Voltage

1.220

1.245

1.270

V

Reference Line Regulation

Reference Load Regulation

Enable/UVLO

6V < V_IN< 32V

1

2

mV

0µA < I_REF< 100µA

Enable Input Threshold

UVLO Threshold

0.6

2.2

1.1

2.5

0.1

1.6

2.8

5

V

Enable Input Current

Soft Start

V_EN/UVLO= 5V

µA

Soft-Start Current

V_SS= 0V

–3.5

–5

100

20

–6.5

µA

Current Limit

Current-Limit Threshold Voltage

Error Amplifier

V_CSH= V_OUT

75

135

mV

Error Sense Amplifier Gain

Current Amp

Current Sense Amplifier Gain

Oscillator Section

Oscillator Frequency

Maximum Duty Cycle

Minimum On-Time

2.0

270

300

86

330

kHz

%

V_OUT= V_OUT(nominal)+ 200mV

140

1.3

0.1

250

1.9

5

ns

SYNC Threshold Level

SYNC Input Current

SYNC Minimum Pulse Width

SYNC Capture Range

Frequency Foldback Threshold

Foldback Frequency

Gate Drivers

0.7

V

V_SYNC= 5V

µA

ns

200

330

0.75

Note 6

kHz

V

measured at VOUT pin

0.95

60

1.15

kHz

Rise/Fall Time

C_L= 3000pF

60

ns

Output Driver Impedance

source

sink

5

3.5

8.5

6

Ω

Driver Nonoverlap Time

PWM Input

80

ns

PWM Input Current

V_PWM= 0V

–10

µA

Note 1. Exceeding the absolute maximum rating may damage the device.

Note 2. The device is not guaranteed to function outside its operating rating.

Note 3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.

Note 4. 25°C limits are 100% production tested. Limits over the operating temperature range are guaranteed by design and are not production tested.

Note 5. > 1.3 × V (for the feedback voltage reference and output voltage line and total regulation).

Note 6. See applications information for limitations on the maximum operating frequency.

V

IN

OUT

April 22, 2004

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Micrel

Typical Characteristics

Quiescent Current

vs. Temperature

Quiescent Current

vs. Supply Voltage

Quiescent Current

vs. Temperature

2.0

4.0

3.5

3.0

2.5

2.0

1.5

1.0

0.5

0

1.50

1.00

0.50

0

1.8

PWM

1.6

UVLO Mode

(mA)

1.4

1.2

1.0

0.8

-0.520

0.15

0.10

0.05

0

PWM

Skip

0.6

0.4

0.2

0

SHUTDOWN

(µA)

-40 -20

0 20 40 60 80 100120140

TEMPERATURE (°C)

-40 -20

0

20 40 60 80 100120140

0

4

8

12 16 20 24 28 32

TEMPERATURE (°C)

INPUT VOLTAGE (V)

V

(Fixed Versions)

V

(Fixed Versions)

REF

Load Regulation

Quiescent Current

vs. Supply Voltage

REF

Line Regulation

1.5

1.0

0.5

0

1.256

1.254

1.252

1.250

1.248

1.246

1.244

1.242

1.240

1.238

1.236

1.260

1.250

1.240

1.230

1.220

1.210

1.200

UVLO Mode

(mA)

0.5

0.4

0.3

0.2

0.1

0

-

SHUTDOWN

(µA)

0

4

8

12 16 20 24 28 32

0

4

8

12 16 20 24 28 32

200 400 600 800 1000

SUPPLY VOLTAGE (V)

LOAD CURRENT (µA)

V

(Fixed Versions)

REF

V

DD

vs. Temperature

Line Regulation

Load Regulation

1.260

1.255

1.250

1.245

1.240

5.0

4.8

4.6

4.4

4.2

4.0

5.00

4.95

4.90

4.85

4.80

-40 -20

0 20 40 60 80 100120140

0

4

8

12 16 20 24 28 32

5

10

15

20

25

TEMPERATURE (°C)

SUPPLY VOLTAGE (V)

LOAD CURRENT (mA)

V

Oscillator Frequency

vs. Supply Voltage

Oscillator Frequency

vs. Temperature

DD

vs. Temperature

10

8

1.0

0.8

0.6

0.4

0.2

0

4.98

4.96

4.94

4.92

4.90

4.88

4.86

4.84

4.82

6

4

2

0

-2

-4

-6

-8

-10

-0.2

-0.4

-0.6

-0.8

-1.0

-40 -20

0 20 40 60 80 100120140

4

8

12 16 20 24 28 32

-40 -20

0

20 40 60 80 100120140

TEMPERATURE (°C)

SUPPLY VOLTAGE (V)

TEMPERATURE (°C)

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April 22, 2004

MIC2182

Micrel

Soft-Start Current

vs. Temperature

Overcurrent Threshold

vs. Temperature

Current-Limit

Foldback

5

4

3

2

1

0

5.0

0.12

0.11

0.10

0.09

0.08

4.8

4.6

4.4

4.2

4.0

V_IN= 5V

_OUT= 3.3V

_CS= 15mΩ

V

R

0

1

2

3

4

5

6

7

8

-40 -20

0

20 40 60 80 100120140

-40 -20 0 20 40 60 80 100120140

OUTPUT CURRENT (A)

TEMPERATURE (°C)

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Block Diagrams

V_IN

C_IN

V_DD

V_IN

EN/UVLO

VDD

6

11

Reference

4.7µF

VIN

10

D2

SS

1

VBST

14

Control

Logic

HSD

C_BST

L1

16

Q2

R_CS

V_OUT

C_OUT

PWM

VSW

15

2

LSD

13

D1

Q1

PGND

12

Current

Limit

PWM Mode

to Skip

Mode

0.024V

Skip-Mode

Current

Limit

0.07V

Low

Comp

–2%V_BG

Hysteresis

Comp

Current

Sense

Amp

CSH

8

PWM

CORRECTIVE

RAMP

RESET

VOUT

9

A_V= 2

R1

R2

SYNC

Oscillator

Error

Amp

5

FB

7

COMP

3

SGND

4

C_COMP

100k

R1

G_m= 0.2×10^-3

V

= 1.245V 1+

= 6.0V

R_COMP

OUT

R2

MIC2182 [adj.]

V

OUT(max)

Figure 2a. Adjustable Output Voltage Version

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April 22, 2004

MIC2182

Micrel

V_IN

C_IN

V_DD

V_IN

EN/UVLO

VDD

6

11

Reference

4.7µF

VIN

10

D2

SS

1

VBST

14

Control

Logic

HSD

C_BST

L1

16

Q2

R_CS

V_OUT

C_OUT

PWM

VSW

15

2

LSD

13

D1

Q1

PGND

12

Current

Limit

PWM Mode

to Skip

Mode

0.024V

Skip-Mode

Current

Limit

0.07V

Low

Comp

–2%V_BG

Hysteresis

Comp

Current

Sense

Amp

CSH

8

PWM

CORRECTIVE

RAMP

RESET

VOUT

9

A_V= 2

*82.5k for 3.3V Output

150k for 5V Output

SYNC

R1*

Oscillator

Error

Amp

5

SGND

4

COMP

R2

50k

3

VREF

7

C_COMP

100k

G_m= 0.2×10^-3

R_COMP

MIC2182-x.x

Figure 2b. Fixed Output Voltage Versions

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MIC2182

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Control Loop

Functional Description

See “Applications Information” following this section for com-

ponent selection information and Figure 14 and Tables 1

through 5 for predesigned circuits.

PWM and Skip Modes of Operation

The MIC2182 operates in PWM (pulse-width-modulation)

mode at heavier output load conditions. At lighter load condi-

tions, the controller can be configured to automatically switch

to a pulse-skipping mode to improve efficiency. The potential

disadvantage of skip mode is the variable switching fre-

quency that accompanies this mode of operation. The occur-

rence of switching pulses depends on component values as

well as line and load conditions. There is an external sync

function that is disabled in skip mode. In PWM mode, the

synchronous buck converter forces continuous current to

flowintheinductor.Inskipmode,currentthroughtheinductor

can settle to zero, causing voltage ringing across the induc-

tor. Pulling the PWM pin (pin 2) low will force the controller to

operate in PWM mode for all load conditions, which will

improve cross regulation of transformer-coupled, multiple

output configurations.

The MIC2182 is a BiCMOS, switched-mode, synchronous

step-down (buck) converter controller. Current-mode control

is used to achieve superior transient line and load regulation.

An internal corrective ramp provides slope compensation for

stable operation above a 50% duty cycle. The controller is

optimized for high-efficiency, high-performance dc-dc con-

verter applications.

The MIC2182 block diagrams are shown in Figure 2a and

Figure 2b.

The MIC2182 controller is divided into 6 functions.

• Control loop

- PWM operation

- Skip-mode operation

PWM Control Loop

• Current limit

The MIC2182 uses current-mode control to regulate the

output voltage. This method senses the output voltage (outer

loop) and the inductor current (inner loop). It uses inductor

current and output voltage to determine the duty cycle of the

buck converter. Sampling the inductor current removes the

inductor from the control loop, which simplifies compensa-

tion.

• Reference, enable, and UVLO

• MOSFET gate drive

• Oscillator and sync

• Soft start

V_IN

C_IN

V_DD

V_IN

VDD

11

Reference

4.7µF

VIN

10

D2

VBST

14

CONTROL LOGIC AND

PULSE-WIDTH MODULATOR

HSD

C_BST

16

Q2

Q1

L1

R_CS

V_OUT

C_OUT

VSW

15

LSD

13

D1

PWM Mode

to Skip

Mode

PGND

12

Q

LOW

FORCES

SKIP MODE

0.024V

R

S

Current

Sense

Amp

CSH

8

PWM

COMPARATOR

VOUT

9

CORRECTIVE

RAMP

RESET

A_V= 2

R1

R2

Oscillator

Error

Amp

FB

7

COMP

3

R1

C_COMP

R_COMP

100k

V

= 1.245V 1+

OUT

G_m= 0.2×10^-3

R2

MIC2182 [adj.] PWM Mode

Figure 3. PWM Operation

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MIC2182

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Skip-Mode Control Loop

A block diagram of the MIC2182 PWM current-mode control

loop is shown in Figure 3 and the PWM mode voltage and

current waveforms are shown in figure 5A. The inductor

current is sensed by measuring the voltage across the

This control method is used to improve efficiency at light

output loads. At light output currents, the power drawn by the

MIC2182 is equal to the input voltage times the IC supply

resistor, R . A ramp is added to the amplified current-sense

current (I ). At light output currents, the power dissipated by

CS

Q

signal to provide slope compensation, which is required to

prevent unstable operation at duty cycles greater than 50%.

the IC can be a significant portion of the total output power,

whichlowerstheefficiencyofthepowersupply.TheMIC2182

draws less supply current in skip mode by disabling portions

of the control and drive circuitry when the IC is not switching.

The disadvantage of this method is greater output voltage

ripple and variable switching frequency.

A transconductance amplifier is used for the error amplifier,

which compares an attenuated sample of the output voltage

with a reference voltage. The output of the error amplifier is

the COMP (compensation) pin, which is compared to the

current-sensewaveforminthePWMblock. Whenthecurrent

signal becomes greater than the error signal, the comparator

turns off the high-side drive. The COMP pin (pin 3) provides

access to the output of the error amplifier and allows the use

of external components to stabilize the voltage loop.

AblockdiagramoftheMIC2182skipmodeisshowninFigure

4. Skip mode voltage and current waveforms are shown in

figure 5B.

V_IN

C_IN

V_DD

VDD

11

Reference

V_IN

4.7µF

VIN

10

D2

VBST

14

CONTROL LOGIC AND

SKIP-MODE LOGIC

HSD

C_BST

16

Q2

Q1

L1

R_CS

V_OUT

C_OUT

VSW

15

LOW-SIDE DRIVER

ONE SHOT

LSD

13

PGND

12

Q

R

S

Skip-Mode

Current

Limit

0.07V

Low

Comp

ONE SHOT

–2%V_BG

LOW

FORCES

PWM MODE

Hysteresis

Comp

±1%

Current

Sense

Amp

CSH

8

VOUT

9

A_V= 2

R1

R2

FB

7

R1

MIC2182 [adj.] Skip Mode

V

= 1.245V 1+

OUT

R2

Figure 4. Skip-Mode Operation

April 22, 2004

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M9999-042204

MIC2182

Micrel

V_IN

0V

V_SW

I_LOAD

I_L1

0A

V_DD

Reset

Pulse

0V

V_IN+ V_DD

V_HSD

0V

V_DD

0V

V_LSD

Figure 5a. PWM-Mode Timing

V_DD

V_HSD

0V

V_DD

V_LSD

0V

V_IN

V_OUT

V_SW

0V

I_LIM(skip)

I_L1

0A

V_DD

V_one-shot

0V

+1%

V_NOMINAL

V_OUT

–1%

0V

I_OUT

0A

Figure 5b. Skip-Mode Timing

A hysteretic comparator is used in place of the PWM error

amplifier and a current-limit comparator senses the inductor

current. A one-shot starts the switching cycle by momentarily

turning on the low side MOSFET to insure the high-side drive

boost capacitor, Cbst, is fully charged. The high-side MOS-

FET is turned on and current ramps up in the inductor, L1.

The high-side drive is turned off when either the peak voltage

on the input of the current-sense comparator exceeds the

threshold, typically 35mV, or the output voltage rises above

the hysteretic threshold of the output voltage comparator.

Once the high-side MOSFET is turned off, the load current

Figure 6 shows the improvement in efficiency that skip mode

makes when at lower output currents.

100

PWM

80

Skip

60

40

20

0

0.01

0.1

1

10

100

discharges the output capacitor, causing V

to fall. The

OUT

OUTPUT CURRENT (A)

cycle repeats when V

1%.

falls below the lower threshold, –

OUT

Figure 6. Efficiency

The maximum peak inductor current depends on the skip-

mode current-limit threshold and the value of the current-

sense resistor, R

.

CS

35mV

I

=

inductor(peak)

R

sense

M9999-042204

12

April 22, 2004

MIC2182

Micrel

Switching from PWM to Skip Mode

rent-limit threshold is 100mV+35mV –25mV. The current-

sense resistor must be sized using the minimum current-limit

threshold. The external components must be designed to

withstand the maximum current limit. The current-sense

resistor value is calculated by the equation below:

The current sense amplifier in Figure 3 monitors the average

voltage across the current-sense resistor. The controller will

switch from PWM to skip mode when the average voltage

across the current-sense resistor drops below approximately

12mV.ThisisshowninFigure7b.Theaverageoutputcurrent

at this transition level for is calculated below.

75mV

R_CS

=

I_OUT(max)

0.012

I

=

The maximum output current is:

135mV

OUT(skipmode)

R

CS

I_OUT(max)

=

where:

R_CS

0.012 = threshold voltage of the internal comparator

= current-sense resistor value

The current-sense pins CSH (pin 8) and V

(pin 9) are

OUT

R

CS

noise sensitive due to the low signal level and high input

impedance.ThePCBtracesshouldbeshortandroutedclose

toeachother.Asmall(1nFto0.1µF)capacitoracrossthepins

will attenuate high frequency switching noise.

Switching from Skip to PWM Mode

The frequency of occurrence of the skip-mode current pulses

increase as the output current increases until the hysteretic

dutycyclereaches100%(continuouspulses). Increasingthe

current past this point will cause the output voltage will drop.

The low limit comparator senses the output voltage when it

drops below 2% of the set output and automatically switches

the converter to PWM mode.

When the peak inductor current exceeds the current-limit

threshold, the current-limit comparator, in Figure 2, turns off

the high-side MOSFET for the remainder of the cycle. The

output voltage drops as additional load current is pulled from

the converter. When the output voltage reaches approxi-

mately 0.95V, the circuit enters frequency-foldback mode

andtheoscillatorfrequencywilldropto60kHzwhilemaintain-

ing the peak inductor current equal to the nominal 100mV

across the external current-sense resistor. This limits the

maximum output power delivered to the load under a short

circuit condition.

The inductor current in skip mode is a triangular wave shape

a minimum value of 0 and a maximum value of 35mV/R

(seeFigure7b). Themaximumaverageoutputcurrentinskip

mode is the average value of the inductor waveform:

CS

35mV

I

= 0.5 ×

OUT(maxskipmode)

R

CS

Reference, Enable, and UVLO Circuits

The capacitor on the PWM pin (pin 2) is discharged when the

IC transitions from skip to PWM mode. This forces the IC to

remain in PWM mode for a fixed period of time. The added

delay prevents unwanted switching between PWM and skip

mode. The capacitor is charged with a 10uA current source

onpin2. Thethresholdonpin2is2.5V. Thedelayforatypical

1nF capacitor is:

The output drivers are enabled when the following conditions

are satisfied:

• The V voltage (pin 11) is greater than its

DD

undervoltage threshold (typically 4.2V).

• The voltage on the enable pin is greater than the

enable UVLO threshold (typically 2.5V)

The internal bias circuit generates a 1.245V bandgap refer-

C

× V

threshold

1nF × 2.5V

10µA

PWM

ence voltage for the voltage error amplifier and a 5V V

t

=

= 250µs

DD

delay

I

source

voltage for the gate drive circuit. The reference voltage in the

fixed-output-voltage versions of the MIC2182 is buffered and

where:

brought to pin 7. The V

pin should be bypassed to GND

REF

C

= capacitor connected to pin 2

PWM

(pin 4) with a 0.1µF capacitor. The adjustable version of the

MIC2182 uses pin 7 for output voltage sensing. A decoupling

capacitor on pin 7 is not used in the adjustable output voltage

version.

Current Limit

The current-limit circuit operates during PWM mode. The

output current is detected by the voltage drop across the

external current-sense resistor (R in Figure 2.). The cur-

CS

35mV THRESHOLD

ACROSS R_CS

.

I_LIM(skip)

Inductor

Current

0A

Figure 7a. Maximum Skip-Mode-Load Inductor Current

I_MIN(PWM)

0A

12mV THRESHOLD

OF AVERAGE VOLTAGE

Inductor

Current

ACROSS R_CS

.

Figure 7b. Minimum PWM-Mode-Load Inductor Current for PWM Operation

April 22, 2004

13

M9999-042204

MIC2182

Micrel

The enable pin (pin 6) has two threshold levels, allowing the

MIC2182toshutdowninalowcurrentmode,orturnoffoutput

switching in UVLO mode. An enable pin voltage lower than

the shutdown threshold turns off all the internal circuitry and

reduces the input current to typically 0.1µA.

Oscillator and Sync

Theinternaloscillatorisfreerunningandrequiresnoexternal

components. The nominal oscillator frequency is 300kHz. If

the output voltage is below approximately 0.95V, the oscilla-

tor operates in a frequency-foldback mode and the switching

frequency is reduced to 60kHz.

If the enable pin voltage is between the shutdown and UVLO

thresholds, theinternalbias, V , andreferencevoltagesare

The SYNC input (pin 5) allows the MIC2182 to synchronize

with an external clock signal. The rising edge of the sync

signal generates a reset signal in the oscillator, which turns

off the low-side gate drive output. The high-side drive then

turns on, restarting the switching cycle. The sync signal is

inhibited when the controller operates in skip mode or during

frequency foldback. The sync signal frequency must be

greater than the maximum specified free running frequency

of the MIC2182. If the synchronizing frequency is lower,

double pulsing of the gate drive outputs will occur. When not

used, the sync pin must be connected to ground.

DD

turned on. The soft-start pin is forced low by an internal

discharge MOSFET. The output drivers are inhibited from

switching and remain in a low state. Raising the enable

voltage above the UVLO threshold of 2.5V allows the soft-

start capacitor to charge and enables the output drivers.

Either of two UVLO conditions will pull the soft-start capacitor

low.

• When the V drops below 4.1V

DD

• When the enable pin drops below the 2.5V

threshold

Figure 8 shows the timing between the external sync signal

(trace 2), the low-side drive (trace 1) and the high-side drive

(trace R1). There is a delay of approximately 250ns between

the rising edge of the external sync signal and turnoff of the

low-side MOSFET gate drive.

MOSFET Gate Drive

The MIC2182 high-side drive circuit is designed to switch an

N-channelMOSFET.ReferringtotheblockdiagraminFigure

2, a bootstrap circuit, consisting of D2 and C

energy to the high-side drive circuit. Capacitor C

charged while the low-side MOSFET is on and the voltage on

, supplies

BST

is

BST

Some concerns of operating at higher frequencies are:

• Higher power dissipation in the internal V

the V pin (pin 15) is approximately 0V. When the high-side

DD

SW

regulator. This occurs because the MOSFET

gates require charge to turn on the device. The

average current required by the MOSFET gate

increases with switching frequency. This in-

creases the power dissipated by the internal

MOSFETdriveristurnedon, energyfromC

isusedtoturn

BST

theMOSFETon. AstheMOSFETturnson, thevoltageonthe

pin increases to approximately V . Diode D2 is re-

V

SW

IN

versed biased and C

floats high while continuing to keep

BST

the high-side MOSFET on. When the low-side switch is

turned back on, C is recharged through D2.

V

regulator. Figure 10 shows the total gate

DD

BST

charge which can be driven by the MIC2182

over the input voltage range, for different values

of switching frequency. The total gate charge

includes both the high- and low-side MOSFETs.

The larger SOP package is capable of dissipat-

ing more power than the SSOP package and

can drive larger MOSFETs with higher gate

drive requirements.

The drive voltage is derived from the internal 5V V

bias

DD

supply. The nominal low-side gate drive voltage is 5V and the

nominal high-side gate drive voltage is approximately 4.5V

due the voltage drop across D2. A fixed 80ns delay between

the high- and low-side driver transitions is used to prevent

currentfromsimultaneouslyflowingunimpededthroughboth

MOSFETs.

TIME

Figure 8. Sync Waveforms

Figure 9. Startup Waveforms

M9999-042204

14

April 22, 2004

MIC2182

• Reduced maximum duty cycle due to switching

Micrel

Thesoft-startvoltageisapplieddirectlytothePWMcompara-

tor. A 5uA internal current source is used to charge up the

soft-start capacitor. The capacitor is discharged when either

the enable voltage drops below the UVLO threshold (2.5V) or

transition times and constant delay times in the

controller. As the switching frequency increased,

the switching period decreases. The switching

transition times and constant delays in the

MIC2182 start to become noticeable. The effect

is to reduce the maximum duty cycle of the

controller. This will cause the minimum input to

output differential voltage (dropout voltage) to

increase.

the V voltage drops below the UVLO level (4.1V).

DD

Thepartswitchesataminimumdutycyclewhenthesoft-start

pin voltage is less than 0.4V. This maintains a charge on the

bootstrap capacitor and insures high-side gate drive voltage.

As the soft-start voltage rises above 0.4V, the duty cycle

increases from the minimum duty cycle to the operating duty

cycle. The oscillator runs at the foldback frequency of 60kHz

until the output voltage rises above 0.95V. Above 0.95V, the

switching frequency increases to 300kHz (or the sync’d

frequency), causing the output voltage to rise a greater rate.

The rise time of the output is dependent on the soft-start

capacitor, output capacitance, output voltage, and load cur-

rent. The oscilloscope photo in Figure 9 show the output

voltage and the soft-start pin voltage at startup.

100

SOP

80

60

40

20

0

300kHz

400kHz

500kHz

Minimum Pulse Width

0

4

8

12 16 20 24 28 32

SUPPLY VOLTAGE (V)

The MIC2182 has a specified minimum pulse width. This

minimum pulse width places a lower limit on the minimum

duty cycle of the buck converter. When the MIC2182 is

operating in forced PWM mode (pin 2 low) and when the

output current is very low or zero, there is a limit on the ratio

Figure 10a. SOP Gate Charge vs. Input Voltage

100

SSOP

80

60

of V

/V . If this limit is exceeded, the output voltage will

OUT IN

rise above the regulated voltage level. A minimum load is

required to prevent the output from rising up. This will not

occur for output voltages greater than 3V.

300kHz

40

400kHz

20

Figure 11 should be used as a guide when the MIC2182 is

forced into PWM-only mode. The actual maximum input

voltage will depend on the exact external components used

(MOSFETs, inductors, etc.).

500kHz

0

4

8

12 16 20 24 28 32

SUPPLY VOLTAGE (V)

35

30

25

20

15

10

Figure 10b. SSOP Gate Charge vs. Input Voltage

It is recommended that the user limits the maximum synchro-

nized frequency to 600kHz. If a higher synchronized fre-

quency is required, it may be possible and will be design

dependent. Please consult Micrel applications for assis-

tance.

Soft Start

Soft start reduces the power supply input surge current at

startup by controlling the output voltage rise time. The input

surge appears while the output capacitance is charged up. A

slower output rise time will draw a lower input surge current.

Soft start may also be used for power supply sequencing.

0

1

2

3

4

5

6

OUTPUT VOLTAGE (V)

Figure 11. Max. Input Voltage in Forced-PWM Mode

This restriction does not occur when the MIC2182 is set to

automatic mode (pin 2 connected to a capacitor) since the

converter operates in skip mode at low output current.

April 22, 2004

15

M9999-042204

MIC2182

Micrel

output currents, the core losses can be a significant contribu-

tor. Core loss information is usually available from the mag-

netics vendor.

Applications Information

The following applications information includes component

selection and design guidelines. See Figure 14 and Tables 1

through 5 for predesigned circuits.

Copper loss in the inductor is calculated by the equation

below:

Inductor Selection

P

= I_inductor(rms)²×R_winding

Values for inductance, peak, and RMS currents are required

to select the output inductor. The input and output voltages

andtheinductancevaluedeterminethepeaktopeakinductor

ripple current. Generally, higher inductance values are used

withhigherinputvoltages.Largerpeaktopeakripplecurrents

will increase the power dissipation in the inductor and

MOSFETs. Larger output ripple currents will also require

more output capacitance to smooth out the larger ripple

current. Smaller peak to peak ripple currents require a larger

inductance value and therefore a larger and more expensive

inductor. A good compromise between size, loss and cost is

to set the inductor ripple current to be equal to 20% of the

maximum output current.

inductorCu

The resistance of the copper wire, R

, increases with

winding

temperature.Thevalueofthewindingresistanceusedshould

be at the operating temperature.

R

= R

× 1+ 0.0042 ×(T − T

)

(

winding(hot)

winding(20°C)

hot

20°C

where:

T

= temperature of the wire

HOT

under operating load

T

= ambient temperature

20°C

R

is room temperature winding resistance

winding(20°C)

The inductance value is calculated by the equation below.

(usually specified by the manufacturer)

Current-Sense Resistor Selection

V

×(V

− V

)

OUT

IN(max)

OUT

L =

Low inductance power resistors, such as metal film resistors

should be used. Most resistor manufacturers make low

inductance resistors with low temperature coefficients, de-

signedspecificallyforcurrent-senseapplications. Bothresis-

tance and power dissipation must be calculated before the

V

× f × 0.2 ×I

S OUT(max)

IN(max)

where:

f = switching frequency

S

0.2 = ratio of ac ripple current to dc output current

= maximum input voltage

resistor is selected. The value of R

is chosen based on

SENSE

V

IN(max)

the maximum output current and the maximum threshold

level. The power dissipated is based on the maximum peak

output current at the minimum overcurrent threshold limit.

The peak-to-peak inductor current (ac ripple current) is:

V_OUT×(V_IN(max)− V_OUT

)

I_PP

=

75mV

V

_IN(max)× f_S×L

R

=

SENSE

I

OUT(max)

The peak inductor current is equal to the average output

current plus one half of the peak to peak inductor ripple

current.

The maximum overcurrent threshold is:

135mV

I_{overcurrent(max)}

=

I_PK= I_OUT(max)+ 0.5 ×I_PP

R_CS

2

The RMS inductor current is used to calculate the I ·R losses

The maximum power dissipated in the sense resistor is:

in the inductor.

2

P

= I

×R

CS

D(R_SENSE

)

overcurrent(max)

2

I_PP

1

3 I_OUT(max)

MOSFET Selection

I_inductor(rms) = I_OUT(max)× 1+

External N-channel logic-level power MOSFETs must be

used for the high- and low-side switches. The MOSFET gate-

to-source drive voltage of the MIC2182 is regulated by an

Maximizing efficiency requires the proper selection of core

material and minimizing the winding resistance. The high

frequencyoperationoftheMIC2182requirestheuseofferrite

materials for all but the most cost sensitive applications.

Lower cost iron powder cores may be used but the increase

incorelosswillreducetheefficiencyofthepowersupply.This

is especially noticeable at low output power. The winding

resistance decreases efficiency at the higher output current

levels. The winding resistance must be minimized although

this usually comes at the expense of a larger inductor.

internal 5V V

operation is specified at V = 4.5V must be used.

regulator. Logic-level MOSFETs, whose

DD

GS

It is important to note the on-resistance of a MOSFET

increases with increasing temperature. A 75°C rise in junc-

tion temperature will increase the channel resistance of the

MOSFET by 50% to 75% of the resistance specified at 25°C.

This change in resistance must be accounted for when

calculating MOSFET power dissipation.

Total gate charge is the charge required to turn the MOSFET

on and off under specified operating conditions (V

Thepowerdissipatedintheinductorisequaltothesumofthe

core and copper losses. At higher output loads, the core

losses are usually insignificant and can be ignored. At lower

and

DS

V

). The gate charge is supplied by the MIC2182 gate drive

GS

circuit. At 300kHz switching frequency and above, the gate

M9999-042204

16 April 22, 2004

MIC2182

Micrel

charge can be a significant source of power dissipation in the

MIC2182. At low output load this power dissipation is notice-

able as a reduction in efficiency. The average current re-

quired to drive the high-side MOSFET is:

C

× V + C

× V

OSS

IN

ISS

GS

t

=

T

I

G

where:

C

and C

are measured at V = 0.

OSS DS

I_{G[high-side](avg)}= Q_G× f_S

ISS

I = gate drive current (1A for the MIC2182)

G

where:

The total high-side MOSFET switching loss is:

I

=

G[high-side](avg)

average high-side MOSFET gate current

P

= (V +V )×I × t × f

AC S

IN D PK T

Q = total gate charge for the high-side MOSFET

where:

G

taken from manufacturer’s data sheet

t = switching transition time

T

with V = 5V.

GS

(typically 20ns to 50ns)

The low-side MOSFET is turned on and off at V

= 0

DS

V = freewheeling diode drop, typically 0.5V.

D

because the freewheeling diode is conducting during this

time. The switching losses for the low-side MOSFET is

usually negligable. Also, the gate drive current for the low-

f it the switching frequency, nominally 300kHz

S

The low-side MOSFET switching losses are negligible and

can be ignored for these calculations.

side MOSFET is more accurately calculated using C

at

ISS

V

= 0 instead of gate charge.

RMS Current and MOSFET Power Dissipation Calculation

DS

For the low-side MOSFET:

= C

Under normal operation, the high-side MOSFET’s RMS

currentisgreatestwhenV islow(maximumdutycycle).The

IN

I

× V × f

GS S

G[low-side](avg)

ISS

low-sideMOSFET’sRMScurrentisgreatestwhenV ishigh

IN

Since the current from the gate drive comes from the input

voltage, the power dissipated in the MIC2182 due to gate

drive is:

(minimum duty cycle). However, the maximum stress the

MOSFETs see occurs during short circuit conditions, where

the output current is equal to I

. (See the Sense

overcurrent(max)

Resistor section). The calculations below are for normal

operation. To calculate the stress under short circuit condi-

P

= V

I

+I

G[low-side](avg)

(

)

gatedrive

IN G[high-side](avg)

tions, substituteI

forI

. Usetheformula

overcurrent(max)

OUT(max)

AconvenientfigureofmeritforswitchingMOSFETsistheon-

resistance times the total gate charge (R

below to calculate D under short circuit conditions.

× Q ). Lower

DS(on)

G

D_shortcircuit= 0.063 −1.8 ×10⁻³× V

numbers translate into higher efficiency. Low gate-charge

logic-level MOSFETs are a good choice for use with the

MIC2182. Power dissipation in the MIC2182 package limits

the maximum gate drive current. Refer to Figure 10 for the

MIC2182 gate drive limits.

IN

The RMS value of the high-side switch current is:

2

I_PP

2

I_SW(highside)(rms) = D × I_OUT(max)

+

12

Parameters that are important to MOSFET switch selection

are:

2

• Voltage rating

• On-resistance

• Total gate charge

I

2

PP

I

(rms) = 1−D I

+

(

)

SW(lowside)

OUT(max)

12

where:

The voltage rating of the MOSFETs are essentially equal to

the input voltage. A safety factor of 20% should be added to

D = duty cycle of the converter

the V

of the MOSFETs to account for voltage spikes

DS(max)

V

OUT

due to circuit parasitics.

D =

η × V

IN

The power dissipated in the switching transistor is the sum of

η = efficiency of the converter.

theconductionlossesduringtheon-time(P

)andthe

conduction

switchinglossesthatoccurduringtheperiodoftimewhenthe

Converter efficiency depends on component parameters,

which have not yet been selected. For design purposes, an

MOSFETs turn on and off (P ).

AC

efficiency of 90% can be used for V less than 10V and 85%

P

= P

+P

IN

SW

conduction

AC

can be used for V greater than 10V. The efficiency can be

IN

where:

P_conduction= I_SW(rms)²×R_SW

= P +P

moreaccuratelycalculatedoncethedesigniscomplete.Ifthe

assumed efficiency is grossly inaccurate, a second iteration

through the design procedure can be made.

P

For the high-side switch, the maximum dc power dissipation

is:

AC

AC(off)

AC(on)

R

= on-resistance of the MOSFET switch.

SW

2

Making the assumption the turn-on and turnoff transition

times are equal, the transition time can be approximated by:

P

= R

×I

(rms)

switch1(dc)

DS(on)1 SW1

April 22, 2004

17

M9999-042204

MIC2182

Micrel

For the low-side switch (N-channel MOSFET), the dc power

dissipation is:

circuit inductance will cause ringing during the high-side

MOSFET turn-on.

An external Schottky diode conducts at a lower forward

voltage preventing the body diode in the MOSFET from

turning on. The lower forward voltage drop dissipates less

power than the body diode. The lack of a reverse recovery

mechanism in a Schottky diode causes less ringing and less

power loss. Depending on the circuit components and oper-

ating conditions, an external Schottky diode will give a 1/2%

to 1% improvement in efficiency. Figure 12 illustrates the

difference in noise on the VSW pin with and without a

Schottky diode.

2

P

= R

×I

(rms)

switch2(dc)

DS(on)2

SW2

Since the ac switching losses for the low side MOSFET is

near zero, the total power dissipation is:

P

= P

switch2(dc)

low-side MOSFET(max)

The total power dissipation for the high-side MOSFET is:

= P +P

P

highsideMOSFET(max)

SWITCH 1(dc)

AC

External Schottky Diode

Output Capacitor Selection

An external freewheeling diode is used to keep the inductor

current flow continuous while both MOSFETs are turned off.

This dead time prevents current from flowing unimpeded

through both MOSFETs and is typically 80ns The diode

conducts twice during each switching cycle. Although the

averagecurrentthroughthisdiodeissmall, thediodemustbe

able to handle the peak current.

The output capacitor values are usually determined by the

capacitorsESR(equivalentseriesresistance).Voltagerating

and RMS current capability are two other important factors in

selectingtheoutputcapacitor.Recommendedcapacitorsare

tantalum, low-ESR aluminum electrolytics, and OS-CON.

The output capacitor’s ESR is usually the main cause of

output ripple. The maximum value of ESR is calculated by:

I_D(avg)= I_OUT× 2 × 80ns × f_S

∆V

OUT

The reverse voltage requirement of the diode is:

R

≤

ESR

I

PP

V

(rrm) = V

IN

diode

where:

The power dissipated by the Schottky diode is:

V

I

= peak to peak output voltage ripple

OUT

P_diode= I_D(avg)× V_F

= peak to peak inductor ripple current

PP

where:

The total output ripple is a combination of the ESR and the

output capacitance. The total ripple is calculated below:

V = forward voltage at the peak diode current

F

The external Schottky diode, D2, is not necessary for circuit

operation since the low-side MOSFET contains a parasitic

body diode. The external diode will improve efficiency and

decrease high frequency noise. If the MOSFET body diode is

used,itmustberatedtohandlethepeakandaveragecurrent.

The body diode has a relatively slow reverse recovery time

and a relatively high forward voltage drop. The power lost in

the diode is proportional to the forward voltage drop of the

diode. As the high-side MOSFET starts to turn on, the body

diodebecomesashortcircuitforthereverserecoveryperiod,

dissipating additional power. The diode recovery and the

2

I

_PP×(1−D)

2

∆V_OUT

=

+ I_PP×R_ESR

(

)

C_OUT× f_S

where:

D = duty cycle

= output capacitance value

C

OUT

f = switching frequency

S

The voltage rating of capacitor should be twice the output

voltage for a tantalum and 20% greater for an aluminum

electrolytic or OS-CON.

The output capacitor RMS current is calculated below:

I

PP

I

(rms) =

C_OUT

12

The power dissipated in the output capacitor is:

2

P

= I

(rms) ×R

DISS(C_OUT

)

C_OUT

ESR(C_OUT

)

Input Capacitor Selection

The input capacitor should be selected for ripple current

rating and voltage rating. Tantalum input capacitors may fail

whensubjectedtohighinrushcurrents,causedbyturningthe

input supply on. Tantalum input capacitor voltage rating

should be at least 2 times the maximum input voltage to

maximize reliability. Aluminum electrolytic, OS-CON, and

multilayer polymer film capacitors can handle the higher

inrush currents without voltage derating.

TIME

Figure 12. Switch Output Noise

With and Without Shottky Diode

M9999-042204

18

April 22, 2004

MIC2182

Micrel

The input voltage ripple will primarily depend on the input

capacitors ESR. The peak input current is equal to the peak

inductor current, so:

• Supply current to the MIC2182

• MOSFET gate-charge power (included in the IC

supply current)

∆V = I

×R

ESR(C_IN

• Core losses in the output inductor

IN

inductor(peak)

)

To maximize efficiency at light loads:

The input capacitor must be rated for the input current ripple.

TheRMSvalueofinputcapacitorcurrentisdeterminedatthe

maximum output current. Assuming the peak to peak induc-

tor ripple current is low:

• Use a low gate-charge MOSFET or use the

smallest MOSFET, which is still adequate for

maximum output current.

• Allow the MIC2182 to run in skip mode at lower

I_C(rms) ≈ I_OUT(max)

×

D ×(1−D)

currents.

IN

• Use a ferrite material for the inductor core, which

has less core loss than an MPP or iron power

core.

The power dissipated in the input capacitor is:

2

P

= I (rms) ×R

DISS(C_IN

)

C_IN

ESR(C_IN

)

Under heavy output loads the significant contributors to

power loss are (in approximate order of magnitude):

Voltage Setting Components

The MIC2182-3.3 and MIC2182-5.0 ICs contain internal

voltage dividers that set the output voltage. The MIC2182

adjustable version requires two resistors to set the output

voltage as shown in Figure 13.

• Resistive on-time losses in the MOSFETs

• Switching transition losses in the MOSFETs

• Inductor resistive losses

• Current-sense resistor losses

R1

• Input capacitor resistive losses (due to the

Error

Amp

FB

7

capacitors ESR)

R2

To minimize power loss under heavy loads:

• Use logic-level, low on-resistance MOSFETs.

Multiplying the gate charge by the on-resistance

gives a Figure of merit, providing a good bal-

ance between low and high load efficiency.

V_REF

1.245V

MIC2182 [adj.]

Figure 13. Voltage-Divider Configuration

• Slow transition times and oscillations on the

voltage and current waveforms dissipate more

power during turn-on and turnoff of the

MOSFETs. A clean layout will minimize parasitic

inductance and capacitance in the gate drive

and high current paths. This will allow the fastest

transition times and waveforms without oscilla-

tions. Low gate-charge MOSFETs will transition

faster than those with higher gate-charge

requirements.

The output voltage is determined by the equation:

R1

V_O= V_REF× 1+

R2

Where: V

for the MIC2182 is typically 1.245V.

REF

A typical value of R1 can be between 3k and 10k. If R1 is too

large it may allow noise to be introduced into the voltage

feedback loop. If R1 is too small in value it will decrease the

efficiency of the power supply, especially at low output loads.

• For the same size inductor, a lower value will

have fewer turns and therefore, lower winding

resistance. However, using too small of a value

will require more output capacitors to filter the

output ripple, which will force a smaller band-

width, slower transient response and possible

instability under certain conditions.

Once R1 is selected, R2 can be calculated using:

V

×R1

REF

R2 =

V − V

O

REF

Voltage Divider Power Dissipation

The reference voltage and R2 set the current through the

voltage divider.

• Lowering the current-sense resistor value will

decrease the power dissipated in the resistor.

However, it will also increase the overcurrent

limit and will require larger MOSFETs and

inductor components.

V_REF

I_divider

=

R2

The power dissipated by the divider resistors is:

• Use low-ESR input capacitors to minimize the

2

P_divider= (R1+R2)×I_divider

power dissipated in the capacitors ESR.

Efficiency Calculation and Considerations

Decoupling Capacitor Selection

Efficiency is the ratio of output power to input power. The

difference is dissipated as heat in the buck converter. Under

light output load, the significant contributors are:

The 4.7µF decoupling capacitor is used to minimize noise on

the VDD pin. The placement of this capacitor is critical to the

proper operation of the IC. It must be placed right next to the

April 22, 2004

19

M9999-042204

MIC2182

Micrel

• When the high-side MOSFET is switched on, the

pins and routed with a wide trace. The capacitor should be a

good quality tantalum. An additional 1µF ceramic capacitor

may be necessary when driving large MOSFETs with high

gatecapacitance. IncorrectplacementoftheV decoupling

capacitor will cause jitter or oscillations in the switching

waveform and large variations in the overcurrent limit.

critical flow of current is from the input capacitor

through the MOSFET, inductor, sense resistor,

output capacitor, and back to the input capacitor.

These paths must be made with short, wide

pieces of trace. It is good practice to locate the

ground terminals of the input and output capaci-

tors close to each.

DD

A 0.1µF ceramic capacitor is required to decouple the VIN.

The capacitor should be placed near the IC and connected

directly to between pin 10 (Vcc) and pin 12 (PGND).

• When the low-side MOSFET is switched on,

current flows through the inductor, sense

resistor, output capacitor, and MOSFET. The

source of the low-side MOSFET should be

located close to the output capacitor.

PCB Layout and Checklist

PCB layout is critical to achieve reliable, stable and efficient

performance. A ground plane is required to control EMI and

minimize the inductance in power, signal and return paths.

• The freewheeling diode, D1 in Figure 2, con-

ducts current during the dead time, when both

MOSFETs are off. The anode of the diode

should be located close to the output capacitor

ground terminal and the cathode should be

located close to the input side of the inductor.

The following guidelines should be followed to insure proper

operation of the circuit.

• Signal and power grounds should be kept

separate and connected at only one location.

Large currents or high di/dt signals that occur

when the MOSFETs turn on and off must be

kept away from the small signal connections.

• The 4.7µF capacitor, which connects to the VDD

terminal (pin 11) must be located right at the IC.

The VDD terminal is very noise sensitive and

placement of this capacitor is very critical.

Connections must be made with wide trace. The

capacitor may be located on the bottom layer of

the board and connected to the IC with multiple

vias.

• The connection between the current-sense

resistor and the MIC2182 current-sense inputs

(pin 8 and 9) should have separate traces,

routed from the terminals directly to the IC pins.

The traces should be routed as closely as

possible to each other and their length should be

minimized. Avoid running the traces under the

inductor and other switching components. A 1nF

to 0.1µF capacitor placed between pins 8 and 9

will help attenuate switching noise on the current

sense traces. This capacitor should be placed

close to pins 8 and 9.

• The V bypass capacitor should be located

IN

close to the IC and connected between pins 10

and 12. Connections should be made with a

ground and power plane or with short, wide

trace.

M9999-042204

20

April 22, 2004

MIC2182

Micrel

Predesigned Circuits

Power supplies larger than 10A can also be constructed

using the MIC2182 using larger power-handling compo-

nents.

Asingleschematicdiagram,showninFigure 14,canbeused

tobuildpowersuppliesrangingfrom3Ato10Aatthecommon

outputvoltagesof1.8V,2.5V,3.3V,and5V.Componentsthat

vary, depending upon output current and voltage, are listed

in the accompanying Tables 3 through 6.

The“PowerSupplyOperatingCharacteristics”graphsfollow-

ing the component and vendor tables provide useful informa-

tion about the actual performance of some of these circuits.

D2

MIC2182

VDD

BST

EN/UVLO HSD

C11

V_IN

SD103BWS

(table)

VIN

C9

4.7µF

16V

C5

0.1µF

R7

100k

C6

0.1µF

Q2

L1

(table)

R2

(table)

V_OUT

PWM

VSW

LSD

C4

1nF

Q1

(table)

D1

(table)

C7

(table)

C12

0.1µF

50V

SS

PGND

CSH

GND

COMP

SYNC

C13, 1nF

R1

2k

C3

0.1µF

VOUT

VREF

C2

2.2nF

C1

0.1µF

50V

SGND

GND

Figure 14. Basic Circuit Diagram for Use with Tables 3 through 6

Specification

Switching frequency ripple

Maximum ambient temperature

Short-circuit capability

Limit

1% of output voltage

85°C

Continuous

300kHz

Switching frequency

Table 1. Specifications for Figure 14 and Tables 3 through 6

Manufacturer

AVX

Telephone Number (USA)

(803) 946-0690

(516) 435-1110

(561) 241-7876

(704) 264-8861

(310) 322-3331

(408) 944-0800

(805) 446-4800

Web Address

www.avxcorp.com

Central Semiconductor

www.centralsemi.com

www.coiltronics.com

Coiltronics

IRC

IR

www.irf.com

www.micrel.com

Micrel

Vishay/Lite On

(diodes)

www.vishay-liteon.com

Vishay/Siliconix

(MOSFETs)

(800) 554-5665

(800) 487-9437

www.siliconix.com

Vishay/Dale

www.vishaytechno.com

(inductors and resistors)

Sumida

(847) 956-0666

www.japanlink.com/sumida

Table 2. Component Suppliers

April 22, 2004

21

M9999-042204

MIC2182

Micrel

3A (6.5V–30V)

4A (6.5V–30V)

5A (6.5V–30V)

10A (6.5V–10V)

Reference

Part No. / Description

C7

qty: 2

TPSE227M010R0100

AVX, 220µF 10V,

0.1Ω ESR,

TPSE227M010R0100

AVX, 220µF 10V,

0.1Ω ESR,

TPSV227M010R0060

AVX, 220µF 10V,

0.06Ω ESR,

TPSV337M010R0060

AVX, 330µF 10V,

0.06Ω ESR,

output filter capacitor

C11

qty: 2

qty: 3

qty: 4

TPSE226M035R0300

AVX, 22µF 35V,

0.3Ω ESR,

TPSE226M035R0300

AVX, 22µF 35V,

0.3Ω ESR,

TPSE226M035R0300

AVX, 22µF 35V,

0.3Ω ESR,

TPSV107M020R0085

AVX, 100µF 20V,

0.06Ω ESR,

input filter capacitor

D1

L1

qty: 1 B140, Vishay,

freewheeling diode

qty: 1 B140, Vishay,

freewheeling diode

qty: 1 B140, Vishay,

freewheeling diode

qty: 1 B330, Vishay,

freewheeling diode

qty: 1 CDRH125-100,

Sumida Inductor,

10µH 4A,

qty: 1 CDRH127-100,

Sumida Inductor,

10µH 5A,

qty: 1 CDRH127-100

Sumida,

10µH 5A,

qty: 1 UP4B-3R3,

Coiltronics,

3.3µH 11A,

output inductor

Q1

Q2

R2

qty: 1 Si4800, Siliconix,

low-side MOSFET

qty: 1 Si4800, Siliconix,

low-side MOSFET

qty: 1 Si4884, Siliconix,

low-side MOSFET

qty: 2 Si4884, Siliconix

low-side MOSFET

qty: 1 Si4800, Siliconix,

high-side MOSFET

qty: 1 Si4800, Siliconix,

high-side MOSFET

qty: 1 Si4884, Siliconix,

high-side MOSFET

qty: 2 Si4884, Siliconix,

high-side MOSFET

qty: 1

qty: 2

WSL-2010 .025 1%,

Vishay, 0.025, 1%, 0.5W,

current sense resistor

WSL-2010 .020 1%,

Vishay, 0.02, 1%, 0.5W,

current sense resistor

WSL-2512 .015 1%,

Vishay, 0.015, 1%, 1W,

current sense resistor

WSL-2512 .015 1% ,

Vishay, 0.015, 1%, 1W,

current sense resistor

U1

MIC2182-5.0BSM or

MIC2182-5.0BM

MIC2182-5.0BSM or

MIC2182-5.0BM

MIC2182-5.0BSM or

MIC2182-5.0BM

Table 3. Components for 5V Output

3A (4.5V–30V)

4A (4.5V–30V)

5A (4.5V–30V)

10A (4.5V–5.5V)

Reference

C7

Part No. / Description

qty: 2

TPSE227M010R0100

AVX, 220µF 10V,

0.1Ω ESR,

TPSE227M010R0100

AVX, 220µF 10V,

0.1Ω ESR,

TPSV227M010R0060

AVX, 220µF 10V,

0.06Ω ESR,

TPSV477M006R0055

AVX, 470µF 6.3V,

0.055Ω ESR,

output filter capacitor

C11

qty: 2

qty: 3

TPSE226M035R0300

AVX, 22µF 35V,

0.3Ω ESR,

TPSE226M035R0300

AVX, 22µF 35V,

0.3Ω ESR,

TPSE226M035R0300

AVX, 22µF 35V,

0.3Ω ESR,

TPSV227M016R0075

AVX, 220µF 16V,

0.075Ω ESR,

filter capacitor

input filter capacitor

D1

L1

qty: 1 B140, Vishay,

freewheeling diode

qty: 1 B140, Vishay,

freewheeling diode

qty: 1 B140, Vishay,

freewheeling diode

qty: 1 B330, Vishay,

freewheeling diode

qty: 1 CDRH125-100,

Sumida Inductor,

10µH 4A,

qty: 1 CDRH127-100,

Sumida Inductor,

10µH 5A,

qty: 1 CDRH127-100

Sumida,

10µH 5A,

qty: 1 UP4B-3R3,

Coiltronics,

3.3µH 11A,

output inductor

Q1

Q2

R2

qty: 1 Si4800, Siliconix,

low-side MOSFET

qty: 1 Si4800, Siliconix,

low-side MOSFET

qty: 1 Si4800, Siliconix,

low-side MOSFET

qty: 2 Si4884, Siliconix,

low-side MOSFET

qty: 1 Si4800, Siliconix,

high-side MOSFET

qty: 1 Si4800, Siliconix,

high-side MOSFET

qty: 1 Si4884, Siliconix,

high-side MOSFET

qty: 2 Si4884, Siliconix,

high-side MOSFET

qty: 1

qty: 2

WSL-2010 .025 1%,

Vishay, 0.025, 1%, 0.5W,

current sense resistor

WSL-2010 .020 1%,

Vishay, 0.02, 1%, 0.5W,

current sense resistor

WSL-2512 .015 1%,

Vishay, 0.015, 1%, 1W,

current sense resistor

WSL-2512 .015 1% ,

Vishay, 0.015, 1%, 1W,

current sense resistor

U1

MIC2182-3.3BSM or

MIC2182-3.3BM

MIC2182-3.3BM or

MIC2182-3.3BSM

MIC2182-3.3BM or

MIC2182-3.3BSM

MIC2182-3.3BM

Table 4. Components for 3.3V Output

M9999-042204

22

April 22, 2004

MIC2182

Micrel

3A (4.5V–30V)

4A (4.5V–30V)

5A (4.5V–30V)

10A (4.5V–5.5V)

Reference

Part No. / Description

C7

qty: 2

TPSE227M010R0100

AVX, 220µF 10V,

0.1Ω ESR,

TPSE227M010R0100

AVX, 220µF 10V,

0.1Ω ESR,

TPSV227M010R0060

AVX, 220µF 10V,

0.06Ω ESR,

TPSV447M006R0055

AVX, 470µF 6.3V,

0.06Ω ESR,

output filter capacitor

C11

qty: 2

qty: 3

TPSE226M035R0300

AVX, 22µF 35V,

0.3Ω ESR,

TPSE226M035R0300

AVX, 22µF 35V,

0.3Ω ESR,

TPSE226M035R0300

AVX, 22µF 35V,

0.3Ω ESR,

TPSV227M016R0075

AVX, 220µF 16V,

0.06Ω ESR,

input filter capacitor

D1

L1

qty: 1 B140, Vishay,

freewheeling diode

qty: 1 B140, Vishay,

freewheeling diode

qty: 1 B140, Vishay,

freewheeling diode

qty: 1 B330, Vishay,

freewheeling diode

qty: 1 CDRH125-100,

Sumida Inductor,

10µH 4A,

qty: 1 CDRH127-100,

Sumida Inductor,

10µH 5A,

qty: 1 CDRH127-100

Sumida,

10µH 5A,

qty: 1 UP4B-3R3,

Coiltronics,

3.3µH 11A,

output inductor

Q1

Q2

R2

qty: 1 Si4800, Siliconix,

low-side MOSFET

qty: 1 Si4884, Siliconix,

low-side MOSFET

qty: 1 Si4884, Siliconix,

low-side MOSFET

qty: 2 Si4884, Siliconix

low-side MOSFET

qty: 1 Si4800, Siliconix,

high-side MOSFET

qty: 1 Si4800, Siliconix,

high-side MOSFET

qty: 1 Si4800, Siliconix,

high-side MOSFET

qty: 2 Si4884, Siliconix,

high-side MOSFET

qty: 1

WSL-2010 .025 1%,

Vishay, 0.025, 1%, 0.5W,

current sense resistor

WSL-2010 .020 1%,

Vishay, 0.02, 1%, 0.5W,

current sense resistor

WSL-2512 .015 1%,

Vishay, 0.015, 1%, 1W,

current sense resistor

WSL-2512 .015 1% ,

Vishay, 0.015, 1%, 1W,

current sense resistor

U1

MIC2182BSM or

MIC2182BM

Table 5. Components for 2.5V Output

3A (4.5V–30V)

4A (4.5V–30V)

5A (4.5V–8V)

10A (4.5V–5.5V)

Reference

Part No. / Description

C7

qty: 2

TPSE227M010R0100

AVX, 220µF 10V,

0.1Ω ESR,

TPSE227M010R0100

AVX, 220µF 10V,

0.1Ω ESR,

TPSV227M010R0060

AVX, 220µF 10V,

0.06Ω ESR,

TPSV447M006R0055

AVX, 470µF 6.3V,

0.06Ω ESR,

output filter capacitor

C11

qty: 2

TPSE226M035R0300

AVX, 22µF 35V,

0.3Ω ESR,

TPSE226M035R0300

AVX, 22µF 35V,

0.3Ω ESR,

TPSE226M035R0300

AVX, 22µF 35V,

0.3Ω ESR,

TPSV227M016R0075

AVX, 220µF 16V,

0.06Ω ESR,

input filter capacitor

D1

L1

qty: 1 B140, Vishay,

freewheeling diode

qty: 1 B140, Vishay,

freewheeling diode

qty: 1 B140, Vishay,

freewheeling diode

qty: 1 B330, Vishay,

freewheeling diode

qty: 1 CDRH125-100,

Sumida Inductor,

10µH 4A,

qty: 1 CDRH127-100,

Sumida Inductor,

10µH 5A,

qty: 1 CDRH127-100

Sumida,

10µH 5A,

qty: 1 UP4B-3R3,

Coiltronics,

3.3µH 11A,

output inductor

Q1

Q2

R2

qty: 1 Si4800, Siliconix,

low-side MOSFET

qty: 1 Si4884, Siliconix,

low-side MOSFET

qty: 1 Si4884, Siliconix,

low-side MOSFET

qty: 2 Si4884, Siliconix

low-side MOSFET

qty: 1 Si4800, Siliconix,

high-side MOSFET

qty: 1 Si4800, Siliconix,

high-side MOSFET

qty: 1 Si4800, Siliconix,

high-side MOSFET

qty: 2 Si4884, Siliconix,

high-side MOSFET

qty: 1

qty: 2

WSL-2010 .025 1%,

Vishay, 0.025, 1%, 0.5W,

current sense resistor

WSL-2010 .020 1%,

Vishay, 0.02, 1%, 0.5W,

current sense resistor

WSL-2512 .015 1%,

Vishay, 0.015, 1%, 1W,

current sense resistor

WSL-2512 .015 1% ,

Vishay, 0.015, 1%, 1W,

current sense resistor

U1

MIC2182BSM or

MIC2182BM

Table 6. Components for 1.8V Output

April 22, 2004

23

M9999-042204

MIC2182

Micrel

Power Supply Operating Characteristics

Effect of Soft-Start Capacitor (C_SS) Value

On Output Voltage Waveforms

During Turn-On

Effect of Soft-Start Capacitor (C_SS) Value

On Output Voltage Waveforms

During Turn-On

(10A Power Supply Configuration)

(4A Power Supply Configuration)

Normal (300kHz Switching Frequency) and

Output Short-Circuit (60kHz) Conditions

Switch Node (Pin 15) Waveforms

Converter Waveforms

V_IN= 7V

SWITCH-NODE

VOLTAGE

L1 = 3.3µH

V_OUT= 3.3V

I_OUT= 10A

HIGH-SIDE

DRIVE VOLTAGE

REFERENCED TO GROUND

QTY: 2

Si4884

HIGH-SIDE MOSFET

GATE-TO-SOURCE VOLTAGE

HIGH-SIDE

MOSFETS

QTY: 2

Si4884

LOW-SIDE

MOSFETS

LOW-SIDE MOSFET

GATE-TO-SOURCE VOLTAGE

10Amps

INDUCTOR CURRENT

Typical Skip-Mode Waveforms

Typical PWM-Mode Waveforms

M9999-042204

24

April 22, 2004

MIC2182

Micrel

Load Transient Response

and Bode Plot

Load Transient Response

and Bode Plot

(4A Power Supply Configuration)

(10A Power Supply Configuration)

V

_IN= 12V

V

_IN= 6V

V_OUT= 3.3V

L1 = 10µH

R2 = 20mΩ

V_OUT= 3.3V

L1 = 3.3µH

R2 = 7.5mΩ

Bode Plot

(4A Power Supply Configuration)

Bode Plot

(10A Power Supply Configuration)

5V Efficiency

(4A Power Supply Configuration)

100

80

60

40

20

0

210

180

150

120

90

100

80

60

40

20

0

210

180

150

120

90

100

Skip

PWM

80

GAIN

60

PHASE

V_IN= 5V

40

R2 = 15mΩ

L1 = 10µH

60

20

PHASE

-20

-40

30

-20

-40

30

1 high-side MOSFET: Si4800

1 low-side MOSFET: Si4800

0

0.01

0.1

1

4

OUTPUT CURRENT (A)

FREQUENCY (Hz)

12V Efficiency

(4A Power Supply Configuration)

24V Efficiency

(4A Power Supply Configuration)

Efficiency

(10A Power Supply Configuration)

100

Skip

80

PWM

80

Skip

PWM

Skip

PWM

60

40

20

60

40

20

0

V_IN= 12V

40

V_IN= 24V

R2 = 15mΩ

L1 = 10µH

R2 = 7.5mΩ

R2 = 15mΩ

L1 = 10µH

L1 = 3.3µH

20

1 high-side MOSFET: Si4800

1 low-side MOSFET: Si4800

2 high-side MOSFETs: Si4884

2 low-side MOSFETs: Si4884

1 high-side MOSFET: Si4800

1 low-side MOSFET: Si4800

0

0.01

0.1

1

4

0.01

0.1

1

4

0.01

0.1

1

10

OUTPUT CURRENT (A)

April 22, 2004

25

M9999-042204

MIC2182

Micrel

Package Information

PIN 1

0.157 (3.99)

0.150 (3.81)

DIMENSIONS:

INCHES (MM)

0.020 (0.51)

REF

0.020 (0.51)

0.013 (0.33)

0.050 (1.27)

BSC

45°

0.0098 (0.249)

0.0040 (0.102)

0°–8°

0.050 (1.27)

0.016 (0.40)

0.394 (10.00)

0.386 (9.80)

SEATING

PLANE

0.0648 (1.646)

0.0434 (1.102)

0.244 (6.20)

0.228 (5.79)

16-pin SOP (M)

5.40 (0.213)

5.20 (0.205)

7.90 (0.311)

DIMENSIONS:

MM (INCH)

7.65 (0.301)

0.875

(0.034) REF

6.33 (0.239)

6.07 (0.249)

2.00 (0.079)

1.73 (0.068)

10°

4°

0.22 (0.009)

0.13 (0.005)

0.38 (0.015)

0.25 (0.010)

1.25 (0.049) REF

0.21 (0.008)

0.05 (0.002)

COPLANARITY:

0.10 (0.004) MAX

0°

–8°

0.95 (0.037)

0.55 (0.022)

0.65 (0.0260)

BSC

16-Pin SSOP (SM)

M9999-042204

26

April 22, 2004

MIC2182

Micrel

April 22, 2004

27

M9999-042204

MIC2182

Micrel

MICREL, INC. 1849 FORTUNE DRIVE SAN JOSE, CA 95131 USA

TEL + 1 (408) 944-0800 FAX + 1 (408) 474-1000 WEB http://www.micrel.com

The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use.

Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.

Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can

reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into

the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s

use or sale of Micrel Products for use in life support appliances, devices or systems is at Purchaser’s own risk and Purchaser agrees to fully indemnify

Micrel for any damages resulting from such use or sale.

M9999-042204

28

April 22, 2004


型号：	MIC2182-3.3YSM
厂家：	MICREL SEMICONDUCTOR
描述：	High-Efficiency Synchronous Buck Controller 高效率同步降压控制器开关光电二极管信息通信管理控制器
文件：	总28页 (文件大小：224K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

MIC2182-3.3YSM [MICREL]

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