MIC5191_06 [MICREL]
Ultra High-Speed, High-Current Active Filter / LDO Controller; 超高速，大电流有源滤波器/ LDO控制器
| 型号： | MIC5191_06 |
| 厂家： | 
MICREL SEMICONDUCTOR |
| 描述： | Ultra High-Speed, High-Current Active Filter / LDO Controller |
| 文件： | 总15页 (文件大小：845K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
MIC5191
Ultra High-Speed, High-Current
Active Filter / LDO Controller
General Description
Features
The MIC5191 is an ultra high-speed linear regulator. It
uses an external N-Channel FET as its power device.
• Input voltage range: VIN = 1.0V to 5.5V
• +1.0% initial output tolerance
The MIC5191's ultra high-speed abilities can handle the
fast load demands of microprocessor cores, ASICs, and
other high-speed devices. Signal bandwidths of greater
than 500 kHz can be achieved with a minimum amount of
capacitance while at the same time keeping the output
voltage clean, regardless of load demand. A powerful
output driver delivers large MOSFETs into their linear
regions, achieving ultra-low dropout voltage.
• Dropout down to 25mV@10A
• Filters out switching frequency noise on input
• Very high large signal bandwidth >500kHz
• PSRR >40dB at 500kHz
• Adjustable output voltage down to 1.0V
• Stable with any output capacitor
• Excellent line and load regulation specifications
• Logic controlled shutdown
1.25VIN ±10% can be turned into 1V ±1% without the use
of a large amount of capacitance.
• Current limit protection
MIC5191 (1.0V reference) is optimized for output voltages
of 1.0V and higher.
The MIC5191 is offered in 10-pin 3mm×3mm MLF® and
10-pin MSOP-10 packages and has an operating junction
temperature range of –40°C to +125°C.
• 10-pin MLF® and MSOP-10 packages
• Available –40°C to +125°C junction temperature
Applications
Data sheets and support documentation can be found on
Micrel’s web site at www.micrel.com.
• Distributed power supplies
• ASIC power supplies
• DSP, µP, and µC power supplies
___________________________________________________________________________________________________________
Typical Application
VCC =12V
C1
0.01µF
VIN 1.2V
V
OUT 1.0V@7A
IR3716S
MIC5191
IS
OUT
FB
VIN
VCC1
VCC2
EN
C2
10µF
PGND
SGND
C3
0.01µF
COMP
R3
GND
GND
PowerPAK is a trademark of Siliconix, Inc.
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
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Micrel, Inc.
MIC5191
Ordering Information
Part Number
Junction
Temperature Range
Voltage
Package
Standard
Pb-Free
MIC5191BML
MIC5191BMM
MIC5191YML
MIC5191YMM
Adj.
Adj.
–40° to +125°C
–40° to +125°C
10-Pin 3mm x 3mm MLF®
10-Pin MSOP
Pin Configuration
VIN
FB
1
2
3
4
5
10 IS
VIN
FB
1
2
3
4
5
10 IS
9
8
7
6
PGND
OUT
VCC2
EN
9
8
7
6
PGND
OUT
VCC2
EN
SGND
VCC1
COMP
SGND
VCC1
COMP
10-Pin 3mm x 3mm MLF® (ML)
Top View
10-Pin MSOP (MM)
Top View
Pin Description
Pin Number
Pin Name
VIN
Pin Function
1
2
3
4
5
6
Input voltage (current sense +).
Feedback input to error amplifier.
Signal ground.
FB
SGND
VCC1
COMP
EN
Supply to the internal voltage regulator.
Error amplifier output for external compensation.
Enable (Input): CMOS-compatible.
Logic high = Enable, Logic low = Shutdown. Do not float pin.
7
8
VCC2
OUT
PGND
IS
Power to output driver.
Output drive to gate of power MOSFET.
Power ground
9
10
Current sense.
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MIC5191
Absolute Maximum Ratings(1)
Operating Ratings(2)
Supply Voltage (VIN)....................................................+6.0V
Enable Voltage (VEN)....................................................+14V
VCC1, VCC2....................................................................+14V
Junction Temperature (TJ) ..................–40°C ≤ TJ ≤ +125°C
ESD Rating(3)
Supply Voltage (VIN)..................................... +1.0V to +5.5V
Enable Voltage (VEN)............................................. 0V to VCC
VCC1, VCC2.................................................. +4.5V to +13.2V
Junction Temperature (TJ) ..................–40°C ≤ TJ ≤ +125°C
Package Thermal Resistance
3x3 MLF-10 (θJA)(4).............................................60°C/W
MSOP-10 (θJA)(5) ..............................................200°C/W
Electrical Characteristics(6)
TA = 25°C with VIN = 1.2V; VCC = 12V; VOUT = 1.0V; bold values indicate –40°C< TJ < +125°C, unless noted.
Parameter
Condition
Min
–1
Typ
Max
+1
Units
%
Output Voltage Accuracy
At 25°C
Over temperature range
VIN = 1.2V to 5.5V
–2
+2
%
Output Voltage Line Regulation
Feedback Voltage
–0.1
0.99
0.005
1
+0.1
1.01
0.5
%/V
V
Output Voltage Load Regulation
VCC Pin Current (VCC1 + VCC2)
VCC Pin Current (VCCsig + VCCdrv)
VIN Pin Current
IL = 10mA to 1A
Enable = 0V
0.02
40
%
µA
mA
µA
µA
mV
µs
Enable = 5V
15
20
15
Current from VIN
10
FB Bias Current
13
30
Current Limit Threshold
Start-up Time
35
50
70
VEN = VIN
25
100
Enable Input Threshold
Regulator enable
Regulator shutdown
0.8
0.6
0.5
100
100
100
V
0.2
V
Enable Hysteresis
mV
nA
nA
Enable Pin Input Current
VIL < 0.2V (Regulator shutdown)
V
IH > 0.8V (Regulator enabled)
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. The device is not guaranteed to function outside its operating rating.
3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF.
4. Per JESD 51-5 (1S2P Direct Attach Method).
5. Per JESD 51-3 (1S0P).
6. Specification for packaged product only.
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MIC5191
Typical Characteristics
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MIC5191
Functional Characteristics
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MIC5191
Functional Diagram
INTERNAL
VOLTAGE
REGULATOR
VCC1
VIN
IS
50mV
CURRENT LIMIT
AMPLIFIER
VCC2
OUT
OUTPUT
CONTROL
AND
EN
ENABLE
LEVEL
SHIFT
PGND
FB
0.5V
ERROR
AMPLIFIER
SGND
COMP
Figure 1. MIC5191 Block Diagram
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MIC5191
Enable
Functional Description
The MIC5191 comes with an active-high enable pin that
allows the regulator to be disabled. Forcing the enable
pin low disables the regulator and sends it into a low off-
mode-current state. Forcing the enable pin high enables
the output voltage. The enable pin cannot be left floating;
a floating enable pin may cause an indeterminate state
on the output.
VIN
The VIN pin is connected to the N-Channel drain. VIN is
the input power being supplied to the output. This pin is
also used to power the internal current limit comparator
and compare the ISENSE voltage for current limit. The
voltage range is from 1.0V min to 5.5V max.
ISENSE
FB
The ISENSE pin is the other input to the current limit
comparator. The output current is limited when the
ISENSE pin's voltage is 50mV less than the VIN pin. In
cases where there is a current limited source and there
isn’t a need for current limit, this pin can be tied directly
to VIN. Its operating voltage range, like the VIN pin, is
1.0V min to 5.5V max.
The feedback pin is used to sense the output voltage for
regulation. The feedback pin is compared to an internal
1.0V reference and the output adjusts the gate voltage
accordingly to maintain regulation. Since the feedback
biasing current is typically 13µA, smaller feedback
resistors should be used to minimize output voltage
error.
VCC1, VCC2
COMP
VCC1 supplies the error amplifier and internal reference,
while VCC2 supplies the output gate drive. For this
reason, ensure these pins have good input capacitor
bypassing for better performance. The operating range
is from 4.5V to 13.2V and both VCC pins should be tied
together. Ensure that the voltage supplied is greater than
a gate-source threshold above the output voltage for the
N-Channel MOSFET selected.
COMP is the external compensation pin. This allows
complete control over the loop to allow stability for any
type of output capacitor, load currents and output
voltage. A detailed explanation of how to compensate
the MIC5191 is in the “Designing with the MIC5191”
section.
SGND, PGND
SGND is the internal signal ground which provides an
isolated ground path from the high current output driver.
The signal ground provides the grounding for noise
sensitive circuits such as the current limit comparator,
error amplifier and the internal reference voltage.
Output
The output drives the external N-Channel MOSFET and
is powered from VCC. The output can sink and source
over 150mA of current to drive either an N-Channel
MOSFET or an external NPN transistor. The output drive
also has short-circuit current protection.
PGND is the power ground and is the grounding path for
the output driver.
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MIC5191
di
Application Information
Designing with the MIC5191
∆V ↓= L
dt ↑
Placing multiple small capacitors with low ESL in parallel
can help reduce the total ESL and reduce voltage droop
during high speed transients. For high speed transients,
the greatest voltage deviation will generally be caused
by output capacitor ESL and parasitic inductance.
Anatomy of a transient response
A voltage regulator can maintain a set output voltage
while its exterior world is pushing and pulling in its
demand for power. The measure of a regulator is
generally how accurately and effectively it can maintain
that voltage, regardless of how the load demands power.
One measure of regulator response is the load step.
This is an intuitive look at how the regulator responds to
a change in load current. Figure 2 is a look at the
transient response to a load step.
di
∆V ↓= L ↓
dt
After the current has overcome the effects of the ESL,
the output voltage will begin to drop proportionally to
time and inversely proportional to output capacitance.
1
∆V =
idt
∫
C
The relationship to output voltage variation will depend
on two aspects, loop bandwidth and output capacitance.
The output capacitance will determine how far the
voltage will fall over a given time. With more capacity-
ance, the drop in voltage will fall at a decreased rate.
This is the reason that for the same bandwidth, more
capacitance provides a better transient response.
1
C
V
it
1
∆V ↓=
idt
∫
↑ C
Output voltage vs. Time
during recovery is
directly proportional to
gain vs. frequency
Also, the time it takes for the regulator to respond is
directly proportional to its gain bandwidth. Higher
bandwidth control loops respond quicker causing a
reduced droop on the supply for the same amount of
capacitance.
1
BW
Time
Figure 2. Typical Transient Response
1
∆V ↓=
idt ↓
∫
C
At the start of a circuit's power demand, the output
voltage is regulated to its set point, while the load current
runs at a constant rate. For many different reasons, a
load may ask for more current without warning. When
this happens, the regulator needs some time to
determine the output voltage drop. This is determined by
the speed of the control loop. So, until enough time has
elapsed, the control loop is oblivious to the voltage
change. The output capacitor must bear the burden of
maintaining the output voltage.
Final recovery back to the regulated voltage is the final
phase of transient response and the most important
factors are gain and time. Higher gain at higher
frequency will get the output voltage closer to its
regulation point quicker. The final settling point will be
determined by the load regulation, which in proportional
to DC (0Hz) gain and the associated loss terms.
There are other factors that contribute to large signal
transient response, such as source impedance, phase
margin and PSRR. For example, if the input voltage
drops due to source impedance during a load transient,
this will contribute to the output voltage deviation by
filtering through to the output reduced by the loops
PSRR at the frequency of the voltage transient. It is
straightforward: good input capacitance reduces the
source impedance at high frequencies. Having between
35° and 45° of phase margin will help speed up the
recovery time. This is caused by the initial overshoot in
response to the loop sensing a low voltage.
di
∆V = L
dt
Since this is a sudden change in voltage, the capacitor
will try to maintain voltage by discharging current to the
output. The first voltage drop is due to the output
capacitor's ESL (equivalent series inductance). The ESL
will resist a sudden change in current from the capacitor
and drop the voltage quickly. The amount of voltage
drop during this time will be proportional to the output
capacitor's ESL and the speed at which the load steps.
Slower load current transients will reduce this effect.
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MIC5191
Internal
Error Amplifier
Compensation
The MIC5191 allows the flexibility of externally
controlling the gain and bandwidth. This allows the
MIC5191 design to be tailored to each individual design.
Driver
20pF
In designing the MIC5191, it is important to maintain
adequate phase margin. This is generally achieved by
having the gain cross the 0dB point with a single pole
20dB/decad roll-off. The compensation pin is configured
as Figure 3 demonstrates.
External
Comp
RCOMP
CCOMP
Internal
Error Amplifier
Driver
Figure 5. External Compensation
20pF
Placing an external capacitor (CCOMP) and resistor
(RCOMP) for the external pole-zero combination. Where
the dominant pole can be calculated as follows:
External
Comp
1
Figure 3. Internal Compensation
FP
=
2π × 3.42MΩ × CCOMP
This places a pole at 2.3 kHz at 80dB and calculates as
follows.
And the zero can be calculated as follows:
1
1
FZ
=
FP
=
2π × RCOMP × CCOMP
2π × 3.42MΩ × 20pF
This allows for high DC gain, and high bandwidth with
the output capacitor and the load providing the final pole.
FP = 2.32kHz
100
225
180
135
100
80
60
40
20
0
225
180
135
90
The Dominant Pole
1
80
60
Fp
2 × 3.42M × Ccomp
External Zero
1
40
20
0
90
45
0
Fz
2 × Rcomp × Ccomp
R load
x C out Pole
45
0
-20
-45
0.01
0.1
1
10
100
1000
10000 100000
-20
-45
Frequency (KHz)
0.01
0.1
1
10
100
1000
10000 100000
Frequency (KHz)
Figure 4. Internal Compensation
Frequency Response
Figure 6. External Compensation
Frequency Response
There is single pole roll off. For most applications, an
output capacitor is required. The output capacitor and
load resistance create another pole. This causes a two-
pole system and can potentially cause design instability
with inadequate phase margin. What should we do?
Answer: we compensate it externally. By providing a
dominant pole and zero–allowing the output capacitor
and load to provide the final pole–a net single pole roll
off is created, with the zero canceling the dominant pole.
Figure 5 demonstrates:
It is recommended that the gain bandwidth should be
designed to be less than 1 MHz. This is because most
capacitors lose capacitance at high frequency and
becoming resistive or inductive. This can be difficult to
compensate for and can create high frequency ringing or
worse, oscillations.
By increasing the amount of output capacitance,
transient response can be improved in multiple ways.
First, the rate of voltage drop vs. time is decreased.
Also, by increasing the output capacitor, the pole formed
by the load and the output capacitor decreases in
frequency. This allows for the increasing of the
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MIC5191
compensation resistor, creating a higher mid-band gain.
Now that we know the amount of power we will be
dissipating, we will need to know the maximum ambient
air temperature. For our case we’re going to assume a
maximum of 65°C ambient temperature, though different
MOSFETs have different maximum operating junction
temperatures. Most MOSFETs are rated to 150°C, while
others are rated as high as 175°C. In this case, we’re
going to limit our maximum junction temperature to
125°C. The MIC5191 has no internal thermal protection
for the MOSFET so it is important that the design
provides margin for the maximum junction temperature.
Our design will maintain better than 125°C junction
temperature with 1.95W of power dissipation at an
ambient temperature of 65°C. Our thermal resistance
calculates as follows:
100
80
60
40
20
0
225
180
135
90
Increasing Cout reduces
the load resistance and
output capacitor pole
allowing for an increase
in mid-band gain
45
0
-20
-45
0.01
0.1
1
10
100
1000
10000 100000
Frequency (KHz)
Figure 7. Increasing Output Capacitance
TJ(max) − TJ(ambient)
θJA
=
=
PD
This will have the effect of both decreasing the voltage
drop as well as returning closer and faster to the
regulated voltage during the recovery time.
125°C − 65°C
θJA
1.95W
MOSFET Selection
θ
JA = 31°C /W
The typical pass element for the MIC5191 is an N-
Channel MOSFET. There are multiple considerations
when choosing a MOSFET. These include:
So our package must have a thermal resistance less
than 31°C /W. Table 1 shows a good approximation of
power dissipation and package recommendation.
•
•
•
•
•
VIN to VOUT differential
Output Current
Package
Power Dissipation
<850mW
<950mW
<1W
Case Size/Thermal Characteristics
Gate Capacitance (CISS<10nF)
Gate to Source threshold
TSOP-6
TSSOP-8
TSSOP-8
The VIN(min) to VOUT ratio and current will determine the
maximum RDSON required. For example, for a 1.8V (±5%)
to 1.5V conversion at 5A of load current, dropout voltage
PowerPAK™ 1212-8
SO-8
<1.1W
<1.125W
<1.4W
PowerPAK™ SO-8 D-Pack
TO-220/TO-263 (D2pack)
can be calculated as follows (using VIN(min)
:
>1.4W
(
VIN − VOUT
)
RDSON
=
=
IOUT
Table 1. Power Dissipation and
Package Recommendation
(
1.71V − 1.5V
)
RDSON
In our example, our power dissipation is greater than
1.4W, so we’ll choose a TO-263 (D2Pack) N-Channel
MOSFET. θJA is calculated as follows:
5A
R
DSON = 42mΩ
θ
JA = θJC + θCS + θSA
For performance reasons, we do not want to run the N-
Channel in dropout. This will seriously affect transient
response and PSRR (power supply ripple rejection). For
this reason, we want to select a MOSFET that has lower
than 42mꢀ for our example application.
Where θJC is the junction to case resistance, θCS is the
case-to-sink resistance and the θSA is the sink-to-ambi-
ent air resistance.
In the D2 package we’ve selected, the θJC is 2°C/W. The
θCS, assuming we are using the PCB as the heat sink,
can be approximated to 0.2°C/W. This allows us to
calculate the minimum θSA:
Size is another important consideration. Most import-
antly, the design must be able to handle the amount of
power being dissipated.
The amount of power dissipated can be calculated as
follows (using VIN(max)):
θ
θ
θ
SA = θJA– θCS – θJC
SA = 31°C/W – 0.2°C/W – 2°C/W
SA = 28.8°C/W
PD = (VIN – VOUT) × IOUT
PD = (1.89V – 1.5V) × 5A
PD = 1.95W
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MIC5191
Referring to Application Hint 17, Designing PCB Heat
Sinks, the minimum amount of copper area for a D2pack
at 28.8°C/W is 2750mm2 (or 0.426in2). The solid line
denotes convection heating only (2 oz. copper) and the
dotted line shows thermal resistance with 250LFM air-
flow. The copper area can be significantly reduced by
increasing airflow or by adding external heat sinks.
source voltage) will be less than the fully enhanced VGS,
it is recommended the VCC voltage has 2V over the
minimum VGS and output voltage. This is due to the
saturation voltage of the MIC5191 output driver.
V
CC1, 2 ≥ 2V + VGS + VOUT
For our example, with a 1.5V output voltage, our
MOSFET is fully enhanced at 4.5VGS, our VCC voltage
should be greater or equal to 8V.
Input Capacitor
Good input bypassing is important for improved perfor-
mance. Low ESR and low ESL input capacitors reduce
both the drain of the N-Channel MOSFET, as well as the
source impedance to the MIC5191. When a load
transient on the output occurs, the load step will also
appear on the input. Deviations on the input voltage will
be reduced by the MIC5191’s PSRR, but nonetheless
appear on the output. There is no minimum input
capacitance, but for optimal performance it is
recommended that the input capacitance be equal to or
greater than the output capacitance.
Output Capacitor
Figure 8. PC Board Heat Sink
The MIC5191 is stable with any type or value of output
capacitor (even without any output capacitor!). This
allows the output capacitor to select which parameters of
the regulator are important. In cases where transient
response is the most important, low ESR and low ESL
ceramic capacitors are recommended. Also, the more
capacitance on the output, the better the transient
response.
Another important characteristic is the amount of gate
capacitance. Large gate capacitance can reduce
transient performance by reducing the ability of the
MIC5190 to slew the gate. It is recommended that the
MOSFET used has an input capacitance <10nF (CISS).
Source threshold specified in most MOSFET data sheets
refers to the minimum voltage needed to fully enhance
the MOSFET. Although for the most part, the MOSFET
will be operating in the linear region and the VGS (gate-
VIN
J1
+VIN
330µF
16V
10µF 10µF
10µF
12V
100k
22µF
U1 MIC2198-BML
L1
1µF
25V
CSH
VOUT
VOUT
6
2
12
11
IRF7821
1.8µH
J2
EN
VIN
HSD
VSW
CDEP134-1R8MC-H
1VOUT @10A
10k
EN/UVLO
CSH
CSH
0.1µF
10µF
10µF
4
10
8
BST
VOUT
VOUT
100pF
MIC5191
VCC1
5
3
IRF7821
VOUT
FB
OUT
1µF
LSD
VDD
330µF
Tantalum
10k
D2
VIN
VCC2
FB
D1
1N5819HW
SD103BWS
ISENSE
GND
1
7
COMP
COMP
560pF
GND
2.2µF
10V
10nF
12.4k
9
11.5k
8.06k
Figure 9. Post Regulator
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MIC5191
Feedback Resistors
Applying the MIC5191
Linear Regulator
VOUT
IR3716S
The primary purpose of the MIC5191 is as a linear
regulator, which enables an input supply voltage to drop
down through the resistance of the pass element to a
regulated output voltage.
MIC5191
R1
R2
FB
COUT
Active Filter
Another application for the MIC5191 is as an active filter
on the output of a switching regulator. This improves the
power supply in several ways.
First, using the MIC5191 as a filter on the output can
significantly reduce high frequency noise. Switching
power supplies tends to create noise at the switching
frequency in the form of a triangular voltage ripple. High
frequency noise is also created by the high-speed
switching transitions. A lot of time, effort, and money are
thrown into the design of switching regulators to
minimize these effects as much as possible. Figure 9
shows the MIC5191 as a post regulator.
GND
Figure 10. Adjustable Output
The feedback resistors adjust the output to the desired
voltage and can be calculated as follows:
R1
R2
⎛
⎞
⎟
VOUT = VREF 1+
⎜
⎝
⎠
VREF is equal to 1.0V for the MIC5191. The minimum
output voltage (R1=0) is 0.5V. For output voltages less
than 1V, use the MIC5190.The resistor tolerance adds
error to the output voltage. These errors are
accumulative for both R1 and R2. For example, our
resistors selected have a ±1% tolerance. This will
contribute to a ±2% additional error on the output
voltage. The feedback resistors must also be small
enough to allow enough current to the feedback node.
Large feedback resistors will contribute to output voltage
error.
V
V
V
ERROR = R1 x 1FB
ERROR = 1kꢀ x 12µA
ERROR = 12mV
Figure 11. Ripple Reduction
Figure 11 shows the amount of ripple reduction for a 500
KHz switching regulator. The fundamental switching
frequency is reduced from greater than 100mV to less
than 10mV.
For our example application, this will cause an increase
in output voltage of 12mV. For the percentage increase,
VERROR
VERROR % =
VERROR % =
×100
VOUT
12mV
1.5V
×100
V
ERROR% = 0.8%
By reducing R1 to 100ꢀ, the error contribution by the
feed-back resistors and feedback current is reduced to
less than 0.1%. This is the reason R1 should not be
greater than 100ꢀ.
Figure 12. 10A Load Transient
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MIC5191
The transient response also contributes to the overall AC
output voltage deviation. Figure 12 shows a 1A to 10A
load transient. The top trace is the output of the
switching regulator (same circuit as Figure10). The
output voltage undershoots by 100mV. Just by their
topology, linear regulators have the ability to respond at
much higher speeds than a switching regulator. Linear
regulators do not have the limitation or restrictions of
switching regulators which must reduce their bandwidth
to less than their switching frequency.
If a large circuit board has multiple small-geometry
ASICs, it will require the powering of multiple loads with
its one power source. Assuming that the ASICs are
dispersed throughout the board and that the core voltage
requires a regulated 1V, Figure 14 shows the long traces
from the power supply to the loads. Not only do we have
to contend with the tolerance of the supply (line
regulation, load regulation, output accuracy and
temperature tolerances), but the trace lengths create
additional issues with resistance and inductance. With
lower voltages these parasitic values can easily bump
the output voltage out of a usable tolerance.
Using the MIC5191 as a filter for a switching regulator
reduces output noise due to ripple and high frequency
switching noise. It also reduces undershoot (Figure 12)
and over-shoot (Figure 13) due to load transients with
decreased capacitance.
Circuit Board
Load
Load
Load
Long Traces
Switching
Power
Supply
Load
Figure 14. Board Layout
But by placing multiple, small MIC5191 circuits close to
each load, the parasitic trace elements caused by
distance to the power supply are almost completely
negated. By adjusting the switching supply voltage to
1.2V, for example, the MIC5191 will provide accurate 1V
output, efficiently and with very little noise.
Figure 13. Transient Response
Due to the high DC gain (80dB) of the MIC5191, it also
adds increased output accuracy and extremely high load
regulation.
Distributed Power Supply
Circuit Board
As technology advances and processes move to smaller
and smaller geometries, voltage requirements go down
and current requirements go up. This creates unique
challenges when trying to supply power to multiple
devices on a board. When there is one load to power,
the difficulties are not quite as complex; trying to
distribute power to multiple loads from one supply is
much more problematic.
Load
MIC5191
MIC5191
Load
Load
MIC5191
Switching
Power
Supply
MIC5191
Load
Figure 15. Improved Distributed Supplies
M9999-122206
December 2006
13
Micrel, Inc.
MIC5191
Package Information
10-Pin MSOP (MM)
10-Pin 3mm x 3mm MLF® (ML)
M9999-122206
December 2006
14
Micrel, Inc.
MIC5191
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its
use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2004 Micrel, Incorporated.
M9999-122206
December 2006
15
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