ML4761 [MICRO-LINEAR]
Adjustable Output Low Voltage Boost Regulator; 可调输出低电压升压稳压器
| 型号： | ML4761 |
| 厂家： | 
MICRO LINEAR CORPORATION |
| 描述： | Adjustable Output Low Voltage Boost Regulator |
| 文件： | 总8页 (文件大小：203K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
July 2000
ML4761*
Adjustable Output Low Voltage Boost Regulator
GENERAL DESCRIPTION
FEATURES
The ML4761 is a boost regulator designed for DC to DC
conversion in 1 to 3 cell battery powered systems. The
combination of BiCMOS process technology, internal
synchronous rectification, variable frequency operation,
and low supply current make the ML4761 ideal for 1 cell
applications. The ML4761 is capable of start-up with input
voltages as low as 1V, and the output voltage can be set
anywhere between 2.5V and 6V by an external resistor
divider connected to the SENSE pin.
■ Guaranteed full load start-up and operation at 1V input
■ Pulse Frequency Modulation and Internal Synchronous
Rectification for high efficiency
■ Minimum external components
■ Low ON resistance internal switching FETs
■ Micropower operation
■ Adjustable output voltage (2.5V to 6V)
An integrated synchronous rectifier eliminates the need for
an external Schottky diode and provides a lower forward
voltage drop, resulting in higher conversion efficiency. In
addition, low quiescent battery current and variable
frequency operation result in high efficiency even at light
loads. The ML4761 requires a minimum number of
external components to build a very small adjustable
regulator circuit capable of achieving conversion
efficiencies in excess of 90%.
The circuit also contains a RESET output which goes low
when the IC can no longer function due to low input
voltage (UVLO).
(* Indicates Part is End Of Life as of July 1, 2000)
BLOCK DIAGRAM
L1
C
*
IN
1
6
C
FF
*
V
V
I
L
N
V
OUT
+
5
4
UVLO
R1
R2
SENSE
+
–
V
REF
BOOST
CONTROL
2
3
+
–
C
OUT
V
REF
PWR
GND
GND
–
8
RESET
7
*OPTIONAL
TO MICROPROCESSOR
1
ML4761
PIN CONNECTION
ML4761
8-Pin SOIC (S08)
V
1
2
3
4
8
7
6
5
PWR GND
RESET
IN
V
REF
GND
V
V
L
SENSE
OUT
TOP VIEW
PIN DESCRIPTION
PIN
PIN
NO.
NAME
FUNCTION
NO.
NAME
FUNCTION
1
2
3
4
V
Battery input voltage
5
6
7
V
Boost regulator output
IN
OUT
L
V
200mV reference output
Analog signal ground
V
Boost inductor connection
REF
GND
RESET
Output goes low when regulation
cannot be achieved
SENSE
Programming pin for setting the
output voltage
8
PWR GND Return for the NMOS output
transistor.
2
ML4761
ABSOLUTE MAXIMUM RATINGS
OPERATING CONDITIONS
Absolute maximum ratings are those values beyond which
the device could be permanently damaged. Absolute
maximum ratings are stress ratings only and functional
device operation is not implied.
Temperature Range
ML4761CS .................................................0°C to 70°C
ML4761ES .............................................. –20°C to 70°C
ML4761IS ............................................... –40°C to 85°C
V
V
Operating Range
IN
Voltage on any pin ....................................................... 7V
ML4761CS .................................... 1.0V to V
ML4761ES, ML4761IS ................... 1.1V to V
–0.2V
–0.2V
OUT
OUT
Peak Switch Current, I
.......................................... 2A
(AVG)
(PEAK)
Average Switch Current, I
............................... 500mA
Operating Range................................. 2.5V to 6.0V
OUT
Junction Temperature ............................................. 150°C
Storage Temperature Range ...................... –65°C to 150°C
Lead Temperature (Soldering 10 sec.) ..................... 260°C
Thermal Resistance (q ) ..................................... 160°C/W
JA
ELECTRICAL CHARACTERISTICS
Unless otherwise specified, VIN = Operating Voltage Range, T = Operating Temperature Range (Note 1)
A
PARAMETER
CONDITIONS
MIN
TYP.
MAX
UNITS
Supply
VIN Current
VIN = VOUT – 0.2V
45
3
55
5
µA
µA
µA
VOUT Quiescent Current
VL Quiescent Current
Reference
1
Output Voltage (VREF
)
0 < IPIN2 < –5µA
VIN = 2.4V
194
200
206
mV
PFM Regulator
Pulse Width (TON
)
C/E Suffix
I Suffix
9
10
10
11
µs
µs
8.5
190
11.5
210
SENSE Comparator
200
mV
Threshold Voltage (VSENSE
)
Load Regulation
See Figure 1
V
IN = 1.2V, IOUT - 25mA
4.85
4.85
5.0
5.0
5.15
5.15
V
V
VIN = 2.4V, IOUT - 135mA
Undervoltage Lockout Threshold
C/E Suffix
I Suffix
0.85
0.95
0.95
1.05
V
V
RESETComparator
RESET Output High Voltage (VOH
)
IOH = –100µA
IOL = 100µA
VOUT – 0.2
V
V
RESET Output Low Voltage (VOL
)
0.2
Note 1: Limits are guaranteed by 100% testing, sampling, or correlation with worst case test conditions.
3
ML4761
27µH
(Sumida CD75)
V
IN
100µF
0.1µF
ML4761
V
V
PWR GND
RESET
IN
REF
GND
V
L
I
OUT
SENSE
V
OUT
97.6KΩ
0.1%
100µF
4.02KΩ
0.1%
Figure 1. Application Test Circuit.
L1
V
IN
6
V
L
Q2
START-UP
V
OUT
+
–
+
5
4
A2
C1
V
OUT
–
Q1
R1
R2
10µs
ONE SHOT
R
S
Q
SENSE
–
+
A1
200mV
Figure 2. PFM Regulator Block Diagram.
4
ML4761
RESETCOMPARATOR
FUNCTIONAL DESCRIPTION
An additional comparator is provided to detect low V .
IN
The ML4761 combines Pulse Frequency Modulation
(PFM) and synchronous rectification to create a boost
converter that is both highly efficient and simple to use.
A PFM regulator charges a single inductor for a fixed
period of time and then completely discharges before
another cycle begins, simplifying the design by
eliminating the need for conventional current limiting
circuitry. Synchronous rectification is accomplished by
replacing an external Schottky diode with an on-chip
PMOS device, reducing switching losses and external
component count.
The inverting input of the comparator is internally
connected to V , while the non-inverting input is
REF
connected to the undervoltage lockout circuit. The output
of the comparator is the RESET pin, which swings from
V
to GND when an undervoltage condition is
OUT
detected.
DESIGN CONSIDERATIONS
INDUCTOR
Selecting the proper inductor for a specific application
usually involves a trade-off between efficiency and
maximum output current. Choosing too high a value will
keep the regulator from delivering the required output
current under worst case conditions. Choosing too low a
value causes efficiency to suffer. It is necessary to know
the maximum required output current and the input
voltage range to select the proper inductor value. The
maximum inductor value can be estimated using the
following formula:
REGULATOR OPERATION
A block diagram of the boost converter is shown in Figure
2. The circuit remains idle when V
is at or above the
OUT
desired output voltage, drawing 45µA from V , and 8µA
IN
from V
through the feedback resistors R1 and R2.
OUT
OUT
When V
drops below the desired output level, the
output of amplifier A1 goes high, signaling the regulator to
deliver charge to the output. Since the output of amplifier
A2 is normally high, the flip-flop captures the A1 set signal
and creates a pulse at the gate of the NMOS transistor Q1.
The NMOS transistor will charge the inductor L1 for 10µs,
resulting in a peak current given by:
V
2 × TON(MIN) × η
IN(MIN)
LMAX
=
(2)
2 × VOUT ×IOUT(MAX)
T
× V
10µs × V
IN
ON
IN
I
=
≈
(1)
L(PEAK)
where h is the efficiency, typically between 0.8 and 0.9.
Note that this is the value of inductance that just barely
delivers the required output current under worst case
conditions. A lower value may be required to cover
inductor tolerance, the effect of lower peak inductor
currents caused by resistive losses, and minimum dead
time between pulses.
L1
L1
For reliable operation, L1 should be chosen so that I
does not exceed 2A.
L(PEAK)
When the one-shot times out, the NMOS FET releases the
V pin, allowing the inductor to fly-back and momentarily
L
charge the output through the body diode of PMOS
transistor Q2. But, as the voltage across the PMOS
transistor changes polarity, its gate will be driven low by
the current sense amplifier A2, causing Q2 to short out its
body diode. The inductor then discharges into the load
through Q2. The output of A2 also serves to reset the flip-
flop and one-shot in preparation for the next charging
cycle. A2 releases the gate of Q2 when its current falls to
Another method of determining the appropriate inductor
value is to make an estimate based on the typical
performance curves given in Figures 4 and 5. Figure 4
shows maximum output current as a function of input
voltage for several inductor values. These are typical
performance curves and leave no margin for inductance
and ON-time variations. To accommodate worst case
conditions, it is necessary to derate these curves by at
least 10% in addition to inductor tolerance. Interpolation
between the different curves will give a reasonable
starting point for an inductor value.
zero. If V
is still low, the flip-flop will immediately
OUT
initiate another pulse. The output capacitor (C1) filters the
inductor current, limiting output voltage ripple. Inductor
current and one-shot waveforms are shown in Figure 3.
INDUCTOR
CURRENT
Q2
ON
Q2
ON
Q(ONE SHOT)
Q1 ON
Q1 ON
Q1 & Q2 OFF
Figure 3. PFM Inductor Current Waveforms and Timing.
5
ML4761
ML4761(V
= 3.3V)
ML4761(V
= 5.0V)
OUT
OUT
500
400
300
200
100
500
400
300
200
100
L = 27µH
L = 56µH
L = 10µH
L = 15µH
L = 15µH
L = 27µH
L = 10µH
L = 56µH
0
0
1.0
2.0
(V)
3.0
1.0
2.0
(V)
3.0
4.0
V
V
IN
IN
Figure 4. Output Current vs. Input Voltage.
ML4761-(V
= 3.3V)
OUT
ML4761-(V
= 5.0V)
OUT
95%
90%
95%
90%
85%
80%
75%
70%
L = 57µH
L = 56µH
L = 27µH
L = 27µH
L = 15µH
85%
80%
75%
70%
65%
L = 15µH
L = 10µH
L = 10µH
0
1.0
2.0
3.0
4.0
V
IN
(V)
0
1
2
3
V
IN
Figure 5. Typical Efficiency as a Function of V .
IN
When comparing various inductors, it is important to keep
in mind that suppliers use different criteria to determine
their ratings. Many use a conservative current level, where
inductance has dropped to 90% of its normal level. In any
case, it is a good idea to try inductors of various current
ratings with the ML4761 to determine which inductor is
the best choice. Check efficiency and maximum output
current, and if a current probe is available, look at the
inductor current to see if it looks like the waveform shown
in Figure 3. For additional information, see Application
Note 29.
Figure 5 shows efficiency under the conditions used to
create Figure 4. It can be seen that efficiency is mostly
independent of input voltage and is closely related to
inductor value. This illustrates the need to keep the
inductor value as high as possible to attain peak system
efficiency. As the inductor value goes down to 10µH, the
efficiency drops to between 70% and 75%. With 56µH,
the efficiency exceeds 90% and there is little room for
improvement. At values greater than 100µH, the operation
of the synchronous rectifier becomes unreliable because
the inductor current is so small that it is difficult for the
control circuitry to detect.
Suitable inductors can be purchased from the following
suppliers:
After the appropriate inductor value is chosen, it is
necessary to find the minimum inductor current rating
required. Peak inductor current is determined from the
following formula:
Coilcraft
Coiltronics
Dale
(708) 639-6400
(407) 241-7876
(605) 665-9301
(708) 956-0666
T
× V
IN(MAX)
ON(MAX)
I
=
(3)
Sumida
L(PEAK)
L
MIN
6
ML4761
OUTPUT CAPACITOR
SETTING THE OUTPUT VOLTAGE
The choice of output capacitor is also important, as it
controls the output ripple and optimizes the efficiency of
the circuit. Output ripple is influenced by three capacitor
parameters: capacitance, ESR, and ESL. The contribution
due to capacitance can be determined by looking at the
change in capacitor voltage required to store the energy
delivered by the inductor in a single charge-discharge
cycle, as determined by the following formula:
The adjustable output can be set to any voltage between
2.5V and 6V by connecting a resistor divider to the
SENSE pin as shown in the block diagram. The resistor
values R and R can be calculated using the following
1
2
equation:
(R +R )
1
2
V
= 0.2 ×
(5)
OUT
R
2
The value of R should be 40ký or less to minimize bias
2
2
TON2 × V
current errors. R is then found by rearranging the
1
IN
∆VOUT
=
(4)
equation:
2 ×L × C × (VOUT − V )
IN
V
0.2
For a 2.4V input, and 5V output, a 27µH inductor, and a
47µF capacitor, the expected output ripple due to
capacitor value is 87mV.
OUT
R = R ×
−1
(6)
1
2
It is important to note that the accuracy of these resistors
directly affects the accuracy of the output voltage. The
SENSE pin threshold variation is ±3%, and the tolerances
Capacitor Equivalent Series Resistance (ESR) and
Equivalent Series Inductance (ESL), also contribute to the
output ripple due to the inductor discharge current
waveform. Just after the NMOS transistor turns off, the
output current ramps quickly to match the peak inductor
current. This fast change in current through the output
capacitor’s ESL causes a high frequency (5ns) spike that
can be over 1V in magnitude. After the ESL spike settles,
the output voltage still has a ripple component equal to
the inductor discharge current times the ESR. This
component will have a sawtooth shape and a peak value
equal to the peak inductor current times the ESR. ESR also
has a negative effect on efficiency by contributing
I-squared R losses during the discharge cycle.
of R and R will add to this to determine the total output
1
2
variation.
REFERENCE CAPACITOR
Under some circumstances input ripple cannot be
reduced effectively. This occurs primarily in applications
where inductor currents are high, causing excess output
ripple due to “pulse grouping”, where the charge-
discharge pulses are not evenly spaced in time. In such
cases it may be necessary to decouple the reference pin
(V ) with a small 10nF to 100nF ceramic capacitor. This
REF
is particularly true if the ripple voltage at V is greater
IN
An output capacitor with a capacitance of 100µF, an ESR
of less than 0.1ý, and an ESL of less than 5nH is a good
general purpose choice. Tantalum capacitors which meet
these requirements can be obtained from the following
suppliers:
than 100mV.
In some applications, input noise may cause output ripple
to become excessive due to “pulse grouping”, where the
charge-discharge pulses are not evenly spaced in time. In
such cases it may be necessary to add a small 20pF to
AVX
(207) 282-5111
(207) 324-4140
100pF ceramic feedforward capacitor (C ) from the V
FF
IN
pin to the SENSE pin.
Sprague
If ESL spikes are causing output noise problems, an EMI
filter can be added in series with the output.
LAYOUT
Good PC board layout practices will ensure the proper
operation of the ML4761. Important layout considerations
include:
INPUT CAPACITOR
Unless the input source is a very low impedance battery, it
will be necessary to decouple the input with a capacitor
with a value of between 47µF and 100µF. This provides
the benefits of preventing input ripple from affecting the
ML4761 control circuitry, and it also improves efficiency
by reducing I-squared R losses during the charge and
discharge cycles of the inductor. Again, a low ESR
capacitor (such as tantalum) is recommended.
• Use adequate ground and power traces or planes
• Keep components as close as possible to the ML4761
• Use short trace lengths from the inductor to the V pin
L
and from the output capacitor to the V
pin
OUT
• Use a single point ground for the ML4761 ground pins,
and the input and output capacitors
7
ML4761
PHYSICAL DIMENSIONS inches (millimeters)
Package: S08
8-Pin SOIC
0.189 - 0.199
(4.80 - 5.06)
8
0.148 - 0.158 0.228 - 0.244
(3.76 - 4.01) (5.79 - 6.20)
PIN 1 ID
1
0.017 - 0.027
(0.43 - 0.69)
(4 PLACES)
0.050 BSC
(1.27 BSC)
0.059 - 0.069
(1.49 - 1.75)
0º - 8º
0.012 - 0.020
(0.30 - 0.51)
0.015 - 0.035
(0.38 - 0.89)
0.006 - 0.010
(0.15 - 0.26)
0.055 - 0.061
(1.40 - 1.55)
0.004 - 0.010
(0.10 - 0.26)
SEATING PLANE
ORDERING INFORMATION
PART NUMBER
TEMPERATURE RANGE
0°C to 70°C
PACKAGE
8-Pin SOIC (S08)
8-Pin SOIC (S08)
8-Pin SOIC (S08)
ML4761CS (End Of Life)
ML4761ES (Obsolete)
ML4761IS (Obsolete)
–20°C to 70°C
–40°C to 85°C
© Micro Linear 1996
is a registered trademark of Micro Linear Corporation
Products described in this document may be covered by one or more of the following patents: 4,897,611; 4,964,026; 5,027,116;
5,281,862; 5,283,483; 5,418,502; 5,508,570; 5,510,727; 5,523,940; 5,546,017, 5,559,470. Other patents are pending.
Micro Linear reserves the right to make changes to any product herein to improve reliability, function or design.
Micro Linear does not assume any liability arising out of the application or use of any product described herein,
neither does it convey any license under its patent right nor the rights of others. The circuits contained in this
data sheet are offered as possible applications only. Micro Linear makes no warranties or representations as to
whether the illustrated circuits infringe any intellectual property rights of others, and will accept no responsibility
or liability for use of any application herein. The customer is urged to consult with appropriate legal counsel
before deciding on a particular application.
2092 Concourse Drive
San Jose, CA 95131
Tel: 408/433-5200
Fax: 408/432-0295
DS4761-01
8
相关型号：
©2020 ICPDF网 联系我们和版权申明