AN912 [MICROCHIP]

Designing LF Talkback for a Magnetic Base Station; 设计LF对讲的磁性基站


型号：	AN912
厂家：	MICROCHIP
描述：	Designing LF Talkback for a Magnetic Base Station 设计LF对讲的磁性基站
文件：	总16页 (文件大小：429K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

AN912

Designing LF Talkback for a Magnetic Base Station

MAIN BUILDING BLOCKS

Author:

Ruan Lourens

Microchip Technology Inc.

Figure 1 shows the main building blocks that make up

the LF Talkback system described in this document.

The base station generates a strong magnetic field by

setting up resonance in a serial resonant tank. The

circulating energy in the resonant tank typically gener-

ates 300V peak-to-peak voltage across the transmitting

antenna coil at 125 kHz. The transponder, whether

active or passive, is magnetically coupled to the base

station’s transmitting coil and the transponder’s

magnetic loading has a small effect on the quality factor

(Q) of the transmitter resonant tank. Talkback is

accomplished by changing or modulating the magnetic

loading and can be observed as small voltage changes

across the base station's resonant transmitter coil. The

difficulty is to detect a few mV of modulation on the

300V peak-to-peak carrier.

INTRODUCTION

This application note builds on application note AN232

Low Frequency Magnetic Transmitter Design

(DS00232). It covers the design process to implement

LF Talkback functionality. AN232 covers some of the

magnetism basics and design principles to implement

the drive circuitry. LF Talkback generally refers to the

process in which a transponder can communicate back

to a magnetic transmitter base station by loading the

generated magnetic field. By measuring the small

changes in the transmitter coil's voltage, used to gener-

ate the field, the communications’ data is extracted. LF

Talkback is commonly used in RFID, automotive

transponders, active transponders, and many other

bidirectional LF communications topologies.

This document will cover the different stages needed to

implement a typical LF Talkback system and explain

the process in choosing the different stage characteris-

tics. It explains the various performance and cost trade-

offs made for the reference design and how it can be

adapted to better suit the readers needs.

FIGURE 1:

Peak

Detector

Decouple

Low Pass

Filter

Data

Slicer

Base Station

Transponder

VREF

Data

 2004 Microchip Technology Inc.

DS00912A-page 1

AN912

A high voltage peak detector is used to extract the

basic envelope of the base station's resonant tank. The

output of the peak detector will be 150 VDC with about

2V peak-to-peak of carrier ripple at 125 kHz and then

about 2 mV of modulated signal. The modulation signal

strength is mostly dependent on the distance between

the transponder and the transmitter coil as the

magnetic coupling decreases to the third power of the

distance between the two devices. The next stage is a

passive high-pass filter to decouple or block the high

DC voltage. The DC extracted voltage is then fed into a

low-pass filter, leaving the required modulating signal.

The last stage is the data slicer that compares the

modulating signal to some reference to extract the

original signal sent by the transponder.

THE PEAK DETECTOR

There are a number of aspects to consider in designing

a peak detector for this application:

1. The peak detector has to be able to operate at

the high voltages of the resonant tank.

2. Maintain a good tank Q or, in other words, it

should not add unnecessary loading on the main

resonant tank. If it does load the tank, it will

result in a lower modulation voltage induced by

the transponder.

3. Reduce carrier ripple as far as possible.

4. Maintain the modulation signal.

5. Have a fast large swing dynamic response and

be able to settle quickly after the field is turned

on.

LF Talkback receiver can be thought of as detecting

and decoding an amplitude modulation (AM) signal that

has a very low modulation index on a relatively large

carrier.

6. Cost of the system.

Some of the peak detector requirements are conflicting

and as a result, the designer has to find an acceptable

compromise with the final system performance in mind.

One can sacrifice a specific parameter and make up for

it in a later stage where optimization of that aspect is

easily accomplished.

SYSTEM ASSUMPTIONS

The LF Talkback system designed in this document is

targeted for a LF base station that has the following

characteristics and is based on the design as per

AN232:

As an example to optimize requirement 3, one needs to

increase the size of the capacitor C2 (Figure 2), but

that will negatively affect requirements 2, 4 and 5 if a

passive peak detector is used. An active peak detector

could have solved the conflict, but at the 600V swing,

one has little choice but to use a passive peak detector

while maintaining a low-cost design. A relatively low

capacitance value is chosen for C1 of 1 nF. This

maintains the dynamic response requirement for

settling quickly after the field is applied and does not

load the tank unnecessarily. Capacitor C2 should have

at least a 300 VDC peak rating and a high tolerance

capacity is acceptable to save cost. An ultra fast diode

is required in the peak detector with a 400V or better

rating and low junction capacitance. A UF1005 diode

was chosen, it has a 600V rating and 10 pF of junction

capacitance.

• The LF Talkback signal is amplitude modulated at

200 µs multiples. This is also referred to as the

basic pulse element period or TE.

• The tank is driven by a 12V half-bridge driver.

• The tank inductance is 162 µH and the resonant

capacitor is 10 nF with a resonant frequency of

125 kHz.

• The tank Q is 25. As a result, the tank or carrier

voltage is 300V peak-to-peak or 150V 0-to-peak.

• Transponder induced modulation of 2 mV in

magnitude needs to be detected.

To get an understanding of the impedances involved,

lets consider the following: using Equation 1, the

equivalent parallel resistance of the tank is 3.18 kΩ.

The additional parallel impedance that a transponder

represents to induce a 2 mV signal on the tank is in the

order of 500 MΩ. What the LF Talkback system detects

is the result of a 500 MΩ resistor being switched in and

out in parallel with the tank at the data rate. Therefore,

it is very important that the peak detector have a high-

impedance at the data rate to maintain good sensitivity.

FIGURE 2:

HV-Env

EQUATION 1:

THE EFFECTIVE PARALLEL

IMPEDANCE OF A

RESONANT TANK

RPARALLEL

= 2πLFCQ

L = Tank inductance in H = 162 µH

Fc = Center frequency of tank = 125 kHz

Q = Tank quality factor = 25

DS00912A-page 2

 2004 Microchip Technology Inc.

AN912

The envelope detector with only D1 and C2 has a

greatly different response to increasing and decreasing

voltage amplitudes of the resonant tank. The voltage

designated by the signal HV_Env (Figure 2) rises

quickly with increasing tank amplitudes because D1

has a low-impedance in forward conduction. The tank

voltage decreases slower when the tank amplitude is

lowered because C2 can only discharge through D1,

which has a high-impedance in the reverse direction.

The situation can be remedied to some extent by the

introduction of R1 which helps to discharge C2, but the

value of R1 should be high enough to maintain a good

tank Q as per requirement 2 above. A 10 MΩ value for

R1 works well, but note that R1 needs to be

implemented as a series of two resistors. This is done

to stay within the safe voltage range of 0805 resistors

are used.

FIGURE 3:

HV-Env

LP Filter

The system can be simplified as shown in Figure 3.

The output of the peak detector can be simplified as the

step response source with a 150V amplitude that also

has the carrier and data signals superimposed on it as

described earlier. The output response of the

decoupling stage is given by Equation 2. This is also

the input signal to the low-pass filter.

The 125 kHz carrier ripple voltage, without R1, is about

2V peak-to-peak and is due to the junction capacitance

and reverse leakage of D1. The addition of R1 has little

effect on the ripple voltage, but does improve the

detectors dynamic performance at the data rate. The

carrier ripple voltage will be filtered out at a later stage

where a more effective solution can be implemented.

EQUATION 2:

V = 150e^-t/τ

τ = RC

It is useful to think in terms of τ (RC time constant)

because the voltage across the resistor reduces by a

factor of 0.368 as every τ second elapses. The

exponential decay curve, for the voltage across R, is

shown in Figure 4 and indicates that the initial voltage

decays rapidly, but settles out slower as the voltage is

reduced across the resistor. The system must be

allowed to settle for a long enough period so that the

step response voltage has reduced to a voltage that is

smaller than the modulation voltage.

THE DC DECOUPLER CONFLICTS

The HV_Env signal (Figure 2) consists of three main

components:

1. A 150V DC signal, as a result of the peak

detector.

2. 2V peak-to-peak ripple voltage at the carrier

frequency.

3. The modulated data signal at a TE of 200 µs and

a 2 mV peak-to-peak amplitude, highest funda-

mental harmonic content is at 2.5 kHz [1/(2*200

uS)], irrespective of the modulation scheme

used (i.e., Manchester, PWM etc.).

The required value for RC, or τ, can be calculated using

Equation 3, based on the following assumptions:

• The system needs to be able to start LF communi-

cations 200 µs after the resonant tank has

stabilized.

The aim of the decoupling stage is to reject the high DC

voltage without adding unnecessary loading to the tank

via the peak detector. It should also have a fast dynamic

response and stabilize quickly after the tank is ener-

gized. The dynamic response of the LF Talkback system

is the major design hurdle to overcome as far as the

decoupling stage is concerned. The problem is aggra-

vated when the transponder needs to communicate on

the LF link soon after the base station communicated

with the transponder.

• The decoupler should settle to at least half the

data modulation voltage.

The base station typically uses On Off Keying (OOK)

modulation to communicate to the transponder. This

means the tank resonance is completely halted and

then started up to transfer data via the magnetic link.

The decoupling stage experiences large “step”

responses as data is transmitted to the transponder.

The tank can ramp up to its full resonant amplitude in

100 µs to 400 µs depending on the drive system used.

 2004 Microchip Technology Inc.

DS00912A-page 3

AN912

FIGURE 4:

160

140

120

100

EQUATION 3:

EQUATION 4:

tSETTLE

FC =

ln(Vo/VSETTLE)

tSETTLE = 200 µS

VO = 150V

2πτ

From a data signal conservation, or high-pass filter

point-of-view, τ should be at least 64 µs. The conflicting

τ requirement shows that a basic high-pass filter is not

sufficient as a decoupler unless either dynamic

response or data signal strength is sacrificed.

VSETTLE = 1 mV

Using Equation 3, τ was calculated to be 16.78 µs. The

question now is how will the data signal be affected by

the decoupling stage? The decoupling stage, shown in

Figure 3, is also a high-pass filter and it was calculated

that the RC time constant needs to be 16.78 µs to

satisfy the transient response requirement. The 3 dB

cutoff frequency, for a τ, of 16.78 µs is calculated as

9.48 kHz using Equation 4. This means that the

decoupling stage will only pass one quarter of the

original data signal at 2.5 kHz, which is not desirable

from a signal-to-noise ratio perspective.

DS00912A-page 4

 2004 Microchip Technology Inc.

AN912

FIGURE 6:

AN IMPROVED DECOUPLER

From the previous section, it is clear that a high-pass

filter is needed with either a controllable τ or a nonlinear

τ that is based on the voltage across the output of the

decoupler. Both approaches will be covered and the

latter solution is shown in Figure 5.

2.5V

-2.5V

HV-Env

FIGURE 5:

LP Filter

2.5V

-2.5V

HV-Env

The final part of the decoupling stage is to lower the

output impedance by adding an active buffer in the

form of an inverting amplifier that has an input resis-

tance equal to R1. The use of an inverting amplifier has

the additional advantages that it can add gain and a

single order low-pass filter to the decoupler, as shown

in Figure 7. The gain is equal to the ratio of R3/R1, and

the low pass cutoff frequency is set by R3 and C2, as

per Equation 4. The low pass cutoff frequency should

be chosen at least two decades above the main low-

pass filter, otherwise it will have an undesirable effect

on the envelope response. For single ended 5V

designs, the gain should be limited to about 10 dB to

avoid amplifier saturation due to carrier ripple and data

modulation.

LP Filter

The addition of the two diodes, shown in Figure 5,

results in a nonlinear τ with respect to voltage because

it effectively lowers the R component of τ whenever the

voltage is either above 3.1V or below -3.1V. In a practi-

cal circuit, the diodes will start conducting when the

tank is turned on and the voltage, across the resistor R,

is around 3 volts, after the tank has stabilized.

Previously, τ was calculated with an initial voltage of

150 VDC, but if the calculation is repeated with an initial

voltage of 3 VDC, then the required τ comes to 25 µs.

The RC time constant is improved by a significant

factor from 16.78 µs to 25 µs with the additional diodes,

however, it is still not in the 64 µs ball park. The diodes

have the additional advantage in that they protect the

low-pass filter from the large positive and negative

voltages that develop across the resistor during tank

transient periods.

FIGURE 7:

HV-Env

Output

The final part to solving the time constant problem is to

add an additional resistor via a switch, as shown in

Figure 6. The switch is closed to reduce τ from 64 µs to

25 µs during transient periods and opened while data is

received via the LF Talkback link.

 2004 Microchip Technology Inc.

DS00912A-page 5

AN912

The filter gain is the final aspect to specifying the

Bessel filter. Using Microchip's FilterLab^®program,

one can get the response for a unity gain – 2.5 kHz, 3d

order Bessel filter. At 125 kHz, the filter has 93 dB of

attenuation and the input ripple amplitude is 300 mV

peak-to-peak. Assuming the filter should have an

output ripple of no more then 1 mV peak-to-peak with

12 dB of headroom for noise, coupled through the

supply line, then one needs at least 62 dB of attenua-

tion. This leaves 31 dB of allowable gain from the third

order filter. For the design, a gain of 20 or 26 dB was

chosen, leaving some additional headroom for ripple

rejection. The 3d order low-pass Bessel filter is shown

in Figure 8 and has a Fc = 2.5 kHz and 26 dB of gain.

Please note that the circuit shown in Figure 8 has a

fairly high output impedance at the data rate, but the

output of the filter will be driving a high-impedance

load, and this is therefore acceptable.

THE LOW PASS FILTER STAGE

The output signal from the decoupling stage consists of

the 125 kHz carrier ripple and the modulated data

signal, if one ignores the dynamic response signal. The

carrier ripple is about 300 mV peak-to-peak. The data

is 4 mV peak-to-peak with 6 dB of gain of the decoupler

and a cutoff frequency at about 10 kHz. The aim of the

low-pass filter stage is to amplify the data signal at

2.5 kHz and to filter out the carrier ripple in the most

effective manner.

The three most common active filter topologies used

are the Chebyshev, Butterworth and Bessel filters. The

Chebyshev filter has the steepest transition from pass

band to stop band, but has ripple in the pass band. The

Butterworth filters have the flattest pass band

response, but does not have such a steep transition as

the Chebyshev. The Bessel filter has a linear phase

response with a smooth transition from pass to stop

band. It seems the Chebyshev filter would best be

suited for this application, but the frequency response

does not tell the whole story.

FIGURE 8:

The data signal is amplitude modulated and the tank

has steep transient response dynamics. As a result, the

filter should have a stable and flat transient response.

The Chebyshev filter has a very sharp frequency cutoff

response, but has the worst transient response of the

three filter topologies. The Chebyshev filter also has an

underdamped step response with overshoot and

ringing. The Butterworth filter has a better transient

response, but still some overshoot. The Bessel filter

has the worst response from a frequency perspective,

but has the best transient response as a result of its

linear phase characteristics. There are of course other

active filter topologies such as elliptical, state variable,

biquad and more, but a Bessel filter has adequate

performance for the application.

78.7k

Input

Output

150 pF

16.5k

10 nF

4.87k

3.92k

10 nF

The data signal, in this example, has maximum

modulation frequency of 2.5 kHz or a TE of 200 µs. A

Bessel filter, with a cutoff frequency of 1/(2.2TE) =

2.27 kHz, would be ideal from a noise rejection point of

view, but a 2.5 kHz cutoff was chosen to minimize sym-

bol overlap. The target is to design a filter with sufficient

performance using a single operational amplifier in

order to reduce the system cost. A dual operational

amplifier can then be used because the decoupling

stage also uses an amplifier. A third order Bessel filter

can now be implemented with the remaining amplifier.

DS00912A-page 6

 2004 Microchip Technology Inc.

AN912

DATA SLICER

AN EXAMPLE SYSTEM

The data slicer is essentially a comparator with some

input hysteresis voltage to reduce the influence of

noise. The overall system gain of the decoupler and the

low-pass filter, at 2.5 kHz, is about 29 dB or a factor of

28, and the system should be able to detect the 2 mV

data signal. The headroom between the hysteresis and

data signal was chosen to be about 9 dB or a factor of

2.8. This means that the minimum input voltage to

overcome the data slicer hysteresis is about 700 µV.

This translates to 20 mV of hysteresis for the data

slicer. Most comparators have some deliberate hyster-

esis to improve noise stability and this amount should

be extracted from the required hysteresis when calcu-

lating the amount of required feedback. Figure 9 shows

a typical hysteresis circuit and Equation 5 can be used

to calculate the amount of hysteresis for a single-ended

circuit.

A complete circuit with layout, based on the foregoing

design study, is shown in this section. The circuit

diagram is shown in Figure 11. The top and bottom

layout for the printed circuit board is shown in

Figure 12. The PIC16F648A was chosen for the

application, it has two comparators,

a USART,

EEPROM and 4k of Flash program memory. The

PIC16F648A can be substituted with its smaller

program memory equivalents, the PIC16F627A or

PIC16F628A. The filter examples have been converted

to operate from a single 5 VDC supply. The 2.5 VDC

virtual ground is provided by the voltage divider

consisting of R23 and R24, shown in Figure 11. The

Reference voltage does not have to be actively

buffered, it is lightly loaded. A 0.1 µF decoupling

capacitor C10 is sufficient for noise reduction.

A TC4422 FET driver, U1, drives the resonant tank

consisting of L1 and C2. The tank generates a strong

magnetic field and the voltage at the test pin TP1 can

reach 320V peak-to-peak. The main antenna, L1, is an

air-cored inductor with a 25 mm radius and 41 turns of

26-gage wire, and has a 162 µH inductance. The

inductor L2 and capacitors C3 and C4 are not popu-

lated and are added to the printed circuit board to test

alternative antennas. The peak detector consists of D1,

C5, R1, and R2, and is connected to the decoupling

stage via C6. The RC time constant of the decoupling

tank is set by C6 and R4 to 177 µs, which is substan-

tially longer than the minimum filter requirement of

64 µs. Resistor R3 is used to change the decoupler's

time constant to 11 µs by changing RB7 from a high-

impedance input to an output.

EQUATION 5:

VDD

VHYST =

FIGURE 9:

VREF

Output

Input

The decoupler buffer, U2:A, has a gain of 6 dB and a

low pass cutoff frequency at 9.8 kHz, set by R5 and C8.

The R22 resistor is used to ensure the proper DC bias

of the stage, but does not have a significant effect on

the overall sensitivity. The output of the decoupler is

connected to the input of the low pass Bessel filter and

one of the PIC16F648's comparators. The remaining

op amp, U2:B, is used for the Bessel filter. U2 is a dual

MCP6002 op amp that has a GBWP of 1 MHz. The

filter components should have better tolerances than

the high voltage components and 1% resistors. The 5%

NP0 capacitors are recommended.

For example, if a comparator with 10 mV of offset and

hysteresis is used, then an additional 10 mV of hyster-

esis should be added. The resistor R2 is calculated to

be 5 MΩ for a VDD of 5 VDC and R1 = 10 kΩ.

 2004 Microchip Technology Inc.

DS00912A-page 7

AN912

The PIC16F648A has various comparator options.

Figure 10 shows the topology that was chosen for this

application. The main filter output signal “ENV_IN” is

connected to comparator C1 via RA0. Resistor R10

was placed in series with the output of the filter to have

10 kΩ impedance. Together with the 4.99 MΩ resistor,

R11 adds an additional 10 mV of hysteresis. The

comparator has a combined offset and hysteresis of

10 mV, in the worst case, making for a total of 20 mV of

hysteresis, in the worst case, and about 15 mV on

average. It should be noted that the output of compar-

ator C1 has to be inverted by setting bit C1INV, in the

CMCON register. The output inversion is needed to

result in positive feedback, via R11, as is shown in

Figure 9. At first glance, it seems as if R10 can be

removed and R11 changed to a 2.43 MΩ resistor, but

the capacitor C12 will cause delay and that can lead to

instability.

There are additional aspects around the decoupler that

need to be explained for the system as it is imple-

mented. The port pin RB7 is essentially an open circuit

when it is configured as an input and the input voltage

is between VDD (5V) and ground. All the general

purpose I/O pins have internal ESD protection diodes

that become conductive when a pin voltage is forced

outside the VDD to ground range. This has the effect

that the RC time constant for the decoupling stage is

reduced to 11 µs from 177 µs whenever the “BIAS”

signal is about 0.6V above VDD, or below ground even

if RB7 is configured as an input. The addition of R3

works well, but keep in mind, the stable DC voltage for

signal “A”, shown in Figure 11, is 2.5 VDC and the signal

“BIAS” is either 5V or ground. One can implement one

of two approaches to correctly bias the signal at point

“A”.

The first solution is to toggle RB7 between high and low

with a 50% duty cycle at 20 kHz or more. This is

equivalent to connecting the “BIAS” signal to the

desired 2.5 VDC. This is only done for a short period

after the tank is turned on or off, to force the decoupler

to stabilize faster than it would with just R4. The second

approach is to force the signal “A” in the required

direction. The voltage at “A” will go above VDD if the

tank is turned on after it has been turned off for some

time. The “BIAS” signal can be grounded during the

turn-on transient period until the voltage at point “A”

reaches the desired 2.5 VDC or VREF. By monitoring

either of the comparator output signals, it is possible to

detect when the voltage at point A goes through VREF.

Pin RB7 can be turned into an input as soon as the

cross over is detected resulting in a decoupler RC time

constant of 177 µs. The filters introduce delay that

cause some overshoot of the voltage at point “A”. The

overshoot can be resolved by allowing some additional

stabilizing time with R4, before LF communication is

interpreted as data.

FIGURE 10:

Two common Reference Comparators with Outputs

CM2:CM0 = 110

VIN-

RA0/AN0

C1VOUT

VIN+

RA3/AN3/CMP1

VIN-

RA1/AN1

C2VOUT

VIN+

RA2/AN2/VREF

Open Drain

RA4/T0CKI/CMP2

DS00912A-page 8

 2004 Microchip Technology Inc.

AN912

SYSTEM MODIFICATIONS

INCREASED DATA SENSITIVITY

The system can be modified to better suit the user's

requirements. The first aspect is to change the Bessel

filter for a different LF Talkback TE. The rule of thumb is

to set the filter's 3 dB cutoff frequency to Fc = 1/(2*TE).

The new values for the Bessel filter, with a 400 µs TE,

is given in Table 1.

Increasing the system’s sensitivity to the modulated

data signal can increase the LF Talkback range. A solu-

tion has been partly described in the previous section;

increase TE from 200 µs to 400 µs and then increase

the gain by up to 18 dB. The component values for a

system with a 400 µs TE, or a center frequency of

1.25 kHz, and a gain of 100, or a 14 dB increase, is

described in Table 2 below. This approach decreases

the dynamic range that may or may not be used

depending on how well the transponder loads the

resonant tank.

TABLE 1:

R6 = 3.57k

R7 = 15.0k

R8 = 71.5k

R9 = 4.42k

R10 = 5.62k

C9 = C12 = 22 nF

C11 = 330 pF

In addition to changing the filter cutoff frequency for a

TE of 400 µs, it is possible to increase the gain up to

18 dB and still maintain the carrier rejection chosen. It

is also possible to increase C6 to a 4.7 nF capacitor,

but please note that this will increase the transient

response period. Increasing C6 will not have a

dramatic influence on the overall system performance

and it is not recommended.

TABLE 2:

R6 = 2.26k

R7 = 10.5k

R8 = 226k

R9 = 4.42k

R10 = 5.62k

C9 = 33 nF C11 = 100 pF C12 = 22 nF

Another solution is to remove resistor R11 to get the

maximum sensitivity from the comparator, but this will

also increase noise in the data. Another quick solution

is to increase the gain of the decoupler buffer by up to

10 dB and lower the decoupler cutoff frequency by

about half the gain increase ratio.

LONGER TRANSIENT STABILIZING

PERIOD

The existing design makes use of a 3d order Bessel

filter. For improved noise reduction, increase the order

of the filter and add more gain per stage. This would

typically be done if a TE of 200 µs or 100 µs, is desired

with more sensitivity than can be reliably obtained with

the example system.

The example system was designed with the require-

ment that LF Talkback communications should be able

to start 200 µS after the resonant tank has stabilized.

The tank itself takes 100 µS to 400 µs to stabilize

sufficiently, depending on the drive mechanism. The

example circuit should be able to start LF Talkback

communications with 2 mV of data modulation after

350 µs to 450 µs, from when the tank is turned. The

exact period depends on the residual charge in the

peak detector from previous transmissions.

DRIVE SYSTEM

The example circuit uses a half-bridge driver based on

the TC range of FET drivers from Microchip. To

increase the transient response period of the tank, start

the tank in Full-bridge mode until the desired tank

amplitude is reached and the tank oscillation is main-

tained in Half-bridge mode. This method is described in

AN232 Low Frequency Magnetic Transmitter Design.

The system can be simplified and improved if the

system allows for a longer transient stabilizing period

before LF Talkback communications are initiated. The

peak detector capacitor, C5, can be increased propor-

tionally to the longer stabilizing period, but not by more

than a factor of about 3, otherwise it can influence data

modulation sensitivity. R1 and R2 tank resistors should

be reduced if C5 is increased, but not proportionally, it

will effect sensitivity. The combined value for R1 and

R2 should be no less then 4 MΩ.

CONCLUSION

This LF Talkback Design application note can be used

to implement a cost-effective system to be used in

RFID, passive keyless entry and other bidirectional

transponder based technologies. The example circuit

can be used as a basis for further hardware and firm-

ware development to suit the user's requirements.

Capacitor C6 can also be increased, but it will not have

a dramatic performance increase. The biggest advan-

tage of a longer transient stabilizing period is that bias

resistor R3 can be increased. Increasing the value of

R3 will result in a slower change in the signal at point

“A”, which means the tank can be controlled more

accurately during the transient period.

 2004 Microchip Technology Inc.

DS00912A-page 9

AN912

APPENDIX A: SCHEMATICS

FIGURE 11:

LF BASE STATION

DS00912A-page 10

 2004 Microchip Technology Inc.

AN912

FIGURE 12:

LF BASE STATION (Continued)

G N D

1 5

C C V

1 6

 2004 Microchip Technology Inc.

DS00912A-page 11

AN912

FIGURE 13:

BOTTOM SIDE

05-01 XXXX REV. A BOTTOM SIDE

FIGURE 14:

TOP SIDE

05-01 XXXX REV. A TOP SIDE

DS00912A-page 12

 2004 Microchip Technology Inc.

AN912

FIGURE 15:

TOP MASK

HIGH VOLTAGE

SECTION

R12

C13

C14

VR1

C18 C19 RS232

C24

POWER

FILTER

TP7

TP6

TP5

TP1

R11

R10

R20

R21

RESET

RB0

U3 R9

C12

C11

TP4

TP3

R19

R18

TP2

05-01 XXXX REV. A TOP MASK

 2004 Microchip Technology Inc.

DS00912A-page 13

AN912

NOTES:

DS00912A-page 14

 2004 Microchip Technology Inc.

Note the following details of the code protection feature on Microchip devices:

•

Microchip products meet the specification contained in their particular Microchip Data Sheet.

•

Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the

intended manner and under normal conditions.

•

There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our

knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip's Data

Sheets. Most likely, the person doing so is engaged in theft of intellectual property.

•

Microchip is willing to work with the customer who is concerned about the integrity of their code.

Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not

mean that we are guaranteeing the product as “unbreakable.”

Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our

products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts

allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.

Information contained in this publication regarding device

applications and the like is intended through suggestion only

and may be superseded by updates. It is your responsibility to

ensure that your application meets with your specifications.

No representation or warranty is given and no liability is

assumed by Microchip Technology Incorporated with respect

to the accuracy or use of such information, or infringement of

patents or other intellectual property rights arising from such

use or otherwise. Use of Microchip’s products as critical

components in life support systems is not authorized except

with express written approval by Microchip. No licenses are

conveyed, implicitly or otherwise, under any intellectual

property rights.

Trademarks

The Microchip name and logo, the Microchip logo, Accuron,

dsPIC, KEELOQ, MPLAB, PIC, PICmicro, PICSTART,

PRO MATE and PowerSmart are registered trademarks of

Microchip Technology Incorporated in the U.S.A. and other

countries.

AmpLab, FilterLab, microID, MXDEV, MXLAB, PICMASTER,

SEEVAL, SmartShunt and The Embedded Control Solutions

Company are registered trademarks of Microchip Technology

Incorporated in the U.S.A.

Application Maestro, dsPICDEM, dsPICDEM.net,

dsPICworks, ECAN, ECONOMONITOR, FanSense,

FlexROM, fuzzyLAB, In-Circuit Serial Programming, ICSP,

ICEPIC, microPort, Migratable Memory, MPASM, MPLIB,

MPLINK, MPSIM, PICkit, PICDEM, PICDEM.net, PICtail,

PowerCal, PowerInfo, PowerMate, PowerTool, rfLAB, rfPIC,

Select Mode, SmartSensor, SmartTel and Total Endurance

are trademarks of Microchip Technology Incorporated in the

U.S.A. and other countries.

Serialized Quick Turn Programming (SQTP) is a service mark

of Microchip Technology Incorporated in the U.S.A.

All other trademarks mentioned herein are property of their

respective companies.

Printed on recycled paper.

Microchip received ISO/TS-16949:2002 quality system certification for

its worldwide headquarters, design and wafer fabrication facilities in

Chandler and Tempe, Arizona and Mountain View, California in October

2003. The Company’s quality system processes and procedures are for

its PICmicro^®8-bit MCUs, KEELOQ^®code hopping devices, Serial

EEPROMs, microperipherals, nonvolatile memory and analog

products. In addition, Microchip’s quality system for the design and

manufacture of development systems is ISO 9001:2000 certified.

 2004 Microchip Technology Inc.

DS00912A-page 15

WORLDWIDE SALES AND SERVICE

China - Beijing

Korea

AMERICAS

Corporate Office

2355 West Chandler Blvd.

Chandler, AZ 85224-6199

Tel: 480-792-7200

Fax: 480-792-7277

Technical Support: 480-792-7627

Web Address: http://www.microchip.com

Unit 706B

168-1, Youngbo Bldg. 3 Floor

Samsung-Dong, Kangnam-Ku

Seoul, Korea 135-882

Wan Tai Bei Hai Bldg.

No. 6 Chaoyangmen Bei Str.

Beijing, 100027, China

Tel: 86-10-85282100

Fax: 86-10-85282104

Tel: 82-2-554-7200 Fax: 82-2-558-5932 or

82-2-558-5934

Singapore

200 Middle Road

#07-02 Prime Centre

Singapore, 188980

Tel: 65-6334-8870 Fax: 65-6334-8850

China - Chengdu

Rm. 2401-2402, 24th Floor,

Ming Xing Financial Tower

No. 88 TIDU Street

Chengdu 610016, China

Tel: 86-28-86766200

Atlanta

3780 Mansell Road, Suite 130

Alpharetta, GA 30022

Tel: 770-640-0034

Fax: 770-640-0307

Taiwan

Kaohsiung Branch

30F - 1 No. 8

Fax: 86-28-86766599

Boston

Min Chuan 2nd Road

Kaohsiung 806, Taiwan

Tel: 886-7-536-4818

Fax: 886-7-536-4803

China - Fuzhou

Unit 28F, World Trade Plaza

No. 71 Wusi Road

Fuzhou 350001, China

Tel: 86-591-7503506

Fax: 86-591-7503521

2 Lan Drive, Suite 120

Westford, MA 01886

Tel: 978-692-3848

Fax: 978-692-3821

Taiwan

Taiwan Branch

11F-3, No. 207

Tung Hua North Road

Taipei, 105, Taiwan

Tel: 886-2-2717-7175 Fax: 886-2-2545-0139

Chicago

333 Pierce Road, Suite 180

Itasca, IL 60143

Tel: 630-285-0071

Fax: 630-285-0075

China - Hong Kong SAR

Unit 901-6, Tower 2, Metroplaza

223 Hing Fong Road

Kwai Fong, N.T., Hong Kong

Tel: 852-2401-1200

Fax: 852-2401-3431

Dallas

EUROPE

Austria

Durisolstrasse 2

A-4600 Wels

Austria

Tel: 43-7242-2244-399

Fax: 43-7242-2244-393

Denmark

Regus Business Centre

Lautrup hoj 1-3

4570 Westgrove Drive, Suite 160

Addison, TX 75001

Tel: 972-818-7423

Fax: 972-818-2924

China - Shanghai

Room 701, Bldg. B

Far East International Plaza

No. 317 Xian Xia Road

Shanghai, 200051

Detroit

Tri-Atria Office Building

32255 Northwestern Highway, Suite 190

Farmington Hills, MI 48334

Tel: 248-538-2250

Tel: 86-21-6275-5700

Fax: 86-21-6275-5060

China - Shenzhen

Rm. 1812, 18/F, Building A, United Plaza

No. 5022 Binhe Road, Futian District

Shenzhen 518033, China

Tel: 86-755-82901380

Fax: 86-755-8295-1393

China - Shunde

Fax: 248-538-2260

Ballerup DK-2750 Denmark

Tel: 45-4420-9895 Fax: 45-4420-9910

Kokomo

France

2767 S. Albright Road

Kokomo, IN 46902

Tel: 765-864-8360

Fax: 765-864-8387

Parc d’Activite du Moulin de Massy

43 Rue du Saule Trapu

Batiment A - ler Etage

91300 Massy, France

Tel: 33-1-69-53-63-20

Fax: 33-1-69-30-90-79

Room 401, Hongjian Building, No. 2

Los Angeles

18201 Von Karman, Suite 1090

Irvine, CA 92612

Tel: 949-263-1888

Fax: 949-263-1338

Fengxiangnan Road, Ronggui Town, Shunde

District, Foshan City, Guangdong 528303, China

Tel: 86-757-28395507 Fax: 86-757-28395571

Germany

China - Qingdao

Rm. B505A, Fullhope Plaza,

No. 12 Hong Kong Central Rd.

Qingdao 266071, China

Tel: 86-532-5027355 Fax: 86-532-5027205

Steinheilstrasse 10

D-85737 Ismaning, Germany

Tel: 49-89-627-144-0

Fax: 49-89-627-144-44

San Jose

1300 Terra Bella Avenue

Mountain View, CA 94043

Tel: 650-215-1444

Italy

India

Via Quasimodo, 12

20025 Legnano (MI)

Milan, Italy

Divyasree Chambers

1 Floor, Wing A (A3/A4)

No. 11, O’Shaugnessey Road

Bangalore, 560 025, India

Tel: 91-80-2290061 Fax: 91-80-2290062

Japan

Fax: 650-961-0286

Toronto

Tel: 39-0331-742611

Fax: 39-0331-466781

Netherlands

P. A. De Biesbosch 14

NL-5152 SC Drunen, Netherlands

Tel: 31-416-690399

6285 Northam Drive, Suite 108

Mississauga, Ontario L4V 1X5, Canada

Tel: 905-673-0699

Fax: 905-673-6509

Benex S-1 6F

3-18-20, Shinyokohama

Kohoku-Ku, Yokohama-shi

Kanagawa, 222-0033, Japan

Tel: 81-45-471- 6166 Fax: 81-45-471-6122

ASIA/PACIFIC

Australia

Suite 22, 41 Rawson Street

Epping 2121, NSW

Australia

Tel: 61-2-9868-6733

Fax: 61-2-9868-6755

Fax: 31-416-690340

United Kingdom

505 Eskdale Road

Winnersh Triangle

Wokingham

Berkshire, England RG41 5TU

Tel: 44-118-921-5869

Fax: 44-118-921-5820

01/26/04

DS00912A-page 16

 2004 Microchip Technology Inc.

AN912 [MICROCHIP]

相关型号：

AN91A10S

AN91A13S

AN920031

AN920032

AN920035

AN920036

AN925

AN929

AN93008/EIE

AN93013/EIE

AN939

AN93B06K