MIC2169YMM-TR [MICROCHIP]
SWITCHING CONTROLLER, 550kHz SWITCHING FREQ-MAX, PDSO10;
| 型号： | MIC2169YMM-TR |
| 厂家： | 
MICROCHIP |
| 描述： | SWITCHING CONTROLLER, 550kHz SWITCHING FREQ-MAX, PDSO10 开关 光电二极管 输出元件 |
| 文件： | 总15页 (文件大小：202K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
MIC2169
500kHz PWM Synchronous Buck Control IC
General Description
Features
TheMIC2169isahigh-efficiency,simpletouse500kHzPWM
synchronous buck control IC housed in a small MSOP-10
package. The MIC2169 allows compact DC/DC solutions
with a minimal external component count and cost.
• 3V to 14.5V input voltage range
• Adjustable output voltage down to 0.8V
• Up to 95% efficiency
• 500kHz PWM operation
• Adjustable current limit senses high-side N-Channel
MOSFET current
• No external current-sense resistor
• Adaptive gate drive increases efficiency
• Fast transient response
– Externally compensated
• Overvoltage protection protects the load in fault
conditions
The MIC2169 operates using a 3V to 14.5V input, without
the need for any additional bias voltage. The output voltage
can be precisely regulated down to 0.8V. The adaptive all
N-Channel MOSFET drive scheme allows efficiencies, over
95%, across a wide load range.
TheMIC2169sensescurrentacrossthehigh-sideN-Channel
MOSFET, eliminating the need for an expensive and lossy
current-sense resistor. Current limit accuracy is maintained
via a positive temperature coefficient that tracks the increas-
• Dual mode current limit speeds up recovery time
• Hiccup mode short-circuit protection
• Small size MSOP 10-lead package
ing R
of the external MOSFET. Additional cost and
DS(ON)
space are saved by the internal in-rush-current limiting and
digital soft-start.
Applications
The MIC2169 is available in a 10-pin MSOP package, with a
wide junction operating range of –40°C to +125°C.
• Point-of-load DC/DC conversion
• Set-top boxes
• Graphic cards
• LCD power supplies
All support documentation can be found on Micrel’s web site
at: www.micrel.com.
• Telecom power supplies
• Networking power supplies
• Cable modems and routers
Typical Application
VIN = 5V
SD103BWS
MIC2169 Efficiency
100μF
100
95
90
85
80
75
70
65
4.7μF
0.1μF
1kΩ
VDD
BST
CS
IRF7821
2.5μH
VIN
HSD
VSW
MIC2169
3.3V
COMP/EN
10kΩ
IRF7821
VIN = 5V
60
55
50
150pF
100nF
LSD
FB
150μF x 2
VOUT = 3.3V
4kΩ
GND
3.24kΩ
0
2
4
6
I
8
10 12 14 16
(A)
LOAD
MIC2169 Adjustable Output 500kHz Converter
Micrel, Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
M9999-032409
March 2009
1
MIC2169
Micrel
Ordering Information
Part Number
Pb-Free Part Number
Frequency
Junction Temp. Range
Package
MIC2169BMM
MIC2169YMM
500kHz
–40°C to +125°C
10-lead MSOP
Pin Configuration
VIN
VDD
CS
1
2
3
4
5
10 BST
9
8
7
6
HSD
VSW
LSD
COMP
FB
GND
10-Pin MSOP (MM)
Pin Description
Pin Number
Pin Name
VIN
Pin Function
Supply Voltage (Input): 3V to 14.5V.
5V Internal Linear Regulator (Output): VDD is the external MOSFET gate
1
2
VDD
drive supply voltage and an internal supply bus for the IC. When VIN is <5V,
this regulator operates in dropout mode.
3
CS
Current Sense (Input): Current-limit comparator noninverting input. The cur-
rent limit is sensed across the MOSFET during the ON time. The current can
be set by the resistor in series with the CS pin.
4
5
6
7
COMP
FB
Compensation (Input): Pin for external compensation. .
Feedback (Input): Input to error amplifier. Regulates error amplifier to 0.8V.
Ground (Return).
GND
LSD
Low-Side Drive (Output): High-current driver output for external synchro-
nous MOSFET.
8
9
VSW
HSD
Switch (Return): High-side MOSFET driver return.
High-Side Drive (Output): High-current output-driver for the high-side MOS-
FET. When VIN is between 3.0V to 5V, 2.5V threshold MOSFETs should be
used. At VIN > 5V, 5V threshold MOSFETs should be used.
10
BST
Boost (Input): Provides the drive voltage for the high-side MOSFET driver.
The gate-drive voltage is higher than the source voltage by VIN minus a
diode drop.
M9999-032409
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March 2009
MIC2169
Micrel
Absolute Maximum Ratings(1)
Operating Ratings(2)
Supply Voltage (V )...................................................15.5V
Supply Voltage (V )..................................... +3V to +14.5V
IN
IN
Booststrapped Voltage (V
).................................V +5V
Output Voltage Range ..........................0.8V to V × D
BST
IN
IN
MAX
Junction Temperature (T )..................–40°C ≤ T ≤ +125°C
Package Thermal Resistance
10-lead MSOP.............................................180°C/W
J
J
Storage Temperature (T ) ........................ –65°C to +150°C
θ
JA
S
Electrical Characteristics(3)
TJ = 25°C, VIN = 5V; bold values indicate –40°C < TJ < +125°C; unless otherwise specified.
Parameter
Condition
(± 1%)
Min
Typ
0.8
0.8
30
Max
0.808
0.816
100
Units
V
Feedback Voltage Reference
Feedback Voltage Reference
Feedback Bias Current
Output Voltage Line Regulation
Output Voltage Load Regulation
Output Voltage Total Regulation
Oscillator Section
0.792
0.784
(± 2% over temp)
V
nA
% / V
%
0.03
0.5
0.6
3V ≤ VIN ≤ 14.5V; 1A ≤ IOUT ≤ 10A; (VOUT = 2.5V)(4)
%
Oscillator Frequency
450
92
500
30
550
kHz
%
Maximum Duty Cycle
Minimum On-Time(4)
60
ns
Input and VDD Supply
PWM Mode Supply Current
VCS = VIN –0.25V; VFB = 0.7V (output switching but excluding
external MOSFET gate current.)
1.5
5
3
mA
V
Digital Supply Voltage (VDD
Error Amplifier
)
VIN ≥ 6V
4.7
5.3
DC Gain
70
1
dB
ms
Transconductance
Soft-Start
Soft-Start Current
Current Sense
After timeout of internal timer. See “Soft-Start” section.
8.5
μA
μA
CS Over Current Trip Point
VCS = VIN –0.25V
160
200
240
Temperature Coefficient
+1800
ppm/°C
Output Fault Correction Thresholds
Upper Threshold, VFB_OVT
Lower Threshold, VFB_UVT
(relative to VFB
)
)
+3
–3
%
%
(relative to VFB
Notes:
1. Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating
the device outside of its operating ratings. The maximum allowable power dissipation is a function of the maximum junction temperature, T (max),
J
the junction-to-ambient thermal resistance, θ , and the ambient temperature, T . The maximum allowable power dissipation will result in excessive
JA
A
die temperature, and the regulator will go into thermal shutdown.
2. Devices are ESD sensitive, handling precautions required.
3. Specification for packaged product only.
4. Guaranteed by design.
March 2009
3
M9999-032409
MIC2169
Micrel
Electrical Characteristics(5)
Parameter
Condition
Min
Typ
Max
Units
Gate Drivers
Rise/Fall Time
Into 3000pF at VIN > 5V
Source, VIN = 5V
Sink, VIN = 5V
Source, VIN = 3V
Sink, VIN = 3V
Note 6
30
ns
Ω
Output Driver Impedance
6
6
Ω
10
10
Ω
Ω
Driver Non-Overlap Time
10
20
ns
Notes:
5. Specification for packaged product only.
6. Guaranteed by design.
M9999-032409
4
March 2009
MIC2169
Micrel
Typical Characteristics
V
= 5V
IN
PWM Mode Supply Current
vs. Supply Voltage
PWM Mode Supply Current
vs. Temperature
V
Line Regulation
FB
0.8010
0.8005
0.8000
0.7995
0.7990
0.7985
0.7980
2.0
1.5
1.0
0.5
2.9
2.7
2.5
2.3
2.1
1.9
1.7
1.5
1.3
1.1
0.9
0.7
0.5
0
0
0
5
10
15
0
5
10
15
-40 -20
0 20 40 60 80 100120140
V
(V)
TEMPERATURE (°C)
SUPPLY VOLTAGE (V)
IN
V
vs. Temperature
FB
V
Line Regulation
V
Load Regulation
DD
DD
5.02
5.00
4.98
4.96
4.94
4.92
4.90
0.806
0.804
0.802
0.800
0.798
0.796
0.794
0.792
6
5
4
3
2
1
0
5
10 15 20 25 30
-60 -30
0
30 60 90 120 150
0
5
10
15
LOAD CURRENT (mA)
TEMPERATURE (°C)
V
(V)
IN
V
Line Regulation
Oscillator Frequency
vs. Temperature
DD
Oscillator Frequency
vs. Supply Voltage
vs. Temperature
5.0
550
540
530
520
510
500
490
480
470
460
450
1.5
1.0
0.5
0
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0.0
-0.5
-1.0
-1.5
-60 -30
0
30 60 90 120 150
-60 -30
0
30 60 90 120 150
5
10
15
TEMPERATURE (°C)
TEMPERATURE (°C)
V
(V)
IN
Overcurrent Trip Point
vs. Temperature
Current Limit Foldback
4
3
2
1
0
260
240
220
200
180
160
140
120
100
Top MOSFET = Si4800
RCS = 1kΩ
0
2
4
6
8
10
-60 -30
0
30 60 90 120 150
I
(A)
TEMPERATURE (°C)
LOAD
March 2009
5
M9999-032409
MIC2169
Micrel
Functional Diagram
CIN
RCS
CS
VIN
VDD
D1
Current Limit
Comparator
VDD
5V
5V LDO
HSD
High-Side
Driver
Q1
5V
BOOST
Current Limit
Reference
0.8V
BG Valid
Bandgap
Reference
CBST
2Ω
RSW
VOUT
L1
SW
Driver
Logic
1.4Ω
1000pF
5V
COUT
Clamp &
Startup
Current
5V
Soft-Start &
Digital Delay
Counter
LSD
Low-Side
Driver
Q2
Ramp
Clock
PWM
Comparator
Enable
Error
Loop
0.8V
FB
VREF +3%
VREF 3%
Hys
Comparator
Error
Amp
R3
R2
MIC2169
COMP
GND
C1
R1
C2
MIC2169 Block Diagram
voltage. This causes the output voltage of the error amplifier
Functional Description
to go high. This will also increase the PWM comparator t
ON
The MIC2169 is a voltage mode, synchronous step-down
switchingregulatorcontrollerdesignedforhighpowerwithout
the use of an external sense resistor. It includes an internal
soft-start function (which reduces the power supply input
surge current at start-up by controlling the output voltage rise
time), a PWM generator, a reference voltage, two MOSFET
drivers, and short-circuit current limiting circuitry to form a
complete 500kHz switching regulator.
time of the top side MOSFET, causing the output voltage to
go up and bringing V
back in regulation.
OUT
Soft-Start
The COMP pin on the MIC2169 is used for the following two
functions:
1. External compensation to stabilize the voltage
control loop.
Theory of Operation
2. Soft-start.
The MIC2169 is a voltage mode step-down regulator. The
blockdiagram,above,illustratesthevoltagecontrolloop.The
output voltage variation due, to load or line changes, will be
sensed by the inverting input of the transconductance error
amplifierviathefeedbackresistorsR3,andR2andcompared
toareferencevoltageatthenon-invertinginput.Thiswillcause
a small change in the DC voltage level at the output of the
error amplifier which is the input to the PWM comparator. The
otherinputtothecomparatorisa5Vtriangularwaveform.The
comparator generates a rectangular waveform whose width
Forbetterunderstandingofthesoft-startfeature, assumeV
IN
= 12V. The COMP pin has an internal 6.5μA current source
thatchargestheexternalcompensationcapacitor.Assoonas
this voltage rises to 180mV (t = Cap_COMP × 0.18V/8.5μA),
theMIC2169allowstheinternalV linearregulatortopower
DD
upandassoonasitcrossestheundervoltagelockoutof2.6V,
the chip’s internal oscillator starts switching.At this point, the
COMP pin current source increases to 40μA and an internal
11-bit counter starts counting. This takes approximately 2ms
to complete. During counting, the COMP voltage is clamped
at 0.65V. After this counting cycle, the COMP current source
is reduced to 8.5μA and the COMP pin voltage rises from
0.65V to 0.95V, the bottom edge of the saw-tooth oscillator.
This is the beginning of 0% duty cycle which increases slowly
causing the output voltage to rise slowly. The MIC2169 has
t
is equal to the time from the start of the clock cycle t until
ON
0
t , the time the triangle crosses the output waveform of the
1
erroramplifier.Toillustratethecontrolloop,assumetheoutput
voltage drops due to sudden load turn-on, this would cause
the inverting input of the error amplifier which is a divided
down version of V
to be slightly less than the reference
OUT
M9999-032409
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March 2009
MIC2169
Micrel
two hysteretic comparators that are enabled when V
is
where:
OUT
within ±3% of steady state. When the output voltage reaches
Inductor Ripple Current =
97%ofprogrammedoutputvoltage,thentheg erroramplifier
is enabled along with the hysteretic comparator. From this
m
V
–V
(
)
IN
OUT
V
×
OUT
V
×F
× L
= 500kHz
IN
SWITCHING
point onwards, the voltage control loop (g error amplifier) is
m
F
fully in control and will regulate the output voltage.
SWITCHING
200μA is the internal sink current to program the MIC2169
current limit.
Soft-start time can be calculated approximately by adding
the following four time frames:
The MOSFET R
therefore, it is recommended that a 50% margin be added
to the load current (I ) in the above equation to avoid
false current limiting due to increased MOSFET junction
temperature rise. It is also recommended to connect the
varies 30% to 40% with temperature;
t1 = Cap_COMP × 0.18V/8.5μA
t2 = 12 bit counter, approx 2ms
t3 = Cap_COMP × 0.3V/8.5μA
DS(ON)
LOAD
V
Cap_COMP
⎛
⎞
OUT
t4 =
× 0.5 ×
⎜
⎝
⎟
⎠
R
resistor directly to the drain of the top MOSFET Q1,
V
8.5μA
CS
IN
and the R resistor to the source of Q1 to accurately sense
SW
Soft-Start Time(Cap_COMP=100nF) = t1 + t2 + t3 +
t4 = 2.1ms + 2ms + 3.5ms + 1.8ms = 10ms
the MOSFETs R
. To make the MIC2169 insensitive to
DS(ON)
board layout and noise generated by the switch node. For
this a 1.4Ω resistor and a 1000pF capacitor is recommended
between the switch node and ground. A 0.1μF capacitor, in
Current Limit
The MIC2169 uses the R
of the top power MOSFET
DS(ON)
parallel with R , should be connected to filter some of the
CS
to measure output current. Since it uses the drain to source
resistance of the power MOSFET, it is not very accurate.
However,thisschemeisadequatetoprotectthepowersupply
and external components during a fault condition by cutting
back the time the top MOSFET is on if the feedback voltage
is greater than 0.67V. In case of a hard short when feedback
voltageislessthan0.67V,theMIC2169dischargestheCOMP
capacitorto0.65V,resetsthedigitalcounterandautomatically
switching noise.
Internal V Supply
DD
TheMIC2169controllerinternallygeneratesV forselfbias-
ing and to provide power to the gate drives. This V supply
is generated through a low-dropout regulator and generates
DD
DD
5V from V supply greater than 5V. For supply voltage less
IN
than 5V, the V linear regulator is approximately 200mV in
DD
shuts off the top gate drive, and the g error amplifier and the
dropout.Therefore,itisrecommendedtoshorttheV supply
m
DD
–3% hysteretic comparators are completely disabled and the
soft-start cycles restarts. This mode of operation is called the
“hiccup mode” and its purpose is to protect the down stream
load in case of a hard short. The circuit in Figure 1 illustrates
the MIC2169 current limiting circuit.
to the input supply through a 5Ω resistor for input supplies
between 2.9V to 5V.
MOSFET Gate Drive
The MIC2169 high-side drive circuit is designed to switch an
N-Channel MOSFET. The block diagram on page 6 shows a
bootstrapcircuit,consistingofD1andCBST.Itsuppliesenergy
tothehigh-sidedrivecircuit.CapacitorCBSTischargedwhile
the low-side MOSFET is on and the voltage on the VSW pin
is approximately 0V. When the high-side MOSFET driver is
turned on, energy from CBST is used to turn the MOSFET
on. As the MOSFET turns on, the voltage on the VSW pin
VIN
C2
CIN
HSD
LSD
Q1
MOSFET N
0.1μF
2Ω
VOUT
L1 Inductor
1.4Ω
RCS
CS
C1
COUT
Q2
MOSFET N
1000pF
increases to approximately V . Diode D1 is reversed biased
IN
and CBST floats high while continuing to keep the high-side
MOSFET on. When the low-side switch is turned back on,
CBST is recharged through D1. The drive voltage is derived
200μA
from the internal 5V V bias supply. The nominal low-side
DD
gate drive voltage is 5V and the nominal high-side gate drive
voltageisapproximately4.5VduethevoltagedropacrossD1.
An approximate 20ns delay between the high- and low-side
driver transition is used to prevent current from simultane-
ously flowing unimpeded through both MOSFETs.
Figure 1. The MIC2169 Current Limiting Circuit
The current limiting resistor R is calculated by the follow-
CS
ing equation:
MOSFET Selection
RDS(ON) Q1 × IL
The MIC2169 controller works from input voltages of 3V to
13.2V and has an internal 5V regulator to provide power to
turn the external N-Channel power MOSFETs for high- and
RCS
=
200μA
Equation (1)
1
low-sideswitches.ForapplicationswhereV <5V,theinternal
I =I
+
IN
L
LOAD
2 Inductor Ripple Current
(
)
V
regulator operates in dropout mode, and it is necessary
DD
that the power MOSFETs used are sub-logic level and are in
full conduction mode for V of 2.5V. For applications when
GS
V > 5V; logic-level MOSFETs, whose operation is specified
IN
March 2009
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M9999-032409
MIC2169
Micrel
at V = 4.5V must be used.
where:
GS
Itisimportanttonotetheon-resistanceofaMOSFETincreases
with rising temperature. A 75°C rise in junction temperature
will increase the channel resistance of the MOSFET by 50%
to 75% of the resistance specified at 25°C. This change in
resistance must be accounted for when calculating MOSFET
powerdissipationandincalculatingthevalueofcurrent-sense
(CS) resistor. Total gate charge is the charge required to turn
the MOSFET on and off under specified operating conditions
(V and V ). The gate charge is supplied by the MIC2169
gate-drive circuit.At 500kHz switching frequency and above,
the gate charge can be a significant source of power dissipa-
tionintheMIC2169.Atlowoutputload, thispowerdissipation
is noticeable as a reduction in efficiency.The average current
required to drive the high-side MOSFET is:
P
=I
×R
SW(rms)2
SW
CONDUCTION
P
= P
+P
AC
AC(off) AC(on)
R
= on-resistance of the MOSFET switch
SW
⎛
⎞
V
O
D = duty cycle
⎜
⎟
V
⎝
⎠
IN
Makingtheassumptiontheturn-onandturn-offtransitiontimes
are equal; the transition times can be approximated by:
DS
GS
C
× V + C
× V
OSS IN
ISS
GS
t
=
T
I
G
where:
I
= Q × f
G S
G[high-side](avg)
C
and C
are measured at V = 0
OSS DS
ISS
I = gate-drive current (1A for the MIC2169)
G
where:
I
The total high-side MOSFET switching loss is:
= average high-side MOSFET gate
G[high-side](avg)
P
=(V +V )× I × t × f
IN D PK T S
current.
AC
Q = total gate charge for the high-side MOSFET taken from
G
where:
manufacturer’s data sheet for V = 5V.
GS
t = switching transition time (typically 20ns to 50ns)
T
Thelow-sideMOSFETisturnedonandoffatV =0because
DS
V = freewheeling diode drop, typically 0.5V
the freewheeling diode is conducting during this time. The
switching loss for the low-side MOSFET is usually negligible.
Also, the gate-drive current for the low-side MOSFET is
D
f it the switching frequency, nominally 500kHz
S
The low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
more accurately calculated using CISS at V = 0 instead
DS
of gate charge.
Inductor Selection
For the low-side MOSFET:
Values for inductance, peak, and RMS currents are required
to select the output inductor. The input and output voltages
and the inductance value determine the peak-to-peak induc-
tor ripple current. Generally, higher inductance values are
used with higher input voltages. Larger peak-to-peak ripple
currents will increase the power dissipation in the inductor
and MOSFETs. Larger output ripple currents will also require
more output capacitance to smooth out the larger ripple cur-
rent. Smaller peak-to-peak ripple currents require a larger
inductance value and therefore, a larger and more expensive
inductor. A good compromise between size, loss and cost is
to set the inductor ripple current to be equal to 20% of the
maximum output current. The inductance value is calculated
by the equation below.
I
= C
× V × f
G[low-side](avg)
ISS GS S
Since the current from the gate drive comes from the input
voltage, the power dissipated in the MIC2169, due to gate
drive, is:
P
= V
I
(
+ I
G[low-side](avg)
)
GATEDRIVE
IN G[high-side](avg)
Aconvenient figure of merit for switching MOSFETs is the on
resistance times the total gate charge R
numbers translate into higher efficiency. Low gate-charge
logic-level MOSFETs are a good choice for use with the
MIC2169.
×Q . Lower
DS(ON)
G
Parameters that are important to MOSFET switch selection
are:
VOUT ×(VIN(max) − VOUT
)
L =
V (max) × fS × 0.2 ×IOUT(max)
IN
• Voltage rating
• On-resistance
• Total gate charge
where:
f = switching frequency, 500kHz
S
The voltage ratings for the top and bottom MOSFET are
essentially equal to the input voltage. A safety factor of 20%
0.2 = ratio of AC ripple current to DC output current
V (max) = maximum input voltage
IN
shouldbeaddedtotheV (max)oftheMOSFETstoaccount
DS
The peak-to-peak inductor current (AC ripple current) is:
for voltage spikes due to circuit parasitics.
VOUT ×(V (max) − VOUT
)
IN
The power dissipated in the switching transistor is the sum
IPP
=
of the conduction losses during the on-time (P
)
V (max) × fS ×L
IN
CONDUCTION
and the switching losses that occur during the period of time
Thepeakinductorcurrentisequaltotheaverageoutputcurrent
plus one half of the peak-to-peak inductor ripple current.
when the MOSFETs turn on and off (P ).
AC
P
= P
+P
SW
CONDUCTION AC
M9999-032409
8
March 2009
MIC2169
IPK = IOUT(max) + 0.5 ×IPP
Micrel
I
= peak-to-peak inductor ripple current
PP
The total output ripple is a combination of the ESR output
capacitance. The total ripple is calculated below:
2
The RMS inductor current is used to calculate the I × R
losses in the inductor.
2
2
2
⎛I × (1− D)⎞
PP
ΔV
=
+ I × R
1⎛
IP
⎞
(
)
⎜
⎝
⎟
⎠
OUT
PP
ESR
I
= IOUT(max) × 1+
C
× f
S
INDUCTOR(rms)
⎜
⎟
OUT
3 I (max)
⎝
⎠
OUT
where:
D = duty cycle
= output capacitance value
Maximizing efficiency requires the proper selection of core
material and minimizing the winding resistance. The high
frequency operation of the MIC2169 requires the use of fer-
rite materials for all but the most cost sensitive applications.
Lower cost iron powder cores may be used but the increase
in core loss will reduce the efficiency of the power supply.
Thisisespeciallynoticeableatlowoutputpower.Thewinding
resistance decreases efficiency at the higher output current
levels. The winding resistance must be minimized although
this usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of the
core and copper losses. At higher output loads, the core
losses are usually insignificant and can be ignored. At lower
output currents, the core losses can be a significant con-
tributor. Core loss information is usually available from the
magnetics vendor. Copper loss in the inductor is calculated
by the equation below:
C
OUT
f = switching frequency
S
Thevoltageratingofcapacitorshouldbetwicethevoltagefor
a tantalum and 20% greater for an aluminum electrolytic.
The output capacitor RMS current is calculated below:
I
PP
I
=
COUT(rms)
12
The power dissipated in the output capacitor is:
=I ×R
P
DISS(COUT
)
COUT(rms)
ESR(COUT )
2
Input Capacitor Selection
Theinputcapacitorshouldbeselectedforripplecurrentrating
and voltage rating. Tantalum input capacitors may fail when
subjected to high inrush currents, caused by turning the input
supply on. To maximize reliability, tantalum input capacitor
voltage rating should be at least two times the maximum in-
put voltage. Aluminum electrolytic, OS-CON, and multilayer
polymerfilmcapacitorscanhandlethehigherinrushcurrents
without voltage derating. The input voltage ripple will primar-
ily depends upon the input capacitor’s ESR. The peak input
current is equal to the peak inductor current, so:
P
= I ×R
INDUCTOR(rms)2
WINDING
INDUCTORCu
The resistance of the copper wire, R
, increases with
WINDING
temperature.Thevalueofthewindingresistanceusedshould
be at the operating temperature:
RWINDING(hot) = RWINDING(20°C) × 1+ 0.0042 × (T
− T20°C
)
)
(
HOT
where:
ΔV =I
×R
ESR(CIN
IN
INDUCTOR(peak)
)
T
= temperature of the wire under operating load
= ambient temperature
HOT
T
The input capacitor must be rated for the input current ripple.
The RMS value of input capacitor current is determined at
the maximum output current. Assuming the peak-to-peak
inductor ripple current is low:
20°C
R
isroomtemperaturewindingresistance(usu-
WINDING(20°C)
ally specified by the manufacturer)
Output Capacitor Selection
Theoutputcapacitorvaluesareusuallydeterminedcapacitors
ESR(equivalentseriesresistance).VoltageandRMScurrent
capability are two other important factors to consider when
selectingtheoutputcapacitor. Recommendedcapacitorsare
tantalum, low-ESR aluminum electrolytics, and POSCAPS.
The output capacitor’s ESR is usually the main cause of
output ripple. The output capacitor ESR also affects the
overallvoltagefeedbackloopfromstabilitypointofview.See:
“FeedbackLoopCompensation” sectionformoreinformation.
The maximum value of ESR is calculated:
IC (rms)≈ IOUT(max) × D× (1−D)
IN
The power dissipated in the input capacitor is:
P
= I ×R
CIN(rms)2
ESR(CIN
DISS(CIN
)
)
Voltage Setting Components
The MIC2169 requires two resistors to set the output voltage
as shown in Figure 2.
ΔV
OUT
R
≤
ESR
I
PP
where:
V
= peak-to-peak output voltage ripple
OUT
March 2009
9
M9999-032409
MIC2169
Micrel
lost in the diode is proportional to the forward voltage drop
of the diode. As the high-side MOSFET starts to turn on, the
body diode becomes a short circuit for the reverse recovery
period, dissipating additional power. The diode recovery and
the circuit inductance will cause ringing during the high-side
MOSFET turn-on. An external Schottky diode conducts at
a lower forward voltage preventing the body diode in the
MOSFET from turning on. The lower forward voltage drop
dissipates less power than the body diode. The lack of a
reverse recovery mechanism in a Schottky diode causes
less ringing and less power loss. Depending upon the circuit
components and operating conditions, an external Schottky
R1
R2
Error
Amp
FB
7
VREF
0.8V
MIC2169 [adj.]
Figure 2. Voltage-Divider Configuration
Where:
V
1
diode will give a / % to 1% improvement in efficiency.
2
for the MIC2169 is typically 0.8V
REF
Feedback Loop Compensation
The output voltage is determined by the equation:
The MIC2169 controller comes with an internal transcon-
ductance error amplifier used for compensating the voltage
feedback loop by placing a capacitor (C1) in series with a
resistor (R1) and another capacitor C2 in parallel from the
COMP pin-to-ground. See “Functional Block Diagram.”
R1
R2
⎛
⎞
V
= V
× 1+
O
REF
⎜
⎟
⎝
⎠
A typical value of R1 can be between 3kΩ and 10kΩ. If R1 is
too large, it may allow noise to be introduced into the voltage
feedback loop. If R1 is too small, in value, it will decrease the
efficiency of the power supply, especially at light loads. Once
R1 is selected, R2 can be calculated using:
Power Stage
The power stage of a voltage mode controller has an induc-
tor, L1, with its winding resistance (DCR) connected to the
output capacitor, C
(ESR) as shown in Figure 3. The transfer function G(s), for
such a system is:
, with its electrical series resistance
OUT
V
× R1
REF
R2 =
V − V
O
REF
L
DCR
External Schottky Diode
VO
An external freewheeling diode is used to keep the inductor
current flow continuous while both MOSFETs are turned off.
This dead time prevents current from flowing unimpeded
through both MOSFETs and is typically 15ns. The diode
conducts twice during each switching cycle. Although the
average current through this diode is small, the diode must
be able to handle the peak current.
ESR
COUT
Figure 3. The Output LC Filter in a Voltage Mode
Buck Converter
I
= I
× 2 × 15ns × f
D(avg)
OUT S
⎛
1+ ESR × s × C
⎞
⎟
(
)
G(s) =
⎜
The reverse voltage requirement of the diode is:
= V
2
DCR × s × C+ s × L× C + 1+ESR × s × C
⎝
⎠
V
DIODE(rrm)
IN
Plottingthistransferfunctionwiththefollowingassumedvalues
(L=2 μH, DCR=0.009Ω, C =1000μF, ESR=0.025Ω) gives
lot of insight as to why one needs to compensate the loop by
adding resistor and capacitors on the COMP pin. Figures 4
and 5 show the gain curve and phase curve for the above
transfer function.
The power dissipated by the Schottky diode is:
= I × V
OUT
P
DIODE
D(avg)
F
where:
V = forward voltage at the peak diode current
F
30
30
The external Schottky diode, D1, is not necessary for circuit
operation since the low-side MOSFET contains a parasitic
body diode. The external diode will improve efficiency and
decrease high frequency noise. If the MOSFET body diode
is used, it must be rated to handle the peak and average cur-
rent. The body diode has a relatively slow reverse recovery
time and a relatively high forward voltage drop. The power
7.5
-15
-37.5
-80
-80
3
4
5
6
.
.
.
.
100
100
1 10
1 10
f
1 10
1 10
1000000
Figure 4. The Gain Curve for G(s)
March 2009
M9999-032409
10
MIC2169
Micrel
g
Error Amplifier
m
It is undesirable to have high error amplifier gain at high
frequencies because high frequency noise spikes would be
picked up and transmitted at large amplitude to the output,
thus,gainshouldbepermittedtofalloffathighfrequencies.At
low frequency, it is desireable to have high open-loop gain to
attenuate the power line ripple. Thus, the error amplifier gain
should be allowed to increase rapidly at low frequencies.
The transfer function with R1, C1, and C2 for the internal
g
error amplifier can be approximated by the following
m
equation:
⎡
⎢
⎢
⎢
⎤
⎥
⎥
⎥
1+ R1 × S× C1
C1× C2 × S
Error Amplifier(z) = gm
×
Figure 5. Phase Curve for G(s)
⎛
⎞
⎟
⎠
s × C1+C2 1+R1×
(
)
⎜
⎢
⎣
⎥
⎦
⎝
C1+ C2
It can be seen from the transfer function G(s) and the gain
curve that the output inductor and capacitor create a two pole
system with a break frequency at:
The above equation can be simplified by assuming
C2<<C1,
1
f
=
⎡
⎢
⎣
⎤
⎥
⎦
1+ R1 × S× C1
LC
2 × π L × C
Error Amplifier(z) = g ×
OUT
m
s × C1 1+R1× C2 × S
(
)(
)
Therefore, f = 3.6kHz
LC
From the above transfer function, one can see that R1 and
C1 introduce a zero and R1 and C2 a pole at the following
frequencies:
By looking at the phase curve, it can be seen that the output
capacitorESR(0.025Ω)cancelsoneofthetwopoles(LC
system by introducing a zero at:
)
OUT
1
Fzero= / π × R1 × C1
2
1
1
f
=
Fpole = / π × C2 × R1
ZERO
2
2 × π ×ESR × C
OUT
1
Fpole@origin = / π × C1
2
Therefore, F
= 6.36kHz.
ZERO
Figures7and8showthegainandphasecurvesfortheabove
transfer function with R1 = 9.3k, C1 = 1000pF, C2 = 100pF,
and g = .005Ω . It can be seen that at 50kHz, the error
From the point of view of compensating the voltage loop, it
is recommended to use higher ESR output capacitors since
they provide a 90° phase gain in the power path. For com-
parison purposes, Figure 6 shows the same phase curve
with an ESR value of 0.002Ω.
–1
m
amplifier exhibits approximately 45° of phase margin.
Figure 7. Error Amplifier Gain Curve
Figure 6. The Phase Curve with ESR = 0.002Ω
It can be seen from Figure 5 that at 50kHz, the phase is
approximately –90° versus Figure 6 where the number is
–150°. This means that the transconductance error ampli-
fier has to provide a phase boost of about 45° to achieve a
closed-loop phase margin of 45° at a crossover frequency
of 50kHz for Figure 4, versus 105° for Figure 6. The simple
RC and C2 compensation scheme allows a maximum error
amplifier phase boost of about 90°. Therefore, it is easier to
stabilize the MIC2169 voltage control loop by using high ESR
value output capacitors.
March 2009
11
M9999-032409
MIC2169
Micrel
100
71.607
50
0
42.933
50
3
4
5
6
.
.
.
.
100
100
1 10
1 10
f
1 10
1 10
1000000
Figure 9. Open-Loop Gain Margin
Figure 8. Error Amplifier Phase Curve
Total Open-Loop Response
250
269.097
The open-loop response for the MIC2169 controller is easily
obtained by adding the power path and the error amplifier
gains together, since they already are in Log scale. It is
desirable to have the gain curve intersect zero dB at tens of
kilohertz, this is commonly called crossover frequency; the
phase margin at crossover frequency should be at least 45°.
Phase margins of 30° or less cause the power supply to have
substantial ringing when subjected to transients, and have
little tolerance for component or environmental variations.
Figures9and10showtheopen-loopgainandphasemargin.
It can be seen from Figure 9 that the gain curve intersects
the 0dB at approximately 50kHz, and from Figure 10, that at
50kHz, the phase shows approximately 50° of margin.
300
350
360
3
4
5
6
.
.
.
.
10
10
100
110
1 10
1 10
1 10
f
1000000
Figure 10. Open-Loop Phase Margin
M9999-032409
12
March 2009
MIC2169
Micrel
Design Example
Layout and Checklist:
7. Low gate charge MOSFETs should be used to
1. Connect the current limiting (R2) resistor directly
to the drain of top MOSFET Q3.
maximize efficiency, such as Si4800, Si4804BDY,
IRF7821, IRF8910, FDS6680A and FDS6912A,
etc.
2. Use a 5Ω resistor from the input supply to the VIN
pin on the MIC2169. Also, place a 1μF ceramic
capacitor from this pin to GND, preferably not thru
a via.
8. Compensation component GND, feedback resistor
ground, chip ground should all run together and
connect to the output capacitor ground. See demo
board layout, top layer.
3. The feedback resistors R3 and R4/R5/R6 should
be placed close to the FB pin. The top side of R3
should connect directly to the output node. Run
this trace away from the switch node (junction of
Q3, Q2, and L1). The bottom side of R3 should
connect to the GND pin on the MIC2169.
9. The 10μF ceramic capacitor should be placed
between the drain of the top MOSFET and the
source of the bottom MOSFET.
10.The10μFceramiccapacitorshouldbeplacedright
on the VDD pin without any vias.
4. The compensation resistor and capacitors should
be placed right next to the COMPpin and the other
side should connect directly to the GND pin on the
MIC2169 rather than going to the plane.
11.ThesourceofthebottomMOSFETshouldconnect
directly to the input capacitor GND with a thick
trace. The output capacitor and the input capacitor
should connect directly to the GND plane.
5. Add a 1.4Ω resistor and a 1000pF capacitor from
theswitchnodetogroundpin. Seepage7, Current
Limiting section for more detail.
12.Placea0.01μFto0.1μFceramiccapacitorinparallel
with the CS resistor to filter any switching noise.
6. Add place holders for gate resistors on the top and
bottom MOSFET gate drives. If necessary, gate
resistors of 10Ω or less should be used.
J1
+Vin 5-12V
Cin=AVX TPSD686M020R0070
+VIN
1
C2
+
C3
68uF
20V
+
C1
10uF/16V
R9
10
R2
470 ohm
C4
10uF/6V
68uF/20V
D1
SD103BWS
C16
0.1uF
C5
0.1uF/25V
4
C13
1uF/16V
Q3
IRF7821
10
BST
L1
Cout=AVX TPSD337M006R0045
CDRH127 / LD-1R0-MC
1.0uH
J4
2
1
1
R11
+
+
+
9
8
HSD
C7
330uF
C6
Vout
C8
Open
RES
R10
330uF/6.3V
VSW
R3
10K
R14
Open
U1
4R02 Ohm
R12
47
D2
MIC2169
C12
0.1uF/25V
C14
DIN
Q2
IRF7821
R13
RES
C11
Open
7
5
4
LSD
FB
C15
100pF
4
COMP/EN
Q1
2N7002E
J2
SHDN
C9
C10
R4
R5
R6
R1
3.16k
4.64K 11.3K
Open 0.1uF
1
1
0 Ohm
R7
100K
R8
4.02K
C
3.3V
B A
2.5V
1.5V
J3
1
JP2
HEADER 3X2
GND
J5
1
GND
MIC2169BMM Evaluation Board Schematic
March 2009
13
M9999-032409
MIC2169
Micrel
MIC2169BMM Bill of Materials
Item
Part Number
Manufacturer
Micrel, Inc.
IR
Description
Qty.
1
2
0
1
1
0
0
1
0
0
1
2
0
1
0
3
2
0
0
1
0
0
0
1
1
1
1
1
1
1
1
2
1
1
0
4
U1
MIC2169-YMM
IRF7821-TR
Buck controller
Q2, Q3
30V, N channel HEXFET , Power MOSFET
OR
SI4390DY
Vishay
D1
D2
SD103BWS
Vishay
30V , Schottky Diode
40V , Schottky Diode
OR
1N5819HW
Diodes Inc.
Vishay
SL04
CMMSH1-40
Central Semi
Sumida
OR
L1
CDRH127LDNP-1R0NC
HC5-1R0
1.0uH, 10A inductor
OR
Cooper Electronic
Coilcraft
SER1360-1R0
OR
C1
C3225X7R1C106M
TPSD686M020R0070
594D686X0020D2T
C2012X5R0J106M
CM21X5R106M06AT
VJ1206Y104KXXAT
TPSD337M006R0045
594D337X06R3D2T
TDK
10uF/16V, X7R Ceramic cap.
68uF, 20V Tantalum
OR
C2 , C3.
AVX
Vishay/Sprague
TDK
C4
10uF/6.3V, 0805 Ceramic cap.
OR
AVX
C5, C10 , C12
C6, C7
C8
Vishay Victramon
AVX
0.1uF/25V Ceramic cap.
330uF, 6.3V, Tantalum
Open
Vishay/Sprague
Vishay Dale
TDK
C9 ,C11.
C13
open
C2012X7R1C105K
1uF/16V, 0805 Ceramic cap.
OR
GRM21BR71C105KA01B. muRata
VJ1206S105KXJAT
Vishay Victramon
OR
C14
C15
C16
R2
DIN
VJ0603A102KXXAT
Vishay Victramon
Vishay Victramon
Vishay
1000pF /25V, 0603 , NPO
0.1uF/25V Ceramic cap.
470 Ohm , 0603, 1/16W, 5%.
10K / 0805 1/10W, 1%
3.16K /0805, 1/10W , 1%
4.64K /0805, 1/10W , 1%
11.3K / 0805, 1/10W, 1%
4.02K ,0603,1/16W, 1%
5 ohm , 1/8W , 1206 , 1%
2 Ohm , 1/8 W , 1206 , 1%
1.4 Ohm , 1/8 W , 1206 , 1%
Open
VJ0603Y104KXXAT
CRCW06034700JRT1
CRCW08051002FRT1
CRCW08053161FRT1
CRCW08054641FRT1
CRCW08051132FRT1
CRCW06034021FRT1
CRCW12065R00FRT1
CRCW12062R00FRT1
CRCW12061R40FRT1
R3
Vishay
R4
Vishay
R5
Vishay
R6
Vishay
R8
Vishay
R9,
Vishay
R10
R12
R14
J1, J3, J4, J5
Vishay
Vishay
2551-2-00-01-00-00-07-0
MilMax
Turret Pins
Notes:
1. Micrel.Inc
2. Vishay corp
3. Diodes. Inc
4. Sumida
408-944-0800
206-452-5664
805-446-4800
408-321-9660
847-803-6100
800-831-9172
843-448-9411
847-803-6100
5. TDK
6. muRata
7. AVX
8. International Rectifier
9. Fairchild Semiconductor
10. Cooper Electronic
11. Coilcraft
207-775-8100
561-752-5000
1-800-322-2645
631-435-1110
12. Central Semi
March 2009
14
M9999-032409
MIC2169
Micrel
Package Information
10-Pin MSOP (MM)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL + 1 (408) 944-0800 FAX + 1 (408) 944-0970 WEB http://www.micrel.com
The information furnished by Micrel in this datasheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use.
Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can
reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into
the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s
use or sale of Micrel Products for use in life support appliances, devices or systems is at Purchaser’s own risk and Purchaser agrees to fully indemnify
Micrel for any damages resulting from such use or sale.
© 2005 Micrel, Incorporated.
March 2009
15
M9999-032409
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