MIC26603-ZAYJL-TR [MICROCHIP]
IC REG BUCK ADJ 6A SYNC 28MLF;
| 型号： | MIC26603-ZAYJL-TR |
| 厂家： | 
MICROCHIP |
| 描述： | IC REG BUCK ADJ 6A SYNC 28MLF 开关 |
| 文件： | 总29页 (文件大小：1350K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
MIC26603-ZA
28V, 6A Hyper Speed Control
Synchronous DC-to-DC Buck Regulator
SuperSwitcher™ II
General Description
The Micrel MIC26603-ZA is
a
constant-frequency,
synchronous buck regulator featuring a unique adaptive
ON-time control architecture. The MIC26603-ZA operates
over an input supply range of 4.5V to 28V and provides a
regulated output of up to 6A of output current. The output
voltage is adjustable down to 0.6V with a guaranteed
accuracy of ±1%, and the device operates at a switching
frequency of 600kHz.
SuperSwitcher™ II
Features
• Hyper Speed Control architecture enables
− High Delta V operation (VIN = 28V and VOUT = 0.6V)
− Small output capacitance
Micrel’s Hyper Speed Control architecture allows for
ultra-fast transient response while reducing the output
• 4.5V to 28V voltage input
capacitance and also makes (High VIN)/(Low VOUT
)
• 6A output current capability, up to 95% efficiency
• Adjustable output from 0.6V to 5.5V
• ±1% feedback accuracy
operation possible. This adaptive tON ripple control
architecture combines the advantages of fixed-frequency
operation and fast transient response in a single device.
• Any Capacitor stable - zero-to-high ESR
• 600kHz switching frequency
• No external compensation
The MIC26603-ZA offers a full suite of features to ensure
protection of the IC during fault conditions. These include
undervoltage lockout to ensure proper operation under
power-sag conditions, internal soft-start to reduce inrush
current, foldback current limit, “hiccup mode” short-circuit
protection, and thermal shutdown. An open-drain Power
Good (PG) pin is provided.
• Power Good (PG) output
• Foldback current limit and short-circuit protection
• Supports safe startup into a pre-biased load
• –40°C to +125°C junction temperature range
• 28-pin 5mm × 6mm QFN package
Datasheets and support documentation are available on
Micrel’s web site at: www.micrel.com.
Applications
• Distributed power systems
• Communications/networking infrastructure
• Set-top box, gateways, and routers
• Printers, scanners, graphic cards, and video cards
Typical Application
Efficiency (VIN = 12V)
vs. Output Current
100
5.0V
95
3.3V
90
2.5V
1.8V
85
1.5V
80
75
70
65
60
55
50
1.2V
1.0V
0.9V
0.8V
0
1
2
3
4
5
6
7
8
OUTPUT CURRENT (A)
Hyper Speed Control, SuperSwitcher, and Any Capacitor are trademarks of Micrel, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
Revision 1.1
July 16, 2014
Micrel, Inc.
MIC26603-ZA
Ordering Information
Switching
Frequency
Junction Temperature
Range
Lead
Finish
Part Number
Voltage
Package
MIC26603-ZAYJL
Adjustable
600kHz
28-Pin 5mm × 6mm QFN
–40°C to +125°C
Pb-Free
Pin Configuration
28-Pin 5mm × 6mm QFN (JL)
(Top View)
Pin Description
Pin Number
Pin Name
Pin Function
5V Internal Linear Regulator output: PVDD supply is the power MOSFET gate drive supply voltage
created by internal LDO from VIN. When VIN < +5.5V, PVDD should be tied to the PVIN pins. A 2.2µF
ceramic capacitor from the PVDD pin to PGND (pin 2) must be placed next to the IC.
1
PVDD
Power Ground: PGND is the ground path for the buck converter power stage. The PGND pins connect
to the low-side N-Channel internal MOSFET gate drive supply ground, the sources of the MOSFETs,
the negative terminals of input capacitors, and the negative terminals of output capacitors. The loop
for the power ground should be as small as possible and separate from the signal ground (SGND)
loop.
2, 5, 6, 7,
8, 21
PGND
3
NC
SW
No Connect.
Switch Node output: Internal connection for the high-side MOSFET source and low-side MOSFET
drain. Because of the high-speed switching on this pin, the SW pin should be routed away from
sensitive nodes.
4, 9, 10,
11, 12
High-Side N-internal MOSFET Drain Connection input: The PVIN operating voltage range is from 4.5V
to 28V. Input capacitors between the PVIN pins and the power ground (PGND) are required and keep
the connection short.
13,14,15,16,
17,18,19
PVIN
BST
Boost output: Bootstrapped voltage to the high-side N-channel MOSFET driver. A Schottky diode is
connected between the PVDD pin and the BST pin. A boost capacitor of 0.1μF is connected between
the BST pin and the SW pin. Adding a small resistor at the BST pin can reduce the turn-on time of
high-side N-Channel MOSFETs.
20
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Micrel, Inc.
MIC26603-ZA
Pin Description (Continued)
Pin Number
Pin Name
Pin Function
Current Sense input: The CS pin senses current by monitoring the voltage across the low-side
MOSFET during the OFF-time. The current sensing is necessary for short circuit protection. To sense
the current accurately, connect the low-side MOSFET drain to SW using a Kelvin connection. The CS
pin is also the high-side MOSFET’s output driver return.
22
CS
Signal Ground: SGND must be connected directly to the ground planes. Do not route the SGND pin to
the PGND pad on the top layer (see “PCB Layout Guidelines” for details).
23
24
25
SGND
FB
Feedback input: Input to the transconductance amplifier of the control loop. The FB pin is regulated to
0.6V. A resistor divider connecting the feedback to the output is used to adjust the desired output
voltage.
Power Good output: Open drain output. The PG pin is externally tied with a resistor to VDD. A high
output is asserted when VOUT > 92% of nominal.
PG
Enable input: A logic level control of the output. The EN pin is CMOS-compatible. Logic high = enable,
logic low = shutdown. In the off state, the supply current of the device is greatly reduced (typically
5µA). Do not leave the EN pin floating.
26
27
EN
VIN
Power Supply Voltage input: Requires a bypass capacitor to SGND.
5V Internal Linear Regulator output: VDD supply is the power MOSFET gate drive supply voltage and
the supply bus for the IC. VDD is created by internal LDO from VIN. When VIN < +5.5V, VDD should
be tied to PVIN pins. A 1µF ceramic capacitor from the VDD pin to SGND pins must be placed next to
the IC.
28
VDD
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Micrel, Inc.
MIC26603-ZA
Absolute Maximum Ratings(1)
Operating Ratings(2)
Supply Voltage (PVIN, VIN) .............................. 4.5V to 28V
PVDD, VDD Supply Voltage ............................ 4.5V to 5.5V
Enable Input (VEN).................................................. 0V to VIN
Junction Temperature (TJ) ........................−40°C to +125°C
Maximum Power Dissipation......................................Note 4
Package Thermal Resistance(4)
PVIN to PGND............................................... −0.3V to +29V
VIN to PGND................................................. −0.3V to PVIN
PVDD, VDD to PGND ..................................... −0.3V to +6V
VSW, VCS to PGND.............................−0.3V to (PVIN +0.3V)
VBST to VSW ........................................................ −0.3V to 6V
VBST to PGND.................................................. −0.3V to 35V
5mm × 6mm QFN (θJA) .....................................28°C/W
VFB, VPG to PGND............................. −0.3V to (VDD + 0.3V)
EN to PGND .......................................−0.3V to (VIN +0.3V)
V
PGND to SGND............................................ −0.3V to +0.3V
Junction Temperature ..............................................+150°C
Storage Temperature (TS).........................−65°C to +150°C
Lead Temperature (soldering, 10s)............................ 260°C
ESD Rating(3)................................................. ESD Sensitive
Electrical Characteristics(5)
PVIN = VIN = VEN = 12V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
Power Supply Input
Input Voltage Range (VIN, PVIN)
Quiescent Supply Current
Shutdown Supply Current
VDD Supply Voltage
VDD Output Voltage
4.5
28
1500
10
V
VFB = 1.5V (non-switching)
VEN = 0V
730
5
µA
µA
4.8
3.7
5.4
4.5
VIN = 7V to 28V, IDD = 40mA
VDD Rising
5
V
V
VDD UVLO Threshold
VDD UVLO Hysteresis
Dropout Voltage (VIN – VDD)
DC-to-DC Controller
Output Voltage Adjust Range (VOUT
Reference
4.2
400
380
mV
mV
600
5.5
IDD = 25mA
)
0.6
V
V
−40°C ≤ TJ ≤ 85°C
0.594
0.591
0.606
0.609
0.6
0.6
0°C ≤ TJ ≤ 85°C, ±1.0%
−40°C ≤ TJ ≤ 125°C, ±1.5%
IOUT = 0A to 6A (continuous mode)
VIN = 4.5V to 28V
Feedback Voltage
Load Regulation
Line Regulation
FB Bias Current
Notes:
0.25
0.25
50
%
%
VFB = 0.6V
nA
1. Exceeding the absolute maximum ratings may damage the device.
2. The device is not guaranteed to function outside its operating ratings.
3. Devices are ESD sensitive. Handling precautions are recommended. Human body model, 1.5k in series with 100pF.
4. PD(MAX) = (TJ(MAX) – TA)/ θJA, where θJA depends upon the printed circuit layout. A 5-in2 4 layer, 0.62”, FR-4 PCB with 2oz finish copper weight per
layer is used for the θJA.
5. Specification is for packaged product only.
Revision 1.1
July 16, 2014
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Micrel, Inc.
MIC26603-ZA
Electrical Characteristics(5) (Continued)
PVIN = VIN = VEN = 12V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Units
Max.
Enable Control
1.8
EN Logic Level High
EN Logic Level Low
EN Bias Current
V
V
0.6
30
VEN = 12V
6
µA
Oscillator
Switching Frequency(6)
Maximum Duty Cycle(7)
Minimum Duty Cycle
Minimum Off-Time
Soft-Start
600
82
kHz
%
450
750
VFB = 0V
VFB = 1.0V
0
%
300
ns
Soft-Start Time
5
ms
Short-Circuit Protection
Current-Limit Threshold
Current-Limit Threshold
Short-Circuit Current
Internal FETs
7.5
6.6
13
13
17
17
A
A
A
VFB = 0.6V, TJ = 25°C
VFB = 0.6V, TJ = 125°C
VFB = 0V
2.7
Top-MOSFET RDS (ON)
Bottom-MOSFET RDS (ON)
SW Leakage Current
VIN Leakage Current
Power Good (PG)
PG Threshold Voltage
PG Hysteresis
ISW = 1A
ISW = 1A
VEN = 0V
VEN = 0V
42
mΩ
mΩ
µA
12.5
60
25
µA
85
95
Sweep VFB from low to high
Sweep VFB from high to low
Sweep VFB from low to high
Sweep VFB < 0.9 × VNOM, IPG = 1mA
92
5.5
100
70
%VOUT
%VOUT
µs
PG Delay Time
200
PG Low Voltage
mV
Thermal Protection
Overtemperature Shutdown
Overtemperature Shutdown Hysteresis
TJ Rising
160
15
°C
°C
Notes:
6. Measured in test mode.
7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF, typically 300ns.
Revision 1.1
July 16, 2014
5
Micrel, Inc.
MIC26603-ZA
Typical Characteristics
VIN Operating Supply Current
vs. Input Voltage
VDD Output Voltage
vs. Input Voltage
VIN Shutdown Current
vs. Input Voltage
20
16
12
8
10
8
60
45
30
15
0
VEN = 0V
REN = OPEN
6
4
VOUT = 1.8V
IOUT = 0A
SWITCHING
4
2
VFB = 0.9V
IDD = 10mA
0
0
4
10
16
22
28
4
10
16
22
28
4
10
16
22
28
28
28
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Feedback Voltage
vs. Input Voltage
Total Regulation
vs. Input Voltage
Current Limit
vs. Input Voltage
0.608
0.604
0.600
0.596
0.592
1.0%
0.5%
0.0%
-0.5%
-1.0%
20
15
10
5
VOUT = 1.8V
IOUT = 0A
VOUT = 1.8V
IOUT = 0A to 6A
VOUT = 1.8V
0
4
10
16
22
28
4
10
16
22
28
4
10
16
22
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Switching Frequency
vs. Input Voltage
Enable Input Current
vs. Input Voltage
PG/VREF Ratio
vs. Input Voltage
700
650
600
550
500
16
12
8
100%
95%
90%
85%
80%
VOUT = 1.8V
IOUT = 0A
4
VEN = VIN
VREF = 0.6V
0
4
4
10
16
22
28
10
16
22
4.0
10.0
16.0
22.0
28.0
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Revision 1.1
July 16, 2014
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Micrel, Inc.
MIC26603-ZA
Typical Characteristics (Continued)
VIN Operating Supply Current
vs. Temperature
VIN Shutdown Current
vs. Temperature
VDD UVLO Threshold
vs. Temperature
20
16
12
8
20
5
4
3
2
1
0
VIN = 12V
IOUT = 0A
VEN = 0V
Rising
Falling
15
10
5
VIN = 12V
VOUT = 1.8V
IOUT = 0A
4
Hyst
SWITCHING
0
0
-50
-25
0
25
50
75
100
125
125
125
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
125
125
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
Feedback Voltage
vs. Temperature
Load Regulation
vs. Temperature
Line Regulation
vs. Temperature
0.608
0.604
0.600
0.596
0.592
1.0%
0.5%
0.0%
-0.5%
-1.0%
0.4%
0.3%
0.2%
0.1%
0.0%
VIN = 4.5V to 28V
VOUT = 1.8V
IOUT = 0A
VIN = 12V
VOUT = 1.8V
IOUT = 0A
VIN = 12V
VOUT = 1.8V
IOUT = 0A to 6A
-50
-25
0
25
50
75
100
-50
-25
0
25
50
75
100 125
-50
-25
0
25
50
75
100
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
Switching Frequency
vs. Temperature
VDD
Current Limit
vs. Temperature
vs. Temperature
700
650
600
550
500
6
5
4
3
25
20
15
10
5
VIN = 12V
VOUT = 1.8V
IOUT = 0A
VIN = 12V
IOUT = 0A
VIN = 12V
VOUT = 1.8V
2
0
-50
-25
0
25
50
75
100
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
Revision 1.1
July 16, 2014
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Micrel, Inc.
MIC26603-ZA
Typical Characteristics (Continued)
Efficiency
vs. Output Current
Feedback Voltage
vs. Output Current
Output Voltage
vs. Output Current
95
90
85
80
75
70
65
60
55
50
0.608
1.819
1.814
1.810
1.805
1.800
1.796
1.791
1.787
1.782
12VIN
0.604
0.600
0.596
0.592
24VIN
VIN = 12V
VOUT = 1.8V
VIN = 12V
VOUT = 1.8V
VOUT = 1.8V
0
1
2
3
4
5
6
6
8
0
1
2
3
4
5
6
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Switching Frequency
vs. Output Current
Line Regulation
vs. Output Current
Output Voltage (VIN = 5V)
vs. Output Current
1.0%
0.5%
0.0%
-0.5%
-1.0%
700
650
600
550
500
5
4.6
4.2
3.8
3.4
3
VIN = 5V
VFB < 0.6V
TA
25ºC
85ºC
125ºC
VIN = 4.5V to 28V
VOUT = 1.8V
VIN = 12V
VOUT = 1.8V
0
1
2
3
4
5
6
7
8
0
1
2
3
4
5
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Efficiency (VIN = 5V)
Die Temperature* (VIN = 5V)
vs. Output Current
IC Power Dissipation (VIN = 5V)
vs. Output Current
vs. Output Current (D01)
100
95
90
85
80
75
70
65
60
55
60
50
40
30
20
10
0
2.5
2.0
1.5
1.0
0.5
0.0
V
IN = 5V
VOUT = 0.8,V, 1.0V, 1.2V, 1.5V, 1.8V,2.5V, 3.3V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
3.3V
0.8V
VIN = 5V
VOUT = 1.8V
50
0
0
1
2
3
4
5
6
1
2
3
4
5
6
7
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Die Temperature* : The temperature measurement was taken at the hottest point on the MIC26603-ZA while it was case mounted on a 5in2 4-layer,
0.62”, FR-4 PCB, with 2oz finish copper weight per layer. See the “Thermal Measurements” section for more details. Actual results will depend on the
size of the PCB, ambient temperature, and proximity to other heat emitting components.
Revision 1.1
July 16, 2014
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Micrel, Inc.
MIC26603-ZA
Typical Characteristics (Continued)
Efficiency (VIN = 12V)
vs. Output Current
IC Power Dissipation (VIN = 12V)
vs. Output Current
Die Temperature* (VIN = 12V)
vs. Output Current
100
95
90
85
80
75
70
65
60
55
50
2.5
60
50
40
30
20
10
0
VIN = 12V
5.0V
3.3V
VOUT = 0.8V, 1.0V, 1.2V, 1.5V, 1.8V, 2.5V, 3.3V, 5.0V
2.0
1.5
1.0
0.5
0.0
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
5.0V
0.8V
VIN = 12V
VOUT = 1.8V
0
1
2
3
4
5
6
7
8
0
1
2
3
4
5
6
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Efficiency (VIN = 24V)
vs. Output Current
IC Power Dissipation (VIN = 24V)
vs. Output Current
Die Temperature* (VIN = 24V)
vs. Output Current
95
90
85
80
75
70
65
60
55
50
2.5
2.0
1.5
1.0
0.5
0.0
60
50
40
30
20
10
0
VIN = 24V
5.0V
VOUT = 0.8V, 1.0V, 1.2V, 1.5V, 1.8V, 2.5V, 3.3V, 5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
5.0V
0.8V
VIN = 24V
VOUT = 1.8V
0
1
2
3
4
5
6
7
8
0
1
2
3
4
5
6
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Thermal Derating*
Thermal Derating*
Thermal Derating*
vs. Ambient Temperature
vs. Ambient Temperature
vs. Ambient Temperature
12
10
8
12
10
8
12
10
8
0.8V
1.8V
0.8V
1.8V
1.5V
6
6
6
3.3V
4
4
4
VIN = 12V
OUT = 0.8, 1.2, 1.8V
VIN = 5V
OUT = 0.8, 1.2, 1.5V
VIN = 5V
V
V
VOUT = 1.8, 2.5, 3.3V
2
2
2
0
0
0
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
Die Temperature* : The temperature measurement was taken at the hottest point on the MIC26603-ZA while it was case mounted on a 5in2 4-layer,
0.62”, FR-4 PCB, with 2oz finish copper weight per layer. See the “Thermal Measurements” section for more details. Actual results will depend on the
size of the PCB, ambient temperature, and proximity to other heat emitting components.
Revision 1.1
July 16, 2014
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Micrel, Inc.
MIC26603-ZA
Typical Characteristics (Continued)
Thermal Derating*
Thermal Derating*
vs. Ambient Temperature
vs. Ambient Temperature
12
12
10
8
10
2.5V
0.8V
8
5V
6
6
2.5V
4
4
VIN = 12V
VIN = 24V
V
OUT = 2.5, 3.3, 5V
V
OUT = 0.8, 1.2, 2.5V
2
0
2
0
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
Die Temperature* : The temperature measurement was taken at the hottest point on the MIC26603-ZA while it was case mounted on a 5in2 4-layer,
0.62”, FR-4 PCB, with 2oz finish copper weight per layer. See the “Thermal Measurements” section for more details. Actual results will depend on the
size of the PCB, ambient temperature, and proximity to other heat emitting components.
Revision 1.1
July 16, 2014
10
Micrel, Inc.
MIC26603-ZA
Functional Characteristics
Revision 1.1
July 16, 2014
11
Micrel, Inc.
MIC26603-ZA
Functional Characteristics (Continued)
Revision 1.1
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Micrel, Inc.
MIC26603-ZA
Functional Characteristics (Continued)
Revision 1.1
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MIC26603-ZA
Functional Diagram
Figure 1. MIC26603-ZA Block Diagram
Revision 1.1
July 16, 2014
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Micrel, Inc.
MIC26603-ZA
Functional Description
The MIC26603-ZA is an adaptive ON-time synchronous
step-down DC-DC regulator with an internal 5V linear
regulator and a Power Good (PG) output. It is designed
to operate over a wide input voltage range from 4.5V to
28V and provides a regulated output voltage at up to 7A
of output current. An adaptive ON-time control scheme is
used to get a constant switching frequency and to
simplify the control compensation. Overcurrent protection
is implemented without using an external sense resistor.
The device includes an internal soft-start function that
reduces the power supply input surge current at start-up
by controlling the output voltage rise time.
The maximum duty cycle is derived from the 300ns
tOFF(min)
:
tS − tOFF(MIN)
300ns
tS
DMAX
=
= 1−
Eq. 2
tS
where tS = 1/600kHz = 1.66µs.
Using MIC26603-ZA with an OFF-time close to tOFF(min)
during steady-state operation is not recommended. Also,
as VOUT increases, the internal ripple injection increases
and reduces the line regulation performance. Therefore,
the maximum output voltage of the MIC26603-ZA should
be limited to 5.5V and the maximum external ripple
injection should be limited to 200mV. Please refer to the
“Setting Output Voltage” subsection in Application
Information for more details.
Theory of Operation
The MIC26603-ZA operates in a continuous mode, as
shown in Figure 1.
Continuous Mode
In continuous mode, the output voltage is sensed by the
MIC26603-ZA feedback pin FB through the voltage
divider R1 and R2. It is then compared to a 0.6V
reference voltage VREF at the error comparator through a
low-gain transconductance (gm) amplifier. If the feedback
voltage decreases and the output of the gm amplifier is
below 0.6V, then the error comparator will trigger the
control logic and generate an ON-time period. The ON-
time period length is predetermined by the “FIXED tON
ESTIMATION” circuitry:
The actual ON-time and resulting switching frequency will
vary with the part-to-part variation in the rise and fall
times of the internal MOSFETs, the output load current,
and variations in the VDD voltage. Also, the minimum tON
results in a lower switching frequency in high VIN to VOUT
applications, such as 24V to 1.0V. The minimum tON
measured on the MIC26603-ZA evaluation board is about
100ns. During load transients, the switching frequency is
changed because of the varying OFF-time.
To illustrate the control loop operation, the datasheet will
discuss both the steady-state and load transient
scenarios. Figure 2 shows the MIC26603-ZA control loop
timing during steady-state operation. During steady-state
operation, the gm amplifier senses the feedback voltage
ripple, which is proportional to the output voltage ripple
and the inductor current ripple, to trigger the ON-time
period. The ON-time is predetermined by the tON
estimator. The termination of the OFF-time is controlled
by the feedback voltage. At the valley of the feedback
VOUT
tON(ESTIMATED)
=
Eq. 1
V × 600kHz
IN
where VOUT is the output voltage and VIN is the power
stage input voltage.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends on the feedback voltage in most
cases. When the feedback voltage decreases and the
output of the gm amplifier is below 0.6V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
300ns, the MIC26603-ZA control logic will apply the
tOFF(min) instead. tOFF(min) is required to maintain enough
energy in the boost capacitor (CBST) to drive the high-side
MOSFET.
voltage ripple, which occurs when VFB falls below VREF
,
the OFF-time period ends and the next ON-time period is
triggered through the control logic circuitry.
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MIC26603-ZA
Unlike true current-mode control, the MIC26603-ZA uses
the output voltage ripple to trigger an ON-time period.
The output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough. The MIC26603-ZA control loop has the advantage
of eliminating the need for slope compensation.
To meet the stability requirements, the MIC26603-ZA
feedback voltage ripple should be in phase with the
inductor current ripple and large enough to be sensed by
the gm amplifier and the error comparator. The
recommended feedback voltage ripple is 20mV~100mV.
If a low-ESR output capacitor is selected, then the
feedback voltage ripple may be too small to be sensed by
the gm amplifier and the error comparator. Also, the
output voltage ripple and the feedback voltage ripple are
not necessarily in phase with the inductor current ripple if
the ESR of the output capacitor is very low. In these
cases, ripple injection is required to ensure proper
operation. Please refer to the “Ripple Injection”
subsection in Application Information for more details
about the ripple injection technique.
Figure 2. MIC26603-ZA Control Loop Timing
Figure 3 shows the operation of the MIC26603-ZA during
a load transient. The output voltage drops because of the
sudden load increase, which makes the VFB less than
VREF. This causes the error comparator to trigger an ON-
time period. At the end of the ON-time period, a minimum
OFF-time tOFF(min) is generated to charge CBST because
the feedback voltage is still below VREF. Then, the next
ON-time period is triggered by the low feedback voltage.
Therefore, the switching frequency changes during the
load transient, but returns to the nominal fixed frequency
after the output has stabilized at the new load current
level. With the varying duty cycle and switching
frequency, the output recovery time is fast and the output
voltage deviation is small in a MIC26603-ZA converter.
VDD Regulator
The MIC26603-ZA provides a 5V regulated output for
input voltage VIN ranging from 5.5V to 28V. When
VIN < 5.5V, VDD should be tied to PVIN pins to bypass
the internal linear regulator.
Soft-Start
Soft-start reduces the power supply input surge current at
startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is charged
up. A slower output rise time draws a lower input surge
current.
The MIC26603-ZA implements an internal digital soft-
start by making the 0.6V reference voltage VREF ramp
from 0 to 100% in about 6ms in 9.7mV steps. Therefore,
the output voltage is controlled to increase slowly by a
stair-case VFB ramp. After the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function correctly.
Current Limit
The MIC26603-ZA uses the RDS(ON) of the internal low-
side power MOSFET to sense overcurrent conditions.
This method avoids adding cost, board space, and power
losses taken by a discrete current sense resistor. The
low-side MOSFET is used because it displays much
lower parasitic oscillations during switching than the high-
side MOSFET.
Figure 3. MIC26603-ZA Load Transient Response
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MIC26603-ZA
In each switching cycle of the MIC26603-ZA converter,
the inductor current is sensed by monitoring the low-side
MOSFET in the OFF-time period. If the peak inductor
current is greater than 13A, then the MIC26603-ZA turns
off the high-side MOSFET and a soft-start sequence is
triggered. This mode of operation is called “hiccup mode”
and its purpose is to protect the downstream load in case
of a hard short. The load current-limit threshold has a
foldback characteristic related to the feedback voltage, as
shown in Figure 4.
MOSFET Gate Drive
The block diagram (Figure 1) shows a bootstrap circuit,
consisting of D1 (a Schottky diode is recommended) and
CBST. This circuit supplies energy to the high-side drive
circuit. Capacitor CBST is charged, while the low-side
MOSFET is on, and the voltage on the SW pin is
approximately 0V. When the high-side MOSFET driver is
turned on, energy from CBST is used to turn the MOSFET
on. As the high-side MOSFET turns on, the voltage on
the SW pin increases to approximately VIN. Diode D1 is
reverse biased and CBST floats high while continuing to
keep the high-side MOSFET on. The bias current of the
high-side driver is less than 10mA so a 0.1μF to 1μF
capacitor is enough to hold the gate voltage with minimal
droop for the power stroke (high-side switching) cycle,
that is, ΔBST = 10mA × 1.67μs/0.1μF = 167mV. When
the low-side MOSFET is turned back on, CBST is
recharged through D1. A small resistor RG, in series with
CBST, can be used to slow down the turn-on time of the
high-side N-channel MOSFET.
Current Limit Threshold
vs. Feedback Voltage
20
16
12
8
The drive voltage is derived from the VDD supply voltage.
The nominal low-side gate drive voltage is VDD and the
nominal high-side gate drive voltage is approximately
VDD – VDIODE, where VDIODE is the voltage drop across D1.
An approximate 30ns delay between the high-side and
low-side driver transitions is used to prevent current from
simultaneously flowing unimpeded through both
MOSFETs.
4
0
0.0
0.2
0.4
0.6
0.8
1.0
FEEDBACK VOLTAGE (V)
Figure 4. MIC26603-ZA Current-Limit
Foldback Characteristic
Power Good (PG)
The Power Good (PG) pin is an open-drain output that
indicates logic high when the output is nominally 92% of
its steady state voltage. A pull-up resistor of more than
10kΩ should be connected from PG to VDD.
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MIC26603-ZA
Application Information
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents increase the power
dissipation in the inductor and MOSFETs. Larger output
ripple currents also require more output capacitance to
smooth out the larger ripple current. Smaller peak-to-
peak ripple currents require a larger inductance value
and therefore a larger and more expensive inductor. A
good compromise between size, loss, and cost is to set
the inductor ripple current equal to 20% of the maximum
output current. The inductance value is calculated in
Equation 3.
Maximizing efficiency requires the selecting the proper
core material and minimizing the winding resistance. The
high-frequency operation of the MIC26603-ZA requires
the use of ferrite materials for all but the most cost-
sensitive applications. Lower-cost iron powder cores may
be used but the increase in core loss reduces the
efficiency of the power supply. This is especially
noticeable at low output power. The winding resistance
decreases efficiency at the higher output current levels.
The winding resistance must be minimized, although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of the
core and copper losses. At higher output loads, the core
losses are usually insignificant and can be ignored. At
lower output currents, the core losses can be a significant
contributor. Core loss information is usually available
from the magnetics vendor. Copper loss in the inductor is
calculated by Equation 7.
VOUT × (VIN(MAX) − VOUT
)
L =
Eq. 3
VIN(MAX) × fSW × 20%×IOUT(MAX)
2
P
= IL(RMS × RWINDING
Eq. 7
INDUCTOR(CU)
)
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding resistance
used should be at the operating temperature.
where:
fSW = switching frequency, 600kHz
20% = ratio of AC ripple current to DC output current
VIN(MAX) = maximum power stage input voltage
PWINDING(Ht) = RWINDING(20°C) × (1+ 0.0042 × (TH − T20°C ))
Eq. 8
The peak-to-peak inductor current ripple is:
where:
TH = temperature of wire under full load
T20°C = ambient temperature
VOUT × (VIN(MAX) − VOUT
VIN(MAX) × fSW × L
)
∆IL(PP)
=
Eq. 4
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
Output Capacitor Selection
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
The type of output capacitor is usually determined by its
equivalent series resistance (ESR). Voltage and RMS
current capability are two other important factors.
Recommended capacitor types are ceramic, low-ESR
aluminum electrolytic, OS-CON and POSCAP. The
output capacitor’s ESR is usually the main cause of the
output ripple. The output capacitor ESR also affects the
stability of the control loop.
IL(PK) = IOUT(MAX) + 0.5 × ∆IL(PP)
Eq. 5
The RMS inductor current is used to calculate the I2R
losses in the inductor.
2
∆IL(PP)
2
IL(RMS) = IOUT(MAX)
+
Eq. 6
12
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MIC26603-ZA
The power dissipated in the output capacitor is:
The maximum value of ESR is calculated using Equation 9.
2
∆VOUT(PP)
PDISS(C
= IC
× ESRC
OUT (RMS)
Eq. 12
)
OUT
OUT
ESRC
≤
Eq. 9
OUT
∆IL(PP)
Input Capacitor Selection
where:
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage derating. The input
voltage ripple primarily depends on the input capacitor’s
ESR. The peak input current is equal to the peak inductor
current, so:
ΔVOUT(pp) = peak-to-peak output voltage ripple
ΔIL(PP) = peak-to-peak inductor current ripple
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 10.
2
∆IL(PP)
2
∆VOUT(PP)
=
+
(
∆IL(PP) × ESRC
)
OUT
COUT ×fSW × 8
Eq. 10
∆VIN = IL(PK) × ESRC
Eq. 13
IN
where:
D = duty cycle
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming the
peak-to-peak inductor current ripple is low:
COUT = output capacitance value
fSW = switching frequency
As described in the “Theory of Operation” subsection in
Functional Description, the MIC26603-ZA requires at
least 20mV peak-to-peak ripple at the FB pin to make the
gm amplifier and the error comparator behave properly.
Also, the output voltage ripple should be in phase with
the inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide enough feedback
voltage ripple. Please refer to the “Ripple Injection”
subsection for more details.
IC (RMS) ≈ IOUT(MAX) × D × (1− D)
Eq. 14
IN
The power dissipated in the input capacitor is:
PDISS(C = IC
× ESRC
IN
Eq. 15
)
IN(RMS)
IN
The voltage rating of the capacitor should be twice the
output voltage for tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated in Equation 11.
Ripple Injection
The VFB ripple required for proper operation of the
MIC26603-ZA gm amplifier and error comparator is 20mV
to 100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. For a low
output voltage, such as a 1V, the output voltage ripple is
only 10mV to 20mV, and the feedback voltage ripple is
less than 20mV. If the feedback voltage ripple is so small
that the gm amplifier and error comparator can’t sense it,
then the MIC26603-ZA will lose control and the output
voltage is not regulated.
∆IL(PP)
IC
=
Eq. 11
OUT (RMS)
12
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MIC26603-ZA
In order to have some amount of VFB ripple, a ripple
injection method is applied for low output voltage ripple
applications.
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1. Enough ripple at the feedback voltage caused by the
large ESR of the output capacitors.
As shown in Figure 5, the converter is stable without
any ripple injection. The feedback voltage ripple is:
Figure 6. Inadequate Ripple at FB
R2
∆VFB(PP)
=
× ESRC
× ∆IL(PP)
Eq. 16
OUT
R1+ R2
where ΔIL(pp) is the peak-to-peak value of the inductor
current ripple.
2. Inadequate ripple at the feedback voltage caused by
the small ESR of the output capacitors.
Figure 7. Invisible Ripple at FB
The output voltage ripple is fed into the FB pin
through a feedforward capacitor Cff in this situation,
as shown in Figure 6. The typical Cff value is between
1nF and 100nF. With the feedforward capacitor, the
feedback voltage ripple is very close to the output
voltage ripple:
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor RINJ and a
capacitor CINJ, as shown in Figure 7. The injected ripple
is:
∆VFB(PP) ≈ ESR × ∆IL(PP)
Eq. 17
1
∆VFB(PP) = VIN × KDIV × D × (1− D)×
Eq. 18
Eq. 19
fSW × τ
3. Virtually no ripple at the FB pin voltage due to the
very-low ESR of the output capacitors.
R1/R2
KDIV
=
RINJ + R1//R2
where:
VIN = power stage input voltage
D = duty cycle
fSW = switching frequency
τ = (R1//R2//RINJ) × Cff
Figure 5. Enough Ripple at FB
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MIC26603-ZA
In Equations 18 and 19, it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
A typical value of R1 can be between 3kΩ and 10kΩ. If
R1 is too large, it may allow noise to be introduced into
the voltage feedback loop. If R1 is too small, it will
decrease the efficiency of the power supply, especially at
light loads. Once R1 is selected, R2 can be calculated
using Equation 24.
1
T
=
<< 1
Eq. 20
fsw × τ
τ
VFB × R1
VOUT − VFB
R2 =
Eq. 24
If the voltage divider resistors R1 and R2 are in the kΩ
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor CINJ is used, which could be considered as
short for a wide range of the frequencies.
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in kΩ range.
Step 2. Select RINJ according to the expected feedback
voltage ripple using Equation 19.
Figure 8. Voltage-Divider Configuration
∆VFB(PP)
fSW × τ
KDIV
=
×
Eq. 21
VIN
D × (1− D)
In addition to the external ripple injection added at the FB
pin, internal ripple injection is added at the inverting input
of the comparator inside the MIC26603-ZA, as shown in
Figure 9. The inverting input voltage VINJ is clamped to
1.2V. As VOUT increases, the swing of VINJ is clamped.
The clamped VINJ reduces the line regulation because it is
reflected as a DC error on the FB terminal. Therefore, the
maximum output voltage of the MIC26603-ZA should be
limited to 5.5V to avoid this problem.
Then the value of RINJ is obtained as:
1
RINJ = (R1//R2)×
− 1
Eq. 22
KDIV
Step 3. Select CINJ as 100nF, which could be considered
as short for a wide range of the frequencies.
Setting Output Voltage
The MIC26603-ZA requires two resistors to set the output
voltage, as shown in Figure 8.
The output voltage is determined by Equation 23:
R1
R2
VOUT = VFB × 1+
Eq. 23
where VFB = 0.6V.
Figure 9. Internal Ripple Injection
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MIC26603-ZA
Thermal Measurements
Measuring the IC’s case temperature is recommended to
ensure that it is within its operating limits. Although this
might seem like an elementary task, it is easy to get false
results. The most common mistake is to use the standard
thermal couple that comes with a thermal meter. This
thermal couple wire gauge is large, typically 22 gauge,
and behaves like a heatsink, resulting in a lower case
measurement.
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared thermometer. If
a thermal couple wire is used, it must be constructed of
36 gauge wire or higher (smaller wire size) to minimize
the wire heat-sinking effect. In addition, the thermal
couple tip must be covered in either thermal grease or
thermal glue to make sure that the thermal couple
junction is making good contact with the case of the IC.
Omega brand thermal couple (5SC-TT-K-36-36) is
adequate for most applications.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor IC. However, an IR
thermometer from Optris has a 1mm spot size, which
makes it a good choice for measuring the hottest point on
the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
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MIC26603-ZA
PCB Layout Guidelines
Note: To minimize EMI and output noise, follow these
layout recommendations.
Inductor
•
•
•
•
Keep the inductor connection to the switch node
(SW) short.
PCB layout is critical to achieve reliable, stable, and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power, signal,
and return paths.
Do not route any digital lines underneath or close to
the inductor.
Keep the switch node (SW) away from the feedback
(FB) pin.
Follow these guidelines to ensure proper MIC26603-ZA
regulator operation:
Connect the CS pin directly to the SW pin to
accurately sense the voltage across the low-side
MOSFET.
IC
•
A 2.2µF ceramic capacitor, which is connected to the
PVDD pin, must be located right at the IC. The PVDD
pin is very noise sensitive, so placement of the
capacitor is critical. Use wide traces to connect to the
PVDD and PGND pins.
•
•
To minimize noise, place a ground plane underneath
the inductor.
The inductor can be placed on the opposite side of
the PCB with respect to the IC. It does not matter
whether the IC or inductor is on the top or bottom as
long as there is enough air flow to keep the power
components within their temperature limits. The input
and output capacitors must be placed on the same
side of the board as the IC.
•
A 1µF ceramic capacitor must be placed right
between VDD and the signal ground (SGND). SGND
must be connected directly to the ground planes. Do
not route the SGND pin to the PGND pad on the top
layer.
•
•
Place the IC close to the point-of-load (POL).
Output Capacitor
Use fat traces to route the input and output power
lines.
•
•
•
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
•
Keep signal and power grounds separate and
connected at only one location.
Phase margin changes as the output capacitor value
and ESR changes. Contact the factory if the output
capacitor is different from what is shown in the BOM.
Input Capacitor
•
•
Place the input capacitor next.
The feedback trace should be separate from the
power trace and connected as near as possible to the
output capacitor. Sensing a long high current load
trace can degrade the DC load regulation.
Place the input capacitor on the same side of the
board and as close to the IC as possible.
•
•
•
•
Keep both the PVIN pin and PGND connections
short.
Optional RC Snubber
Place several vias to the ground plane close to the
input capacitor ground terminal.
•
Place the RC snubber on either side of the board and
as close to the SW pin as possible.
Use either X7R or X5R dielectric input capacitors. Do
not use Y5V or Z5U type capacitors.
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
•
•
If a Tantalum input capacitor is placed in parallel with
the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the
overvoltage spike seen on the input supply when
power is suddenly applied.
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Micrel, Inc.
MIC26603-ZA
Evaluation Board Schematic
Figure 10. Schematic of MIC26603-ZA Evaluation Board
(J11, R13, R15 are for testing purposes)
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Micrel, Inc.
MIC26603-ZA
Bill of Materials
Item
Part Number
Manufacturer
Description
Qty.
C1
Open
12105C475KAZ2A
GRM32ER71H475KA88L
C3225X7R1H475K
Open
AVX(8)
Murata(9)
TDK(10)
C2, C3
4.7µF Ceramic Capacitor, X7R, Size 1210, 50V
2
C4, C13, C15
C5
12106D107MAT2A
GRM32ER60J107ME20L
C3225X5R0J107M
06035C104KAT2A
GRM188R71H104KA93D
C1608X7R1H104K
0603ZC105KAT2A
GRM188R71A105KA61D
C1608X7R1A105K
0603ZD225KAT2A
GRM188R61A225KE34D
C1608X5R1A225K
06035C472KAZ2A
GRM188R71H472K
C1608X7R1H472K
B41851F7227M
AVX
Murata
TDK
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
1.0µF Ceramic Capacitor, X7R, Size 0603, 10V
2.2µF Ceramic Capacitor, X5R, Size 0603, 10V
1
3
1
1
AVX
C6, C7, C10
Murata
TDK
AVX
C8
Murata
TDK
AVX
C9
Murata
TDK
AVX
C12
Murata
TDK
EPCOS(11)
4.7nF Ceramic Capacitor, X7R, Size 0603, 50V
220µF Aluminum Capacitor, 35V
1
1
1
C14
C11, C16
Open
SD103AWS
MCC(12)
Diodes Inc.(13)
Vishay(14)
40V, 350mA, Schottky Diode, SOD323
D1
SD103AWS-7
SD103AWS
Cooper
L1
HCF1305-2R2-R
2.2µH Inductor, 15A Saturation Current
1
Bussmann(15)
R1
R2
CRCW06032R21FKEA
CRCW06032R00FKEA
CRCW060319K6FKEA
CRCW06032K49FKEA
CRCW06034K99FKEA
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
2.21Ω Resistor, Size 0603, 1%
2.00Ω Resistor, Size 0603, 1%
19.6kΩ Resistor, Size 0603, 1%
2.49kΩ Resistor, Size 0603, 1%
4.99kΩ Resistor, Size 0603, 1%
1
1
1
1
1
R3
R4
R5
Notes:
8. AVX: www.avx.com.
9. Murata: www.murata.com.
10. TDK: www.tdk.com.
11. EPCOS: www.epcos.com.
12. MCC: www.mccsemi.com.
13. Diodes Inc.: www.diodes.com.
14. Vishay: www.vishay.com.
15. Cooper Bussmann: www.cooperbussmann.com.
Revision 1.1
July 16, 2014
25
Micrel, Inc.
MIC26603-ZA
Bill of Materials (Continued)
Item
R6
Part Number
Manufacturer
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Description
Qty.
1
CRCW06033K74FKEA
CRCW06032K49FKEA
CRCW06031K65FKEA
CRCW06031K24FKEA
CRCW0603787RFKEA
CRCW0603549RFKEA
CRCW0603340RFKEA
CRCW06030000FKEA
CRCW060310K0FKEA
CRCW060349R9FKEA
CRCW06031R21FKEA
Open
3.74kΩ Resistor, Size 0603, 1%
2.49kΩ Resistor, Size 0603, 1%
1.65kΩ Resistor, Size 0603, 1%
1.24kΩ Resistor, Size 0603, 1%
787Ω Resistor, Size 0603, 1%
549Ω Resistor, Size 0603, 1%
340Ω Resistor, Size 0603, 1%
0Ω Resistor, Size 0603, 5%
10.0kΩ Resistor, Size 0603, 1%
49.9Ω Resistor, Size 0603, 1%
1.21Ω Resistor, Size 0603, 1%
R7
1
R8
1
R9
1
R10
1
R11
1
R12
1
R13
1
R14, R17
R15
2
1
R16, R18
R20
2
28V, 6A Hyper Speed Control Synchronous
DC-to-DC Buck Regulator
MIC26603-ZAYJL
Micrel. Inc.(16 )
U1
1
Note:
16. Micrel, Inc.: www.micrel.com.
Revision 1.1
July 16, 2014
26
Micrel, Inc.
MIC26603-ZA
PCB Layout Recommendations
MIC26603-ZA Evaluation Board Top Layer
MIC26603-ZA Evaluation Board Mid-Layer 1 (Ground Plane)
Revision 1.1
July 16, 2014
27
Micrel, Inc.
MIC26603-ZA
PCB Layout Recommendations (Continued)
MIC26603-ZA Evaluation Board Mid-Layer 2
MIC26603-ZA Evaluation Board Bottom Layer
Revision 1.1
July 16, 2014
28
Micrel, Inc.
MIC26603-ZA
Package Information and Recommended Land Pattern(17)
28-Pin 5mm × 6mm QFN (JL)
Note:
17. Package information is correct as of the publication date. For updates and most current information, go to www.micrel.com.
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2013 Micrel, Incorporated.
Revision 1.1
July 16, 2014
29
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